AD8306

a 5 MHz–400 MHz 100 dB High Precision Limiting-Logarithmic Amplifier AD8306 FUNCTIONAL BLOCK DIAGRAM SIX STAGES TOTAL GAIN 72dB INHI 12dB INLO LADR ATTEN 4 DET BIAS CTRL I–V TEN DETECTORS SPACED 12dB 12dB 12dB LIM LMLO LMDR TYP GAIN 18dB LMHI FEATURES Complete, Fully Calibrated Log-Limiting IF Amplifier 100 dB Dynamic Range: –91 dBV to +9 dBV Stable RSSI Scaling Over Temperature and Supplies: 20 mV/dB Slope, –95 dBm Intercept 0.4 dB RSSI Linearity up to 200 MHz Programmable Limiter Gain and Output Current Differential Outputs to 10 mA, 2.4 V p-p Overall Gain 90 dB, Bandwidth 400 MHz Constant Phase (Typical 56 ps Delay Skew) Single Supply of +2.7 V to +6.5 V at 16 mA Typical Fully Differential Inputs, RIN = 1 k , C IN = 2.5 pF 500 ns Power-Up Time, 4.5 V. For frequencies in the range 10 MHz to 200 MHz these high drive levels are easily achieved using a matching network. Using such a network, having an inductor at the input, the input transient is eliminated. Limiter Output Interface The simplified limiter output stage is shown in Figure 22. The bias for this stage is provided by a temperature-stable reference voltage of nominally 400 mV which is forced across the external resistor RLIM connected from Pin 9 (LMDR, or limiter drive) by a special op amp buffer stage. The biasing scheme also introduces a slight “lift” to this voltage to compensate for the finite current gain of the current source Q3 and the output transistors Q1 and Q2. A maximum current of 10 mA is permissible (RLIM = 40 Ω). In special applications, it may be desirable to modulate the bias current; an example of this is provided in the Applications section. Note that while the bias currents are temperature stable, the ac gain of this stage will vary with temperature, by –6 dB over a 120°C range. A pair of supply and temperature stable complementary currents is generated at the differential output LMHI and LMLO (Pins 12 and 13), having a square wave form with rise and fall times of typically 0.6 ns, when load resistors of 50 Ω are used. The voltage at these output pins may swing to 1.2 V below the supply voltage applied to VPS2 (Pin 15). Because of the very high gain bandwidth product of this amplifier considerable care must be exercised in using the limiter outputs. The minimum necessary bias current and voltage swings should be used. These outputs are best utilized in a fully-differential mode. A flux-coupled transformer, a balun, or an output matching network can be selected to transform these voltages to a single-sided form. Equal load resistors are recommended, even when only one output pin is used, and these should always be returned to the same well decoupled node on the PC board. When the AD8306 is used only to generate an RSSI output, the limiter should be completely disabled by omitting RLIM and strapping LMHI and LMLO to VPS2. VPS2 LMHI LMLO 50k COMM 4k Figure 20. Enable Interface Input Interface Figure 21 shows the essentials of the signal input interface. The parasitic capacitances to ground are labeled CP; the differential input capacitance, CD, mainly due to the diffusion capacitance of Q1 and Q2. In most applications both input pins are accoupled. The switch S closes when Enable is asserted. When disabled, the inputs float, bias current IE is shut off, and the coupling capacitors remain charged. If the log amp is disabled for long periods, small leakage currents will discharge these capacitors. If they are poorly matched, charging currents at power-up can generate a transient input voltage which may block the lower reaches of the dynamic range until it has become much less than the signal. VPS1 S 1.78V 3.65k CC SIGNAL INPUT CC INLO INHI 1.725V RIN = 1k 1.725V 2.6k (TOP-END DETECTORS) CP COMM CP 130 GAIN BIAS 1.26V 3.4mA PTAT 3.65k IB = 15mA CD 2.5pF 67 Q1 20e RIN = 3k Q2 20e TO STAGES 1 THRU 5 67 TO 2ND STAGE 1.3k 1.3k Q1 FROM FINAL LIMITER STAGE 4e Q2 4e 400mV Q3 2.6k 1.3k 1.3k COM1 LMDR RLIM OA ZERO-TC Figure 21. Signal Input Interface Figure 22. Limiter Output Interface RSSI Output Interface In most applications, the input signal will be single-sided, and may be applied to either Pin 4 or 5, with the remaining pin accoupled to ground. Under these conditions, the largest input signal that can be handled is –3 dBV (sine amplitude of 1 V) when operating from a 3 V supply; a +3 dBV input may be –8– The outputs from the ten detectors are differential currents, having an average value that is dependent on the signal input level, plus a fluctuation at twice the input frequency. The currents are summed at the internal nodes LGP and LGN shown in Figure 23. A further current IT is added to LGP, to position REV. A AD8306 the intercept to –108 dBV, by raising the RSSI output voltage for zero input, and to provide temperature compensation, resulting in a stable intercept. For zero signal conditions, all the detector output currents are equal. For a finite input, of either polarity, their difference is converted by the output interface to a singlesided voltage nominally scaled 20 mV/dB (400 mV per decade), at the output VLOG (Pin 16). This scaling is controlled by a separate feedback stage, having a tightly controlled transconductance. A small uncertainty in the log slope and intercept remains (see Specifications); the intercept may be adjusted (see Applications). VPS2 CURRENT MIRROR ISOURCE >50mA ON DEMAND FLTR C1 3.5pF CF VLOG 20mV/dB range: a 60 Hz hum, picked up due to poor grounding techniques; spurious coupling from digital logic on the same PC board; a strong EMI source; etc. Very careful shielding is essential to guard against such unwanted signals, and also to minimize the likelihood of instability due to HF feedback from the limiter outputs to the input. With this in mind, the minimum possible limiter gain should be used. Where only the logarithmic amplifier (RSSI) function is required, the limiter should be disabled by omitting RLIM and tying the outputs LMHI and LMLO directly to VPS2. A good ground plane should be used to provide a low impedance connection to the common pins, for the decoupling capacitor(s) used at VPS1 and VPS2, and at the output ground. Note that COM2 is a special ground pin serving just the RSSI output. The four pins labeled PADL tie down directly to the metallic lead frame, and are thus connected to the back of the chip. The process on which the AD8306 is fabricated uses a bonded-wafer technique to provide a silicon-on-insulator isolation, and there is no junction or other dc path from the back side to the circuitry on the surface. These paddle pins must be connected directly to the ground plane using the shortest possible lead lengths to minimize inductance. The voltages at the two supply pins should not be allowed to differ greatly; up to 500 mV is permissible. It is desirable to allow VPS1 to be slightly more negative than VPS2. When the primary supply is greater than 2.7 V, the decoupling resistors R1 and R2 (Figure 24) may be increased to improve the isolation and lower the dissipation in the IC. However, since VPS2 supports the RSSI load current, which may be large, the value of R2 should take this into account. Basic Connections for Log (RSSI) Output SUMMED 1.3k DETECTOR OUTPUTS LGP LGN IT VLOG 1.3k 3.3k 3.3k ISINK FIXED 1mA 250 s TRANSCONDUCTANCE DETERMINES SLOPE 125 A COMM Figure 23. Simplified RSSI Output Interface The RSSI output bandwidth, fLP, is nominally 3.5 MHz. This is controlled by the compensation capacitor C1, which may be increased by adding an external capacitor, CF, between FLTR (Pin 10) and VLOG (Pin 16). An external 33 pF will reduce fLP to 350 kHz, while 360 pF will set it to 35 kHz, in each case with an essentially one-pole response. In general, the relationships (for fLP in MHz) are: CF = 12.7 × 10–10 – 3.5 pF ; f LP f LP = 12.7 × 10−6 C F + 3.5 pF (1) Using a load resistance of 50 Ω or greater, and at any temperature, the peak output voltage may be at least 2.4 V when using a supply of 4.5 V, and at least 2.1 V for a 3 V supply, which is consistent with the maximum permissible input levels. The incremental output resistance is approximately 0.3 Ω at low frequencies, rising to 1 Ω at 150 kHz and 18 Ω at very high frequencies. The output is unconditionally stable with load capacitance, but it should be noted that while the peak sourcing current is over 100 mA, and able to rapidly charge even large capacitances, the internally provided sinking current is only 1 mA. Thus, the fall time from the 2 V level will be as long as 2 µs for a 1 nF load. This may be reduced by adding a grounded load resistance. USING THE AD8306 Figure 24 shows the connections required for most applications. The AD8306 is enabled by connecting ENBL to VPS1. The device is put into the sleep mode by grounding this pin. The inputs are ac-coupled by C1 and C2, which normally should have the same value (CC). The input is, in this case, terminated with a 52.3 Ω resistor that combines with the AD8306’s input resistance of 1000 Ω to give a broadband input impedance of 50 Ω. Alternatively an input matching network can be used (see Input Matching section). R1 10 0.1 F 2 VPS1 C1 0.01 F SIGNAL INPUTS C2 0.01 F RT 52.3 3 PADL 4 INHI 5 INLO 6 PADL 7 COM1 ENABLE 8 ENBL VPS2 15 PADL 14 LMHI 13 LMLO 12 PADL 11 FLTR 10 LMDR 9 CF (OPTIONAL SEE TEXT) R2 10 0.1 F VS (2.7V TO 6.5V) RSSI 1 COM2 VLOG 16 AD8306 The AD8306 exhibits very high gain from 1 MHz to over 1 GHz, at which frequency the gain of the main path is still over 65 dB. Consequently, it is susceptible to all signals, within this very broad frequency range, that find their way to the input terminals. It is important to remember that these are quite indistinguishable from the “wanted” signal, and will have the effect of raising the apparent noise floor (that is, lowering the useful dynamic range). Therefore, while the signal of interest may be an IF of, say, 200 MHz, any of the following could easily be larger than this signal at the lower extremities of its dynamic REV. A –9– Figure 24. Basic Connections for RSSI (Log) Output The 0.01 µF coupling capacitors and the resulting 50 Ω input impedance give a high-pass corner frequency of around 600 kHz. (1/(2 π RC)), where C = (C1)/2. In high frequency applications, this corner frequency should be placed as high as possible, to minimize the coupling of unwanted low frequency signals. In AD8306 low frequency applications, a simple RC network forming a lowpass filter should be added at the input for the same reason. If the limiter output is not required, Pin 9 (LMDR) should be left open and Pins 12 and 13 (LMHI, LMLO) should be tied to VPS2 as shown in Figure 24. Figure 25 shows the output versus the input level in dBV, for sine inputs at 10 MHz, 50 MHz and 100 MHz (add 13 to the dBV number to get dBm Re 50 Ω. Figure 26 shows the typical logarithmic linearity (log conformance) under the same conditions. 2.5 100MHz 2 50MHz where VOUT is the demodulated and filtered RSSI output, VSLOPE is the logarithmic slope, expressed in V/dB, PIN is the input signal, expressed in decibels relative to some reference level (either dBm or dBV in this case) and PO is the logarithmic intercept, expressed in decibels relative to the same reference level. For example, for an input level of –33 dBV (–20 dBm), the output voltage will be VOUT = 0.02 V/dB × (–33 dBV – (–108 dBV)) = 1.5 V (3) The most widely used convention in RF systems is to specify power in dBm, that is, decibels above 1 mW in 50 Ω. Specification of log amp input level in terms of power is strictly a concession to popular convention; they do not respond to power (tacitly “power absorbed at the input”), but to the input voltage. The use of dBV, defined as decibels with respect to a 1 V rms sine wave, is more precise, although this is still not unambiguous because waveform is also involved in the response of a log amp, which, for a complex input (such as a CDMA signal) will not follow the rms value exactly. Since most users specify RF signals in terms of power—more specifically, in dBm/50 Ω—we use both dBV and dBm in specifying the performance of the AD8306, showing equivalent dBm levels for the special case of a 50 Ω environment. Values in dBV are converted to dBm re 50 Ω by adding 13. Output Response Time and CF RSSI OUTPUT – V 10MHz 1.5 1 0.5 0 –120 –100 –80 –60 –40 –20 INPUT LEVEL – dBV 0 20 Figure 25. RSSI Output vs. Input Level at TA = +25 °C for Frequencies of 10 MHz, 50 MHz and 100 MHz 5 4 3 2 DYNAMIC RANGE 10MHz 50MHz 100MHz 1dB 86 90 96 3dB 93 97 100 The RSSI output has a low-pass corner frequency of 3.5 MHz, which results in a 10% to 90% rise time of 73 ns. For low frequency applications, the corner frequency can be reduced by adding an external capacitor, CF, between FLTR (Pin 10) and VLOG (Pin 16) as shown in Figure 24. For example, an external 33 pF will reduce the corner frequency to 350 kHz, while 360 pF will set it to 35 kHz, in each case with an essentially one-pole response. Using the Limiter 1 100MHz 0 10MHz –1 –2 –3 –4 –5 –120 50MHz Figure 27 shows the basic connections for operating the limiter and the log output concurrently. The limiter output is a pair of differential currents of magnitude, IOUT, from high impedance (open-collector) sources. These are converted to equal-amplitude voltages by supply-referenced load resistors, RLOAD. The limiter output current is set by RLIM, the resistor connected between Pin 9 (LMDR) and ground. The limiter output current is set according the equation: IOUT = –400 mV/RLIM (5) and has an absolute accuracy of ± 5%. The supply referenced voltage on each of the limiter pins will thus be given by: VLIM = VS –400 mV × RLOAD/RLIM (6) ERROR – dB –100 –80 –60 –40 –20 INPUT LEVEL – dBV 0 20 Figure 26. Log Linearity vs. Input Level at TA = +25 °C, for Frequencies of 10 MHz, 50 MHz and 100 MHz Transfer Function in Terms of Slope and Intercept The transfer function of the AD8306 is characterized in terms of its Slope and Intercept. The logarithmic slope is defined as the change in the RSSI output voltage for a 1 dB change at the input. For the AD8306 the slope is calibrated to be 20 mV/dB. The intercept is the point at which the extrapolated linear response would intersect the horizontal axis. For the AD8306 the intercept is calibrated to be –108 dBV (–95 dBm). Using the slope and intercept, the output voltage can be calculated for any input level within the specified input range using the equation: VOUT = VSLOPE × (PIN – PO) (2) –10– REV. A AD8306 R1 10 0.1 F 2 VPS1 C1 0.01 F SIGNAL INPUTS C2 0.01 F RT 52.3 3 PADL 4 INHI 5 INLO 6 PADL 7 COM1 ENABLE 8 ENBL VPS2 15 PADL 14 LMHI 13 LMLO 12 PADL 11 FLTR 10 NC RLIM (SEE TEXT) LMDR 9 RLOAD RL 0.01 F 0.01 F 1 COM2 VLOG 16 R2 10 0.1 F VS (2.7V TO 6.5V) RSSI will be a contribution from the input noise current. Thus, the total noise will be reduced by a smaller factor. The intercept at the primary input will be lowered to –121 dBV (–108 dBm). Impedance matching and drive balancing using a flux-coupled transformer is useful whenever broadband coupling is required. However, this may not always be convenient. At high frequencies, it will often be preferable to use a narrow-band matching network, as shown in Figure 28, which has several advantages. First, the same voltage gain can be achieved, providing increased sensitivity, but now a measure of selectively is simultaneously introduced. Second, the component count is low: two capacitors and an inexpensive chip inductor are needed. Third, the network also serves as a balun. Analysis of this network shows that the amplitude of the voltages at INHI and INLO are quite similar when the impedance ratio is fairly high (i.e., 50 Ω to 1000 Ω). VS 10 1 COM2 0.1 F 2 VPS1 C1 = CM ZIN C2 = CM LM 5 INLO 6 PADL 7 COM1 8 ENBL LMLO 12 PADL 11 FLTR 10 NC RLIM LMDR 9 3 PADL 4 INHI VPS2 15 PADL 14 LMHI 13 LIMITER OUTPUT 10 VLOG 16 0.1 F RSSI AD8306 LIMITER OUTPUT NC = NO CONNECT Figure 27. Basic Connections for Operating the Limiter Depending on the application, the resulting voltage may be used in a fully balanced or unbalanced manner. It is good practice to retain both load resistors, even when only one output pin is used. These should always be returned to the same well decoupled node on the PC board (see layout of evaluation board). The unbalanced, or single-sided mode, is more inclined to result in instabilities caused by the very high gain of the signal path. The limiter current may be set as high as 10 mA (which requires RLIM to be 40 Ω) and can be optionally increased somewhat beyond this level. It is generally inadvisable, however, to use a high bias current, since the gain of this wide bandwidth signal path is proportional to the bias current, and the risk of instability is elevated as RLIM is reduced (recommended value is 400 Ω). However, as the size of RLOAD is increased, the bandwidth of the limiter output decreases from 585 MHz for RLOAD = RLIM = 50 Ω to 50 MHz for RLOAD = RLIM = 400 Ω (bandwidth = 210 MHz for RLOAD = RLIM = 100 Ω and 100 MHz for RLOAD = RLIM = 200 Ω). As a result, the minimum necessary limiter output level should be chosen while maintaining the required limiter bandwidth. For RLIM = RLOAD = 50 Ω, the limiter output is specified for input levels between –78 dBV (–65 dBm) and +9 dBV (+22 dBm). The output of the limiter may be unstable for levels below –78 dBV (–65 dBm). However, keeping RLIM above 100 Ω will make instabilities on the output less likely for input levels below –78 dBV. A transformer or a balun (e.g., MACOM part number ETC1-1-13) can be used to convert the differential limiter output voltages to a single-ended signal. Input Matching AD8306 NC = NO CONNECT Figure 28. High Frequency Input Matching Network Figure 29 shows the response for a center frequency of 100 MHz. The response is down by 50 dB at one-tenth the center frequency, falling by 40 dB per decade below this. The very high frequency attenuation is relatively small, however, since in the limiting case it is determined simply by the ratio of the AD8306’s input capacitance to the coupling capacitors. Table I provides solutions for a variety of center frequencies fC and matching from impedances ZIN of nominally 50 Ω and 100 Ω. Exact values are shown, and some judgment is needed in utilizing the nearest standard values. 14 13 12 11 10 9 8 7 6 5 4 3 2 1 0 –1 60 70 80 90 100 110 120 FREQUENCY – MHz 130 140 150 INPUT AT TERMINATION GAIN Where either a higher sensitivity or a better high frequency match is required, an input matching network is valuable. Using a flux-coupled transformer to achieve the impedance transformation also eliminates the need for coupling capacitors, lowers any dc offset voltages generated directly at the input, and usefully balances the drives to INHI and INLO, permitting full utilization of the unusually large input voltage capacity of the AD8306. The choice of turns ratio will depend somewhat on the frequency. At frequencies below 30 MHz, the reactance of the input capacitance is much higher than the real part of the input impedance. In this frequency range, a turns ratio of 2:9 will lower the effective input impedance to 50 Ω while raising the input voltage by 13 dB. However, this does not lower the effect of the short circuit noise voltage by the same factor, since there Figure 29. Response of 100 MHz Matching Network REV. A –11– DECIBELS AD8306 Table I. Step 2: Calculate CO and LO fC MHz 10 10.7 15 20 21.4 25 30 35 40 45 50 60 80 100 120 150 200 250 300 350 400 450 500 Match to 50 (Gain = 13 dB) CM LM pF nH 140 133 95.0 71.0 66.5 57.0 47.5 40.7 35.6 31.6 28.5 23.7 17.8 14.2 11.9 9.5 7.1 5.7 4.75 4.07 3.57 3.16 2.85 3500 3200 2250 1660 1550 1310 1070 904 779 682 604 489 346 262 208 155 104 75.3 57.4 45.3 36.7 30.4 25.6 Match to 100 (Gain = 10 dB) CM LM pF nH 100.7 94.1 67.1 50.3 47.0 40.3 33.5 28.8 25.2 22.4 20.1 16.8 12.6 10.1 8.4 6.7 5.03 4.03 3.36 2.87 2.52 2.24 2.01 4790 4460 3120 2290 2120 1790 1460 1220 1047 912 804 644 448 335 261 191 125 89.1 66.8 52.1 41.8 34.3 28.6 Now having a purely resistive input impedance, we can calculate the nominal coupling elements CO and LO, using CO = 1 ; LO = 2 πfC (R IN RM ) (R IN RM ) (8) 2 πfC For the AD8306, RIN is 1 kΩ. Thus, if a match to 50 Ω is needed, at fC = 100 MHz, CO must be 7.12 pF and LO must be 356 nH. Step 3: Split CO Into Two Parts Since we wish to provide the fully-balanced form of network shown in Figure 28, two capacitors C1 = C2 each of nominally twice CO, shown as CM in the figure, can be used. This requires a value of 14.24 pF in this example. Under these conditions, the voltage amplitudes at INHI and INLO will be similar. A somewhat better balance in the two drives may be achieved when C1 is made slightly larger than C2, which also allows a wider range of choices in selecting from standard values. For example, capacitors of C1 = 15 pF and C2 = 13 pF may be used (making CO = 6.96 pF). Step 4: Calculate LM The matching inductor required to provide both LIN and LO is just the parallel combination of these: LM = LINLO/(LIN + LO) (9) With LIN = 1 µH and LO = 356 nH, the value of LM to complete this example of a match of 50 Ω at 100 MHz is 262.5 nH. The nearest standard value of 270 nH may be used with only a slight loss of matching accuracy. The voltage gain at resonance depends only on the ratio of impedances, as is given by R R  GAIN = 20 log  IN  = 10 log  IN  R  RS   S Altering the Logarithmic Slope General Matching Procedure For other center frequencies and source impedances, the following method can be used to calculate the basic matching parameters. Step 1: Tune Out CIN At a center frequency fC, the shunt impedance of the input capacitance CIN can be made to disappear by resonating with a temporary inductor LIN, whose value is given by LIN = 1/{(2 π fC)2CIN} = 1010/fC2 (7) when CIN = 2.5 pF. For example, at fC = 100 MHz, LIN = 1 µH. (10) Simple schemes can be used to increase and decrease the logarithmic slope as shown in Figure 30. For the AD8306, only power, ground and logarithmic output connections are shown; refer to Figure 24 for complete circuitry. In Figure 30(a), the op amp’s gain of +2 increases the slope to 40 mV/dB. In Figure 30(b), the AD8031 buffers a resistive divider to give a slope of +5V 0.1 F 10 10 0.1 F +5V 0.1 F 10 10 0.1 F 10 0.1 F VPS1 VPS2 VLOG AD8031 40mV/dB VPS1 VPS2 VLOG 10 0.1 F 5k AD8031 AD8306 PADL, COM1, COM2 AD8306 PADL, COM1, COM2 5k 5k 5k 10mV/dB (a) Figure 30. Altering the Logarithmic Slope (b) –12– REV. A AD8306 10 mV/dB The AD8031 rail-to-rail op amp, used in both examples, can swing from 50 mV to 4.95 mV on a single +5 V supply. If high output current is required (> 10 mA), the AD8051, which also has rail-to-rail capability but can deliver up to 45 mA of output current, can be used. APPLICATIONS VS 10 1 COM2 0.1 F 2 VPS1 3 PADL 4 INHI 5 INLO 6 PADL 7 COM1 8 ENBL VPS2 15 0.1 F PADL 14 LMHI 13 LMLO 12 PADL 11 FLTR 10 LMDR 9 0mA TO 10mA 18 VARIABLE OUTPUT 2N3904 0V TO +1V 0.1 F 8.2k 10 VLOG 16 RSSI AD8306 The AD8306 is a versatile and easily applied log-limiting amplifier. Being complete, it can be used with very few external components, and most applications can be accommodated using the simple connections shown in the preceding section. A few examples of more specialized applications are provided here. High Output Limiter Loading AD8031 1.8k The AD8306 can generate a fairly large output power at its differential limiter output interface. This may be coupled into a 50 Ω grounded load using the narrow-band coupling network following similar lines to those provided for input matching. Alternatively, a flux-linked transformer, having a center-tapped primary, may be used. Even higher output powers can be obtained using emitter-followers. In Figure 31, the supply voltage to the AD8306 is dropped from 5 V to about 4.2 V, by the diode. This increases the available swing at each output to about 2 V. Taking both outputs differentially, a square wave output of 4 V p-p can be generated. IN914 APPROX. 4.2V 10 1 COM2 0.1 F 2 VPS1 3 PADL 4 INHI 5 INLO 6 PADL 7 COM1 8 ENBL VPS2 15 PADL 14 LMHI 13 LMLO 12 PADL 11 FLTR 10 LMDR 9 RLIM 3V TO 5V 5V TO 3V DIFFERENTIAL OUTPUT = 4V pk-pk 10 VLOG 16 0.1 F RLOAD RSSI SET RL = 5*RLIM +5V RLOAD Figure 32. Variable Limiter Output Programming Effect of Waveform Type on Intercept The AD8306 fundamentally responds to voltage and not to power. A direct consequence of this characteristic is that input signals of equal rms power, but differing crest factors, will produce different results at the log amp’s output. The effect of differing signal waveforms is to shift the effective value of the log amp’s intercept. Graphically, this looks like a vertical shift in the log amp’s transfer function. The device’s logarithmic slope however is not affected. For example, consider the case of the AD8306 being alternately fed by an unmodulated sine wave and by a single CDMA channel of the same rms power. The AD8306’s output voltage will differ by the equivalent of 3.55 dB (71 mV) over the complete dynamic range of the device (the output for a CDMA input being lower). Table II shows the correction factors that should be applied to measure the rms signal strength of a various signal types. A sine wave input is used as a reference. To measure the rms power of a square wave, for example, the mV equivalent of the dB value given in the table (20 mV/dB times 3.01 dB) should be subtracted from the output voltage of the AD8306. Table II. Shift in AD8306 Output for Signals with Differing Crest Factors AD8306 Figure 31. Increasing Limiter Output Voltage When operating at high output power levels and high frequencies, very careful attention must be paid to the issue of stability. Oscillation is likely to be observed when the input signal level is low, due to the extremely high gain-bandwidth product of the AD8306 under such conditions. These oscillations will be less evident when signal-balancing networks are used, operating at frequencies below 200 MHz, and they will generally be fully quenched by the signal at input levels of a few dB above the noise floor. Modulated Limiter Output Signal Type Sine Wave Square Wave or DC Triangular Wave GSM Channel (All Time Slots On) CDMA Channel (Forward Link, 9 Channels On) CDMA Channel (Reverse Link) PDC Channel (All Time Slots On) Gaussian Noise Evaluation Board Correction Factor (Add to Output Reading) 0 dB –3.01 dB +0.9 dB +0.55 dB +3.55 dB +0.5 dB +0.58 dB +2.51 dB The limiter output stage of the AD8306 also provides an analog multiplication capability: the amplitude of the output square wave can be controlled by the current withdrawn from LMDR (Pin 9). An analog control input of 0 V to +1 V is used to generate an exactly-proportional current of 0 mA to 10 mA in the npn transistor, whose collector is held at a fixed voltage of ∼400 mV by the internal bias in the AD8306. When the input signal is above the limiting threshold, the output will then be a squarewave whose amplitude is proportional to the control bias. An evaluation board, carefully laid out and tested to demonstrate the specified high speed performance of the AD8306 is available. Figure 33 shows the schematic of the evaluation board, which fairly closely follows the basic connections schematic shown in Figure 27. For ordering information, please refer to the Ordering Guide. Links, switches and component settings for different setups are described in Table III. REV. A –13– AD8306 R3 0 1 COM2 2 VPS1 C3 0.1 F 3 PADL 4 INHI 5 INLO 6 PADL 7 COM1 A EXT ENABLE B SW1 8 ENBL VLOG 16 VPS2 15 PADL 14 LMHI 13 C7 (OPEN) LMLO 12 PADL 11 FLTR 10 LK1 LMDR 9 R8 402 L1 (OPEN) C6 0.01 F R11 0 R9 (OPEN) R6 50 C4 0.1 F R7 50 C5 0.01 F R12 0 R4 (OPEN) VRSSI +VS SIG INHI R2 10 R5 10 +VS C1 0.01 F AD8306 R10 52.3 SIG INLO R1 0 C2 0.01 F Figure 33. Evaluation Board Schematic Table III. Evaluation Board Setup Options Component SW1 Function Device Enable. When in Position A, the ENBL pin is connected to +VS and the AD8306 is in normal operating mode. In Position B, the ENBL pin is connected to an SMA connector labeled Ext Enable. A signal can be applied to this connector to enable/disable the AD8306. This pad is used to ac-couple INLO to ground for single-ended input drive. To drive the AD8306 differentially, R1 should be removed. Input Interface. The 52.3 Ω resistor in position R10, along with C1 and C2, create a high-pass input filter whose corner frequency (640 kHz) is equal to 1/(2πRC), where C = (C1)/2 and R is the parallel combination of 52.3 Ω and the AD8306’s input impedance of 1000 Ω. Alternatively, the 52.3 Ω resistor can be replaced by an inductor to form an input matching network. See Input Matching Network section for more details. Slope Adjust. A simple slope adjustment can be implemented by adding a resistive divider at the VLOG output. R3 and R4, whose sum should be about 1 kΩ, and never less than 40 Ω (see specs), set the slope according to the equation: Slope = 20 mV/dB × R4/(R3 + R4). Limiter Output Coupling. C5 and C6 ac-couple the limiter’s differential outputs. By adjusting these values and installing an inductor in L1, an output matching network can be implemented. To convert the limiter’s differential output to singleended, R11 and R12 (nominally 0 Ω) can be replaced with a surface mount balun such as the ETC1-1-13 (Macom). The balun can be grounded by soldering a 0 Ω into Position R9 (nominally open). Limiter Output Current. With LK1 installed, R8 enables and sets the limiter output current. The limiter’s output current is set according to the equation (IOUT = 400 mV/R8). The limiter current can be as high as 10 mA (R8 = 40 Ω). To disable the limiter (recommended if the limiter is not being used), LK1 should be removed. Default Condition SW1 = A R1 R/L, C1, C2 R1 = 0 Ω R10 = 52.3 Ω C1 = C2 = 0.01 µF R3/R4 R3 = 0 Ω R4 = L1, C5, C6 L1 = Open C5 = 0.01 µF C6 = 0.01 µF R9 = Open R10 = R11 = 0 Ω LK1 Installed. R8 = 402 Ω R6, R7 (Limited Load Resistors) = 50 Ω R8, LK1 C7 RSSI Bandwidth Adjust. The addition of C7 (farads) will lower the RSSI bandwidth of C7 = Open the VLOG output according to the equation: fCORNER (Hz) = 12.7 × 10–6/(C7 + 3.5 × 10–12). –14– REV. A AD8306 Figure 34. Layout of Signal Layer Figure 36. Signal Layer Silkscreen Figure 35. Layout of Power Layer Figure 37. Power Layer Silkscreen REV. A –15– AD8306 OUTLINE DIMENSIONS Dimensions shown in inches and (mm). 16-Lead Narrow Body SO (SO-16) 0.3937 (10.00) 0.3859 (9.80) 0.1574 (4.00) 0.1497 (3.80) 16 1 9 8 0.2440 (6.20) 0.2284 (5.80) PIN 1 0.050 (1.27) BSC 0.0688 (1.75) 0.0532 (1.35) 0.0196 (0.50) 0.0099 (0.25) 45 0.0098 (0.25) 0.0040 (0.10) 8 0.0192 (0.49) SEATING 0.0099 (0.25) 0 0.0138 (0.35) PLANE 0.0075 (0.19) 0.0500 (1.27) 0.0160 (0.41) –16– REV. A PRINTED IN U.S.A. C3592a–9–8/99