AOZ3013PI
EZBuck™ 3 A Synchronous Buck Regulator
Not Recommended For New Designs
General Description
Features
The AOZ3013PI is a high efficiency, easy to use, 3 A
synchronous buck regulator. The AOZ3013PI operates
from 4.5 V to 18 V input voltage range, and provides up
to 3 A of continuous output current with an output voltage
adjustable down to 0.8 V.
4.5 V to 18 V operating input voltage range
The AOZ3013PI comes in an exposed pad SO-8
package and is rated over a -40 °C to +85 °C operating
ambient temperature range.
Output voltage adjustable to 0.8 V
65 mΩ internal high-side switch and 36 mΩ internal
low-side switch (at 12 V)
ig
n
s
Up to 95 % efficiency
External soft start
es
3 A continuous output current
D
500 kHz PWM operation
Cycle-by-cycle current limit
Replacement Parts:
AOZ6663DI
AOZ6683CI
ew
Pre-bias start-up
Short-circuit protection
N
Thermal shutdown
Fo
r
Exposed pad SO-8 package
Applications
Point of load DC/DC converters
Typical Application
ec
VIN
Set top boxes
DVD and Blu-ray players/recorders
Cable modems
om
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en
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LCD TV
C1
10µF
R
CSS
VIN
SS
ot
L1 4.7µH
N
EN
AOZ3013PI
R1
COMP
RC
CC
VOUT
LX
C2, C3
22µF
FB
AGND
PGND
R2
Figure 1. 3.3 V, 3 A Synchronous Buck Regulator, Fs = 500 kHz
Rev. 1.0 October 2013
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Page 1 of 14
AOZ3013PI
Ordering Information
Part Number
Ambient Temperature Range
Package
Environmental
AOZ3013PI
-40 °C to +85 °C
EPAD SO-8
Green Product
AOS Green Products use reduced levels of Halogens, and are also RoHS compliant.
Please visit www.aosmd.com/media/AOSGreenPolicy.pdf for additional information.
2
AGND
3
FB
4
PAD
(LX)
LX
7
SS
6
EN
5
COMP
Fo
r
(Top View)
N
Exposed Pad SO-8
es
VIN
8
D
1
ew
PGND
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s
Pin Configuration
1
PGND
2
VIN
3
AGND
4
FB
ec
COMP
R
5
EN
ot
6
Pin Function
Power ground. PGND needs to be electrically connected to AGND.
en
Pin Name
Supply voltage input. When VIN rises above the UVLO threshold and EN is logic high,
the device starts up.
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Pin Number
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d
Pin Description
Analog ground. AGND is the reference point for controller section. AGND needs to be
electrically connected to PGND.
Feedback input. The FB pin is used to set the output voltage via a resistive voltage divider
between the output and AGND.
External loop compensation pin. Connect a RC network between COMP and AGND to
compensate the control loop.
Enable pin. Pull EN to logic high to enable the device. Pull EN to logic low to disable the
device. If on/off control in not needed, connect EN to VIN and do not leave it open.
SS
8
LX (Sense)
Switching node. This pin has to be externally connected to exposed pad LX through PCB.
Exposed pad
LX
Switching node. LX is the drain of the internal PFET. LX is used as the thermal pad of the
power stage.
N
7
Rev. 1.0 October 2013
Soft-start pin. 5 µA current charging current.
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Page 2 of 14
AOZ3013PI
Block Diagram
VIN
UVLO
& POR
EN
Internal
+5V
5V LDO
Regulator
OTP
+
Reference
& Bias
Softstart
ig
n
–
Q1
ILimit
5µA
+
EAmp
FB
–
–
Level
Shifter
+
FET
Driver
PWM
Control
Logic
PWM
Comp
D
+
0.8V
es
SS
SS
ew
+
Over-Voltage
Protection
Comparator
LX
Q2
N
COMP
Fo
r
500kHz
Oscillator
+
–
om
m
Absolute Maximum Ratings
en
de
d
0.96V
s
ISen
Exceeding the Absolute Maximum Ratings may damage the
device.
Parameter
LX to AGND
EN to AGND
R
LX to AGND (20 ns)
ec
Supply Voltage (VIN)
ot
FB, SS, COMP to AGND
N
PGND to AGND
The device is not guaranteed to operate beyond the Maximum
Recommended Operating Conditions.
Parameter
20 V
-0.7 V to VIN +0.3 V
-5 V to 22 V
-0.3 V to VIN +0.3 V
-0.3 V to 6.0 V
PGND
Recommended Operating Conditions
Rating
Supply Voltage (VIN)
Rating
4.5 V to 18 V
Output Voltage Range
0.8 V to 0.85*VIN
Ambient Temperature (TA)
-40 °C to +85 °C
Package Thermal Resistance
Exposed Pad SO-8 (JA)
50 °C/W
-0.3 V to +0.3 V
Junction Temperature (TJ)
+150 °C
Storage Temperature (TS)
-65 °C to +150 °C
ESD Rating(1)
AGND
2.0 kV
Note:
1. Devices are inherently ESD sensitive, handling precautions are
required. Human body model rating: 1.5 kΩ in series with 100 pF.
Rev. 1.0 October 2013
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Page 3 of 14
AOZ3013PI
Electrical Characteristics
TA = 25 °C, VIN = VEN = 12 V, VOUT = 3.3 V unless otherwise specified(2)
VIN
Conditions
Min.
Supply Voltage
VUVLO
IIN
Parameter
4.5
Input Under-Voltage Lockout
Threshold
Supply Current (Quiescent)
VIN Rising
4.1
VIN Falling
3.7
IOUT = 0, VFB = 1.2 V, VEN > 2 V
1.6
IOFF
Shutdown Supply Current
VEN = 0 V
VFB
Feedback Voltage
TA = 25 °C
0.788
IFB
Feedback Voltage Input Current
EN Input Threshold
Off Threshold
On Threshold
D
CSS = 16 nF
N
MODULATOR
Maximum Duty Cycle
TMIN
Controllable Minimum On Time
Fo
r
Frequency
DMAX
de
Error Amplifier Transconductance
PROTECTION
en
Over-Voltage Protection
om
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VOVP
Current Limit
Over-Temperature Shutdown Limit
OUTPUT STAGE
ec
High-Side Switch On-Resistance
V
mA
1
10
µA
0.8
0.812
V
0.5
%
1
%
200
0.6
200
500
µA
ms
600
kHz
150
ns
85
3.5
V
mV
1
400
nA
%
8
A/ V
200
µA / V
4.5
A
Off Threshold
960
On Threshold
800
TJ Rising
150
TJ Falling
100
VIN = 12 V
65
VIN = 5 V
110
VIN = 12 V
36
VIN = 5 V
50
mV
°C
mΩ
mΩ
ot
R
Low-Side Switch On-Resistance
V
2
d
Current Sense Transconductance
ILIM
ew
EN Leakage Current
fO
18
2
EN Input Hysteresis
SS Time
Units
2.5
es
Line Regulation
VEN
Max.
ig
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Load Regulation
VHYS
Typ.
s
Symbol
N
Note:
2. Specification in BOLD indicate an ambient temperature range of -40 °C to +85 °C. These specifications are guaranteed by design.
Rev. 1.0 October 2013
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Page 4 of 14
AOZ3013PI
Typical Performance Characteristics
Circuit of Figure 1. TA = 25 °C, VIN = VEN = 12 V, VOUT = 3.3 V unless otherwise specified.
Full Load Operation
Light Load Operation
Vin ripple
0.5V/div
Vo ripple
0.2V/div
Vo ripple
0.2V/div
s
Vin ripple
0.2V/div
es
ig
n
VLX
10V/div
IL
2A/div
D
IL
2A/div
VLX
10V/div
2µs/div
ew
2µs/div
Short Circuit Protection
N
Start Up to Full Load
VLX
10V/div
de
d
Vo
2V/div
Vo
2V/div
Fo
r
Vin
10V/div
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en
lo
2A/div
IL
2A/div
10ms/div
50% to 100% Load Transient
Short Circuit Recovery
ec
2µs/div
N
ot
R
Vo
2V/div
Vo ripple
0.2V/div
VLX
10V/div
Io
2A/div
IL
2A/div
10ms/div
100µs/div
Rev. 1.0 October 2013
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Page 5 of 14
AOZ3013PI
Efficiency
Efficiency (VIN = 12V, L = 4.7µH) vs. Load Current
100
95
90
75
5V OUTPUT
3.3V OUTPUT
1.8V OUTPUT
70
65
D
1.2V OUTPUT
60
0.5
1.0
1.5
2.0
2.5
3.0
N
0
ew
55
50
ig
n
s
80
es
Efficiency (%)
85
Fo
r
Load Current (A)
d
Detailed Description
om
m
en
de
The AOZ3013PI is a current-mode step down regulator
with an integrated high-side PMOS switch and a low-side
NMOS switch. The AOZ3013PI operates from a 4.5 V to
18 V input voltage range and supplies up to 3 A of load
current. Features include enable control, power-on reset,
input under voltage lockout, output over voltage
protection, external soft-start and thermal shut down.
ec
The AOZ3013PI is available in an exposed pad SO-8
package.
Enable and Soft Start
N
ot
R
The AOZ3013PI has an external soft start feature to limit
in-rush current and ensure the output voltage ramps up
smoothly to regulation voltage. The soft start process
begins when the input voltage rises to 4.1 V and voltage
on the EN pin is HIGH. In the soft start process, the
FB voltage is ramped to follow the voltage of the soft start
pin until it reaches 0.8 V. The voltage of the soft-start pin
is charged by an internal 5 µA current.
The EN pin of the AOZ3013PI is active high. Connect the
EN pin to VIN if the enable function is not used. Pulling
EN to ground will disable the AOZ3013PI. Do not leave
EN open. The voltage on the EN pin must be above 2 V
to enable the AOZ3013PI. When the EN pin voltage falls
below 0.6 V, the AOZ3013PI is disabled.
Rev. 1.0 October 2013
Steady-State Operation
Under heavy load steady-state conditions, the converter
operates in fixed frequency and Continuous-Conduction
Mode (CCM).
The AOZ3013PI integrates an internal P-MOSFET as the
high-side switch. Inductor current is sensed by amplifying
the voltage drop across the drain to source of the high
side power MOSFET. Output voltage is divided down by
the external voltage divider at the FB pin. The difference
of the FB pin voltage and reference voltage is amplified
by the internal transconductance error amplifier. The
error voltage, which shows on the COMP pin, is
compared against the current signal, which is the sum of
inductor current signal and ramp compensation signal, at
the PWM comparator input. If the current signal is less
than the error voltage, the internal high-side switch is on.
The inductor current flows from the input through the
inductor to the output. When the current signal exceeds
the error voltage, the high-side switch is off. The inductor
current is freewheeling through the internal low-side
N-MOSFET switch to output. The internal adaptive FET
driver guarantees no turn on overlap of both the
high-side and the low-side switch.
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Page 6 of 14
AOZ3013PI
Compared with regulators using freewheeling Schottky
diodes, the AOZ3013PI uses a freewheeling NMOSFET
to realize synchronous rectification. This greatly
improves the converter efficiency and reduces power
loss in the low-side switch.
Since the switch duty cycle can be as high as 100 %, the
maximum output voltage can be set as high as the input
voltage minus the voltage drop on the upper PMOS and
the inductor.
Over Current Protection (OCP)
s
The AOZ3013PI has multiple protection features to
prevent system circuit damage under abnormal
conditions.
The sensed inductor current signal is also used for over
current protection. Since the AOZ3013PI employs peak
current mode control, the COMP pin voltage is
proportional to the peak inductor current. The COMP pin
voltage is limited to be between 0.4 V and 2.5 V internally.
The peak inductor current is automatically limited
cycle-by-cycle.
es
V O_MAX = V IN – I O R DS ON
Protection Features
ig
n
The AOZ3013PI uses a P-Channel MOSFET as the
high-side switch. This saves the bootstrap capacitor
normally seen in a circuit using an NMOS switch. It also
allows 100 % turn-on of the high-side switch to achieve
linear regulation mode of operation. The minimum
voltage drop from VIN to VO is the load current times DC
resistance of the MOSFET plus DC resistance of the
buck inductor. It can be calculated by equation below:
D
where;
VO_MAX is the maximum output voltage,
ew
VIN is the input voltage from 4.5 V to 18 V,
IO is the output current from 0 A to 3 A, and
N
RDS(ON) is the on resistance of the internal MOSFET.
When the output is shorted to ground under fault
conditions, the inductor current slowly decays during a
switching cycle because the output voltage is 0 V.
To prevent catastrophic failure, a secondary current limit
is designed inside the AOZ3013PI. The measured
inductor current is compared against a preset voltage
which represents the current limit, between 3.5 A and 5.0
A. When the output current is greater than the current
limit, the high side switch will be turned off. The converter
will initiate a soft start once the over-current condition is
resolved.
de
d
Output voltage can be set by feeding back the output to
the FB pin using a resistor divider network as shown in
Figure 1. The resistor divider network includes R1 and
R2. Usually, a design is started by picking a fixed R2
value and calculating the required R1 with the equation
below:
Fo
r
Output Voltage Programming
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en
R 1
V O = 0.8 1 + -------
R 2
Some standard value of R1 and R2 for the most common
output voltages are listed in Table 1.
R1 (kΩ)
R2 (kΩ)
1.0
Open
4.99
10
10
11.5
1.8
12.7
10.2
2.5
21.5
10
3.3
31.1
10
5.0
52.3
10
ec
VO (V)
1.2
N
ot
1.5
R
0.8
Power-On Reset (POR)
A power-on reset circuit monitors the input voltage. When
the input voltage exceeds 4.1 V, the converter starts
operation. When input voltage falls below 3.7 V, the
converter will be shut down.
Thermal Protection
An internal temperature sensor monitors the junction
temperature. The sensor shuts down the internal control
circuit and high side PMOS if the junction temperature
exceeds 150 ºC. The regulator will restart automatically
under the control of the soft-start circuit when the junction
temperature decreases to 100 ºC.
Table 1.
The combination of R1 and R2 should be large enough to
avoid drawing excessive current from the output, which
will cause power loss.
Rev. 1.0 October 2013
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Page 7 of 14
AOZ3013PI
Application Information
The basic AOZ3013PI application circuit is show in
Figure 1. Component selection is explained below.
Input Capacitor
VO VO
IO
V IN = ----------------- 1 – --------- --------f C IN
V IN V IN
D
ew
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ec
R
High inductance gives low inductor ripple current but
requires larger size inductor to avoid saturation. Low
ripple current reduces inductor core losses. It also
reduces RMS current through inductor and switches,
which results in less conduction loss. Usually, peak to
peak ripple current on the inductor is designed to be
20 % to 40 % of output current.
When selecting the inductor, confirm it is able to handle
the peak current without saturation at the highest
operating temperature.
The inductor takes the highest current in a buck circuit.
The conduction loss on the inductor needs to be checked
for thermal and efficiency requirements.
N
ot
ICIN_RMS(m) 0.3
IO
0.2
0.1
0
N
I L
I Lpeak = I O + -------2
The relationship between the input capacitor RMS
current and voltage conversion ratio is calculated and
shown in Figure 2 below. It can be seen that when VO is
half of VIN, CIN is under the worst current stress. The
worst current stress on CIN is 0.5 x IO.
0
VO
VO
-
I L = ----------- 1 – -------fL
V IN
The peak inductor current is:
if we let m equal the conversion ratio:
0.4
The inductor is used to supply constant current to output
when it is driven by a switching voltage. For a given input
and output voltage, inductance and switching frequency
together decide the inductor ripple current, which is:
Fo
r
VO
VO
- 1 – --------
I CIN_RMS = I O -------V IN
V IN
es
Inductor
Since the input current is discontinuous in a buck
converter, the current stress on the input capacitor is
another concern when selecting the capacitor. For a buck
circuit, the RMS value of input capacitor current can be
calculated by:
0.5
ig
n
The input ripple voltage can be approximated by
equation below:
VO
-------- = m
V IN
Note that the ripple current rating from capacitor
manufactures are based on a certain operating life time.
Further de-rating may need to be considered for long
term reliability.
s
The input capacitor must be connected to the VIN pin and
the PGND pin of AOZ3013PI to maintain steady input
voltage and filter out the pulsing input current. The
voltage rating of input capacitor must be greater than
maximum input voltage plus ripple voltage.
For reliable operation and best performance, the input
capacitors must have a current rating higher than
ICIN_RMS at the worst operating conditions. Ceramic
capacitors are preferred for input capacitors because of
their low ESR and high current rating. Depending on the
application circuits, other low ESR tantalum capacitors
may be used. When selecting ceramic capacitors, X5R or
X7R type dielectric ceramic capacitors should be used
for their better temperature and voltage characteristics.
0.5
m
1
Surface mount inductors in different shape and styles are
available from Coilcraft, Elytone and Murata. Shielded
inductors are small and radiate less EMI noise. However,
they cost more than unshielded inductors. The choice
depends on EMI requirement, price and size.
Figure 2. ICIN vs. Voltage Conversion Ratio
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Page 8 of 14
AOZ3013PI
Output Capacitor
The output capacitor is selected based on the DC output
voltage rating, output ripple voltage specification and
ripple current rating.
The selected output capacitor must have a higher rated
voltage specification than the maximum desired output
voltage including ripple. De-rating needs to be
considered for long term reliability.
Usually, the ripple current rating of the output capacitor is
a smaller issue because of the low current stress. When
the buck inductor is selected to be very small and
inductor ripple current is high, the output capacitor could
be overstressed.
Loop Compensation
The AOZ3013PI employs peak current mode control for
ease of use and fast transient response. Peak current
mode control eliminates the double pole effect of the
output L&C filter. It also greatly simplifies the
compensation loop design.
ig
n
s
Output ripple voltage specification is another important
factor for selecting the output capacitor. In a buck
converter circuit, output ripple voltage is determined by
inductor value, switching frequency, output capacitor
value and ESR. It can be calculated by the equation
below:
es
With peak current mode control, the buck power stage
can be simplified to be a one-pole and one-zero system
in frequency domain. The pole is dominant pole can be
calculated by:
D
1
V O = I L ESR CO + -------------------------
8fC
1
f P1 = ----------------------------------2 C O R L
ew
O
where,
CO is output capacitor value, and
The zero is a ESR zero due to the output capacitor and
its ESR. It is can be calculated by:
N
ESRCO is the equivalent series resistance of the output
capacitor.
Fo
r
1
f Z1 = -----------------------------------------------2 C O ESR CO
where;
CO is the output filter capacitor,
O
en
1
V O = I L ------------------------8fC
de
d
When a low ESR ceramic capacitor is used as the output
capacitor, the impedance of the capacitor at the switching
frequency dominates. Output ripple is mainly caused by
capacitor value and inductor ripple current. The output
ripple voltage calculation can be simplified to:
R
ec
V O = I L ESR CO
om
m
If the impedance of ESR at switching frequency
dominates, the output ripple voltage is mainly decided by
capacitor ESR and inductor ripple current. The output
ripple voltage calculation can be further simplified to:
N
ot
For lower output ripple voltage across the entire
operating temperature range, X5R or X7R dielectric type
of ceramic, or other low ESR tantalum capacitors are
recommended as output capacitors.
In a buck converter, output capacitor current is
continuous. The RMS current of output capacitor is
decided by the peak to peak inductor ripple current. It can
be calculated by:
I L
I CO_RMS = ---------12
RL is load resistor value, and
ESRCO is the equivalent series resistance of output capacitor.
The compensation design shapes the converter control
loop transfer function for the desired gain and phase.
Several different types of compensation networks can be
used with the AOZ3013PI. For most cases, a series
capacitor and resistor network connected to the
COMP pin sets the pole-zero and is adequate for a stable
high-bandwidth control loop.
In the AOZ3013PI, FB and COMP are the inverting input
and the output of the internal error amplifier. A series
R and C compensation network connected to COMP
provides one pole and one zero. The pole is:
G EA
f P2 = ------------------------------------------2 C C G VEA
where;
GEA is the error amplifier transconductance, which is 200 x 10-6
A/V,
GVEA is the error amplifier voltage gain, which is 500 V/V, and
CC is the compensation capacitor in Figure 1.
Rev. 1.0 October 2013
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Page 9 of 14
AOZ3013PI
The zero given by the external compensation network,
capacitor CC and resistor RC, is located at:
Thermal Management and Layout
Considerations
In the AOZ3013PI buck regulator circuit, high pulsing
current flows through two circuit loops. The first loop
starts from the input capacitors, to the VIN pin, to the
LX pad, to the filter inductor, to the output capacitor and
load, and then returns to the input capacitor through
ground. Current flows in the first loop when the high side
switch is on. The second loop starts from the inductor,
to the output capacitors and load, to the low side
NMOSFET. Current flows in the second loop when the
low side NMOSFET is on.
1
f Z2 = ----------------------------------2 C C R C
ig
n
s
To design the compensation circuit, a target crossover
frequency fC to close the loop must be selected. The
system crossover frequency is where the control loop
has unity gain. The crossover is the also called the
converter bandwidth. Generally a higher bandwidth
means faster response to load transients. However, the
bandwidth should not be too high because of system
stability concern. When designing the compensation
loop, converter stability under all line and load condition
must be considered.
es
In PCB layout, minimizing the area of the two loops will
reduce the noise of the circuit and improves efficiency.
A ground plane is strongly recommended to connect the
input capacitor, the output capacitor, and the PGND pin
of the AOZ3013PI.
D
Usually, it is recommended to set the bandwidth to be
equal or less than 1/10 of the switching frequency.
ew
In the AOZ3013PI buck regulator circuit, the major power
dissipating components are the AOZ3013PI and the
output inductor. The total power dissipation of converter
circuit can be measured by input power minus output
power:
Fo
r
VO
2 C C
R C = f C ---------- ----------------------------G G
V
EA
P total_loss = V IN I IN – V O I O
CS
The power dissipation of the inductor can be
approximately calculated by the output current and DCR
value of the inductor:
d
FB
N
The strategy for choosing RC and CC is to set the cross
over frequency with RC and set the compensator zero
with CC. Using selected crossover frequency, fC, to
calculate RC:
de
where;
en
fC is the desired crossover frequency. For best performance,
fC is set to be about 1/10 of the switching frequency;
VFB is 0.8V,
om
m
GEA is the error amplifier transconductance, which is
200 x 10-6 A/V, and
GCS is the current sense circuit transconductance, which is
8 A/V
R
ec
The compensation capacitor CC and resistor RC together
make a zero. This zero is put somewhere close to the
dominate pole fp1 but lower than 1/5 of the selected
crossover frequency. CC can is selected by:
ot
1.5
C C = ----------------------------------2 R C f P1
P inductor_loss = IO2 R inductor 1.1
The actual junction temperature can be calculated by the
power dissipation in the AOZ3013PI and the thermal
impedance from junction to ambient:
T junction = P total_loss – P inductor_loss JA
The maximum junction temperature of the AOZ3013PI is
150 ºC, which limits the maximum load current capability.
Please see the thermal de-rating curves for maximum
load current of the AOZ3013PI under different ambient
temperature.
N
The thermal performance of the AOZ3013PI is strongly
affected by the PCB layout. Care should be taken during
the design process to ensure that the IC will operate
under the recommended environmental conditions.
The above equation can be simplified to:
CO RL
C C = --------------------RC
An easy-to-use application software which helps to
design and simulate the compensation loop can be found
at www.aosmd.com.
Rev. 1.0 October 2013
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Page 10 of 14
AOZ3013PI
Layout Considerations
The AOZ3013PI is an exposed pad SO-8 package.
Several layout tips are listed for the best electric and
thermal performance.
1. The exposed pad (LX) is connected to the internal
PFET and NFET drains. Connected a large copper
plane to the LX pin to help thermal dissipation.
ig
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s
2. Do not use a thermal relief connection to the VIN pin
or the PGND pin. Pour a maximized copper area to
the PGND pin and the VIN pin to help thermal
dissipation.
es
3. The input capacitor should be connected as close as
possible to the VIN pin and the PGND pin.
ew
D
4. A ground plane is preferred. If a ground plane is not
used, separate PGND from AGND and only connect
them at one point to avoid the PGND pin noise
coupling to the AGND pin.
N
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ec
om
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en
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d
7. Keep sensitive signal trace away from the LX pad.
Fo
r
6. Pour copper plane on all unused board area and
connect it to stable DC nodes, like VIN, GND or
VOUT.
N
5. Make the current trace from the LX pad to L to Co to
the PGND as short as possible.
Rev. 1.0 October 2013
www.aosmd.com
Page 11 of 14
AOZ3013PI
Package Dimensions, SO-8 EP1
Gauge plane
0.2500
D0
C
L
L1
E
s
E1
E3
D
es
ig
n
E2
D1
Note 5
ew
D
e
B
A1
A
Fo
r
A2
d
Dimensions in millimeters
2.20
ec
5.74
N
ot
R
2.87
2.71
1.27
0.80
0.635
UNIT: mm
Min.
Min.
1.40
Nom.
1.55
Max.
1.70
Symbols
A
A1
A2
B
0.00
1.40
0.31
0.05
1.50
0.406
0.10
A1
A2
B
0.000
0.055
C
D
D0
D1
E
e
E1
E2
E3
L
y
θ
| L1–L1' |
L1
0.17
4.80
3.20
3.10
5.80
—
3.80
2.21
C
D
D0
D1
E
e
E1
E2
E3
L
y
θ
| L1–L1' |
L1
0.007
0.189
de
om
m
3.70
Dimensions in inches
Symbols
A
en
RECOMMENDED LAND PATTERN
θ
N
7 (4x)
L1'
—
4.96
3.40
3.30
6.00
1.27
3.90
2.41
0.40 REF
0.40
0.95
—
—
0°
—
1.60
0.51
0.25
5.00
3.60
3.50
6.20
—
4.00
2.61
1.27
0.10
3°
8°
0.04
0.12
1.04 REF
0.055
0.012
Nom.
0.061
Max.
0.067
0.002
0.004
0.059
0.016
0.063
0.020
—
0.010
0.195 0.197
0.126 0.134 0.142
0.122 0.130 0.138
0.228 0.236 0.244
—
0.050
—
0.150 0.153 0.157
0.087 0.095 0.103
0.016 REF
0.016 0.037 0.050
—
—
0.004
0°
—
3°
8°
0.002 0.005
0.041 REF
Notes:
1. Package body sizes exclude mold flash and gate burrs.
2. Dimension L is measured in gauge plane.
3. Tolerance 0.10mm unless otherwise specified.
4. Controlling dimension is millimeter, converted inch dimensions are not necessarily exact.
5. Die pad exposure size is according to lead frame design.
6. Followed from JEDEC MS-012
Rev. 1.0 October 2013
www.aosmd.com
Page 12 of 14
AOZ3013PI
Tape and Reel Dimensions, SO-8 EP1
Carrier Tape
P1
D1
P2
T
E1
E2
ig
n
s
E
B0
D0
A0
P0
Feeding Direction
es
K0
UNIT: mm
K0
2.10
±0.10
D0
1.60
±0.10
D1
1.50
±0.10
E
12.00
±0.10
E1
1.75
±0.10
E2
5.50
±0.10
P0
8.00
±0.10
P1
4.00
±0.10
P2
2.00
±0.10
D
B0
5.20
±0.10
ew
A0
6.40
±0.10
T
0.25
±0.10
N
Package
SO-8
(12mm)
Reel
Fo
r
W1
de
N
M
UNIT: mm
om
m
R
K
en
V
S
d
G
H
W
W1
17.40
±1.00
H
K
ø13.00
10.60
+0.50/-0.20
S
2.00
±0.50
G
—
R
—
V
—
ot
R
ec
W
N
Tape Size Reel Size
M
12mm
ø330
ø330.00 ø97.00 13.00
±0.10 ±0.30
±0.50
N
Leader/Trailer and Orientation
Trailer Tape
300mm min. or
75 empty pockets
Rev. 1.0 October 2013
Components Tape
Orientation in Pocket
www.aosmd.com
Leader Tape
500mm min. or
125 empty pockets
Page 13 of 14
AOZ3013PI
Part Marking
Z3013PI
FAYWLT
ig
n
s
Part Number Code
es
Assembly Lot Code
Fab & Assembly Location
de
d
Fo
r
N
ew
D
Year & Week Code
en
LEGAL DISCLAIMER
ec
om
m
Applications or uses as critical components in life support devices or systems are not authorized. AOS does not
assume any liability arising out of such applications or uses of its products. AOS reserves the right to make
changes to product specifications without notice. It is the responsibility of the customer to evaluate suitability of the
product for their intended application. Customer shall comply with applicable legal requirements, including all
applicable export control rules, regulations and limitations.
R
AOS' products are provided subject to AOS' terms and conditions of sale which are set forth at:
http://www.aosmd.com/terms_and_conditions_of_sale
ot
LIFE SUPPORT POLICY
N
ALPHA AND OMEGA SEMICONDUCTOR PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL
COMPONENTS IN LIFE SUPPORT DEVICES OR SYSTEMS.
As used herein:
1. Life support devices or systems are devices or
systems which, (a) are intended for surgical implant into
the body or (b) support or sustain life, and (c) whose
failure to perform when properly used in accordance
with instructions for use provided in the labeling, can be
reasonably expected to result in a significant injury of
the user.
Rev. 1.0 October 2013
2. A critical component in any component of a life
support, device, or system whose failure to perform can
be reasonably expected to cause the failure of the life
support device or system, or to affect its safety or
effectiveness.
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Page 14 of 14
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