LMR23630
SNVSAH2E – DECEMBER 2015 – REVISED AUGUST 2020
LMR23630 SIMPLE SWITCHER® 36-V, 3-A Synchronous Step-Down Converter
1 Features
3 Description
•
•
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•
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The LMR23630 SIMPLE SWITCHER® is an easy-touse 36 V, 3 A synchronous step-down regulator.
With a wide input range from 4 V to 36 V, it is
suitable for various industrial applications for power
conditioning from unregulated sources. Peak-currentmode control is employed to achieve simple controlloop compensation and cycle-by-cycle current limiting.
A quiescent current of 75 μA makes the device
suitable for battery-powered systems. An ultra-low
2 μA shutdown current can further prolong battery
life. Internal loop compensation means that the user
is free from the tedious task of loop compensation
design. This also minimizes the external components.
The device has an option for constant frequency
FPWM mode to achieve small output-voltage ripple at
light load. An extended family (HSOIC) is available in
1-A (LMR23610) and 2.5-A (LMR23625) load-current
options in a pin-to-pin compatible package allowing
simple, optimum PCB layout. A precision enable
input allows simplification of regulator control and
system power sequencing. Protection features include
cycle-by-cycle current limit, hiccup-mode short-circuit
protection, and thermal shutdown due to excessive
power dissipation.
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2 Applications
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Device Information
Factory and building automation systems: PLC
CPU, HVAC control, elevator control
Assest tracking
General purpose wide VIN regulation
PART NUMBER (1)
LMR23630
(1)
PACKAGE
BODY SIZE (NOM)
HSOIC (8)
4.89 mm × 3.90 mm
WSON (12)
3.00 mm × 3.00 mm
For detail part numbers for all available different options, see
the orderable addendum at the end of the data sheet.
100
VIN up to 36 V
CIN
90
VIN
EN/SYNC
BOOT
CBOOT
AGND
L
VOUT
SW
RFBT
COUT
VCC
FB
RFBB
Efficiency (%)
•
•
•
•
•
•
•
•
•
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4-V to 36-V Input range
3-A Continuous output current
Integrated synchronous rectification
Current-mode control
Minimum switch on-time: 60 ns
Internal compensation for ease of use
400-kHz Switching frequency and adjustable
frequency options
PFM and forced PWM mode options
Frequency synchronization to external clock
75-µA Quiescent current at no load for PFM option
Soft start into a prebiased load
High duty-cycle operation supported
Precision enable input
Output short-circuit protection with hiccup mode
Thermal protection
8-Pin HSOIC with PowerPAD™ package
12-Pin WSON wettable flanks package with
PowerPAD™
Use the LMZM33603 module for faster time to
market
Create a custom design using the LMR23625 with
the WEBENCH® Power Designer
80
70
60
CVCC
PGND
50
Simplified Schematic
40
0.0001
VOUT = 5 V
VOUT = 3.3 V
0.001
0.01
0.1
1
IOUT (A)
10
D001
Efficiency vs Load VIN = 12 V, PFM Option
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
LMR23630
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SNVSAH2E – DECEMBER 2015 – REVISED AUGUST 2020
Table of Contents
1 Features............................................................................1
2 Applications..................................................................... 1
3 Description.......................................................................1
4 Revision History.............................................................. 2
5 Device Comparison Table...............................................4
6 Pin Configuration and Functions...................................4
Pin Functions.................................................................... 5
7 Specifications.................................................................. 6
7.1 Absolute Maximum Ratings........................................ 6
7.2 ESD Ratings............................................................... 6
7.3 Recommended Operating Conditions.........................6
7.4 Thermal Information....................................................7
7.5 Electrical Characteristics.............................................7
7.6 Timing Requirements.................................................. 9
7.7 Switching Characteristics............................................9
7.8 Typical Characteristics.............................................. 10
8 Detailed Description......................................................12
8.1 Overview................................................................... 12
8.2 Functional Block Diagram......................................... 12
8.3 Feature Description...................................................13
8.4 Device Functional Modes..........................................20
9 Application and Implementation.................................. 21
9.1 Application Information............................................. 21
9.2 Typical Applications.................................................. 21
10 Power Supply Recommendations..............................28
11 Layout........................................................................... 28
11.1 Layout Guidelines................................................... 28
11.2 Layout Example...................................................... 29
11.3 Compact Layout for EMI Reduction........................ 29
11.4 Ground Plane and Thermal Considerations............30
11.5 Feedback Resistors................................................ 31
12 Device and Documentation Support..........................32
12.1 Device Support....................................................... 32
12.2 Receiving Notification of Documentation Updates..32
12.3 Support Resources................................................. 32
12.4 Trademarks............................................................. 32
12.5 Electrostatic Discharge Caution..............................32
12.6 Glossary..................................................................32
13 Mechanical, Packaging, and Orderable
Information.................................................................... 32
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision D (February 2018) to Revision E (July 2020)
Page
• Updated the numbering format for tables, figures and cross-references throughout the document...................1
• Added LMZM33630 bullet in Section 1 ..............................................................................................................1
Changes from Revision C (June 2017) to Revision D (February 2018)
Page
• Changed Programmable Logic Controller Power Supply to Factory and Building Automation System... in
Applications ....................................................................................................................................................... 1
• Deleted Multi-Function Printers and Industrial Power Supplies and reworded Applications ............................. 1
• Changed HVAC Systems from Applications to General Purpose Wide VIN Regulation ................................... 1
• Added "2.2-µF, 16-V" for VCC pin bypass capacitor ..........................................................................................5
• Change the Max Recommend Operating Condition for Iout to be 3-A from 2.5-A ............................................ 6
• Consolidating all the common EC table characteristic between HSOIC and WSON, for example Operation
Input Voltage, VIN_UVLO, IEN and Mnimum turn-on time ................................................................................. 7
• Changed Typical Value for VIN_UVLO Rising threshold typical from 3.6-V to 3.7-V and minimum Falling
threshold from 3-V to 2.9-V ................................................................................................................................7
• Changed the operating from "4.5-V" ... to "4-V" in Device Functional Modes ..................................................20
• Changed from VOUT = 7 V to 36 V to VIN = 7 V to 36 V on Figure 9-9 .............................................................26
Changes from Revision B (April 2017) to Revision C (June 2017)
Page
• Deleted Deleted "Automotive Battery Regulation" and reworded Applications ................................................. 1
• Added WSON Package and Options .................................................................................................................1
• Added Device Comparison Table ...................................................................................................................... 4
• Change EN Abs Max to EN/SYNC Abs Max ..................................................................................................... 6
• Updating ESD Ratings to include HSOIC and WSON .......................................................................................6
• Corrected Equation 17 denominator from "(VOUT x VOS)" to "(VOUT + VOS)".................................................... 23
• clarified equations Equation 23 and Equation 24 ............................................................................................ 30
Changes from Revision A (July 2016) to Revision B (April 2017)
Page
• Changed spec from 6.0 to 6.2 for max under Current Limit................................................................................7
2
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•
SNVSAH2E – DECEMBER 2015 – REVISED AUGUST 2020
Changed spec from 4.2 to 4.6 for max under Current Limit................................................................................7
Changes from Revision * (December 2015) to Revision A (July 2016)
Page
• Changed from Product Preview to Production Data with all the remaining sections added............................... 1
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5 Device Comparison Table
PACKAGE
PART NUMBER
HSOIC (8)
WSON (12) (Pin 6 is RT)
WSON (12) (Pin 6 is PGOOD)
FIXED 400 kHz
LMR23630ADDA
yes
LMR23630AFDDA
LMR23630DRR
ADJUSTABLE
FREQUENCY
RESISTOR
POWER GOOD
FPWM
no
no
no
yes
no
no
yes
no
yes
no
no
LMR23630FDRR
no
yes
no
yes
LMR23630APDRR
yes
no
yes
no
6 Pin Configuration and Functions
Figure 6-1. DRR Package 12-Pin WSON With RT and Thermal Pad Top View
Figure 6-2. DRR Package 12-Pin WSON With PGOOD and Thermal Pad Top View
SW
1
BOOT
2
VCC
FB
8
PGND
7
VIN
3
6
AGND
4
5
EN/SYNC
Thermal Pad
(9)
Figure 6-3. DDA Package 8-Pin HSOIC Top View
4
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Pin Functions
I/O (1)
PIN
HSOI
C
DESCRIPTION
WSON With
RT
WSON With
PGOOD
NAME
1
1, 2
1, 2
SW
P
Switching output of the regulator. Internally connected to both power
MOSFETs. Connect to power inductor.
2
3
3
BOOT
P
Boot-strap capacitor connection for high-side driver. Connect a high quality
100nF or 470nF capacitor from BOOT to SW.
3
4
4
VCC
P
Internal bias supply output for bypassing. Connect 2.2-µF, 16-V bypass
capacitor from this pin to AGND. Do not connect external loading to this
pin. Never short this pin to ground during operation.
4
5
5
FB
A
Feedback input to regulator, connect the midpoint of feedback resistor
divider to this pin.
N/A
6
N/A
RT
A
Connect a resistor RT from this pin to AGND to program switching
frequency. Leave floating for 400-kHz default switching frequency.
N/A
N/A
6
PGOOD
A
Open drain output for power-good flag. Use a 10-kΩ to 100-kΩ pullup
resistor to logic rail or other DC voltage no higher than 12 V.
5
8
8
EN/SYNC
A
Enable input to regulator. High = On, Low = Off. Can be connected to VIN.
Do not float. Adjust the input undervoltage lockout with two resistors. The
internal oscillator can be synchronized to an external clock by coupling a
positive pulse into this pin through a small coupling capacitor. See Section
8.3.4 for details.
6
7
7
AGND
G
Analog ground pin. Ground reference for internal references and logic.
Connect to system ground.
7
9, 10
9, 10
VIN
P
Input supply voltage.
8
12
12
PGND
G
Power ground pin, connected internally to the low side power FET.
Connect to system ground, PAD, AGND, ground pins of CIN and COUT.
Path to CIN must be as short as possible.
9
13
13
PAD
G
Low impedance connection to AGND. Connect to PGND on PCB. Major
heat dissipation path of the die. Must be used for heat sinking to ground
plane on PCB.
N/A
11
11
NC
N/A
(1)
Not for use. Leave this pin floating.
A = Analog, P = Power, G = Ground.
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7 Specifications
7.1 Absolute Maximum Ratings
Over the recommended operating junction temperature range of –40°C to +125°C (unless otherwise noted) (1)
PARAMETER
Input voltages
MIN
MAX
UNIT
VIN to PGND
–0.3
42
EN/SYNC to AGND
–5.5
VIN+ 0.3
FB to AGND
–0.3
4.5
RT to AGND
–0.3
4.5
PGOOD to AGND
–0.3
15
AGND to PGND
–0.3
0.3
–1
VIN + 0.3
SW to PGND
–5
42
BOOT to SW
–0.3
5.5
VCC to AGND
–0.3
4.5(2)
TJ
Junction temperature
–40
150
°C
Tstg
Storage temperature
–65
150
°C
Output voltages
(1)
(2)
SW to PGND less than 10 ns transients
V
V
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress
ratings only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under
Recommended Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device
reliability.
In shutdown mode, the VCC to AGND maximum value is 5.25 V.
7.2 ESD Ratings
VALUE
V(ESD)
(1)
(2)
Electrostatic discharge
Human-body model (HBM) for HSOIC (1)
±2000
Human-body model (HBM) for WSON with RT and
PGOOD(1)
±2500
Charged-device model (CDM) for HSOIC and WSON RT(2)
±1000
Charged-device model (CDM) for WSON PGOOD(2)
±750
UNIT
V
JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process.
7.3 Recommended Operating Conditions
Over the recommended operating junction temperature range of –40°C to +125°C (unless otherwise noted) (1)
MIN
VIN
Input voltage
4
36
–5
36
FB
–0.3
1.2
PGOOD
–0.3
12
EN/SYNC
V
Input current
PGOOD pin current
0
1
mA
Output voltage
VOUT
1
28
V
Output current
IOUT
Temperature
Operating junction temperature, TJ
(1)
6
MAX UNIT
0
3
A
–40
125
°C
Operating Ratings indicate conditions for which the device is intended to be functional, but do not ensure specific performance limits.
For ensured specifications, see Section 7.5.
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7.4 Thermal Information
THERMAL METRIC (1) (2)
DDA (8 PINS)
DRR (12 PINS)
UNIT
RθJA
Junction-to-ambient thermal resistance
42.0
41.5
°C/W
ψJT
Junction-to-top characterization parameter
5.9
0.3
°C/W
ψJB
Junction-to-board characterization parameter
23.4
16.5
°C/W
RθJC(top)
Junction-to-case (top) thermal resistance
45.8
39.1
°C/W
RθJC(bot)
Junction-to-case (bottom) thermal resistance
3.6
3.4
°C/W
RθJB
Junction-to-board thermal resistance
23.4
16.3
°C/W
(1)
(2)
For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report.
Determine power rating at a specific ambient temperature TA with a maximum junction temperature (TJ) of 125°C (see Section 7.3).
7.5 Electrical Characteristics
Limits apply over the recommended operating junction temperature (TJ) range of –40°C to +125°C, unless otherwise stated.
Minimum and maximum limits are specified through test, design or statistical correlation. Typical values represent the most
likely parametric norm at TJ = 25 °C, and are provided for reference purposes only.
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX UNIT
POWER SUPPLY (VIN PIN)
VIN
Operation input voltage
VIN_UVLO
Undervoltage lockout thresholds
4
36
Rising threshold
3.3
3.7
3.9
Falling threshold
2.9
3.3
3.5
2
4
ISHDN
Shutdown supply current
VEN = 0 V, VIN = 12 V, TJ = –40°C to 125°C
IQ
Operating quiescent current (nonswitching)
VIN =12 V, VFB = 1.1 V, TJ = –40°C to
125°C, PFM mode
75
V
V
μA
μA
ENABLE (EN/SYNC PIN)
VEN_H
Enable rising threshold voltage
VEN_HYS
Enable hysteresis voltage
VWAKE
Wake-up threshold
1.4
1.7
0.4
V
V
0.4
VIN = 4 V to 36 V, VEN= 2 V
IEN
1.55
V
10
100
Input leakage current at EN pin
VIN = 4 V to 36 V, VEN= 36 V
1
nA
nA
μA
μA
VOLTAGE REFERENCE (FB PIN)
VREF
Reference voltage
ILKG_FB
Input leakage current at FB pin
VIN = 4.0 V to 36 V, TJ = 25 °C
VIN = 4.0 V to 36 V, TJ = –40°C to 125°C
VFB= 1 V
0.985
1
1.015
V
0.98
1
1.02
V
10
nA
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Limits apply over the recommended operating junction temperature (TJ) range of –40°C to +125°C, unless otherwise stated.
Minimum and maximum limits are specified through test, design or statistical correlation. Typical values represent the most
likely parametric norm at TJ = 25 °C, and are provided for reference purposes only.
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX UNIT
POWER GOOD (PGOOD PIN)
VPG_OV
Power-good flag overvoltage
tripping threshold
% of reference voltage
104%
107%
110%
VPG_UV
Power-good flag undervoltage
tripping threshold
% of reference voltage
92%
94%
96.5%
VPG_HY S
Power-good flag recovery
hysteresis
% of reference voltage
VIN_PG_MIN
Minimum VIN for valid PGOOD
output
50 μA pullup to PGOOD pin, VEN = 0 V, TJ
= 25°C
1.5
V
50 μA pullup to PGOOD pin, VIN = 1.5 V,
VEN = 0 V
0.4
V
0.5 mA pullup to PGOOD pin, VIN = 13.5 V,
VEN = 0 V
0.4
V
VPG_LOW
PGOOD low level output voltage
1.5%
INTERNAL LDO (VCC PIN)
VCC
Internal LDO output voltage
VCC_UVLO
VCC undervoltage lockout
thresholds
4.1
V
Rising threshold
2.8
3.2
3.6
Falling threshold
2.4
2.8
3.2
HSOIC package
3.8
5
6.2
WSON package
4
5.5
6.6
HSOIC package
2.9
3.6
4.6
WSON package
2.9
3.6
4.2
V
CURRENT LIMIT
IHS_LIMIT
Peak inductor current limit
ILS_LIMIT
Valley inductor current limit
IL_ZC
Zero cross current limit
HSOIC and WSON package
IL_NEG
Negative current limit (FPWM
option)
SOIC and WSON package
–0.04
–2.7
–2
A
A
A
–1.3
A
INTEGRATED MOSFETS
RDS_ON_HS
High-side MOSFET ON-resistance
RDS_ON_LS
Low-side MOSFET ON-resistance
SOIC package, VIN = 12 V, IOUT = 1 A
185
WSON package, VIN = 12 V, IOUT = 1 A
160
SOIC package, VIN = 12 V, IOUT = 1 A
105
WSON package, VIN = 12 V, IOUT = 1 A
mΩ
mΩ
95
THERMAL SHUTDOWN
8
TSHDN
Thermal shutdown threshold
THYS
Hysteresis
162
170
15
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178
°C
°C
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7.6 Timing Requirements
Over the recommended operating junction temperature range of –40°C to +125°C (unless otherwise noted)
MIN
NOM
MAX
UNIT
HICCUP MODE
NOC (1)
Number of cycles that LS current
limit is tripped to enter hiccup mode
TOC
Hiccup retry delay time
64
SOIC package
Cycles
5
WSON package
ms
10
SOFT START
TSS
Internal soft-start time
SOIC package, the time of internal
reference to increase from 0 V to 1 V
2
WSON package, the time of internal
reference to increase from 0 V to 1 V
6
ms
ms
POWER GOOD
TPGOOD_RISE
Power-good flag rising transition
deglitch delay
150
μs
TPGOOD_FALL
Power-good flag falling transition
deglitch delay
18
μs
(1)
Ensured by design.
7.7 Switching Characteristics
Over the recommended operating junction temperature range of –40°C to +125°C (unless otherwise noted)
PARAMETER
MIN
TYP
MAX
60
90
UNIT
SW (SW PIN)
TON_MIN
TOFF_MIN
Minimum turnon time
(1)
WSON package
Minimum turnoff time
100
ns
ns
OSCILLATOR (RT and EN/SYNC PIN)
fSW_DEFAULT
fADJ
Oscillator default frequency
Fixed frequency version or RT pin open
circuit
340
400
460
kHz
Minimum adjustable frequency
RT = 198 kΩ with 1% accuracy
150
200
250
kHz
Maximum adjustable frequency
RT = 17.8 kΩ with 1% accuracy
1750
2150
2425
kHz
fSYNC
SYNC frequency range
200
2200
kHz
VSYNC
Amplitude of SYNC clock AC
signal (measured at SYNC pin)
2.8
5.5
TSYNC_MIN
Minimum sync clock ON-time
and OFF-time
(1)
100
V
ns
Specified by design.
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7.8 Typical Characteristics
100
100
90
90
80
80
70
70
Efficiency (%)
Efficiency (%)
Unless otherwise specified the following conditions apply: VIN = 12 V, fSW = 400 kHz, L = 8.2 µH, COUT = 150 µF,
TA = 25°C.
60
50
40
20
10
0
1E-5
0.0001
0.001
0.01
IOUT (A)
0.1
1
40
20
10
0
1E-5
VOUT = 5 V
90
80
80
70
70
Efficiency (%)
100
60
50
40
20
10
0.01
IOUT (A)
0.1
1
1
10
D002
VOUT = 3.3 V
50
40
PFM, VIN = 12 V
PFM, VIN = 24 V
PFM, VIN = 36 V
FPWM, VIN = 12 V
FPWM, VIN = 24 V
FPWM, VIN = 36 V
30
20
10
0
1E-5
10
0.0001
0.001
D003
fSW = 200 kHz (Sync)
0.1
60
PFM, VIN = 12 V
PFM, VIN = 24 V
PFM, VIN = 36 V
FPWM, VIN = 12 V
FPWM, VIN = 24 V
FPWM, VIN = 36 V
30
0.01
IOUT (A)
Figure 7-2. Efficiency vs Load Current
90
0.001
0.001
fSW = 400 kHz
100
0.0001
0.0001
D001
Figure 7-1. Efficiency vs Load Current
0
1E-5
PFM, VIN = 12 V
PFM, VIN = 24 V
PFM, VIN = 36 V
FPWM, VIN = 12 V
FPWM, VIN = 24 V
FPWM, VIN = 36 V
30
10
fSW = 400 kHz
Efficiency (%)
50
PFM, VIN = 12 V
PFM, VIN = 24 V
PFM, VIN = 36 V
FPWM, VIN = 12 V
FPWM, VIN = 24 V
FPWM, VIN = 36 V
30
VOUT = 5 V
0.01
IOUT (A)
0.1
1
10
D004
fSW = 200 kHz (Sync)
Figure 7-3. Efficiency vs Load Current
VOUT = 3.3 V
Figure 7-4. Efficiency vs Load Current
5.09
5.015
VIN = 12 V
VIN = 24 V
VIN = 36 V
VIN = 12 V
VIN = 24 V
VIN = 36 V
5.08
5.07
5.06
5.01
5.05
VOUT (V)
VOUT (V)
60
5.04
5.03
5.005
5.02
5.01
5
5
4.99
0
0.5
1
1.5
IOUT (A)
2
PFM version
2.5
3
0
D004
VOUT = 5 V
1
1.5
IOUT (A)
2
FPWM Version
2.5
3
D005
VOUT = 5 V
Figure 7-6. Load Regulation
Figure 7-5. Load Regulation
10
0.5
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5.5
3.6
5
4.5
VOUT (V)
VOUT (V)
3.3
4
2.7
IOUT = 0.5 A
IOUT = 1.0 A
IOUT = 2.0 A
IOUT = 3.0 A
3.5
2.4
3.3
3
4
4.5
5
VIN (V)
5.5
3
6
IOUT = 0.5 A
IOUT = 1.0 A
IOUT = 2.0 A
IOUT = 3.0 A
3.5
3.7
D006
4.1
4.3
4.5
D007
VOUT = 3.3 V
VOUT = 5 V
Figure 7-8. Dropout Curve
Figure 7-7. Dropout Curve
80
VIN UVLO Rising Threshold (V)
3.67
75
IQ (µA)
3.9
VIN (V)
70
65
60
-50
0
50
Temperature (°C)
100
3.66
3.65
3.64
3.63
3.62
3.61
-50
150
0
50
Temperature (°C)
100
150
D009
D008
VIN = 12 V
VFB = 1.1 V
Figure 7-10. VIN UVLO Rising Threshold vs
Junction Temperature
Figure 7-9. IQ vs Junction Temperature
5.5
0.425
5
0.42
Current Limit (A)
VIN UVLO Hysteresis (V)
LS Limit
HS Limit
0.415
4.5
4
3.5
0.41
-50
0
50
Temperature (°C)
100
150
D010
Figure 7-11. VIN UVLO Hysteresis vs Junction
Temperature
3
-50
0
50
Temperature (°C)
100
150
D011
VIN = 12 V
Figure 7-12. HS and LS Current Limit vs Junction
Temperature
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8 Detailed Description
8.1 Overview
The LMR23630 SIMPLE SWITCHER® regulator is an easy-to-use synchronous step-down DC-DC converter
operating from 4-V to 36-V supply voltage. The device delivers up to 3-A DC load current with good thermal
performance in a small solution size. For both the HSOIC and WSON packages, an extended family is available
in multiple current options from 1 A to 3 A in pin-to-pin compatible packages.
The LMR23630 employs constant frequency peak-current-mode control. The device enters PFM mode at light
load to achieve high efficiency. A user-selectable FPWM version is provided to achieve low output voltage
ripple, tight output voltage regulation, and constant switching frequency. The switching frequency is 400 kHz for
the fixed-frequency version. For the version which has RT pin, the switching frequency is adjustable from 200
kHz to 2.2 MHz. The device is internally compensated, which reduces design time and requires few external
components. The LMR23630 is capable of synchronization to an external clock within the range of 200 kHz to
2.2 MHz.
Additional features such as precision enable, power-good flag, and internal soft-start provide a flexible and
easy-to-use solution for a wide range of applications. Protection features include thermal shutdown, VIN and
VCC undervoltage lockout, cycle-by-cycle current limit, and hiccup-mode short-circuit protection.
The family requires very few external components and has a pinout designed for simple, optimum PCB layout.
8.2 Functional Block Diagram
VCC
EN/SYNC
SYNC Signal
SYNC
Detector
VCC
Enable
LDO
VIN
Precision
Enable
Internal
SS
CBOOT
HS I Sense
EA
REF
Rc
TSD
UVLO
Cc
(PGOOD)
PWM CONTROL LOGIC
PFM
Detector
OV/UV
Detector
SW
FB
Slope
Comp
Freq
Foldback
AGND
Zero
Cross
HICCUP
Detector
SYNC Signal
(RT)
Oscillator
LS I Sense
FB
PGND
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8.3 Feature Description
8.3.1 Fixed Frequency Peak Current Mode Control
The following operating description of the LMR23630 refers to Section 8.2 and to the waveforms in Figure 8-1.
The LMR23630 is a step-down synchronous buck regulator with integrated high-side (HS) and low-side (LS)
switches (synchronous rectifier). The LMR23630 supplies a regulated output voltage by turning on the HS and
LS NMOS switches with controlled duty cycle. During high-side switch ON-time, the SW pin voltage swings up
to approximately VIN, and the inductor current iL increase with linear slope (VIN – V OUT) / L. When the HS switch
is turned off by the control logic, the LS switch is turned on after an anti-shoot-through dead time. Inductor
current discharges through the LS switch with a slope of –VOUT / L. The control parameter of a buck converter is
defined as duty cycle D = tON / TSW, where tON is the high-side switch ON-time and TSW is the switching period.
The regulator control loop maintains a constant output voltage by adjusting the duty cycle D. In an ideal buck
converter, where losses are ignored, D is proportional to the output voltage and inversely proportional to the
input voltage: D = VOUT / VIN.
VSW
SW Voltage
D = tON/ TSW
VIN
tON
tOFF
t
0
-VD
Inductor Current
iL
TSW
ILPK
IOUT
'iL
t
0
Figure 8-1. SW Node and Inductor Current Waveforms in Continuous Conduction Mode (CCM)
The LMR23630 employs fixed-frequency peak-current-mode control. A voltage feedback loop is used to get
accurate DC voltage regulation by adjusting the peak current command based on voltage offset. The peak
inductor current is sensed from the high-side switch and compared to the peak current threshold to control the
ON-time of the high-side switch. The voltage feedback loop is internally compensated, which allows for fewer
external components, makes it easy to design, and provides stable operation with almost any combination of
output capacitors. The regulator operates with fixed switching frequency at normal load condition. At light load
condition, the LMR23630 operates in PFM mode to maintain high efficiency (PFM option) or in FPWM mode for
low output-voltage ripple, tight output-voltage regulation, and constant switching frequency (FPWM option).
8.3.2 Adjustable Frequency
The switching frequency can be programmed for the adjustable-switching-frequency version of LMR23630 by
the impedance RT from the RT pin to ground. The frequency is inversely proportional to the RT resistance. The
RT pin can be left floating and the LMR23630 operates at 400-kHz default switching frequency. The RT pin is
not designed to be shorted to ground. For a desired frequency, typical RT resistance can be found by Equation 1.
Table 8-1 gives typical RT values for a given fSW.
RT(kΩ) = 40200 / fSW(kHz) – 0.6
(1)
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250
RT Resistance (kŸ)
200
150
100
50
0
0
500
1000
1500
2000
Switching Frequency (kHz)
2500
C008
Figure 8-2. RT vs Frequency Curve
Table 8-1. Typical Frequency Setting RT Resistance
fSW (kHz)
RT (kΩ)
200
200
350
115
500
78.7
750
53.6
1000
39.2
1500
26.1
2000
19.6
2200
17.8
8.3.3 Adjustable Output Voltage
A precision 1-V reference voltage is used to maintain a tightly regulated output voltage over the entire operating
temperature range. The output voltage is set by a resistor divider from output voltage to the FB pin. TI
recommends using 1% tolerance resistors with a low temperature coefficient for the FB divider. Select the
lowside resistor RFBB for the desired divider current and use Equation 2 to calculate high-side RFBT. RFBT in the
range from 10 kΩ to 100 kΩ is recommended for most applications. A lower RFBT value can be used if static
loading is desired to reduce VOUT offset in PFM operation. Lower RFBT will reduce efficiency at very light load.
Less static current goes through a larger RFBT and might be more desirable when light load efficiency is critical.
However, RFBT larger than 1 MΩ is not recommended because it makes the feedback path more susceptible
to noise. Larger RFBT value requires more carefully designed feedback path on the PCB. The tolerance and
temperature variation of the resistor dividers affect the output voltage regulation.
VOUT
RFBT
FB
RFBB
Figure 8-3. Output Voltage Setting
RFBT
14
VOUT VREF
u RFBB
VREF
(2)
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8.3.4 Enable/Sync
The voltage on the EN/SYNC pin controls the ON or OFF operation of LMR23630. A voltage less than 1 V
(typical) shuts down the device while a voltage higher than 1.6 V (typical) is required to start the regulator.
The EN/SYNC pin is an input and cannot be left open or floating. The simplest way to enable the operation of
the LMR23630 is to connect the EN to VIN. This allows self-start-up of the LMR23630 when VIN is within the
operation range.
Many applications can benefit from the employment of an enable divider RENT and RENB (Figure 8-4) to establish
a precision system UVLO level for the converter. System UVLO can be used for supplies operating from utility
power as well as battery power. It can be used for sequencing, ensuring reliable operation, or supply protection,
such as a battery discharge level. An external logic signal can also be used to drive EN input for system
sequencing and protection.
VIN
RENT
EN/SYNC
RENB
Figure 8-4. System UVLO by Enable Divider
The EN pin also can be used to synchronize the internal oscillator to an external clock. The internal oscillator
can be synchronized by AC-coupling a positive edge into the EN pin. The AC-coupled peak-to-peak voltage
at the EN pin must exceed the SYNC amplitude threshold of 2.8 V (typical) to trip the internal synchronization
pulse detector, and the minimum SYNC clock ON-time and OFF-time must be longer than 100 ns (typical).
A 3.3-V or a higher amplitude pulse signal coupled through a 1-nF capacitor CSYNC is a good starting point.
Keeping RENT // RENB (RENT parallel with RENB) in the 100 kΩ range is a good choice. RENT is required for this
synchronization circuit, but RENB can be left unmounted if system UVLO is not needed. LMR23630 switching
action can be synchronized to an external clock from 200 kHz to 2.2 MHz. Figure 8-6 and Figure 8-7 show the
device synchronized to an external system clock.
VIN
CSYNC
RENT
EN/SYNC
RENB
Clock
Source
Figure 8-5. Synchronize to External Clock
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Figure 8-6. Synchronizing in PWM Mode
Figure 8-7. Synchronizing in PFM Mode
8.3.5 VCC, UVLO
The LMR23630 integrates an internal LDO to generate VCC for control circuitry and MOSFET drivers. The
nominal voltage for VCC is 4.1 V. The VCC pin is the output of an LDO and must be properly bypassed. Place a
high-quality ceramic capacitor with a value of 2.2 µF to 10 µF, 16 V or higher rated voltage as close as possible
to VCC and grounded to the exposed PAD and ground pins. The VCC output pin must not be loaded, or shorted
to ground during operation. Shorting VCC to ground during operation may cause damage to the LMR23630.
VCC undervoltage lockout (UVLO) prevents the LMR23630 from operating until the VCC voltage exceeds 3.3 V
(typical). The VCC UVLO threshold has 400 mV (typical) of hysteresis to prevent undesired shutdown due to
temporary VIN drops.
8.3.6 Minimum ON-time, Minimum OFF-time and Frequency Foldback at Dropout Conditions
Minimum ON-time, TON_MIN, is the smallest duration of time that the HS switch can be on. TON_MIN is typically
60 ns in the LMR23630. Minimum OFF-time, TOFF_MIN, is the smallest duration that the HS switch can be
off. TOFF_MIN is typically 100 ns in the LMR23630. In CCM operation, TON_MIN and TOFF_MIN limit the voltage
conversion range given a selected switching frequency.
The minimum duty cycle allowed is:
DMIN = TON_MIN × fSW
(3)
And the maximum duty cycle allowed is:
DMAX = 1 – TOFF_MIN × fSW
(4)
Given fixed TON_MIN and TOFF_MIN, the higher the switching frequency the narrower the range of the allowed
duty cycle. In the LMR23630, a frequency foldback scheme is employed to extend the maximum duty cycle
when TOFF_MIN is reached. The switching frequency decreases once longer duty cycle is needed under low VIN
conditions. Wide range of frequency foldback allows the LMR23630 output voltage stay in regulation with a much
lower supply voltage VIN. This leads to a lower effective dropout voltage.
Given an output voltage, the choice of the switching frequency affects the allowed input voltage range, solution
size and efficiency. The maximum operation supply voltage can be found by:
VIN _ MAX
VOUT
fSW u TON _ MIN
(5)
At lower supply voltage, the switching frequency decreases once TOFF_MIN is tripped. The minimum VIN without
frequency foldback can be approximated by:
16
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VOUT
1 fSW u TOFF _ MIN
(6)
Taking considerations of power losses in the system with heavy load operation, VIN_MAX is higher than the result
calculated in Equation 5. With frequency foldback, VIN_MIN is lowered by decreased fSW.
450
400
Frequency (kHz)
350
300
250
200
150
IOUT = 0.5 A
IOUT = 1.0 A
IOUT = 2.0 A
IOUT = 3.0 A
100
50
0
4.6
4.8
5
5.2
5.4
5.6
VIN (V)
5.8
6
6.2
6.4
D013
Figure 8-8. Frequency Foldback at Dropout (VOUT = 5 V, fSW = 400 kHz)
8.3.7 Power Good (PGOOD)
The power-good version of LMR23630 has a built in power-good flag shown on PGOOD pin to indicate whether
the output voltage is within its regulation level. The PGOOD signal can be used for start-up sequencing of
multiple rails or fault protection. The PGOOD pin is an open-drain output that requires a pullup resistor to an
appropriate DC voltage. Voltage detected by the PGOOD pin must never exceed 15 V, and limit the maximum
current into this pin to 1 mA. A typical range of pullup resistor value is 10 kΩ to 100 kΩ.
When the FB voltage is within the power-good band, +6% above and –6% below the internal reference voltage
VREF typically, the PGOOD switch is turned off, and the PGOOD voltage is as high as the pulled-up voltage.
When the FB voltage is outside of the tolerance band, +7% above or –7% below VREF typically, the PGOOD
switch is turned on, and the PGOOD pin voltage is pulled low to indicate power bad. A glitch filter prevents false
flag operation for short excursions in the output voltage, such as during line and load transients. The values for
the various filter and delay times can be found in Section 7.6. Power-good operation can best be understood by
reference to Figure 8-9.
VREF
107%
106%
94%
93%
PGOOD
High
Low
Figure 8-9. Power-Good Flag
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8.3.8 Internal Compensation and CFF
The LMR23630 is internally compensated as shown in Section 8.2. The internal compensation is designed so
that the loop response is stable over the entire operating frequency and output voltage range. Depending on the
output voltage, the compensation loop phase margin can be low with all ceramic capacitors. TI recommends an
external feed-forward capacitor CFF be placed in parallel with the top resistor divider RFBT for optimum transient
performance.
VOUT
RFBT
CFF
FB
RFBB
Figure 8-10. Feed-forward Capacitor for Loop Compensation
The feed-forward capacitor CFF in parallel with RFBT places an additional zero before the crossover frequency of
the control loop to boost phase margin. The zero frequency can be found by:
fZ _ CFF
1
2S u CFF u RFBT
(7)
An additional pole is also introduced with CFF at the frequency of:
fP _ CFF
1
2S u CFF u RFBT //RFBB
(8)
The zero fZ_CFF adds phase boost at the crossover frequency and improves transient response. The pole fP-CFF
helps maintaining proper gain margin at frequency beyond the crossover. Table 9-1 lists the combination of
COUT, CFF and RFBT for typical applications, designs with similar COUT but RFBT other than recommended value,
adjust CFF such that (CFF × RFBT) is unchanged and adjust RFBB such that (RFBT / RFBB) is unchanged.
Designs with different combinations of output capacitors need different CFF. Different types of capacitors have
different equivalent series resistance (ESR). Ceramic capacitors have the smallest ESR and need the most CFF.
Electrolytic capacitors have much larger ESR than ceramic, and the ESR zero frequency location would be low
enough to boost the phase up around the crossover frequency. Designs that use mostly electrolytic capacitors at
the output may not need any CFF. The location of this ESR zero frequency can be calculated with Equation 9:
fZ _ESR
1
2S u COUT u ESR
(9)
The CFF creates a time constant with RFBT that couples in the attenuate output voltage ripple to the FB node.
If the CFF value is too large, it can couple too much ripple to the FB and affect VOUT regulation. Therefore,
calculate CFF based on output capacitors used in the system. At cold temperatures, the value of CFF might
change based on the tolerance of the chosen component. This may reduce its impedance and ease noise
coupling on the FB node. To avoid this, more capacitance can be added to the output or the value of CFF can be
reduced.
8.3.9 Bootstrap Voltage (BOOT)
The LMR23630 provides an integrated bootstrap voltage regulator. A small capacitor between the BOOT and
SW pins provides the gate drive voltage for the high-side MOSFET. The BOOT capacitor is refreshed when the
high-side MOSFET is off and the low-side switch conducts. The recommended value of the BOOT capacitor is
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0.1 μF to 0.47 μF. TI recommends a ceramic capacitor with an X7R or X5R grade dielectric with a voltage rating
of 16 V or higherfor stable performance over temperature and voltage.
8.3.10 Overcurrent and Short-Circuit Protection
The LMR23630 is protected from over-current conditions by cycle-by-cycle current limit on both the peak and
valley of the inductor current. Hiccup mode will be activated if a fault condition persists to prevent over-heating.
High-side MOSFET overcurrent protection is implemented by the nature of the peak-current-mode control. The
HS switch current is sensed when the HS is turned on after a set blanking time. The HS switch current is
compared to the output of the error amplifier (EA) minus slope compensation every switching cycle. See the
Section 8.2 for more details. The peak current of HS switch is limited by a clamped maximum peak current
threshold IHS_LIMIT which is constant. Thus, the peak current limit of the high-side switch is not affected by the
slope compensation and remains constant over the full duty cycle range.
The current going through LS MOSFET is also sensed and monitored. When the LS switch turns on, the inductor
current begins to ramp down. The LS switch is not turned OFF at the end of a switching cycle if its current is
above the LS current limit ILS_LIMIT. The LS switch is kept ON so that inductor current keeps ramping down, until
the inductor current ramps below the LS current limit ILS_LIMIT. Then the LS switch is turned OFF, and the HS
switch is turned on after a dead time. This is somewhat different than the more typical peak current limit and
results in Equation 10 for the maximum load current.
IOUT _ MAX
ILS _ LIMIT
VIN
VOUT
2 u fSW u L
u
VOUT
VIN
(10)
If the current of the LS switch is higher than the LS current limit for 64 consecutive cycles, hiccup current
protection mode is activated. In hiccup mode, the regulator is shut down and kept off for 5 ms typically before
the LMR23630 tries to start again. If overcurrent or short-circuit fault condition still exist, hiccup will repeat until
the fault condition is removed. Hiccup mode reduces power dissipation under severe overcurrent conditions,
prevents over-heating and potential damage to the device.
For FPWM version, the inductor current is allowed to go negative. If this current exceed IL_NEG, the LS switch is
turned off until the next clock cycle. This is used to protect the LS switch from excessive negative current.
8.3.11 Thermal Shutdown
The LMR23630 provides an internal thermal shutdown to protect the device when the junction temperature
exceeds 170°C (typical). The device is turned off when thermal shutdown activates. Once the die temperature
falls below 155°C (typical), the device reinitiates the power-up sequence controlled by the internal soft-start
circuitry.
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8.4 Device Functional Modes
8.4.1 Shutdown Mode
The EN pin provides electrical ON and OFF control for the LMR23630. When VEN is below 1 V (typical), the
device is in shutdown mode. The LMR23630 also employs VIN and VCC UVLO protection. If VIN or V CC voltage
is below their respective UVLO level, the regulator is turned off.
8.4.2 Active Mode
The LMR23630 is in active mode when VEN is above the precision enable threshold, VIN and VCC are above
their respective UVLO level. The simplest way to enable the LMR23630 is to connect the EN pin to VIN pin. This
allows self startup when the input voltage is in the operating range 4 V to 36 V. See Section 8.3.5 and Section
8.3.4 for details on setting these operating levels.
In active mode, depending on the load current, the LMR23630 is in one of four modes:
1. Continuous conduction mode (CCM) with fixed switching frequency when load current is above half of the
peak-to-peak inductor current ripple (for both PFM and FPWM options).
2. Discontinuous conduction mode (DCM) with fixed switching frequency when load current is lower than half of
the peak-to-peak inductor current ripple in CCM operation (only for PFM option).
3. Pulse frequency modulation mode (PFM) when switching frequency is decreased at very light load (only for
PFM option).
4. Forced pulse width modulation mode (FPWM) with fixed switching frequency even at light load (only for
FPWM option).
8.4.3 CCM Mode
CCM operation is employed in the LMR23630 when the load current is higher than half of the peak-to-peak
inductor current. In CCM operation, the frequency of operation is fixed, output voltage ripple is at a minimum in
this mode, and the maximum output current of 3 A can be supplied by the LMR23630.
8.4.4 Light Load Operation (PFM Version)
For PFM version, when the load current is lower than half of the peak-to-peak inductor current in CCM, the
LMR23630 operates in DCM, also known as diode emulation mode (DEM). In DCM, the LS switch is turned
off when the inductor current drops to IL_ZC (–40 mA typical). Both switching losses and conduction losses are
reduced in DCM, compared to forced PWM operation at light load.
At even lighter current loads, PFM is activated to maintain high efficiency operation. When either the minimum
HS switch ON-time (tON_MIN ) or the minimum peak inductor current IPEAK_MIN (300 mA typ) is reached, the
switching frequency decreases to maintain regulation. In PFM, switching frequency is decreased by the control
loop when load current reduces to maintain output voltage regulation. Switching loss is further reduced in
PFM operation due to less frequent switching actions. The external clock synchronizing is not valid when the
LMR23630 device enters into PFM mode.
8.4.5 Light Load Operation (FPWM Version)
For FPWM version, LMR23630 is locked in PWM mode at full load range. This operation is maintained, even at
no-load, by allowing the inductor current to reverse its normal direction. This mode trades off reduced light load
efficiency for low output voltage ripple, tight output voltage regulation, and constant switching frequency. In this
mode, a negative current limit of IL_NEG is imposed to prevent damage to the regulators low side FET. When in
FPWM mode the converter synchronizes to any valid clock signal on the EN/SYNC input.
20
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9 Application and Implementation
Note
Information in the following applications sections is not part of the TI component specification, and
TI does not warrant its accuracy or completeness. TI’s customers are responsible for determining
suitability of components for their purposes. Customers should validate and test their design
implementation to confirm system functionality.
9.1 Application Information
The LMR23630 is a step-down DC-to-DC regulator. It is typically used to convert a higher DC voltage to a
lower DC voltage with a maximum output current of 3 A. The following design procedure can be used to
select components for the LMR23630. Alternately, the WEBENCH® software may be used to generate complete
designs. When generating a design, the WEBENCH® software utilizes iterative design procedure and accesses
comprehensive databases of components. See www.ti.com for more details.
9.2 Typical Applications
The LMR23630 only requires a few external components to convert from a wide voltage-range supply to a fixed
output voltage. Figure 9-1 shows a basic schematic.
VIN 12 V
BOOT
VIN
CBOOT
0.47 F
L
10 H
CIN
10 F
EN/
SYNC
SW
PAD
CFF
47 pF
FB
CVCC
2.2 F
VCC
VOUT
5 V/3 A
RFBT
88.7 NŸ
COUT
100 F
RFBB
22.1 NŸ
PGND
AGND
Figure 9-1. LM23630 Application Circuit
The external components must fulfill the needs of the application, but also the stability criteria of the device
control loop. Table 9-1 can be used to simplify the output filter component selection.
Table 9-1. L, COUT, and CFF Typical Values
fSW (kHz)
200
400
1000
2200
(1)
(2)
VOUT (V)
L (µH) (2)
COUT (µF) (3)
CFF (pF)
RFBT (kΩ)(4) (5)
3.3
15
300
150
51
5
18
200
100
88.7
12
33
100
See(1)
243
510
24
33
47
See(1)
3.3
6.8
150
75
51
5
10
100
47
88.7
12
15
68
See(1)
243
510
24
15
47
See(1)
3.3
3.3
68
39
51
5
4.7
47
25
88.7
3.3
2.2
47
33
51
5
2.2
33
18
88.7
High ESR COUT gives enough phase boost and CFF not needed.
Inductance value is calculated based on VIN = 36 V.
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(3)
(4)
(5)
All the COUT values are after derating. Add more when using ceramic capacitors.
RFBT = 0 Ω for VOUT = 1 V. RFBB = 22.1 kΩ for all other VOUT setting.
For designs with RFBT other than recommended value, please adjust CFF such that (CFF × RFBT) is unchanged and adjust RFBB such
that (RFBT / RFBB) is unchanged.
9.2.1 Design Requirements
Detailed design procedure is described based on a design example. For this design example, use the
parameters listed in Table 9-2 as the input parameters.
Table 9-2. Design Example Parameters
DESIGN PARAMETER
EXAMPLE VALUE
Input voltage, VIN
12 V typical, range from 8 V to 28 V
Output voltage, VOUT
5V
Maximum output current IO_MAX
3A
Transient response 0.2 A to 2.5 A
5%
Output voltage ripple
50 mV
Input voltage ripple
400 mV
Switching frequency fSW
400 kHz
9.2.2 Detailed Design Procedure
9.2.2.1 Custom Design With WEBENCH® Tools
Click here to create a custom design using the LMR23625 device with the WEBENCH® Power Designer.
1. Start by entering the input voltage (VIN), output voltage (VOUT), and output current (IOUT) requirements.
2. Optimize the design for key parameters such as efficiency, footprint, and cost using the optimizer dial.
3. Compare the generated design with other possible solutions from Texas Instruments.
The WEBENCH Power Designer provides a customized schematic along with a list of materials with real-time
pricing and component availability.
In most cases, these actions are available:
• Run electrical simulations to see important waveforms and circuit performance
• Run thermal simulations to understand board thermal performance
• Export customized schematic and layout into popular CAD formats
• Print PDF reports for the design, and share the design with colleagues
Get more information about WEBENCH tools at www.ti.com/WEBENCH.
9.2.2.2 Output Voltage Setpoint
The output voltage of LMR23630 is externally adjustable using a resistor divider network. The divider network is
comprised of top feedback resistor RFBT and bottom feedback resistor RFBB. Equation 11 is used to determine
the output voltage:
RFBT
VOUT VREF
u RFBB
VREF
(11)
Choose the value of RFBB to be 22.1 kΩ. With the desired output voltage set to 5 V and the VREF = 1 V, the RFBB
value can then be calculated using Equation 11. The formula yields to a value 88.7 kΩ.
9.2.2.3 Switching Frequency
The default switching frequency of the LMR23630 is 400 kHz. For other switching frequency, the device must be
synchronized to an external clock, see Section 8.3.4 for more details.
9.2.2.4 Inductor Selection
The most critical parameters for the inductor are the inductance, saturation current, and the rated current. The
inductance is based on the desired peak-to-peak ripple current ΔiL. Because the ripple current increases with
22
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the input voltage, the maximum input voltage is always used to calculate the minimum inductance LMIN. Use
Equation 12 to calculate the minimum value of the output inductor. KIND is a coefficient that represents the
amount of inductor ripple current relative to the maximum output current of the device. A reasonable value of
KIND should be 20% to 40%. During an instantaneous short or overcurrent operation event, the RMS and peak
inductor current can be high. The inductor current rating should be higher than the current limit of the device.
'iL
LMIN
VOUT u VIN _ MAX
VOUT
VIN _ MAX u L u fSW
VIN _ MAX
VOUT
IOUT u KIND
u
(12)
VOUT
VIN _ MAX u fSW
(13)
In general, it is preferable to choose lower inductance in switching power supplies, because it usually
corresponds to faster transient response, smaller DCR, and reduced size for more compact designs. However,
inductance that is too low can generate an inductor current ripple that is too high so that overcurrent protection
at the full load could be falsely triggered. It also generates more conduction loss and inductor core loss. Larger
inductor current ripple also implies larger output voltage ripple with same output capacitors. With peak current
mode control, TI does not recommend having an inductor current ripple that is too small. A larger peak-current
ripple improves the comparator signal-to-noise ratio.
For this design example, choose KIND = 0.4, the minimum inductor value is calculated to be 8.56 µH. Choose the
nearest standard 8.2 μH ferrite inductor with a capability of 4-A RMS current and 6-A saturation current.
9.2.2.5 Output Capacitor Selection
Choose the output capacitor(s), COUT, with care because it directly affects the steady-state output-voltage ripple,
loop stability, and the voltage over/undershoot during load-current transients.
The output ripple is essentially composed of two parts. One is caused by the inductor current ripple going
through the equivalent series resistance (ESR) of the output capacitors:
'VOUT_ESR
'iL u ESR
KIND u IOUT u ESR
(14)
The other is caused by the inductor current ripple charging and discharging the output capacitors:
'VOUT _ C
'iL
8 u fSW u COUT
KIND u IOUT
8 u fSW u COUT
(15)
The two components in the voltage ripple are not in phase, so the actual peak-to-peak ripple is smaller than the
sum of two peaks.
Output capacitance is usually limited by transient performance specifications if the system requires tight voltage
regulation with presence of large current steps and fast slew rate. When a fast large load increase happens,
output capacitors provide the required charge before the inductor current can slew up to the appropriate level.
The control loop of the regulator usually needs four or more clock cycles to respond to the output voltage droop.
The output capacitance must be large enough to supply the current difference for four clock cycles to maintain
the output voltage within the specified range. Equation 17 shows the minimum output capacitance needed
for specified output undershoot. When a sudden large load decrease happens, the output capacitors absorb
energy stored in the inductor. which results in an output voltage overshoot. Equation 14 calculates the minimum
capacitance required to keep the voltage overshoot within a specified range.
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COUT !
4 u IOH IOL
fSW u VUS
(16)
2
2
IOH
IOL
COUT !
VOUT
VOS
2
2
VOUT
uL
(17)
where
•
•
•
•
IOL = Low level output current during load transient
IOH = High level output current during load transient
VUS = Target output voltage undershoot
VOS = Target output voltage overshoot
For this design example, the target output ripple is 50 mV. Presuppose ΔVOUT_ESR = ΔVOUT_C = 50 mV, and
chose KIND = 0.4. Equation 16 yields ESR no larger than 41.7 mΩ and Equation 17 yields COUT no smaller than
7.5 μF. For the target over/undershoot range of this design, VUS = VOS = 5% × VOUT = 250 mV. The COUT can be
calculated to be no smaller than 108 μF and 28.5 μF by Equation 15 and Equation 17, respectively. Consider of
derating, one 47-μF, 16-V and one 100-μF, 10-V ceramic capacitor with 5-mΩ ESR are used in parallel.
9.2.2.6 Feed-Forward Capacitor
The LMR23630 is internally compensated. Depending on the VOUT and frequency fSW, if the output capacitor
COUT is dominated by low-ESR (ceramic types) capacitors, it could result in low phase margin. To improve the
phase boost an external feed-forward capacitor CFF can be added in parallel with RFBT. CFF is chosen such
that phase margin is boosted at the crossover frequency without CFF. A simple estimation for the crossover
frequency (fX) without CFF is shown in Equation 18, assuming COUT has very small ESR, and COUT value is after
derating.
fX
8.32
VOUT u COUT
(18)
Equation 19 for CFF was tested:
CFF
1
4S u fX u RFBT
(19)
For designs with higher ESR, CFF is not needed when COUT has very high ESR and CFF calculated from
Equation 19 must reduced with medium ESR. Table 9-1 can be used as a quick starting point.
For the application in this design example, a 47-pF, 50-V, COG capacitor is selected.
9.2.2.7 Input Capacitor Selection
The LMR23630 device requires high-frequency input decoupling capacitor(s) and a bulk input capacitor,
depending on the application. The typical recommended value for the high-frequency decoupling capacitor is
4.7 μF to 10 μF. TI recommends a high-quality ceramic capacitor type X5R or X7R with sufficiency voltage
rating. To compensate the derating of ceramic capacitors, a voltage rating twice the maximum input voltage is
recommended. Additionally, some bulk capacitance can be required, especially if the LMR23630 circuit is not
located within approximately 5 cm from the input voltage source. This capacitor is used to provide damping to
the voltage spike due to the lead inductance of the cable or the trace. For this design, two 4.7-μF, 50-V, X7R
ceramic capacitors are used. A 0.1-μF for high-frequency filtering and place it as close as possible to the device
pins.
9.2.2.8 Bootstrap Capacitor Selection
Every LMR23630 design requires a bootstrap capacitor (CBOOT). TI recommends a capacitor of 0.47 μF, ated 16
V or higher. The bootstrap capacitor is located between the SW pin and the BOOT pin. The bootstrap capacitor
must be a high-quality ceramic type with an X7R or X5R grade dielectric for temperature stability.
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9.2.2.9 VCC Capacitor Selection
The VCC pin is the output of an internal LDO for LMR23630. To insure stability of the device, place a minimum of
2.2-μF, 16V, X7R capacitor from this pin to ground.
9.2.2.10 Undervoltage Lockout Setpoint
The system undervoltage lockout (UVLO) is adjusted using the external voltage divider network of RENT and
RENB. The UVLO has two thresholds, one for power up when the input voltage is rising and one for power down
or brown outs when the input voltage is falling. The following equation can be used to determine the VIN UVLO
level.
VIN _ RISING
VENH u
RENT RENB
RENB
(20)
The EN rising threshold (VENH) for LMR23630 is set to be 1.55 V (typical). Choose the value of RENB to be 287
kΩ to minimize input current from the supply. If the desired VIN UVLO level is at 6 V, then the value of RENT can
be calculated using Equation 21:
RENT
§ VIN _ RISING
¨¨
© VENH
·
1¸¸ u RENB
¹
(21)
Equation 21 yields a value of 820 kΩ. The resulting falling UVLO threshold, equals 4.4 V, can be calculated by
Equation 22, where EN hysteresis (VEN_HYS) is 0.4 V (typical).
VIN _ FALLING
VENH
VEN _ HYS u
RENT RENB
RENB
(22)
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9.2.3 Application Curves
Unless otherwise specified the following conditions apply: VIN = 12 V, fSW = 400 kHz, L = 8.2 µH, COUT = 150 µF, TA = 25 °C.
VOUT = 5 V
IOUT = 3 A
fSW = 400 kHz
VOUT = 5 V
Figure 9-2. CCM Mode
VOUT = 5 V
IOUT = 0 mA
VOUT = 5 V
fSW = 400 kHz
VOUT = 5 V
IOUT = 0 mA
fSW = 400 kHz
Figure 9-5. FPWM Mode
IOUT = 2 A
VIN = 12 V
Figure 9-6. Start Up by VIN
26
fSW = 400 kHz
Figure 9-3. DCM Mode
Figure 9-4. PFM Mode
VIN = 12 V
IOUT = 150 mA
VOUT = 5 V
IOUT = 2 A
Figure 9-7. Start Up by EN
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VIN = 12 V
VOUT = 5 V
IOUT = 0.3 A to 3 A, 100 mA / μs
VOUT = 5 V
IOUT = 3 A
Figure 9-8. Load Transient
VOUT = 5 V
VIN = 7 V to 36 V, 2 V / μs
Figure 9-9. Line Transient
IOUT = 1 A to short
VOUT = 5 V
Figure 9-10. Short Protection
IOUT = short to 1 A
Figure 9-11. Short Recovery
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10 Power Supply Recommendations
The LMR23630 is designed to operate from an input voltage supply range between 4.5 V and 36 V for the
HSOIC package and 4 V to 36 V for the WSON package. This input supply must be able to withstand the
maximum input current and maintain a stable voltage. The resistance of the input supply rail must be low enough
that an input current transient does not cause a high enough drop at the LMR23630 supply voltage that can
cause a false UVLO fault triggering and system reset. If the input supply is located more than a few inches from
the LMR23630, additional bulk capacitance may be required in addition to the ceramic input capacitors. The
amount of bulk capacitance is not critical, but a 47-μF or 100-μF electrolytic capacitor is a typical choice.
11 Layout
11.1 Layout Guidelines
Layout is a critical portion of good power supply design. The following guidelines will help users design a PCB
with the best power conversion performance, thermal performance, and minimized generation of unwanted EMI.
1. The input bypass capacitor CIN must be placed as close as possible to the VIN and PGND pins. Grounding
for both the input and output capacitors should consist of localized top side planes that connect to the PGND
pin and PAD.
2. Place bypass capacitors for VCC close to the VCC pin and ground the bypass capacitor to device ground.
3. Minimize trace length to the FB pin net. Both feedback resistors, RFBT and RFBB should be located close to
the FB pin. Place CFF directly in parallel with RFBT. If VOUT accuracy at the load is important, make sure VOUT
sense is made at the load. Route VOUT sense path away from noisy nodes and preferably through a layer on
the other side of a shielded layer.
4. Use ground plane in one of the middle layers as noise shielding and heat dissipation path.
5. Have a single point ground connection to the plane. The ground connections for the feedback and enable
components should be routed to the ground plane. This prevents any switched or load currents from flowing
in the analog ground traces. If not properly handled, poor grounding can result in degraded load regulation or
erratic output voltage ripple behavior.
6. Make VIN, VOUT and ground bus connections as wide as possible. This reduces any voltage drops on the
input or output paths of the converter and maximizes efficiency.
7. Provide adequate device heat-sinking. Use an array of heat-sinking vias to connect the exposed pad to the
ground plane on the bottom PCB layer. If the PCB has multiple copper layers, these thermal vias can also be
connected to inner layer heat-spreading ground planes. Ensure enough copper area is used for heat-sinking
to keep the junction temperature below 125 °C.
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11.2 Layout Example
Output Bypass
Capacitor
Output Inductor
SW
Input Bypass
Capacitor
PGND
BOOT Capacitor
BOOT
VCC
Capacitor
VIN
VCC
AGND
FB
EN/
SYNC
UVLO Adjust Resistor
Output Voltage Set
Resistor
Thermal VIA
VIA (Connect to GND Plane)
Figure 11-1. Sample HSOIC Package Layout
Output
Inductor
Output Bypass
Capacitor
BOOT
Capacitor
VCC
Capacitor
SW
PGND
SW
NC
BOOT
VIN
VCC
VIN
FB
EN/SYNC
RT
AGND
Input Bypass
Capacitor
UVLO Adjust
Resistor
RT
Thermal VIA
Output Voltage
Set Resistor
VIA (Connect to GND Plane)
Figure 11-2. Sample WSON Package Layout
11.3 Compact Layout for EMI Reduction
Radiated EMI is generated by the high di/dt components in pulsing currents in switching converters. The larger
area covered by the path of a pulsing current, the more EMI is generated. High frequency ceramic bypass
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capacitors at the input side provide primary path for the high di/dt components of the pulsing current. Placing
ceramic bypass capacitor(s) as close as possible to the VIN and PGND pins is the key to EMI reduction.
The SW pin connecting to the inductor must be as short as possible and just wide enough to carry the load
current without excessive heating. Use short, thick traces or copper pours (shapes) high current conduction path
to minimize parasitic resistance. Place the output capacitors close to the VOUT end of the inductor and closely
grounded to PGND pin and exposed PAD.
Place the bypass capacitors on VCC as close as possible to the pin and closely grounded to PGND and the
exposed PAD.
11.4 Ground Plane and Thermal Considerations
TI recommends using one of the middle layers as a solid ground plane. Ground plane provides shielding
for sensitive circuits and traces. It also provides a quiet reference potential for the control circuitry. Connect
the AGND and PGND pins to the ground plane using vias right next to the bypass capacitors. PGND pin is
connected to the source of the internal LS switch. They must be connected directly to the grounds of the input
and output capacitors. The PGND net contains noise at switching frequency and may bounce due to load
variations. PGND trace, as well as VIN and SW traces, must be constrained to one side of the ground plane. The
other side of the ground plane contains much less noise and should be used for sensitive routes.
TI also recommends providing adequate device heat sinking by utilizing the PAD of the device as the primary
thermal path. Use a minimum 4 by 2 array of 12 mil thermal vias to connect the PAD to the system ground plane
heat sink. The vias should be evenly distributed under the PAD. Use as much copper as possible, for system
ground plane, on the top and bottom layers for the best heat dissipation. Use a four-layer board with the copper
thickness for the four layers, starting from the top of, 2 oz / 1 oz / 1 oz / 2 oz. Four-layer boards with enough
copper thickness provides low current conduction impedance, proper shielding, and lower thermal resistance.
The thermal characteristics of the LMR23630 are specified using the parameter RθJA, which characterize the
junction temperature of silicon to the ambient temperature in a specific system. Although the value of RθJA is
dependent on many variables, it still can be used to approximate the operating junction temperature of the
device. To obtain an estimate of the device junction temperature, one may use the following relationship:
TJ = PD × RθJA + TA
(23)
PD = VIN × IIN × (1 – Efficiency) – 1.1 × IOUT 2 × DCR in watt
(24)
where
•
•
•
•
•
TJ = junction temperature in °C
PD = device power dissipation in watt
RθJA = junction-to-ambient thermal resistance of the device in °C/W
TA = ambient temperature in °C
DCR = inductor DC parasitic resistance in ohm
The maximum operating junction temperature of the LMR23630 is 125°C. RθJA is highly related to PCB size and
layout, as well as environmental factors such as heat sinking and air flow.
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11.5 Feedback Resistors
To reduce noise sensitivity of the output voltage feedback path, it is important to place the resistor divider and
CFF close to the FB pin, rather than close to the load. The FB pin is the input to the error amplifier, so it is a high
impedance node and very sensitive to noise. Placing the resistor divider and CFF closer to the FB pin reduces
the trace length of FB signal and reduces noise coupling. The output node is a low impedance node, so the trace
from VOUT to the resistor divider can be long if short path is not available.
If voltage accuracy at the load is important, make sure voltage sense is made at the load. Doing so corrects
for voltage drops along the traces and provide the best output accuracy. The voltage sense trace from the
load to the feedback resistor divider should be routed away from the SW node path and the inductor to
avoid contaminating the feedback signal with switch noise, while also minimizing the trace length. This is most
important when high value resistors are used to set the output voltage. TI recommends routing the voltage sense
trace and place the resistor divider on a different layer than the inductor and SW node path, such that there is a
ground plane in between the feedback trace and inductor/SW node polygon. This provides further shielding for
the voltage feedback path from EMI noises.
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12 Device and Documentation Support
12.1 Device Support
12.1.1 Development Support
12.1.1.1 Custom Design With WEBENCH® Tools
Click here to create a custom design using the LMR23625 device with the WEBENCH® Power Designer.
1. Start by entering the input voltage (VIN), output voltage (VOUT), and output current (IOUT) requirements.
2. Optimize the design for key parameters such as efficiency, footprint, and cost using the optimizer dial.
3. Compare the generated design with other possible solutions from Texas Instruments.
The WEBENCH Power Designer provides a customized schematic along with a list of materials with real-time
pricing and component availability.
In most cases, these actions are available:
• Run electrical simulations to see important waveforms and circuit performance
• Run thermal simulations to understand board thermal performance
• Export customized schematic and layout into popular CAD formats
• Print PDF reports for the design, and share the design with colleagues
Get more information about WEBENCH tools at www.ti.com/WEBENCH.
12.2 Receiving Notification of Documentation Updates
To receive notification of documentation updates, navigate to the device product folder on ti.com. Click on
Subscribe to updates to register and receive a weekly digest of any product information that has changed. For
change details, review the revision history included in any revised document.
12.3 Support Resources
TI E2E™ support forums are an engineer's go-to source for fast, verified answers and design help — straight
from the experts. Search existing answers or ask your own question to get the quick design help you need.
Linked content is provided "AS IS" by the respective contributors. They do not constitute TI specifications and do
not necessarily reflect TI's views; see TI's Terms of Use.
12.4 Trademarks
PowerPAD™ is a trademark of TI.
TI E2E™ is a trademark of Texas Instruments.
SIMPLE SWITCHER® are registered trademarks of TI.
WEBENCH® is a registered trademark of Texas Instruments.
All trademarks are the property of their respective owners.
12.5 Electrostatic Discharge Caution
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled
with appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may
be more susceptible to damage because very small parametric changes could cause the device not to meet its published
specifications.
12.6 Glossary
TI Glossary
This glossary lists and explains terms, acronyms, and definitions.
13 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
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25-Jan-2021
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
LMR23630ADDA
ACTIVE SO PowerPAD
DDA
8
75
RoHS & Green
NIPDAUAG
Level-2-260C-1 YEAR
-40 to 125
F30A
LMR23630ADDAR
ACTIVE SO PowerPAD
DDA
8
2500
RoHS & Green
NIPDAUAG
Level-2-260C-1 YEAR
-40 to 125
F30A
LMR23630AFDDA
ACTIVE SO PowerPAD
DDA
8
75
RoHS & Green
NIPDAUAG
Level-2-260C-1 YEAR
-40 to 125
F30AF
LMR23630AFDDAR
ACTIVE SO PowerPAD
DDA
8
2500
RoHS & Green
NIPDAUAG
Level-2-260C-1 YEAR
-40 to 125
F30AF
LMR23630APDRRR
ACTIVE
WSON
DRR
12
3000
RoHS & Green
SN
Level-2-260C-1 YEAR
-40 to 125
3630P
LMR23630APDRRT
ACTIVE
WSON
DRR
12
250
RoHS & Green
SN
Level-2-260C-1 YEAR
-40 to 125
3630P
LMR23630DRRR
ACTIVE
WSON
DRR
12
3000
RoHS & Green
SN
Level-2-260C-1 YEAR
-40 to 125
23630
LMR23630DRRT
ACTIVE
WSON
DRR
12
250
RoHS & Green
SN
Level-2-260C-1 YEAR
-40 to 125
23630
LMR23630FDRRR
ACTIVE
WSON
DRR
12
3000
RoHS & Green
SN
Level-2-260C-1 YEAR
-40 to 125
3630F
LMR23630FDRRT
ACTIVE
WSON
DRR
12
250
RoHS & Green
SN
Level-2-260C-1 YEAR
-40 to 125
3630F
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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