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OPA1641-Q1, OPA1642-Q1
SBOS791A – JUNE 2017 – REVISED NOVEMBER 2017
OPA164x-Q1 SoundPlus™ JFET-Input, Automotive-Grade, Audio Operational Amplifiers
1 Features
3 Description
•
•
The OPA1641-Q1 (single) and OPA1642-Q1 (dual)
series are JFET-input, ultra-low distortion, low-noise
operational amplifiers (op amps) fully specified for
audio applications.
1
•
•
•
•
•
•
•
•
•
Qualified for Automotive Applications
AEC-Q100 Qualified With the Following Results:
– Device Temperature Grade 1: –40°C to
+125°C Ambient Operating Temperature
– Device HBM ESD Classification Level 2
– Device CDM ESD Classification Level C6
Low Noise: 5.1 nV/√Hz at 1 kHz
Ultra-low Distortion: 0.00005% at 1 kHz
High Slew Rate: 20 V/μs
Unity Gain Stable
No Phase Reversal
Low Quiescent Current:
1.8 mA per Channel
Rail-to-Rail Output
Wide Supply Range: ±2.25 V to ±18 V
Single, Dual, and Quad Versions Available
The rail-to-rail output swing allows increased
headroom, making these devices ideal for use in any
audio circuit. Features include 5.1-nV/√Hz noise, low
total harmonic distortion + noise (THD+N)
(0.00005%), a low input bias current of 2 pA, and low
quiescent current of 1.8 mA per channel.
These devices operate over a very wide supply
voltage range of ±2.25 V to ±18 V. The OPA164x-Q1
series of operational amplifiers are unity-gain stable
and provide excellent dynamic behavior over a wide
range of load conditions.
The dual version features completely independent
circuitry for lowest crosstalk and freedom from
interactions between channels, even when overdriven
or overloaded.
2 Applications
•
•
•
•
•
Device Information(1)
PART NUMBER
Automotive
HEV and EV Power Train
Advanced Driver Assist (ADAS)
Infotainment Systems
In-Cabin Microphones
PACKAGE
OPA1641-Q1
OPA1642-Q1
BODY SIZE (NOM)
SOIC (8)
4.90 mm × 3.90 mm
VSSOP (8)
3.00 mm × 3.00 mm
SOIC (8)
4.90 mm × 3.90 mm
VSSOP (8)
3.00 mm × 3.00 mm
(1) For all available packages, see the orderable addendum at
the end of the data sheet.
space
Simplified Internal Schematic
Extremely Stable Input Capacitance
7.5
Pre-Output Driver
IN–
OUT
IN+
V–
Copyright © 2017 Texas Instruments Inc.
Common-Mode Capacitance (pF)
V+
7
Traditional JFET-Input Amplifier
6.5
OPA164x Family
6
5.5
5
4.5
4
–10
–8
–6
–4
–2
0
2
4
6
8
10
Common-Mode Voltage (V)
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
OPA1641-Q1, OPA1642-Q1
SBOS791A – JUNE 2017 – REVISED NOVEMBER 2017
www.ti.com
Table of Contents
1
2
3
4
5
6
7
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Pin Configuration and Functions .........................
Specifications.........................................................
1
1
1
2
3
4
6.1
6.2
6.3
6.4
6.5
6.6
4
4
4
4
5
7
Absolute Maximum Ratings .....................................
ESD Ratings..............................................................
Recommended Operating Conditions.......................
Thermal Information ..................................................
Electrical Characteristics...........................................
Typical Characteristics ..............................................
Detailed Description ............................................ 12
7.1
7.2
7.3
7.4
Overview .................................................................
Functional Block Diagram .......................................
Feature Description.................................................
Device Functional Modes........................................
12
12
13
15
8
Application and Implementation ........................ 16
8.1 Application Information............................................ 16
8.2 Typical Application ................................................. 21
9 Power Supply Recommendations...................... 23
10 Layout................................................................... 24
10.1 Layout Guidelines ................................................. 24
10.2 Layout Example .................................................... 24
11 Device and Documentation Support ................. 25
11.1
11.2
11.3
11.4
11.5
11.6
11.7
11.8
Device Support ....................................................
Documentation Support .......................................
Related Links ........................................................
Receiving Notification of Documentation Updates
Community Resources..........................................
Trademarks ...........................................................
Electrostatic Discharge Caution ............................
Glossary ................................................................
25
25
26
26
26
26
26
26
12 Mechanical, Packaging, and Orderable
Information ........................................................... 26
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Original (June 2017) to Revision A
•
2
Page
Added separate AOL values for OPA1642-Q1 ........................................................................................................................ 6
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SBOS791A – JUNE 2017 – REVISED NOVEMBER 2017
5 Pin Configuration and Functions
OPA1641-Q1: D and DGK Packages
8-Pin SOIC and VSSOP
Top View
NC
(1)
(1)
1
8
NC
(1)
–IN
2
–
7
V+
+IN
3
+
6
OUT
V–
4
5
NC
(1)
NC - no internal connection
Pin Functions: OPA1641-Q1
PIN
NAME
NO.
I/O
DESCRIPTION
–IN
2
I
Inverting input
+IN
3
I
Noninverting input
NC
1, 5, 8
—
No connection
OUT
6
O
Output
V–
4
—
Negative (lowest) power supply
V+
7
—
Positive (highest) power supply
OPA1642-Q1: D and DGK Packages
8-Pin SOIC and VSSOP
Top View
+IN A
3
V–
4
B
–
2
+
A
–
–IN A
1
+
OUT A
8
V+
7
OUT B
6
–IN B
5
+IN B
Pin Functions: OPA1642-Q1
PIN
I/O
DESCRIPTION
NAME
NO.
–IN A
2
I
Inverting input, channel A
–IN B
6
I
Inverting input, channel B
+IN A
3
I
Noninverting input, channel A
+IN B
5
I
Noninverting input, channel B
OUT A
1
O
Output, channel A
OUT B
7
O
Output, channel B
V–
4
—
Negative (lowest) power supply
V+
8
—
Positive (highest) power supply
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6 Specifications
6.1 Absolute Maximum Ratings
over operating free-air temperature range (unless otherwise noted) (1)
MIN
VS
MAX
UNIT
40
V
Supply voltage
(2)
VIN
Input voltage
IIN
Input current (2)
(V–) – 0.5
(V+) + 0.5
V
±10
mA
±VS
V
VIN(DIFF) Differential input voltage
(3)
IO
Output short-circuit
TA
Operating temperature
–55
125
°C
TJ
Junction temperature
–65
150
°C
Tstg
Storage temperature
–65
150
°C
(1)
(2)
(3)
Continuous
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended
Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
Input pins are diode-clamped to the power-supply rails. Input signals that can swing more than 0.5 V beyond the supply rails must be
current-limited to 10 mA or less. The input voltage and output negative-voltage ratings can be exceeded if the input and output current
ratings are followed.
Short-circuit to VS / 2 (ground in symmetrical dual-supply setups), one amplifier per package.
6.2 ESD Ratings
VALUE
V(ESD)
(1)
Electrostatic discharge
Human-body model (HBM), per AEC Q100-002
(1)
±3000
Charged-device model (CDM), per AEC Q100-011
±1000
UNIT
V
AEC Q100-002 indicates that HBM stressing shall be in accordance with the ANSI/ESDA/JEDEC JS-001 specification.
6.3 Recommended Operating Conditions
over operating free-air temperature range (unless otherwise noted)
MIN
Supply voltage (V+, V–)
Single supply
Dual supply
NOM
MAX
4.5
36
±2.25
±18
–40
125
Specified temperature
UNIT
V
°C
6.4 Thermal Information
OPA1641-Q1, OPA1642-Q1
THERMAL METRIC (1)
D (SOIC)
DGK (VSSOP)
8 PINS
8 PINS
UNIT
180
°C/W
RθJA
Junction-to-ambient thermal resistance
160
RθJC(top)
Junction-to-case (top) thermal resistance
75
55
°C/W
RθJB
Junction-to-board thermal resistance
60
130
°C/W
ψJT
Junction-to-top characterization parameter
9
n/a
°C/W
ψJB
Junction-to-board characterization parameter
50
120
°C/W
RθJC(bot)
Junction-to-case (bottom) thermal resistance
N/A
N/A
°C/W
(1)
4
For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report.
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SBOS791A – JUNE 2017 – REVISED NOVEMBER 2017
6.5 Electrical Characteristics
at TA = 25°C, VS = 4.5 V to 36 (±2.25 V to ±18 V), RL = 2 kΩ connected to midsupply, and VCM = VOUT = midsupply (unless
otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
AUDIO PERFORMANCE
THD+N
IMD
Total harmonic distortion +
noise
Intermodulation distortion
0.00005%
G = 1, f = 1 kHz, VO = 3 VRMS
–126
SMPTE/DIN two-tone, 4:1
(60 Hz and 7 kHz), G = 1,
VO = 3 VRMS
0.00004%
DIM 30 (3-kHz square wave and
15-kHz sine wave), G = 1,
VO = 3 VRMS
0.00008%
CCIF twin-tone
(19 kHz and 20 kHz), G = 1,
VO = 3 VRMS
0.00007%
dB
–128
dB
–122
dB
–123
dB
FREQUENCY RESPONSE
GBW
Gain-bandwidth product
G=1
11
SR
Slew rate
G=1
20
MHz
V/μs
Full-power bandwidth (1)
VO = 1 VP
3.2
MHz
Overload recovery time (2)
G = –10
600
ns
Channel separation (dual
and quad)
f = 1 kHz
–126
dB
Input voltage noise
f = 20 Hz to 20 kHz
NOISE
4.3
f = 10 Hz
en
Input voltage noise density
In
Input current noise density
μVPP
8
f = 100 Hz
5.8
f = 1 kHz
5.1
f = 1 kHz
0.8
nV/√Hz
fA/√Hz
OFFSET VOLTAGE
VOS
Input offset voltage
VS = ±18 V
PSRR
VOS vs power supply
VS = ±2.25 V to ±18 V
1
3.5
0.14
2
mV
μV/V
INPUT BIAS CURRENT
IB
Input bias current
VCM = 0 V
±2
±20
pA
IOS
Input offset current
VCM = 0 V
±2
±20
pA
(V+) – 3.5
V
INPUT VOLTAGE RANGE
VCM
Common-mode voltage
range
CMRR
Common-mode rejection
ratio
(V–) – 0.1
VCM = (V–) – 0.1 V to (V+) – 3.5 V,
VS = ±18 V
120
126
dB
INPUT IMPEDANCE
Differential
Common-mode
(1)
(2)
VCM = (V–) – 0.1 V to (V+) – 3.5 V
1013 || 8
Ω || pF
1013 || 6
Ω || pF
Full power bandwidth = SR / (2π × VP), where SR = slew rate.
See Figure 19 and Figure 20.
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Electrical Characteristics (continued)
at TA = 25°C, VS = 4.5 V to 36 (±2.25 V to ±18 V), RL = 2 kΩ connected to midsupply, and VCM = VOUT = midsupply (unless
otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
OPA1641-Q1:
(V–) + 0.2 V ≤ VO ≤ (V+) – 0.2 V,
RL = 10 kΩ
120
134
OPA1641-Q1:
(V–) + 0.35 V ≤ VO ≤ (V+) – 0.35 V,
RL = 2 kΩ
114
126
OPA1642-Q1:
(V–) + 0.2 V ≤ VO ≤ (V+) – 0.2 V,
RL = 10 kΩ
114
134
OPA1642-Q1:
(V–) + 0.35 V ≤ VO ≤ (V+) – 0.35 V,
RL = 2 kΩ
111
126
MAX
UNIT
OPEN-LOOP GAIN
AOL
Open-loop voltage gain
dB
OUTPUT
VO
Voltage output swing from
rail
IOUT
Output current
ZO
Open-loop output impedance
ISC
Short-circuit current
CLOAD
Capacitive load drive
RL = 10 kΩ,
AOL ≥ 120 dB (OPA1641-Q1)
AOL ≥ 114 dB (OPA1642-Q1)
(V–) + 0.2
RL = 2 kΩ,
AOL ≥ 114 dB (OPA1641-Q1)
AOL ≥ 111 dB (OPA1642-Q1)
(V–) + 0.35
(V+) – 0.2
V
(V+) – 0.35
See Typical Characteristics
See Typical Characteristics
Source
36
Sink
mA
–30
See Typical Characteristics
POWER SUPPLY
VS
Specified voltage
IQ
Quiescent current (per
amplifier)
±2.25
IOUT = 0 A
1.8
±18
V
2.3
mA
TEMPERATURE RANGE
6
Specified range
–40
125
°C
Operating range
–55
125
°C
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SBOS791A – JUNE 2017 – REVISED NOVEMBER 2017
6.6 Typical Characteristics
at TA = 25°C, RL = 2 kΩ connected to midsupply, and VCM = VOUT = midsupply (unless otherwise noted)
100 nV/div
Voltage Noise Density (nV/ÖHz)
100
10
1
0.1
1
10
100
1k
10k
100k
Time (1 s/div)
Frequency (Hz)
Figure 2. 0.1-Hz to 10-Hz Noise
Figure 1. Input Voltage Noise Density vs Frequency
35
160
25
Common-Mode Rejection Ratio (dB)
Power-Supply Rejection Ratio (dB)
Output Voltage (VPP)
Maximum Output
Voltage Range
Without Slew-Rate
Induced Distortion
VS = ±15 V
30
20
15
VS = ±5 V
10
VS = ±2.25 V
5
0
140
CMRR
120
100
–PSRR
80
+PSRR
60
40
20
0
10k
100k
1M
10M
1
10
100
1k
Frequency (Hz)
10k
100k
1M
10M
100M
Frequency (Hz)
Figure 3. Maximum Output Voltage vs Frequency
140
Figure 4. CMRR and PSRR vs Frequency
(Referred to Input)
30
180
120
Gain
20
135
90
60
40
Gain (dB)
80
G = +10
Phase (°)
Gain (dB)
100
10
G = +1
0
Phase
45
20
–10
G = –1
0
-20
50 100
1k
10k
100k
1M
10M
0
100M
–20
100k
1M
10M
100M
Frequency (Hz)
Frequency (Hz)
Figure 5. Gain and Phase vs Frequency
Figure 6. Closed-Loop Gain vs Frequency
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Typical Characteristics (continued)
G = –1
RL = 600 Ω
0.0001%
–120
G = +1
RL = 2 kΩ
10
100
1k
10k
0.001%
G = +1
RL = 600 Ω
–120
0.0001%
G = +1
RL = 2 kΩ
10
20k
100
1k
VOUT = 3 VRMS
BW = 80 kHz
–120
0.0001%
G = –1, RL = 2 kΩ
G = +1, RL = 2 kΩ
–140
1
10
0.01%
Intermodulation Distortion
–100
0.001%
Total Harmonic Distortion + Noise (dB)
Total Harmonic Distortion + Noise
–80
0.1
BW > 500 kHz
Figure 8. THD+N Ratio vs Frequency
Figure 7. THD+N Ratio vs Frequency
0.01%
0.00001%
-80
SMPTE/DIN
Two-Tone
4:1 (60 Hz and 7 kHz)
0.001%
-100
DIM30
(3 kHz Square Wave
and 15 kHz Sine Wave)
-120
0.0001%
CCIF Twin-Tone
(19 kHz and 20 kHz)
-140
0.00001%
20
0.1
1
Output Amplitude (VRMS)
BW = 80 kHz
–140
100k
10k
Frequency (Hz)
Frequency (Hz)
VOUT = 3 VRMS
–100
G = –1
RL = 2 kΩ
0.00001%
–140
0.00001%
G = –1
RL = 600 Ω
Total Harmonic Distortion + Noise (dB)
G = +1
RL = 600 Ω
–80
0.01%
Intermodulation Distortion (dB)
Total Harmonic Distortion + Noise
G = –1
RL = 2 kΩ
Total Harmonic Distortion + Noise
–100
0.001%
Total Harmonic Distortion + Noise (dB)
at TA = 25°C, RL = 2 kΩ connected to midsupply, and VCM = VOUT = midsupply (unless otherwise noted)
10
20
Output Amplitude (VRMS)
1 kHz Signal
RSOURCE = 0 Ω
G=1
Figure 9. THD+N Ratio vs Output Amplitude
Figure 10. Intermodulation Distortion
vs Output Amplitude
–90
Output
–100
RL = 600 Ω
5 V/div
Channel Separation (dB)
–80
–110
–120
+18 V
RL = 2 kΩ
OPA1641-Q1
Output
–130
-18 V
37 VPP
Sine Wave
(±18.5 V)
RL = 5 kΩ
–140
10
100
1k
10k
100k
Frequency (Hz)
VS = ±15 V
VOUT = 3 VRMS
Time (0.4 ms/div)
G=1
Figure 11. Channel Separation vs Frequency
8
Figure 12. No Phase Reversal
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SBOS791A – JUNE 2017 – REVISED NOVEMBER 2017
Typical Characteristics (continued)
20 mV/div
20 mV/div
at TA = 25°C, RL = 2 kΩ connected to midsupply, and VCM = VOUT = midsupply (unless otherwise noted)
+15 V
OPA1641-Q1
-15 V
RL
RI
= 2 kW
RF
= 2 kW
+15 V
OPA1641-Q1
CL
CL
-15 V
Time (100 ns/div)
Time (100 ns/div)
G=1
G = –1
CL = 100 pF
CL = 100 pF
Figure 14. Small-Signal Step Response (100 mV)
2 V/div
2 V/div
Figure 13. Small-Signal Step Response (100 mV)
Time (400 ns/div)
Time (400 ns/div)
G=1
CL = 100 pF
G = –1
Figure 15. Large-Signal Step Response
CL = 100 pF
Figure 16. Large-Signal Step Response
VOUT
5 V/div
5 V/div
VIN
20 kW
20 kW
2 kW
VIN
2 kW
OPA1641-Q1
VOUT
OPA1641-Q1
VIN
VOUT
VIN
VOUT
Time (0.4 ms/div)
Time (0.4 ms/div)
G = –10
G = –10
Figure 17. Positive Overload Recovery
Figure 18. Negative Overload Recovery
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Typical Characteristics (continued)
at TA = 25°C, RL = 2 kΩ connected to midsupply, and VCM = VOUT = midsupply (unless otherwise noted)
40%
45%
ROUT = 0 W
35%
RI = 2 kW
+15 V
40%
ROUT
+15 V
OPA1641-Q1
RL
-15 V
ROUT
35%
CL
OPA1641-Q1
CL
30%
25%
Overshoot
Overshoot
30%
ROUT = 24 W
20%
ROUT = 0 W
RF = 2 kW
15%
ROUT = 24 W
-15 V
25%
20%
ROUT = 51 W
15%
ROUT = 51 W
10%
10%
5%
5%
0
0
0
100 200 300 400 500 600 700 800 900 1000
0
100 200 300 400 500 600 700 800 900 1000
Capacitive Load (pF)
Capacitive Load (pF)
G=1
G = –1
Figure 19. Small-Signal Overshoot vs Capacitive Load
(100-mV Output Step)
Figure 20. Small-Signal Overshoot vs Capacitive Load
(100-mV Output Step)
0
80
70
10 kΩ
–0.2
60
IB and IOS (pA)
AOL (μV/V)
–0.4
2 kΩ
–0.6
–0.8
+IB
50
40
30
20
–IB
10
0
–1
–10
–1.2
–IOS
–20
-40
-15
35
10
60
85
–40
–15
10
35
60
85
Temperature (°C)
Temperature (°C)
Figure 21. Open-Loop Gain vs Temperature
Figure 22. IB and IOS vs Temperature
2.5
10
8
2
+IB
4
–IB
2
0
IQ (mA)
IB and IOS (pA)
6
IOS
–2
1.5
1
–4
0.5
–6
–8
Common-Mode Range
–10
–18
–12
–6
0
6
12
18
0
–40 –25 –10
5
20
35
50
65
80
95
110 125
Temperature (°C)
Common-Mode Voltage (V)
VS = ±18 V
Figure 23. IB and IOS vs Common-Mode Voltage
10
Figure 24. Quiescent Current vs Temperature
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SBOS791A – JUNE 2017 – REVISED NOVEMBER 2017
Typical Characteristics (continued)
at TA = 25°C, RL = 2 kΩ connected to midsupply, and VCM = VOUT = midsupply (unless otherwise noted)
60
2
1.75
50
ISC-SOURCE
1.5
40
ISC (mA)
IQ (mA)
1.25
1
0.75
30
ISC-SINK
20
0.5
10
0.25
Specified Supply-Voltage Range
0
0
0
4
8
12
16
20
24
28
32
36
–50
–25
0
25
50
75
100
125
Temperature (°C)
Supply Voltage (V)
VOUT = Midsupply (includes self-heating)
Figure 26. Short-Circuit Current vs Temperature
Figure 25. Quiescent Current vs Supply Voltage
18
1k
17.5
100
16
ZO (W)
Output Voltage (V)
17
16.5
+85°C
-40°C
+125°C
+25°C
-16
10
-16.5
-17
-17.5
1
-18
0
10
20
30
40
50
10
100
10k
100k
1M
10M
100M
Frequency (Hz)
Output Current (mA)
Figure 27. Output Voltage vs Output Current
1k
Figure 28. Open-Loop Output Impedance vs Frequency
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7 Detailed Description
7.1 Overview
The OPA164x-Q1 family of operational amplifiers combine an ultra-low noise JFET input stage with a rail-to-rail
output stage to provide high overall performance in audio applications. The internal topology is selected
specifically to deliver extremely low distortion, consume limited power, and accommodate small packages. These
amplifiers are well-suited for analog signal processing applications such as active filter circuits, pre-amplifiers,
and tone controls. The unique input stage design and semiconductor processes used in this device deliver
extremely high performance even in applications with high source impedance and wide common-mode voltage
swings.
7.2 Functional Block Diagram
V+
Pre-Output Driver
IN–
OUT
IN+
V–
12
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7.3 Feature Description
7.3.1 Phase Reversal Protection
The OPA164x-Q1 family has internal phase-reversal protection. Many op amps exhibit phase reversal when the
input is driven beyond the linear common-mode range. This condition is most often encountered in noninverting
circuits when the input is driven beyond the specified common-mode voltage range, causing the output to
reverse into the opposite rail. The input of the OPA164x-Q1 prevents phase reversal with excessive commonmode voltage. Instead, the appropriate rail limits the output voltage. This performance is shown in Figure 29.
5 V/div
Output
+18 V
OPA1641-Q1
Output
-18 V
37 VPP
Sine Wave
(±18.5 V)
Time (0.4 ms/div)
Figure 29. Output Waveform Devoid of Phase Reversal During an Input Overdrive Condition
7.3.2 Output Current Limit
The output current of the OPA164x-Q1 series is limited by internal circuitry to 36 mA and –30 mA (sourcing and
sinking), to protect the device if the output is accidentally shorted. This short-circuit current depends on
temperature; see Figure 26.
Although uncommon for most modern audio applications to require 600-Ω load drive capability, many audio
operational amplifier applications continue to specify the total harmonic distortion (THD+N) at 600-Ω load for
comparative purposes. Figure 7 and Figure 8 provide typical THD+N measurement curves for the OPA164x-Q1
series, where the output drives a 3-VRMS signal into a 600-Ω load. However, correct device operation cannot be
ensured when driving 600-Ω loads at full supply. Depending on supply voltage and temperature, this operating
condition can possibly trigger the output current limit circuitry of the device.
7.3.3 EMI Rejection Ratio (EMIRR)
The electromagnetic interference (EMI) rejection ratio, or EMIRR, describes the EMI immunity of operational
amplifiers. An adverse effect that is common to many operational amplifiers is a change in the offset voltage as a
result of RF signal rectification. An operational amplifier that is more efficient at rejecting this change in offset as
a result of EMI has a higher EMIRR and is quantified by a decibel value. Measuring EMIRR can be performed in
many ways, but this document provides the EMIRR IN+, which specifically describes the EMIRR performance
when the RF signal is applied to the noninverting input pin of the operational amplifier. In general, only the
noninverting input is tested for EMIRR for the following three reasons:
• Operational amplifier input pins are known to be the most sensitive to EMI, and typically rectify RF signals
better than the supply or output pins.
• The noninverting and inverting operational amplifier inputs have symmetrical physical layouts and exhibit
nearly matching EMIRR performance.
• EMIRR is easier to measure on noninverting pins than on other pins because the noninverting input pin can
be isolated on a printed-circuit-board (PCB). This isolation allows the RF signal to be applied directly to the
noninverting input pin with no complex interactions from other components or connecting PCB traces.
A more formal discussion of the EMIRR IN+ definition and test method is provided in application report EMI
Rejection Ratio of Operational Amplifiers, available for download at www.ti.com.
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Feature Description (continued)
The EMIRR IN+ of the OPA164x-Q1 is plotted versus frequency in Figure 30. If available, any dual and quad
operational amplifier device versions have nearly identical EMIRR IN+ performance. The OPA164x-Q1 unity-gain
bandwidth is 11 MHz. EMIRR performance below this frequency denotes interfering signals that fall within the
operational amplifier bandwidth.
140
120
EMIRR (dB)
100
80
60
40
20
0
10M
100M
1G
10G
Frequency (Hz)
C003
Figure 30. OPA164x-Q1 EMIRR vs Frequency
Table 1 lists the EMIRR IN+ values for the OPA164x-Q1 at particular frequencies commonly encountered in realworld applications. Applications listed in Table 1 can be centered on or operated near the particular frequency
shown. This information can be of special interest to designers working with these types of applications, or
working in other fields likely to encounter RF interference from broad sources, such as the industrial, scientific,
and medical (ISM) radio band.
Table 1. OPA164x-Q1 EMIRR IN+ for Frequencies of Interest
14
FREQUENCY
APPLICATION AND ALLOCATION
EMIRR IN+
400 MHz
Mobile radio, mobile satellite, space operation, weather, radar, UHF
53.1 dB
900 MHz
GSM, radio communication and navigation, GPS (to 1.6 GHz), ISM,
aeronautical mobile, UHF
72.2 dB
1.8 GHz
GSM, mobile personal comm. broadband, satellite, L-band
80.7 dB
2.4 GHz
802.11b/g/n, Bluetooth™, mobile personal comm., ISM, amateur radio and satellite, S-band
86.8 dB
3.6 GHz
Radiolocation, aero comm./nav., satellite, mobile, S-band
91.7 dB
5 GHz
802.11a/n, aero communication and navigation, mobile communication,
space and satellite operation, C-band
96.6 dB
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7.3.3.1 EMIRR IN+ Test Configuration
Figure 31 shows the circuit configuration for testing the EMIRR IN+. An RF source is connected to the
operational amplifier noninverting input pin using a transmission line. The operational amplifier is configured in a
unity-gain buffer topology with the output connected to a low-pass filter (LPF) and a digital multimeter (DMM). A
large impedance mismatch at the operational amplifier input causes a voltage reflection; however, this effect is
characterized and accounted for when determining the EMIRR IN+. The resulting dc offset voltage is sampled
and measured by the multimeter. The LPF isolates the multimeter from residual RF signals that can interfere with
multimeter accuracy. See EMI Rejection Ratio of Operational Amplifiers for more details.
Ambient temperature: 25Û&
+VS
±
50
Low-Pass Filter
+
RF source
DC Bias: 0 V
Modulation: None (CW)
Frequency Sweep: 201 pt. Log
-VS
Not shown: 0.1 µF and 10 µF
supply decoupling
Sample /
Averaging
Digital Multimeter
Figure 31. EMIRR IN+ Test Configuration Schematic
7.4 Device Functional Modes
7.4.1 Operating Voltage
The OPA164x-Q1 series of operational amplifiers can be used with single or dual supplies from an operating
range of VS = 4.5 V (±2.25 V) and up to VS = 36 V (±18 V). These devices do not require symmetrical supplies;
only a minimum supply voltage of 4.5 V (±2.25 V) is required. For VS less than ±3.5 V, the common-mode input
range does not include midsupply. Supply voltages higher than 40 V can permanently damage the device; see
Absolute Maximum Ratings for more information. Key parameters are specified over the operating temperature
range, TA = –40°C to +85°C. Key parameters that vary over the supply voltage or temperature range are shown
in Typical Characteristics.
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8 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
8.1 Application Information
The OPA164x-Q1 amplifiers are unity-gain stable, audio operational amplifiers with very low noise, input bias
current, and input offset voltage. Applications with noisy or high-impedance power supplies require decoupling
capacitors placed close to the device pins. In most cases, 0.1-μF capacitors are adequate. Figure 32 shows a
simplified schematic of the OPA1641-Q1.
V+
Pre-Output Driver
IN–
OUT
IN+
V–
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Figure 32. Simplified Internal Schematic
16
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Application Information (continued)
8.1.1 Total Harmonic Distortion Measurements
The OPA164x-Q1 series operational amplifiers have excellent distortion characteristics. THD + noise is below
0.00005% (G = 1, VO = 3 VRMS, BW = 80 kHz) throughout the audio frequency range, 20 Hz to 20 kHz, with a
2-kΩ load (see Figure 7).
The distortion produced by the OPA164x-Q1 series op amps is below the measurement limit of many
commercially available distortion analyzers. However, a special test circuit (such as shown in Figure 33) can be
used to extend the measurement capabilities.
Operational amplifier distortion can be considered an internal error source that can be referred to the input.
Figure 33 shows a circuit that causes the operational amplifier distortion to be 101 times (or approximately
40 dB) greater than that normally produced by the op amp. The addition of R3 to the otherwise standard
noninverting amplifier configuration alters the feedback factor or noise gain of the circuit. The closed-loop gain is
unchanged, but the feedback available for error correction is reduced by a factor of 101, extending the resolution
by 101. The input signal and load applied to the op amp are the same as with conventional feedback without R3.
Keep the value of R3 small to minimize any effect on distortion measurements.
The validity of this technique can be verified by duplicating measurements at high gain or high frequency where
the distortion is within the measurement capability of the test equipment. Measurements for this document were
made with an audio precision system two distortion and noise analyzer that greatly simplifies repetitive
measurements. However, the measurement technique can be performed with manual distortion measurement
instruments.
space
R1
R2
SIGNAL DISTORTION
GAIN
GAIN
VO = 3 VRMS
R3
Signal Gain = 1+
OPA1641-Q1
R2
R1
Distortion Gain = 1+
R2
R1 II R3
Generator
Output
R1
R2
R3
1
101
¥
1 kW
10 W
11
101
100 W
1 kW
11 W
Analyzer
Input
Audio Precision
System Two(1)
with PC Controller
Load
Copyright © 2016, Texas Instruments Incorporated
(1)
For measurement bandwidth, see Figure 7 through Figure 10.
Figure 33. Distortion Test Circuit
8.1.2 Source Impedance and Distortion
In traditional JFET-input op amps, the impedance applied to the positive and negative inputs in noninverting
applications must be matched for lowest distortion. Legacy methods for fabricating the JFETs in the FET input
stage exhibit a varying input capacitance with applied common-mode input voltage. In inverting configurations,
the input does not vary with input voltage because the inverting input is held at virtual ground. However, in
noninverting applications, the inputs do vary, and the gate-to-source voltage is not constant. This effect produces
increased distortion resulting from the varying capacitance for unmatched source impedances. However, the
OPA164x-Q1 family of amplifiers is designed to maintain a constant input capacitance with varying commonmode voltage to prevent this mechanism of distortion. The variation of input capacitance with common-mode
voltage for a traditional amplifier is compared to the OPA164x-Q1 family in Figure 34.
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Common-Mode Capacitance (pF)
7.5
7
Traditional JFET-Input Amplifier
6.5
OPA164x Family
6
5.5
5
4.5
4
–10
–8
–6
–4
–2
0
2
4
6
8
10
Common-Mode Voltage (V)
Figure 34. Input Capacitance of the OPA164x-Q1 Family of Amplifiers Compared to Traditional JFETinput Amplifiers
Total Harmonic Distortion + Noise
1%
–40
10 kΩ
+
–50
–
5 VRMS
0.1%
–60
–70
Traditional JFET-Input Amplifier
0.01%
–80
–90
OPA164x Amplifier
0.001%
–100
–110
0.0001%
–120
10
100
1k
Total Harmonic Distortion + Noise (dB)
By stabilizing the input capacitance, the distortion performance of the amplifier is greatly improved for
noninverting configurations with high source impedances. The measured performance of an OPA164x-Q1
amplifier is compared to a traditional JFET-input amplifier in Figure 35. The unity-gain configuration, high source
impedance, and large-signal amplitude produce additional distortion in the traditional amplifier.
10k 20k
Frequency (Hz)
Figure 35. Measured THD+N of the OPA164x-Q1 Family of Amplifiers Compared to Traditional JFETinput Amplifiers
8.1.3 Capacitive Load and Stability
The dynamic characteristics of the OPA164x-Q1 are optimized for commonly encountered gains, loads, and
operating conditions. The combination of low closed-loop gain and high capacitive loads decreases the phase
margin of the amplifier and can lead to gain peaking or oscillations. As a result, heavier capacitive loads must be
isolated from the output. The simplest way to achieve this isolation is to add a small resistor (ROUT equal to 50 Ω,
for example) in series with the output.
Figure 19 and Figure 20 illustrate graphs of Small-Signal Overshoot vs Capacitive Load for several values of
ROUT. For details of analysis techniques and application circuits, see see Feedback Plots Define Op Amp AC
Performance available for download at www.ti.com
18
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8.1.4 Power Dissipation and Thermal Protection
The OPA164x-Q1 op amps are capable of driving 2-kΩ loads with power-supply voltages of up to ±18 V over the
specified temperature range. In a single-supply configuration, where the load is connected to the negative supply
voltage, the minimum load resistance is 2.8 kΩ at a supply voltage of 36 V. For lower supply voltages (either
single-supply or symmetrical supplies), a lower load resistance can be used, as long as the output current does
not exceed 13 mA; otherwise, the device short-circuit current-protection circuit can activate.
Internal power dissipation increases when operating at high supply voltages. Copper leadframe construction
used in the OPA164x-Q1 series of devices improves heat dissipation compared to conventional materials. PCB
layout can help reduce a possible increase in junction temperature. Wide copper traces help dissipate the heat
by functioning as an additional heat sink. Temperature rise can be further minimized by soldering the devices
directly to the PCB rather than using a socket.
Although the output current is limited by internal protection circuitry, accidental shorting one or more output
channels of a device can result in excessive heating. For instance, when an output is shorted to midsupply, the
typical short-circuit current of 36 mA leads to an internal power dissipation of over 600 mW at a supply of ±18 V.
In case of a dual OPA1642-Q1 in an VSSOP-8 package (thermal resistance RθJA = 180°C/W), such a power
dissipation results in the die temperature to be 220°C above ambient temperature, when both channels are
shorted. This temperature increase destroys the device.
To prevent such excessive heating that can destroy the device, the OPA164x-Q1 series has an internal thermal
shutdown circuit that shuts down the device if the die temperature exceeds approximately 180°C. When this
thermal shutdown circuit activates, a built-in hysteresis of 15°C ensures that the die temperature must drop to
approximately 165°C before the device switches on again.
8.1.5 Electrical Overstress
Designers often ask questions about the capability of an operational amplifier to withstand electrical overstress.
These questions tend to focus on the device inputs, but can involve the supply voltage pins or even the output
pin. Each of these different pin functions have electrical stress limits determined by the voltage breakdown
characteristics of the particular semiconductor fabrication process and specific circuits connected to the pin.
Additionally, internal electrostatic discharge (ESD) protection is built into these circuits to protect them from
accidental ESD events both before and during product assembly.
Having a good understanding of this basic ESD circuitry and the relevance to an electrical overstress event is
helpful. Figure 36 illustrates the ESD circuits contained in the OPA164x-Q1 series (indicated by the dashed line
area). The ESD protection circuitry involves several current-steering diodes connected from the input and output
pins and routed back to the internal power-supply lines where an internal absorption device is connected. This
protection circuitry is intended to remain inactive during normal circuit operation.
An ESD event produces a short-duration, high-voltage pulse that is transformed into a short-duration, highcurrent pulse when discharging through a semiconductor device. The ESD protection circuits are designed to
provide a current path around the operational amplifier core to prevent damage. The energy absorbed by the
protection circuitry is then dissipated as heat.
When an ESD voltage develops across two or more of the amplifier device pins, current flows through one or
more of the steering diodes. Depending on the path that the current takes, the absorption device can activate.
The absorption device has a trigger (or threshold voltage) that is above the normal operating voltage of the
OPA164x-Q1 but below the device breakdown voltage level. When this threshold is exceeded, the absorption
device quickly activates and clamps the voltage across the supply rails to a safe level.
When the operational amplifier connects into a circuit as shown in Figure 36, the ESD protection components are
intended to remain inactive and not become involved in the application circuit operation. However, circumstances
can arise where an applied voltage exceeds the operating voltage range of a given pin. If this condition occurs,
some of the internal ESD protection circuits can be biased on and conduct current. Any such current flow occurs
through steering diode paths and rarely involves the absorption device.
Figure 36 depicts a specific example where the input voltage (VIN) exceeds the positive supply voltage (+VS) by
500 mV or more. Much of what happens in the circuit depends on the supply characteristics. If +VS can sink the
current, one of the upper input steering diodes conducts and directs current to +VS. Excessively high current
levels can flow with increasingly higher VIN. As a result, the data sheet specifications recommend that
applications limit the input current to 10 mA.
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If the supply is not capable of sinking the current, VIN can begin sourcing current to the operational amplifier, and
then take over as the source of positive supply voltage. The danger in this case is that the voltage can rise to
levels that exceed the operational amplifier absolute maximum ratings.
Another common question involves what happens to the amplifier if an input signal is applied to the input when
the power supplies +VS and –VS are at 0 V. The amplifier behavior depends on the supply characteristic when at
0 V, or at a level below the input signal amplitude. If the supplies appear as high impedance, then the operational
amplifier supply current can be supplied by the input source through the current steering diodes. This state is not
a normal bias condition; the amplifier most likely does not operate normally. If the supplies are low impedance,
then the current through the steering diodes can become quite high. The current level depends on the ability of
the input source to deliver current and any resistance in the input path.
If there is an uncertainty about the ability of the supply to absorb this current, external Zener diodes can be
added to the supply pins, as shown in Figure 36. The Zener voltage must be selected so the diode does not turn
on during normal operation. However, the Zener voltage must be low enough so that the Zener diode conducts if
the supply pin begins to rise above the safe operating supply voltage level.
(2)
TVS
RF
+VS
+V
OPA1641-Q1
RI
ESD CurrentSteering Diodes
-In
RS
(3)
+In
Op-Amp
Core
Edge-Triggered ESD
Absorption Circuit
ID
VIN
Out
RL
(1)
-V
-VS
(2)
TVS
Copyright © 2016, Texas Instruments Incorporated
(1)
VIN = +VS + 500 mV.
(2)
TVS: +VS(max) > VTVSBR (Min) > +VS
(3)
Suggested value is approximately 1 kΩ.
Figure 36. Equivalent Internal ESD Circuitry and the Relation to a Typical Circuit Application
20
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8.2 Typical Application
8.2.1 Single-Supply Electret Microphone Preamplifier for Speech
Electret microphones are commonly used in automotive hands-free phone systems because of their small size,
low cost, and relatively good signal-to-noise ratio (SNR). The low noise and distortion of the OPA1641-Q1 makes
the device a good choice for preamplifier circuits for electret microphones. The circuit shown in Figure 37 is a
single-supply preamplifier circuit for electret microphones with a bandwidth from 100 Hz to 3 kHz for capturing
speech.
CFB
+9 V
1 nF
RFB
RBIAS
14
k
61.9 k
CIN
+9 V
0.1 F
0.1 F
Electret
Microphone
+9 V
RDIV
100 k
±
Output
+
RDIV
100 k
VS+
OPA1641-Q1
VS±
CDIV
160 nF
Copyright © 2016, Texas Instruments Incorporated
Figure 37. Preamplifier Circuit for Electret Microphones Using a Single Power Supply Voltage
8.2.1.1 Design Requirements
• 9-V single supply
• 1-VRMS output for 100-dBSPL input
• Approximately 100-Hz to 3-kHz, –3-dB Bandwidth
• Microphone sensitivity: 8 µA / Pa
• Microphone operating voltage: 2 V to 10 V
• Microphone bias current: 500 µA
8.2.1.2 Detailed Design Procedure
In this circuit, the op amp is configured as a transimpedance amplifier which converts the signal current of the
microphone into an output voltage. The bandwidth of this circuit is limited to the vocal range as is common in
telephony systems. The gain of the circuit is determined by the feedback resistor (RFB), which must be calculated
according to the microphone sensitivity. For this design, a microphone output current of 8 µA per Pascal (Pa) of
air pressure was selected. Using this value, the output current for a sound pressure level of 100 dBSPL, or 2 Pa
air pressure, is calculated in Equation 1.
8 PA
imic
u 2 Pa = 16 PA
1 Pa
(1)
RFB is then calculated from this current to produce 1-VRMS output for a 100-dBSPL input signal in Equation 2.
VO 1 VRMS
RFB
62500 o 61.9 k:
imic 16 PA
(2)
The feedback capacitor (CFB) is calculated to limit the bandwidth of the amplifier to 3 kHz in Equation 3.
1
1
CFB
857 u 10 12 o 1 nF
2 ˜ S ˜ RFB ˜ fH 2 ˜ S ˜ (61.9 k:) ˜ (3 kHz)
(3)
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Typical Application (continued)
RBIAS is necessary to divert the microphone signal current through capacitor CIN rather than flowing from the
power supply (VCC). Larger values of RBIAS allow for a smaller capacitor to be used for CIN and reduces the
overall noise of the circuit. However, the maximum value for RBIAS is limited by the microphone bias current and
minimum operating voltage.
The value of RBIAS is calculated in Equation 4.
VCC VMIC 9 V 2 V
RBIAS
14 k:
IBIAS
500 PA
(4)
Input capacitor CIN forms a high-pass filter in combination with resistor RBIAS. The filter corner frequency
calculation is shown in Equation 5 to place the high-pass corner frequency at 100 Hz.
1
1
CIN
113.7 u 10 9 o 100 nF
2 ˜ S ˜ RBIAS ˜ fL 2 ˜ S ˜ (14 k:) ˜ (100 Hz)
(5)
The voltage divider network at the op amp noninverting input is used to bias the op amp output to the midsupply
point (VCC / 2) to maximize the output voltage range of the circuit. This result is easily achieved by selecting the
same value for both resistors in the divider. The absolute value of those resistors is limited by the acceptable
power-supply current drawn by the voltage divider. Choosing 50 µA as an acceptable limit of supply current gives
a value of 100 kΩ for the resistors in the divider, as Equation 6 shows.
V
9V
RDIV t CC t
t 90 k: o 100 k:
2 ˜ IDIV 2 ˜ 50 PA
(6)
Finally, to minimize the additional noise contribution from the voltage divider, a capacitor is placed at the op amp
noninverting input. This capacitor forms a low-pass filter with the parallel combination of the voltage divider
resistors. Selecting a filter corner frequency of 20 Hz minimizes the noise contribution of the voltage divider
inside the amplifier passband; see Equation 7.
1
1
CDIV
1.592 u 10 7 o 160 nF
§ RDIV ·
§ 100 k: ·
2˜S˜¨
¸ ˜ (20 Hz)
¸ ˜ fL 2 ˜ S ˜ ¨
2
©
¹
© 2 ¹
(7)
22
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Typical Application (continued)
8.2.1.3 Application Curve
The transfer function of the microphone preamplifier circuit is shown in Figure 38. The nominal gain of the circuit
is 95.46 dB, or 59,292.5 V per amp of input current. The –3-dB bandwidth limits of the circuit are 105.7 Hz and
2.77 kHz.
98
96
94
Gain (dB, V/A)
92
90
88
86
84
82
80
10
100
1k
10k
20k
Frequency (Hz)
Figure 38. Microphone Preamplifier Transfer Function
9 Power Supply Recommendations
The OPA164x-Q1 devices are specified for operation from 4.5 V to 36 V (±2.25 V to ±18 V); many specifications
apply from –40°C to +125°C. Parameters that can exhibit significant variance with regard to operating voltage or
temperature are presented in Typical Characteristics.
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10 Layout
10.1 Layout Guidelines
For best operational performance of the device, use good PCB layout practices, including:
• Noise can propagate into analog circuitry through the power pins of the circuit as a whole and of the op
amp itself. Bypass capacitors reduce the coupled noise by providing low-impedance power sources local
to the analog circuitry.
– Connect low-ESR, 0.1-µF ceramic bypass capacitors between each supply pin and ground, placed as
close as possible to the device. A single bypass capacitor from V+ to ground is applicable for singlesupply applications.
• Separate grounding for analog and digital portions of circuitry is one of the simplest and most-effective
methods of noise suppression. One or more layers on multilayer PCBs are usually devoted to ground
planes. A ground plane helps distribute heat and reduces EMI noise pickup. Take care to physically
separate digital and analog grounds, paying attention to the flow of the ground current.
• To reduce parasitic coupling, run the input traces as far away as possible from the supply or output
traces. If these traces cannot be kept separate, crossing the sensitive trace perpendicular is much better
as opposed to in parallel with the noisy trace.
• Place the external components as close as possible to the device. As shown in Figure 39, keeping RF
and RG close to the inverting input minimizes parasitic capacitance.
• Keep the length of input traces as short as possible. Always remember that the input traces are the most
sensitive part of the circuit.
• Consider a driven, low-impedance guard ring around the critical traces. A guard ring can significantly
reduce leakage currents from nearby traces that are at different potentials.
• TI recommends cleaning the PCB following board assembly for best performance.
• Any precision integrated circuit can experience performance shifts resulting from moisture ingress into the
plastic package. Following any aqueous PCB cleaning process, TI recommends baking the PCB
assembly to remove moisture introduced into the device packaging during the cleaning process. A low
temperature, post-cleaning bake at 85°C for 30 minutes is sufficient for most circumstances.
10.2 Layout Example
+
VIN
VOUT
RG
RF
(Schematic Representation)
Run the input traces
as far away from
the supply lines
as possible
Place components
close to device and to
each other to reduce
parasitic errors
VS+
RF
N/C
N/C
GND
±IN
V+
VIN
+IN
OUTPUT
V±
N/C
RG
Use low-ESR,
ceramic bypass
capacitor
GND
VS±
GND
Use low-ESR, ceramic
bypass capacitor
VOUT
Ground (GND) plane on another layer
Figure 39. OPA1641-Q1 Layout Example
24
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OPA1641-Q1, OPA1642-Q1
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SBOS791A – JUNE 2017 – REVISED NOVEMBER 2017
11 Device and Documentation Support
11.1 Device Support
11.1.1 Development Support
11.1.1.1 TINA-TI™ (Free Software Download)
TINA™ is a simple, powerful, and easy-to-use circuit simulation program based on a SPICE engine. TINA-TI is a
free, fully-functional version of the TINA software, preloaded with a library of macromodels in addition to a range
of both passive and active models. TINA-TI provides all the conventional dc, transient, and frequency domain
analysis of SPICE, as well as additional design capabilities.
Available as a free download from the Analog eLab Design Center, TINA-TI offers extensive post-processing
capability that allows users to format results in a variety of ways. Virtual instruments offer the ability to select
input waveforms and probe circuit nodes, voltages, and waveforms, creating a dynamic quick-start tool.
NOTE
These files require that either the TINA software (from DesignSoft™) or TINA-TI software
be installed. Download the free TINA-TI software from the TINA-TI folder.
11.1.1.2 TI Precision Designs
TI Precision Designs, available online at http://www.ti.com/ww/en/analog/precision-designs/, are analog solutions
created by TI’s precision analog applications experts and offer the theory of operation, component selection,
simulation, complete PCB schematic and layout, bill of materials, and measured performance of many useful
circuits.
11.1.1.3 WEBENCH® Filter Designer
WEBENCH® Filter Designer is a simple, powerful, and easy-to-use active filter design program. The WEBENCH
Filter Designer lets optimized filter designs to be created using a selection of TI operational amplifiers and
passive components from TI's vendor partners.
Available as a web based tool from the WEBENCH® Design Center, the WEBENCH® Filter Designer allows
complete multistage active filter solutions to be designed, optimized, and simulated within minutes.
11.2 Documentation Support
11.2.1 Related Documentation
For related documentation see the following:
• Operational amplifier gain stability, Part 3: AC gain-error analysis
• Operational amplifier gain stability, Part 2: DC gain-error analysis
• Using infinite-gain, MFB filter topology in fully differential active filters
• Op Amp Performance Analysis
• Single-Supply Operation of Operational Amplifiers
• Tuning in Amplifiers
• Shelf-Life Evaluation of Lead-Free Component Finishes
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25
OPA1641-Q1, OPA1642-Q1
SBOS791A – JUNE 2017 – REVISED NOVEMBER 2017
www.ti.com
11.3 Related Links
Table 2 lists quick access links. Categories include technical documents, support and community resources,
tools and software, and quick access to sample or buy.
Table 2. Related Links
PARTS
PRODUCT FOLDER
SAMPLE & BUY
TECHNICAL
DOCUMENTS
TOOLS &
SOFTWARE
SUPPORT &
COMMUNITY
OPA1641-Q1
Click here
Click here
Click here
Click here
Click here
OPA1642-Q1
Click here
Click here
Click here
Click here
Click here
11.4 Receiving Notification of Documentation Updates
To receive notification of documentation updates, navigate to the device product folder on ti.com. In the upper
right corner, click on Alert me to register and receive a weekly digest of any product information that has
changed. For change details, review the revision history included in any revised document.
11.5 Community Resources
The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective
contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of
Use.
TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration
among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help
solve problems with fellow engineers.
Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and
contact information for technical support.
11.6 Trademarks
E2E is a trademark of Texas Instruments.
TINA-TI is a trademark of Texas Instruments, Inc and DesignSoft, Inc.
TINA, DesignSoft are trademarks of DesignSoft, Inc.
11.7 Electrostatic Discharge Caution
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more
susceptible to damage because very small parametric changes could cause the device not to meet its published specifications.
11.8 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
12 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
26
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Product Folder Links: OPA1641-Q1 OPA1642-Q1
PACKAGE OPTION ADDENDUM
www.ti.com
10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
OPA1641AQDGKRQ1
ACTIVE
VSSOP
DGK
8
2500
RoHS & Green
NIPDAUAG
Level-2-260C-1 YEAR
-40 to 125
1641
OPA1642AQDGKRQ1
ACTIVE
VSSOP
DGK
8
2500
RoHS & Green
NIPDAUAG
Level-2-260C-1 YEAR
-40 to 125
1642
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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