THS3061DGNG4

THS3061 THS3062 www.ti.com SLOS394B – JULY 2002 – REVISED NOVEMBER 2009 LOW-DISTORTION, HIGH SLEW RATE, CURRENT-FEEDBACK AMPLIFIERS Check for Samples: THS3061 THS3062 FEATURES 1 • • • • • • 23 Unity Gain Bandwidth: 300 MHz 0.1-dB Bandwidth: 120 MHz (G = 2) High Slew Rate: 7000 V/μs HD3 at 10 MHz: –81 dBc (G = 2, RL = 150 Ω) High Output Current: ±145 mA into 50 Ω Power-Supply Voltage Range: ±5 V to ±15 V APPLICATIONS • • • • • High-Speed Signal Processing Test and Measurement Systems VDSL Line Driver High-Voltage ADC Preamplifier Video Line Driver The THS3061 and THS3062 provide well-regulated ac performance characteristics with power supplies ranging from ±5-V operation up to ±15-V supplies. Most notably, the 0.1-dB flat bandwidth is exceedingly high, reaching beyond 100 MHz, and the THS306x has less than 0.3 dB of peaking in the frequency response when configured in unity gain. The unity-gain bandwidth of 300 MHz provides excellent distortion characteristics at 10 MHz. The flexibility of the current-feedback design allows a 220-MHz, –3-dB bandwidth in a gain of 10, indicating excellent performance even at high gains. The THS306x consumes 8.3 mA per-channel quiescent current at room temperature, and has the capability of producing up to ±145 mA of output current. The THS3061 is packaged in an 8-pin SOIC and an 8-pin MSOP with PowerPAD™. The THS3062 is available in an 8-pin SOIC with PowerPAD and an 8-pin MSP with PowerPAD. DESCRIPTION The THS3061 (single) and THS3062 (dual) are high-voltage, high slew-rate current feedback amplifiers utilizing Texas Instruments' BICOM-1 process. Designed for low-distortion with a high slew rate of 7000 V/μs, the THS306x amplifiers are ideally suited for applications requiring large, linear output signals such as video line drivers and VDSL line drivers. space RELATED DEVICES AND DESCRIPTIONS DEVICE THS3001 Low Distortion Current-Feedback Amplifier THS3112 Dual Current-Feedback Amplifier With 175 mA Drive THS3122 Dual Current-Feedback Amplifier With 350 mA Drive OPA691 Wideband Current-Feedback Amplifier with 350 mA Drive SLEW RATE vs OUTPUT STEP HARMONIC DISTORTION vs FREQUENCY 8000 –20 G=5 VCC = ±15 Rf = 375 Ω TA = 25°C 6000 G=1 –30 Harmonic Distortion – dB 7000 SR – Slew Rate – V/ µ s DESCRIPTION 5000 VCC = ±15 4000 3000 2000 –40 –50 VCC = ±15 V VCC = ±5 V RL = 1 kΩ Rf = 750 Ω VO = 2VPP –60 2nd HD –70 –80 3rd HD 1000 –90 0 0 5 10 15 Output Step – VPP 20 25 –100 100 k 1M 10 M f – Frequency – Hz 100 M 1 2 3 Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. PowerPAD is a trademark of Texas Instruments. All other trademarks are the property of their respective owners. PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of the Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. Copyright © 2002–2009, Texas Instruments Incorporated THS3061 THS3062 SLOS394B – JULY 2002 – REVISED NOVEMBER 2009 www.ti.com This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with appropriate precautions. Failure to observe proper handling and installation procedures can cause damage. ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more susceptible to damage because very small parametric changes could cause the device not to meet its published specifications. ABSOLUTE MAXIMUM RATINGS Over operating free-air temperature range unless otherwise noted. (1) VS Supply voltage VI Input voltage IO Output current VID Differential input voltage ±16.5 V ±VS 200 mA ±3 V Continuous power dissipation TJ TJ (2) Tstg (1) (2) See Dissipation Ratings Table Maximum junction temperature +150°C Maximum junction temperature, continuous operation, long term reliability +125°C Storage temperature range –65°C to +150°C The absolute maximum ratings under any condition is limited by the constraints of the silicon process. Stresses above these ratings may cause permanent damage. Exposure to absolute maximum conditions for extended periods may degrade device reliability. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those specified is not implied. The maximum junction temperature for continuous operation is limited by package constraints. Operation above this temperature may result in reduced reliability and/or lifetime of the device. PACKAGE DISSIPATION RATINGS (1) (2) (3) POWER RATING (TJ = +125°C) (2) PACKAGE θJC (°C/W) θJA (°C/W) (1) TA ≤ +25°C TA = +85°C D (8 pin) 38.3 97.5 1.02 W 410 mW DDA (8 pin) (3) 9.2 45.8 2.18 W 873 mW DGN (8 pin) (3) 4.7 58.4 1.71 W 680 mW This data was taken using the JEDEC High-K test PCB. This data was taken using 2 oz. trace and copper pad that is soldered directly to a 3 in x 3 in PCB. The THS306x may incorporate a PowerPAD on the underside of the chip. This acts as a heatsink and must be connected to a thermally dissipative plane for proper power dissipation. Failure to do so may result in exceeding the maximum junction temperature which could permanently damage the device. See TI technical brief SLMA002 for more information about utilizing the PowerPAD thermally enhanced package. RECOMMENDED OPERATING CONDITIONS Supply voltage 2 Submit Documentation Feedback MIN MAX Dual supply ±5 ±15 Single supply 10 30 UNIT V Copyright © 2002–2009, Texas Instruments Incorporated Product Folder Link(s): THS3061 THS3062 THS3061 THS3062 www.ti.com SLOS394B – JULY 2002 – REVISED NOVEMBER 2009 ORDERING INFORMATION (1) PART NUMBER PACKAGE TYPE PACKAGE MARKING SOIC-8 — MSOP-8-PP (2) BIB SOIC-8 — SOIC-8-PP (2) — MSOP-8-PP (2) BIC TRANSPORT MEDIA, QUANTITY Single THS3061D THS3061DR THS3061DGN THS3061DGNR Rails, 75 Tape and Reel, 2500 Rails, 80 Tape and Reel, 2500 Dual THS3062D THS3062DR THS3062DDA THS3062DDAR THS3062DGN THS3062DGNR (1) (2) Rails, 75 Tape and Reel, 2500 Rails, 75 Tape and Reel, 2500 Rails, 80 Tape and Reel, 2500 For the most current package and ordering information, see the Package Option Addendum at the end of this document, or see the TI web site at www.ti.com. The PowerPAD is electrically isolated from all other pins. PIN ASSIGNMENTS TOP VIEW D, DGN TOP VIEW D, DDA, DGN THS3061 NC VIN − VIN + VS− 1 8 2 7 3 6 4 5 THS3062 NC VS+ VOUT NC 1VOUT 1VIN − 1VIN + VS− 1 8 2 7 3 6 4 5 VS+ 2VOUT 2VIN − 2VIN+ NC − No internal connection PARAMETER MEASUREMENT INFORMATION Rg Rg Rf Rf VI _ _ VI + 50 Ω RT VO RL RL Figure 1. Noninverting Test Circuit VO + Figure 2. Inverting Test Circuit Copyright © 2002–2009, Texas Instruments Incorporated Product Folder Link(s): THS3061 THS3062 Submit Documentation Feedback 3 THS3061 THS3062 SLOS394B – JULY 2002 – REVISED NOVEMBER 2009 www.ti.com ELECTRICAL CHARACTERISTICS At VS = ±15 V: Rf = 560 Ω, RL = 150 Ω, and G = 2, unless otherwise noted. THS3061, THS3062 TYP PARAMETER OVER TEMPERATURE TEST CONDITIONS +25°C +25°C 0°C to +70°C –40°C to +85°C UNITS MIN/ TYP/ MAX MHz Typ AC PERFORMANCE Small-signal bandwidth (VO = 100 mVPP, Peaking < 0.3 dB) G = 1, Rf = 750Ω 300 G = 2, Rf = 560 Ω 275 G = 5, Rf = 357 Ω 260 G = 10, Rf = 200 Ω 220 Bandwidth for 0.1-dB flatness G = 2, VO = 100mVPP 120 MHz Typ Peaking at a gain of 1 VO = 100 mVPP 0.3 dB Typ Large-signal bandwidth G = 2, VO = 4 VPP 120 MHz Typ G = 5, 20 V-Step 7000 G = 2, 10 V-Step 5700 V/μs Typ 1 ns Typ 30 ns Typ 125 ns Typ dBc Typ dBc Typ Slew rate (25% to 75% level) Rise and fall time Settling time to 0.1% 0.01% G = 2, VO = 10 V-Step G = –2, VO = 2 V-Step Harmonic distortion 2nd order harmonic G = 2, f = 10 MHz, VO = 2 VPP 3rd order harmonic RL = 150 Ω –78 RL = 1 kΩ –73 RL = 150 Ω –81 RL = 1 kΩ –82 3rd order intermodulation distortion G = 2, fc = 10 MHz, VO = 2 VPP(envelope) Δf = 200 kHz –93 dBc Typ Input voltage noise f > 10 kHz 2.6 nV/√Hz Typ 20 pA/√Hz Typ 36 pA/√Hz Typ Input current noise (noninverting) Input current noise (inverting) Differential gain (NTSC, PAL) Differential phase (NTSC, PAL) f > 10 kHz G = 2, RL = 150 Ω 0.02% Typ 0.01° Typ DC PERFORMANCE Open-loop transimpedance gain VO = 0 V, RL = 1 kΩ Input offset voltage 1 0.7 0.6 0.6 MΩ Min ±0.7 ±3.5 ±4.4 ±4.5 mV Max ±10 ±10 μV/°C Typ ±2.0 ±20 ±32 ±35 μA Max ±25 ±30 nA/°C Typ ±38 ±40 A Max ±45 ±50 nA/°C Typ V Min Average offset voltage drift Input bias current (inverting) Average bias current drift (–) VCM = 0 V Input bias current (noninverting) ±6.0 ±25 Average bias current drift (+) INPUT Common-mode input range Common-mode rejection ratio Input resistance Input capacitance 4 Submit Documentation Feedback ±13.9 ±13.1 ±13.1 ±13.1 VCM = ±0.5 V 72 60 58 58 dB Min Noninverting 518 kΩ Typ Inverting 71 Ω Typ Noninverting 1 pF Typ Copyright © 2002–2009, Texas Instruments Incorporated Product Folder Link(s): THS3061 THS3062 THS3061 THS3062 www.ti.com SLOS394B – JULY 2002 – REVISED NOVEMBER 2009 ELECTRICAL CHARACTERISTICS (continued) At VS = ±15 V: Rf = 560 Ω, RL = 150 Ω, and G = 2, unless otherwise noted. THS3061, THS3062 TYP PARAMETER OVER TEMPERATURE TEST CONDITIONS +25°C +25°C 0°C to +70°C –40°C to +85°C ±13.7 ±13.4 ±13.4 ±13.3 ±13 ±12.6 ±12.4 ±12.3 UNITS MIN/ TYP/ MAX V Min OUTPUT Voltage output swing RL = 1 kΩ RL = 150 Ω Current output, sourcing RL = 50 Ω 145 140 135 130 mA Min Current output, sinking RL = 50 Ω –145 –140 –135 –130 mA Min 0.1 Ω Typ ±15 V Typ POWER SUPPLY Closed-loop output impedance G = 1, f = 1 MHz Specified operating voltage Maximum operating voltage ±16.5 ±16.5 ±16.5 V Max Maximum quiescent current/channel 8.3 10 11.7 12 mA Max Minimum quiescent current/channel 8.3 6.1 6 6 mA Min Power-supply rejection (+PSRR) VS+ = 14.50 V to 15.50 V 76 65 63 63 dB Min Power-supply rejection (–PSRR) VS– = –14.50 V to –15.50 V 74 65 63 63 dB Min Copyright © 2002–2009, Texas Instruments Incorporated Product Folder Link(s): THS3061 THS3062 Submit Documentation Feedback 5 THS3061 THS3062 SLOS394B – JULY 2002 – REVISED NOVEMBER 2009 www.ti.com ELECTRICAL CHARACTERISTICS At VS = ±5 V: Rf = 560 Ω, RL = 150 Ω , and G = 2, unless otherwise noted. THS3061, THS3062 TYP PARAMETER OVER TEMPERATURE TEST CONDITIONS +25°C +25°C 0°C to +70°C –40°C to +85°C UNITS MIN/ TYP/ MAX MHz Typ AC PERFORMANCE Small-signal bandwidth (VO = 100 mVPP, peaking < 0.3 dB) G = 1, Rf = 750 Ω 275 G = 2, Rf = 560 Ω 250 G = 5, Rf = 383 Ω 230 G = 10, Rf = 200 Ω 210 Bandwidth for 0.1-dB flatness G = 2, VO = 100 mVPP 100 MHz Typ Peaking at a gain of 1 VO = 100 mVPP < 0.3 dB Typ Large-signal bandwidth G = 2, VO = 4 VPP 100 MHz Typ G = 1, 5-V Step, Rf = 750 Ω 2700 G = 5, 5-V Step, Rf = 357 Ω 1300 V/μs Typ ns Typ ns Typ dBc Typ dBc Typ Slew rate (25% to 75% level) Rise and fall time 0.1% Settling time to 0.01% G = 2, VO = 5-V Step 2 20 G = –2, VO = 2-V Step 160 Harmonic distortion 2nd order harmonic G = 2, f = 10 MHz, VO = 2 VPP 3rd order harmonic RL = 150 Ω –76 RL = 1 kΩ –70 RL = 150 Ω –79 RL = 1 kΩ –77 3rd order intermodulation distortion G = 2, fc = 10 MHz, VO = 2 VPP(envelope) Δf = 200 kHz –91 dBc Typ Input voltage noise f > 10 kHz 2.6 nV/√Hz Typ 20 pA/√Hz Typ 36 pA/√Hz Typ Input current noise (noninverting) Input current noise (inverting) Differential gain (NTSC, PAL) Differential phase (NTSC, PAL) f > 10 kHz G = 2, RL = 150 Ω 0.025% Typ 0.01° Typ DC PERFORMANCE Open-loop transimpedance gain VO = 0 V, RL = 1 kΩ Input offset voltage 0.8 0.6 0.5 0.5 ±0.3 ±3.5 ±4.4 ±9 Average offset voltage drift Input bias current (inverting) Average bias current drift (–) ±2.0 ±20 ±6.0 ±25 VCM = 0 V Input bias current (noninverting) Average bias current drift (+) MΩ Min ±4.5 mV Max ±9 μV/°C Typ ±32 ±35 μA Max ±20 ±25 nA/°C Typ ±38 ±40 μA Max ±30 ±35 nA/°C Typ Min INPUT Common-mode input range Common-mode rejection ratio Input resistance Input capacitance ±3.9 ±3.1 ±3.1 ±3.1 V VCM = ±0.5 V 70 60 58 58 dB Min Noninverting 518 kΩ Typ Inverting 71 Ω Typ Noninverting 1 pF Typ V Min OUTPUT Voltage output swing 6 Submit Documentation Feedback RL = 1 kΩ ±4.1 ±3.8 ±3.8 ±3.7 RL = 150 Ω ±4.0 ±3.6 ±3.6 ±3.5 Copyright © 2002–2009, Texas Instruments Incorporated Product Folder Link(s): THS3061 THS3062 THS3061 THS3062 www.ti.com SLOS394B – JULY 2002 – REVISED NOVEMBER 2009 ELECTRICAL CHARACTERISTICS (continued) At VS = ±5 V: Rf = 560 Ω, RL = 150 Ω , and G = 2, unless otherwise noted. THS3061, THS3062 TYP PARAMETER Current output, sourcing Current output, sinking Closed-loop output impedance OVER TEMPERATURE TEST CONDITIONS RL = 50 Ω G = 1, f = 1 MHz +25°C +25°C 0°C to +70°C –40°C to +85°C UNITS MIN/ TYP/ MAX 63 61 60 59 mA Min –63 –61 –60 –59 mA Min 0.1 Ω Typ ±5 V Typ POWER SUPPLY Specified operating voltage Minimum operating voltage ±4.5 ±4.5 ±4.5 V Min Maximum quiescent current 6.3 8.0 9.2 9.5 mA Max Minimum quiescent current 6.3 5.0 4.7 4.6 mA Min Power-supply rejection (+PSRR) VS+ = 4.50 V to 5.50 V 73 65 63 63 dB Min Power-supply rejection (–PSRR) VS– = –4.50 V to –5.50 V 75 65 63 63 dB Min space space space TYPICAL CHARACTERISTICS Table of Graphs FIGURE Small-signal frequency response 3-14 Large-signal frequency response 15, 16 Harmonic distortion vs Frequency 17-23 Harmonic distortion vs Output voltage 24-29 Output impedance vs Frequency 30 Common-mode rejection ratio vs Frequency 31 Input current noise vs Frequency 32 Voltage noise density vs Frequency 33 Power-supply rejection ratio vs Frequency 34 Common-mode rejection ratio (DC) vs Input common-mode range Supply current vs Power-supply voltage 36, 37 Slew rate vs Output voltage 38, 39 Slew rate vs Output step Input offset voltage vs Output voltage swing Overdrive recovery time 35 40 41 42, 43 Differential gain vs Number of 150-Ω loads 44, 45 Differential phase vs Number of 150-Ω loads 46, 47 Copyright © 2002–2009, Texas Instruments Incorporated Product Folder Link(s): THS3061 THS3062 Submit Documentation Feedback 7 THS3061 THS3062 SLOS394B – JULY 2002 – REVISED NOVEMBER 2009 www.ti.com TYPICAL CHARACTERISTICS SMALL-SIGNAL FREQUENCY RESPONSE SMALL-SIGNAL FREQUENCY RESPONSE 2 1 0 -1 Rf 1 kΩ Rf 750 Ω -2 G=1 VCC = ±5 V RL = 1 kΩ VI = 100 mVPP Rf 500 Ω 1 -1 Rf 1 kΩ 1M 10 M 100M f - Frequency - Hz -4 1G 100 k 1M 10 M 100 M f - Frequency - Hz SMALL-SIGNAL FREQUENCY RESPONSE SMALL-SIGNAL FREQUENCY RESPONSE SMALL-SIGNAL FREQUENCY RESPONSE 9 10 G=1 VCC = ±15 V RL = 1 kΩ VI = 100 mVPP Rf 357 Ω Rf 500 Ω 6 4 Rf 1 kΩ 2 0 Rf 750 Ω -3 G=2 VCC = ±15, ±5 V RL = 150 Ω VI = 100 mVPP -2 -4 1M 10 M 100 M f - Frequency - Hz 4 Rf 1 kΩ G=2 VCC = ±15, ±5 V RL = 1 kΩ VI = 100 mVPP 2 1 Rf 560 Ω 0 100 k 1G 5 3 Rf 560 Ω -4 1M 10 M 100 M 100 k 1G 1M 10 M 100 M Figure 6. Figure 7. Figure 8. SMALL-SIGNAL FREQUENCY RESPONSE SMALL-SIGNAL FREQUENCY RESPONSE SMALL-SIGNAL FREQUENCY RESPONSE 18 Rf 200 Ω 16 14 12 100 k 1M 10 M 100 M 1G f - Frequency - Hz Figure 9. Submit Documentation Feedback 16 Rf 560 Ω 10 Rf 383 Ω Rf 200 Ω 14 12 Rf 560 Ω 10 18 G=5 VCC = ±15 V RL = 1 kΩ VI = 100 mVPP Gain - dB Gain - dB G=5 VCC = ±5 V RL = 150 Ω VI = 100 mVPP 8 1G f - Frequency - Hz f - Frequency - Hz 18 8 Rf 357 Ω 7 6 Rf 1 kΩ 100 k 1G 8 8 Gain - dB -1 16 1M 10 M 100 M f - Frequency - Hz Figure 5. -2 Gain - dB 100 k Figure 4. 0 8 1G Figure 3. 2 1 Rf 750 Ω -3 -4 100 k Rf 1 kΩ -1 Rf 750 Ω -3 -4 Rf 500 Ω -2 -2 -3 G=1 VCC = ±15 V RL = 150 Ω VI = 100 mVPP 0 0 Gain - dB Gain - dB 1 Rf 500 Ω Gain - dB 2 Gain - dB 2 3 G=1 VCC = ±5 V RL = 150 Ω VI = 100 mVPP Gain - dB 3 SMALL-SIGNAL FREQUENCY RESPONSE G=5 VCC = ±15 V RL = 150 Ω VI = 100 mVPP 14 12 Rf 560 Ω 10 Rf 357 Ω Rf 200 Ω Rf 357 Ω 8 100 k 1M 10 M 100 M 1G 100 k 1M 10 M f - Frequency - Hz f - Frequency - Hz Figure 10. Figure 11. 100 M 1G Copyright © 2002–2009, Texas Instruments Incorporated Product Folder Link(s): THS3061 THS3062 THS3061 THS3062 www.ti.com SLOS394B – JULY 2002 – REVISED NOVEMBER 2009 TYPICAL CHARACTERISTICS (continued) SMALL-SIGNAL FREQUENCY RESPONSE SMALL-SIGNAL FREQUENCY RESPONSE 3 3 G = -1 VCC =±5 V RL = 150 Ω VI = 100 mVPP 2 20 1 Gain - dB 19 18 G = 10 VCC =±5 V, ±15 V RL = 150 Ω, 1 kΩ VI = 100 mVPP, 17 Rf 332 Ω 1 0 -1 Rf 560 Ω -2 Rf 511 Ω 1M 10 M 100 M 1G Rf 560 Ω Rf 475 Ω 1M 10 M 100 M 1G 100 k 1M 10 M 100 M f - Frequency - Hz f - Frequency - Hz Figure 12. Figure 13. Figure 14. LARGE-SIGNAL FREQUENCY RESPONSE LARGE-SIGNAL FREQUENCY RESPONSE THS3061 HARMONIC DISTORTION vs FREQUENCY 9 8 Harmonic Distortion - dB 6 Gain - dB Gain - dB Rf = 1 KΩ 5 4 3 G=2 VCC = ±15 V Rf = 560 Ω VI = 2 VPP, 0 100 k 3 G=2 VCC = ±5 V Rf = 604 Ω RL = 150 Ω RL = 1 kΩ VI = 2 VPP, 10 M 100 M 100 k 1G 1M -40 -50 -60 2nd HD -70 -80 3rd HD -90 2 1M VCC = ±15 V G=1 RL = 150 Ω VO = 2 VPP Rf = 750 Ω -30 RL = 150 Ω 6 10 M 100 M -100 1G 100 k f - Frequency - Hz f - Frequency - Hz 1M 10 M Figure 16. Figure 17. THS3062 HARMONIC DISTORTION vs FREQUENCY HARMONIC DISTORTION vs FREQUENCY HARMONIC DISTORTION vs FREQUENCY -20 Harmonic Distortion - dB -60 -30 2nd HD -80 -40 -50 −60 VCC = ±5 V G=1 RL = 150 Ω VO = 2 VPP Rf = 750 Ω 2nd HD -60 -70 -80 -90 3rd HD Harmonic Distortion − dB VCC = ±15 V G=1 RL = 150 Ω VO = 2 VPP Rf = 845 Ω 100 k 1M 10 M f - Frequency - Hz Figure 18. −70 −80 2nd HD −90 3rd HD −100 −110 100 k 100 M VCC = ±5 V, VCC = ±15 V G=2 Rf = 560 Ω RL = 150 Ω VO = 1 VPP 3rd HD -100 -100 100 M f - Frequency - Hz Figure 15. -20 1G -20 7 Harmonic Distortion - dB -1 -4 100 k f - Frequency - Hz -40 0 -3 -4 100 k Rf 332 Ω -2 -3 16 G = -1 VCC =±15 V RL = 150 Ω VI = 100 mVPP 2 Gain - dB 21 Gain - dB SMALL-SIGNAL FREQUENCY RESPONSE 1M 10 M f - Frequency - Hz 100 M Figure 19. Copyright © 2002–2009, Texas Instruments Incorporated Product Folder Link(s): THS3061 THS3062 1M 10 M 100 M f − Frequency − Hz Figure 20. Submit Documentation Feedback 9 THS3061 THS3062 SLOS394B – JULY 2002 – REVISED NOVEMBER 2009 www.ti.com TYPICAL CHARACTERISTICS (continued) HARMONIC DISTORTION vs FREQUENCY -60 2nd HD -70 -80 3rd HD -60 -70 -80 3rd HD 1M 10 M 100 M -100 -60 -70 -80 3rd HD -90 1M 10 M 100 M -100 100 k Figure 23. THS3061 HARMONIC DISTORTION vs OUTPUT VOLTAGE THS3061 HARMONIC DISTORTION vs OUTPUT VOLTAGE THS3061 HARMONIC DISTORTION vs OUTPUT VOLTAGE -60 2nd HD -75 3rd HD -80 -85 VCC = ±5 V G=1 RL = 1 kΩ Rf = 750 Ω f = 10 MHz -90 -95 -70 3rd HD -75 -80 -85 VCC = ±15 V G=1 RL = 1 kΩ Rf = 750 Ω f = 10 MHz -90 -95 -100 2 3 4 5 6 3rd HD -80 -85 -90 1 2 3 4 5 6 0 1 VO - Output Voltage - V 2 3 4 Figure 26. THS3061 HARMONIC DISTORTION vs OUTPUT VOLTAGE HARMONIC DISTORTION vs OUTPUT VOLTAGE HARMONIC DISTORTION vs OUTPUT VOLTAGE -60 -60 -85 -70 -80 2nd HD 8 MHz -90 3rd HD 1 MHz VCC = ±15 V G=2 Rf = 560 Ω RL = 150 Ω -100 -90 -95 3 4 5 VO - Output Voltage - V Figure 27. Submit Documentation Feedback 6 Harmonic Distortion - dBc Harmonic Distortion - dBc 2nd HD 1 MHz -80 2 6 3rd HD 8 MHz 2nd HD 3rd HD 1 5 VO - Output Voltage - V Figure 25. -75 0 -75 Figure 24. VCC = ±5 V G=1 RL = 150 Ω f= 1 MHz -70 2nd HD -100 0 VO - Output Voltage - V -65 -70 100 M -95 -100 1 VCC = ±15 V G=1 RL = 150 Ω f= 1 MHz Rf = 750 Ω -65 Harmonic Distortion - dBc Harmonic Distortion - dBc -65 -70 -60 1M 10 M f - Frequency - Hz Figure 22. -60 0 2nd HD Figure 21. 2nd HD -65 Harmonic Distortion - dBc -50 f - Frequency - Hz -60 Harmonic Distortion - dBc -40 -90 f - Frequency - Hz 10 G=1 VCC = ±15 V VCC = ±5 V RL = 1 kΩ Rf = 750 Ω VO = 2VPP -30 2nd HD -90 -100 -20 VCC = ±15 V G=2 Rf = 560 Ω RL = 150 Ω VO = 2VPP -50 Harmonic Distortion - dB -50 Harmonic Distortion - dB -40 VCC = ±5 V G=2 Rf = 560 Ω RL = 150 Ω VO = 2VPP HARMONIC DISTORTION vs FREQUENCY Harmonic Distortion - dB -40 HARMONIC DISTORTION vs FREQUENCY -110 0 2 4 6 8 10 2nd HD 1 MHz -70 2nd HD f = 8 MHz -80 3rd HD 8 MHz -90 3rd HD f = 1 MHz VCC = ±5 V G=2 Rf = 560 Ω RL = 150 Ω -100 -110 0 12 VO - Output Voltage - V 1 2 3 4 5 6 VO - Output Voltage - V Figure 28. Figure 29. Copyright © 2002–2009, Texas Instruments Incorporated Product Folder Link(s): THS3061 THS3062 THS3061 THS3062 www.ti.com SLOS394B – JULY 2002 – REVISED NOVEMBER 2009 TYPICAL CHARACTERISTICS (continued) OUTPUT IMPEDANCE vs FREQUENCY 10 1 0.1 0.01 1M 10 M 100 M THS3062 50 40 THS3061 20 10 In150 100 50 In+ 1M 10 M 100 M 1G 10 100 k Figure 31. Figure 32. VOLTAGE NOISE DENSITY vs FREQUENCY POWER-SUPPLY REJECTION RATIO vs FREQUENCY COMMON-MODE REJECTION RATIO (DC) vs INPUT COMMON-MODE RANGE 25 VCC = ±5 V 20 15 10 5 VCC = ±15 V 10 100 1k 10 k f - Frequency - Hz 70 CMMR - Common-Mode Rejection Ratio (DC) - dB Figure 30. 0 VCC = ±15 V, ±5 V G=2 Rf = 560 Ω RL = 150 Ω VO = 35 mVPP 60 50 40 PSRR+ 30 20 10 PSRR0 -10 100 k 100 k 1M 10 M 100 M f - Frequency - Hz 80 RL = 150 Ω 70 60 50 40 30 20 10 0 -15 -10 -5 0 5 Figure 34. Figure 35. THS3061 SUPPLY CURRENT vs POWER-SUPPLY VOLTAGE THS3062 SUPPLY CURRENT vs POWER-SUPPLY VOLTAGE SLEW RATE vs OUTPUT VOLTAGE 21 -40°C 6 4 2 0 2.5 85°C G = -1 Rf = 475 Ω RL = 150 Ω TA = 25°C 18 3000 SR - Slew Rate - V/ µ s 8 ICC - Supply Current - mA 25°C 25°C 15 12 -40°C 9 6 6.5 8.5 10.5 12.5 4 Power Supply Voltage - V Figure 36. VCC = ±15 2000 1500 VCC = ±5 1000 0 0 14.5 16.5 2500 500 3 4.5 15 3500 24 85°C 10 10 Input Common-Mode Voltage Range - V Figure 33. 12 I CC - Supply Current - mA 100 1k 10 k f - Frequency - Hz f - Frequency - Hz PSRR - Power Supply Rejection Ratio - dBc Hz nV/ Vn - Voltage Noise Density - 30 200 0 0 100 k 1G 45 35 Hz 60 30 VCC = ±15 V, VCC = ±5 V, VCC = ±2.5 V, 250 pA 70 f - Frequency - Hz 40 300 G=2 VCC = ±15 V, ±5 V RL = 150 Ω Rf = 1 kΩ 80 I n - Input Current Noise - G=2 Rf = 560 Ω VCC = ±15 V 100 k 90 CMRR - Common-Mode Rejection Ratio - dB ZO - Output Impedance - Ω 1000 100 INPUT CURRENT NOISE vs FREQUENCY COMMON-MODE REJECTION RATIO vs FREQUENCY 6 8 10 12 14 Power Supply Voltage - V 16 Figure 37. Copyright © 2002–2009, Texas Instruments Incorporated Product Folder Link(s): THS3061 THS3062 0 2 4 6 VO - Output Voltage - V Figure 38. Submit Documentation Feedback 11 THS3061 THS3062 SLOS394B – JULY 2002 – REVISED NOVEMBER 2009 www.ti.com TYPICAL CHARACTERISTICS (continued) SLEW RATE vs OUTPUT VOLTAGE SLEW RATE vs OUTPUT STEP 8000 G=1 Rf = 750 Ω RL = 150 Ω TA = 25°C SR - Slew Rate - V/ µ s 2000 VCC = ±15 1500 VCC = ±5 1000 500 6000 5000 VCC = ±15 4000 3000 2000 1000 1 2 3 4 VO - Output Voltage - V 5 0 6 5 Figure 39. OVERDRIVE RECOVERY TIME 3 5 0 0 -1 -1 -2 G=2 VCC = ±5 Rf = 604 Ω RL = 150 Ω -2 -3 -3 0.5 0 1 1.5 VI - Input Voltage - V 1 VO - Output Voltage - V 0 0 -4 0.6 10 5 -5 -5 G=5 VCC = ±15 Rf = 560 Ω RL = 150 Ω 0.5 0 1 1.5 0.5 VCC = ±5 0.4 0.3 0.2 -10 0.1 -15 0 2 VCC = ±15 1 2 3 Number of 150-Ω Loads t - Time - µs Figure 42. Figure 43. Figure 44. DIFFERENTIAL GAIN vs NUMBER OF 150-Ω LOADS DIFFERENTIAL PHASE vs NUMBER OF 150-Ω LOADS DIFFERENTIAL PHASE vs NUMBER OF 150-Ω LOADS 0.07 G=2 Rf = 560 Ω PAL Modulation 0.4 VCC = ±5 0.3 VCC = ±15 0.2 0.05 0.03 0.02 0.01 0 0 1 2 3 Number of 150-Ω Loads Figure 45. Submit Documentation Feedback 4 VCC = ±5 0.04 0.1 4 0.08 G=2 Rf = 560 Ω NTSC Modulation 0.06 Differential Phase - Degree 0.5 20 G=2 Rf = 560 Ω NTSC Modulation t - Time - µs 0.6 Differential Gain - % OVERDRIVE RECOVERY TIME -1 2 -15 -10 -5 0 5 10 15 VO - Output Voltage Swing - V DIFFERENTIAL GAIN vs NUMBER OF 150-Ω LOADS 1 -3 -1 Figure 41. 2 -4 85°C Figure 40. 15 -2 25°C -0.5 -20 Differential Phase - Degree VI - Input Voltage - V 2 25 3 3 1 20 4 4 2 10 15 Output Step - VPP Differential Gain - % 0 -40°C -1.5 0 0 VO - Output Voltage - V SR - Slew Rate - V/ µ s 2500 0 G=5 VCC = ±15 Rf = 375 Ω TA = 25°C 7000 VIO- Input Offset Voltage - mV 3000 12 INPUT OFFSET VOLTAGE vs OUTPUT VOLTAGE SWING G=2 Rf = 560 Ω PAL Modulation 0.06 VCC = ±5 0.04 0.02 VCC = ±15 VCC = ±15 1 2 3 0 4 1 2 3 Number of 150-Ω Loads Number of 150-Ω Loads Figure 46. Figure 47. 4 Copyright © 2002–2009, Texas Instruments Incorporated Product Folder Link(s): THS3061 THS3062 THS3061 THS3062 www.ti.com SLOS394B – JULY 2002 – REVISED NOVEMBER 2009 APPLICATION INFORMATION INTRODUCTION The THS306x is a high-speed operational amplifier configured in a current-feedback architecture. The device is built using Texas Instruments' BiCOM-I process, a 30-V, dielectrically isolated, complementary bipolar process with NPN and PNP transistors possessing fTs of several GHz. This configuration implements an exceptionally high-performance amplifier that has a wide bandwidth, high slew rate, fast settling time, and low distortion. MAXIMUM SLEW RATE FOR REPETITIVE SIGNALS The THS3061 and THS3062 are recommended for high slew rate, pulsed applications where the internal nodes of the amplifier have time to stabilize between pulses.It is recommended to have at least a 20-ns delay between pulses. The THS3061 and THS3062 are not recommended for applications with repetitive signals (sine, square, sawtooth, or other types) that exceed 900 V/μs. Using this device in these types of applications results in an excessive current draw from the power supply and possible device damage. For applications with a high slew rate and repetitive signals, the THS3091 and THS3095 (singles) or the THS3092 and THS3096 (duals) are recommended instead. RECOMMENDED FEEDBACK AND GAIN RESISTOR VALUES As with all current-feedback amplifiers, the bandwidth of the THS306x is an inversely proportional function of the value of the feedback resistor. The recommended resistors for optimum frequency response are shown in Table 1. These should be used as a starting point, and once optimum values are found, 1% tolerance resistors should be used to maintain frequency response characteristics. For most applications, a feedback-resistor value of 750 Ω is recommended—a good compromise between bandwidth and phase margin that yields a very stable amplifier. As shown in Table 1, to maintain the highest bandwidth with increasing gain, the feedback resistor is reduced. The advantage of dropping the feedback resistance (and the gain-resistor value) is that the noise of the system is also reduced compared to no reduction of these resistor values (see NOISE CALCULATIONS). Thus, keeping the bandwidth as high as possible maintains very good distortion performance of the amplifier by keeping the excess loop gain as high as possible. Table 1. Recommended Resistor Values for Optimum Frequency Response GAIN RF for VCC = ±15 V RF for VCC = ±5 V 1 750 Ω 750 Ω 2, –1 560 Ω 560 Ω 5 357 Ω 383 Ω 10 200 Ω 200 Ω Care must be taken to not set these values too low. The amplifier's output must drive the feedback resistance (and gain resistance), and this may place a burden on the amplifier. The end result is that distortion may actually increase due to the low-impedance load presented to the amplifier. The designer must carefully manage the amplifier bandwidth and the associated loading effects for optimum performance. The THS3061/62 amplifiers exhibit very good distortion performance and bandwidth, and can use power supplies up to 15 V. The excellent current-drive capability of up to 145 mA into a 50-Ω load allows many versatile applications. One application is driving a twisted pair line (that is, a telephone line). Figure 48 shows a simple circuit for driving a twisted pair differentially. Copyright © 2002–2009, Texas Instruments Incorporated Product Folder Link(s): THS3061 THS3062 Submit Documentation Feedback 13 THS3061 THS3062 SLOS394B – JULY 2002 – REVISED NOVEMBER 2009 www.ti.com THS3062(a) 0.1 µF en Noiseless + _ eni + 10 µF IN+ eno eRf RS + _ VI+ eRs RS +12 V Rf RLine 2n2 499 Ω 0.1 µF Telephone Line 210 Ω Rg RLine THS3062(b) VI– Figure 49. Noise Model RS + _ eRg IN– 1:n RLine space 2n2 499 Ω 0.1 µF 10 µF space + –12 V space Figure 48. Simple Line Driver With THS3062 Due to the high supply voltages and the large current-drive capability, the power dissipation of the amplifier must be carefully considered. To have as much power dissipation as possible in a small package, the THS3062 is available only in a MSOP-8 PowerPAD package (DGN), and an even lower thermal-impedance SOIC-8 PowerPAD package (DDA). The thermal impedance of a standard SOIC package is too large to allow useful applications with up to 30 V across the power-supply terminals with this dual amplifier. But the THS3061 (a single amplifier) can be used in the standard SOIC package. Again, the amplifier power dissipation must be carefully examined, or else the amplifiers could overheat, severely degrading performance. See the Power Dissipation and Thermal Considerations section for more information on thermal management. NOISE CALCULATIONS Noise can cause errors on very small signals. This is especially true for amplifying small signals coming over a transmission line or an antenna. The noise model for current-feedback amplifiers (CFB) is the same as for voltage feedback amplifiers (VFB). The only difference between the two is that CFB amplifiers generally specify different current-noise parameters for each input, while VFB amplifiers usually only specify one noise-current parameter. The noise model is shown in Figure 49. This model includes all of the noise sources as follows: • en = Amplifier internal voltage noise (nV/√Hz) • IN+ = Noninverting current noise (pA/√Hz) • IN– = Inverting current noise (pA/√Hz) • eRx = Thermal voltage noise associated with each resistor (eRx = 4 kTRx) 14 Submit Documentation Feedback space space The total equivalent input noise density (eni) is calculated by using the following equation: e ni + Ǹǒ 2 e nǓ ) IN ) ǒ R S Ǔ 2 ǒ ) IN * ǒR f ø RgǓǓ 2 ǒ Ǔ ) 4 kTR s ) 4 kT R ø R g f where k = Boltzmann’s constant = 1.380658 × 10–23 T = Temperature in degrees Kelvin (273 +°C) Rf || Rg = Parallel resistance of Rf and Rg To calculate the equivalent output noise of the amplifier, multiply the equivalent input noise density (eni) by the overall amplifier gain (AV). e no + e ǒ Ǔ R A + e 1) f ni V ni Rg (Noninverting Case) As the previous equations show, to keep noise at a minimum, small value resistors should be used. As the closed-loop gain is increased (by reducing RF and RG), the input noise is reduced considerably because of the parallel resistance term. This leads to the general conclusion that the most dominant noise sources are the source resistor (RS) and the internal amplifier noise voltage (en). Because noise is summed in a root-mean-squares method, noise sources smaller than 25% of the largest noise source can be effectively ignored. This can greatly simplify the formula and make noise calculations much easier. PCB LAYOUT TECHNIQUES FOR OPTIMAL PERFORMANCE Achieving optimum performance with high-frequency devices in the THS306x family requires careful attention to board layout, parasitic effects, and external component types. Recommendations to optimize performance include: Copyright © 2002–2009, Texas Instruments Incorporated Product Folder Link(s): THS3061 THS3062 THS3061 THS3062 www.ti.com • • • SLOS394B – JULY 2002 – REVISED NOVEMBER 2009 Minimize parasitic capacitance to any ac ground for all signal I/O pins. Parasitic capacitance on the output and input pins can cause instability. To reduce unwanted capacitance, a window around the signal I/O pins should be opened in all of the ground and power planes around those pins. Ground and power planes should be unbroken elsewhere on the board. Minimize the distance (< 0.25”) from the power supply pins to high frequency 0.1-μF decoupling capacitors. At the device pins, the ground and power plane layout should not be in close proximity to the signal I/O pins. Avoid narrow power and ground traces to minimize inductance between the pins and the decoupling capacitors. The power supply connections should always be decoupled with these capacitors. Larger (6.8 μF or more) tantalum decoupling capacitors, effective at lower frequencies, should also be used on the main supply pins. These may be placed somewhat farther from the device and may be shared among several devices in the same area of the printed circuit board (PCB). The primary goal is to minimize the impedance in the differential-current return paths. For driving differential loads with the THS3062, adding a capacitor between the power-supply pins improves 2nd-order harmonic-distortion performance. This also minimizes the current loop formed by the differential drive. Careful selection and placement of external components preserve the high frequency performance of the THS306x family. Resistors should be a very low reactance type. Surface-mount resistors work best and allow a tighter overall layout. Again, keep leads and PCB trace lengths as short as possible. Never use wirebound-type resistors in a high-frequency application. Since the output pin and inverting input pins are the most sensitive to parasitic capacitance, always position the feedback and series-output resistors, if any, as close as possible to the inverting-input pins and output pins. Other network components, such as input-termination resistors, should be placed close to the gain-setting resistors. Even with a low parasitic capacitance shunting the external resistors, excessively high resistor values can create significant time constants that can degrade performance. Good axial metal-film or surface-mount resistors have approximately 0.2 pF in shunt with the resistor. For resistor values > 2.0 kΩ, this parasitic capacitance can add a pole and/or a zero that can affect circuit operation. Keep resistor values as low as possible, consistent with load-driving considerations. • • Connections to other wideband devices on the board may be made with short direct traces or through onboard transmission lines. For short connections, consider the trace and the input to the next device as a lumped capacitive load. Relatively wide traces (50 mils to 100 mils) should be used, preferably with ground and power planes opened up around them. Estimate the total capacitive load and determine if isolation resistors on the outputs are necessary. Low parasitic capacitive loads (< 4 pF) may not need an RS since the THS306x family is nominally compensated to operate with a 2-pF parasitic load. Higher parasitic capacitive loads without an RS are allowed as the signal gain increases (increasing the unloaded phase margin). If a long trace is required, and the 6-dB signal loss intrinsic to a doubly-terminated transmission line is acceptable, implement a matched-impedance transmission line using microstrip or stripline techniques (consult an ECL design handbook for microstrip and stripline layout techniques). A 50-Ω environment is not necessary onboard, and in fact, a higher-impedance environment improves distortion as shown in the distortion-versus-load plots. With a characteristic board-trace impedance based on board material and trace dimensions, a matching series resistor is used in the trace from the output of the THS306x, as well as a terminating shunt resistor at the input of the destination device. Remember also that the terminating impedance is the parallel combination of the shunt resistor and the input impedance of the destination device: this total effective impedance should be set to match the trace impedance. If the 6-dB attenuation of a doubly-terminated transmission line is unacceptable, a long trace can be series-terminated at the source end only. Treat the trace as a capacitive load in this case. This does not preserve signal integrity as well as a doubly-terminated line. If the input impedance of the destination device is low, there is some signal attenuation due to the voltage divider formed by the series output into the terminating impedance. Socketing a high speed part like the THS306x family is not recommended. The additional lead length and pin-to-pin capacitance introduced by the socket can create an extremely troublesome parasitic network that can make it almost impossible to achieve a smooth, stable frequency response. Best results are obtained by soldering the THS306x family parts directly onto the board. PowerPAD DESIGN CONSIDERATIONS The THS306x family is available in a thermally-enhanced PowerPAD family of packages. These packages are constructed using a downset Copyright © 2002–2009, Texas Instruments Incorporated Product Folder Link(s): THS3061 THS3062 Submit Documentation Feedback 15 THS3061 THS3062 SLOS394B – JULY 2002 – REVISED NOVEMBER 2009 www.ti.com leadframe upon which the die is mounted [see Figure 50(a) and Figure 50(b)]. This arrangement results in the lead frame being exposed as a thermal pad on the underside of the package [see Figure 50(c)]. Because this thermal pad has direct thermal contact with the die, excellent thermal performance can be achieved by providing a good thermal path away from the thermal pad. 0.205 0.060 0.017 Pin 1 0.013 0.030 0.075 The PowerPAD package allows for both assembly and thermal management in one manufacturing operation. During the surface-mount solder operation (when the leads are being soldered), the thermal pad can also be soldered to a copper area underneath the package. Through the use of thermal paths within this copper area, heat can be conducted away from the package into either a ground plane or other heat dissipating device. The PowerPAD package represents a breakthrough in combining the small area and ease of assembly of surface mount, compared with awkward mechanical methods of heatsinking. DIE Side View (a) Thermal Pad DIE End View (b) Bottom View (c) Figure 50. Views of Thermally Enhanced Package Although there are many ways to properly heatsink the PowerPAD package, the following steps illustrate the recommended approach. PowerPAD PCB LAYOUT CONSIDERATIONS 1. Prepare the PCB with a top-side etch pattern as shown in Figure 51. There should be etch for the leads as well as etch for the thermal pad. 16 Submit Documentation Feedback 0.025 0.094 0.010 vias 0.035 0.040 Top View Figure 51. DGN PowerPAD PCB Etch and Via Pattern 2. Place five holes in the area of the thermal pad. These holes should be 10 mils in diameter. Keep them small so that solder wicking through the holes is not a problem during reflow. 3. Additional vias may be placed anywhere along the thermal plane outside of the thermal pad area. This helps dissipate the heat generated by the THS306x-family IC. These additional vias may be larger than the 10-mil diameter vias directly under the thermal pad. They can be larger because they are not in the thermal pad area to be soldered, so wicking is not a problem. 4. Connect all holes to the internal ground plane. 5. When connecting these holes to the ground plane, do not use the typical web or spoke via connection methodology. Web connections have a high thermal resistance that is useful for slowing the heat transfer during soldering operations, making the soldering of vias that have plane connections easier. In this application, however, low thermal resistance is desired for the most efficient heat transfer. Therefore, the holes under the THS306x family PowerPAD package should make their connection to the internal ground plane with a complete connection around the entire circumference of the plated-through hole. 6. The top-side solder mask should leave the terminals of the package and the thermal pad area with its five holes exposed. The bottom-side solder mask should cover the five holes of the thermal pad area. This prevents solder from being pulled away from the thermal pad area during the reflow process. 7. Apply solder paste to the exposed thermal pad area and all IC terminals. 8. With these preparatory steps in place, the IC is simply placed in position and run through the Copyright © 2002–2009, Texas Instruments Incorporated Product Folder Link(s): THS3061 THS3062 THS3061 THS3062 www.ti.com SLOS394B – JULY 2002 – REVISED NOVEMBER 2009 POWER DISSIPATION AND THERMAL CONSIDERATIONS To maintain maximum output capability, the THS360x does not incorporate automatic thermal shutoff protection. The designer must ensure that the design does not violate the absolute maximum junction temperature of the device. Failure may result if the absolute maximum junction temperature of +150°C is exceeded. For best performance, design for a maximum junction temperature of +125°C. Between +125°C and +150°C, damage does not occur, but the performance of the amplifier begins to degrade. The thermal characteristics of the device are dictated by the package and the PC board. Maximum power dissipation for a given package can be calculated using Equation 1. P Dmax + Tmax * T A q JA where PDmax is the maximum power dissipation in the amplifier (W). Tmax is the absolute maximum junction temperature (°C). TA is the ambient temperature (°C). θJA = θJC + θCA θJC is the thermal coefficient from the silicon junctions to the case (°C/W). θCA is the thermal coefficient from the case to ambient air (°C/W). (1) For systems where heat dissipation is more critical, the THS306x family of devices is offered in an 8-pin MSOP with PowerPAD, and the THS3062 is available in the SOIC-8 PowerPAD package offering even better thermal performance. The thermal coefficients for the PowerPAD packages are substantially improved over traditional SOICs. Maximum power dissipation levels are given in the graph for the available packages. Data for the PowerPAD packages assumes a board layout that follows the PowerPAD layout guidelines referenced above and detailed in the PowerPAD application note number SLMA002. The following graph also illustrates the effect of not soldering the PowerPAD to a PCB. The thermal impedance increases substantially, which can cause serious heat and performance issues. Always be sure to solder the PowerPAD to the PCB for optimum performance. 4 PD – Maximum Power Dissipation – W solder reflow operation as any standard surface-mount component. This results in a properly-installed device. TJ = 125°C 3.5 θJA = 45.8°C/W 3 θJA = 58.4°C/W 2.5 θJA = 98°C/W 2 1.5 1 0.5 θJA = 158°C/W 0 –40 –20 0 20 40 60 80 TA – Free-Air Temperature – °C 100 Results are With No Air Flow and PCB Size = 3”x3” θJA = 45.8°C/W for 8-Pin SOIC w/PowerPad (DDA) θJA = 58.4°C/W for 8-Pin MSOP w/PowerPad (DGN) θJA = 98°C/W for 8-Pin SOIC High Test PCB (D) θJA = 158°C/W for 8-Pin MSOP w/PowerPad w/o Solder Figure 52. Maximum Power Dissipation vs Ambient Temperature When determining whether or not the device satisfies the maximum power dissipation requirement, it is important not only to consider quiescent power dissipation, but also dynamic power dissipation. Often, this is difficult to quantify because the signal pattern is inconsistent, but an estimate of the RMS power dissipation can provide visibility into a possible problem. DRIVING A CAPACITIVE LOAD Driving capacitive loads with high-performance amplifiers is not a problem as long as certain precautions are taken. The first is to realize that the THS306x has been internally compensated to maximize its bandwidth and slew-rate performance. When the amplifier is compensated in this manner, capacitive loading directly on the output decreases the device's phase margin, leading to high-frequency ringing or oscillations. Therefore, for capacitive loads of greater than 10 pF, it is recommended that a resistor be placed in series with the output of the amplifier as shown in Figure 53. A minimum value of 10 Ω works well for most applications. For example, in 75-Ω transmission systems, setting the series-resistor value to 75 Ω both isolates any capacitive loading and provides the proper line impedance matching at the source end. Copyright © 2002–2009, Texas Instruments Incorporated Product Folder Link(s): THS3061 THS3062 Submit Documentation Feedback 17 THS3061 THS3062 SLOS394B – JULY 2002 – REVISED NOVEMBER 2009 www.ti.com Rf Rg Input _ 10 Ω Output THS306x + CLOAD and should not be considered when using a current-feedback amplifier. Because of this, integrators and simple low-pass filters, which are easily implemented on a VFB amplifier, must be designed slightly differently. If filtering is required, simply place an RC-filter at the noninverting terminal of the operational-amplifier (see Figure 54). Rg Rf f Figure 53. Driving a Capacitive Load V – VI + R1 VO –3dB O + V I ǒ + 1) 1 2pR1C1 Ǔǒ R f Rg Ǔ 1 1 ) sR1C1 C1 General Configurations A common error for the first-time CFB-amplifier user is creating a unity gain buffer amplifier by shorting the output directly to the inverting input. In this configuration, a CFB amplifier oscillates, and is not recommended. The THS306x, like all CFB amplifiers, must have a feedback resistor for stable operation. Additionally, placing capacitors directly from the output to the inverting input is not recommended, because at high frequencies, a capacitor has a very low impedance. This results in an unstable amplifier 18 Submit Documentation Feedback Figure 54. Single-Pole Low-Pass Filter Copyright © 2002–2009, Texas Instruments Incorporated Product Folder Link(s): THS3061 THS3062 THS3061 THS3062 www.ti.com SLOS394B – JULY 2002 – REVISED NOVEMBER 2009 If a multiple-pole filter is required, a Sallen-Key filter can work very well with CFB amplifiers. This is because the filtering elements are not in the negative feedback loop and stability is not compromised. Because of their high slew rates and high bandwidths, CFB amplifiers can pass very accurate signals and help minimize distortion. An example is shown in Figure 55. Rg Rf For Stable Operation: – THS306x VO + R2 R1 || RA ≥ ( 1+ VO ≅ VI R1 R2 Rf Rg Rf Rg sR1C1 ) VI C1 RA C1 + _ VI R1 R1 = R2 = R C1 = C2 = C Q = Peaking Factor (Butterworth Q = 0.707) R2 f C2 Rg –3dB Rg = Rf + ( 2– Rf 1 Q ) Figure 55. 2-Pole Low-Pass Sallen-Key Filter There are two simple ways to create an integrator with a CFB amplifier. The first, shown in Figure 56, adds a resistor in series with the capacitor. This is acceptable, because at high frequencies, the resistor is dominant and the feedback impedance never drops below the resistor value. The second, shown in Figure 57, uses positive feedback to create the integration. Caution is advised because oscillations can occur due to the positive feedback. The THS306x may also be employed as a very good video-distribution amplifier. One characteristic of distribution amplifiers is the fact that the differential phase (dP) and the differential gain (dG) are compromised as the number of lines increases and the closed-loop gain increases. Be sure to use termination resistors throughout the distribution system to minimize reflections and capacitive loading. Rg Rf – 75-Ω Transmission Line VO1 75 Ω 75 Ω THS306x N Lines 75 Ω VON 75 Ω V Rg – + 75 Ω + VI C1 Rf VI Figure 57. Noninverting CFB Integrator 1 2pRC VO O + VI THS306x ǒ R ȡS ) Rf1C1ȣ Ǔȧ S ȧ Ȣ Ȥ f Rg Figure 58. Video Distribution Amplifier Application Figure 56. Inverting CFB Integrator Copyright © 2002–2009, Texas Instruments Incorporated Product Folder Link(s): THS3061 THS3062 Submit Documentation Feedback 19 THS3061 THS3062 SLOS394B – JULY 2002 – REVISED NOVEMBER 2009 www.ti.com REVISION HISTORY NOTE: Page numbers for previous revisions may differ from page numbers in the current version. Changes from Revision A (October, 2002) to Revision B Page • Updated document format to current standards ................................................................................................................... 1 • Deleted lead temperature specification from Absolute Maximum Ratings table .................................................................. 2 • Added Maximum Slew Rate for Repetitive Signals section ................................................................................................ 13 20 Submit Documentation Feedback Copyright © 2002–2009, Texas Instruments Incorporated Product Folder Link(s): THS3061 THS3062 PACKAGE OPTION ADDENDUM www.ti.com 13-Aug-2021 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Pins Package Drawing Qty Eco Plan (2) Lead finish/ Ball material MSL Peak Temp Op Temp (°C) Device Marking (3) (4/5) (6) THS3061D ACTIVE SOIC D 8 75 RoHS & Green NIPDAU Level-1-260C-UNLIM -40 to 85 3061 THS3061DGN ACTIVE HVSSOP DGN 8 80 RoHS & Green NIPDAU Level-1-260C-UNLIM -40 to 85 BIB THS3061DGNR ACTIVE HVSSOP DGN 8 2500 RoHS & Green NIPDAU Level-1-260C-UNLIM -40 to 85 BIB THS3061DGNRG4 ACTIVE HVSSOP DGN 8 2500 RoHS & Green NIPDAU Level-1-260C-UNLIM -40 to 85 BIB THS3061DR ACTIVE SOIC D 8 2500 RoHS & Green NIPDAU Level-1-260C-UNLIM -40 to 85 3061 THS3062D ACTIVE SOIC D 8 75 RoHS & Green NIPDAU Level-1-260C-UNLIM -40 to 85 3062 THS3062DDA ACTIVE SO PowerPAD DDA 8 75 RoHS & Green SN Level-1-260C-UNLIM -40 to 85 3062 THS3062DDAG3 ACTIVE SO PowerPAD DDA 8 75 RoHS & Green SN Level-1-260C-UNLIM -40 to 85 3062 THS3062DGN ACTIVE HVSSOP DGN 8 80 RoHS & Green NIPDAU | NIPDAUAG Level-1-260C-UNLIM -40 to 85 BIC THS3062DGNG4 ACTIVE HVSSOP DGN 8 80 RoHS & Green Level-1-260C-UNLIM -40 to 85 BIC NIPDAU (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may reference these types of products as "Pb-Free". RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption. Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of