SLOS336C − DECEMBER 2000 − REVISED OCTOBER 2002
D Fully Specified for 3.3-V and 5-V Operation
D Wide Power Supply Compatibility
D OR DGN PACKAGE
(TOP VIEW)
2.5 V − 5.5 V
SHUTDOWN
BYPASS
IN+
IN−
D Power Supply Rejection at 217 Hz
D
D
D
D
− 84 dB at VDD = 5 V
− 81 dB at VDD = 3.3 V
Output Power for RL = 8 Ω
− 700 mW at VDD = 5 V
− 250 mW at VDD = 3.3 V
Ultralow Supply Current in Shutdown
Mode . . . 1.5 nA
Thermal and Short-Circuit Protection
Surface-Mount Packaging
− SOIC
− PowerPAD MSOP
− MicroStar Junior (BGA)
1
8
2
7
3
6
4
5
VO −
GND
VDD
VO +
MicroStar Juniort (GQS) Package
(TOP VIEW)
(E2)
SHUTDOWN
(E3)
BYPASS
(E4)
IN+
(E5)
IN−
(A2)
(A3)
(A4)
(A5)
VO−
GND
VDD
VO+
(SIDE VIEW)
NOTE: The shaded terminals are used for thermal
connections to the ground plane.
description
The TPA751 is a bridge-tied load (BTL) audio power amplifier developed especially for low-voltage applications
where internal speakers are required. Operating with a 3.3-V supply, the TPA751 can deliver 250-mW of
continuous power into a BTL 8-Ω load at less than 0.6% THD+N throughout voice band frequencies. Although
this device is characterized out to 20 kHz, its operation is optimized for narrower band applications such as
wireless communications. The BTL configuration eliminates the need for external coupling capacitors on the
output in most applications, which is particularly important for small battery-powered equipment. This device
features a shutdown mode for power-sensitive applications with a supply current of 1.5 nA during shutdown.
The TPA751 is available in a 3.0 × 3.0 mm MicroStar Junior (BGA), 8-pin SOIC surface-mount package and
a surface-mount PowerPAD MSOP.
VDD 6
RF
VDD/2
Audio
Input
RI
CI
4
IN −
3
IN+
2
BYPASS
VDD
CS
−
VO+ 5
+
CB
−
VO− 8
+
700 mW
7
GND
From System Control
1
SHUTDOWN
Bias
Control
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PowerPAD and MicroStar Junior are trademarks of Texas Instruments.
Copyright 2002, Texas Instruments Incorporated
! " #$ %! " &$'(# ! ) !%*
)$#!" # ! "&%# # ! " &% !+% !% " %, " "! $ %!"
"! ) ) - !.* )$#! & #%""/ ) %" ! %#%"" (. #($)%
!%"!/ (( & %!% "*
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
1
SLOS336C − DECEMBER 2000 − REVISED OCTOBER 2002
AVAILABLE OPTIONS
PACKAGED DEVICES
MicroStar-Junior (BGA)‡
(GQS)
SMALL OUTLINE†
(D)
MSOP‡
(DGN)
TPA751GQS
TPA751D
TPA751DGN
Device
Package symbolization
TPA751
TPA751
ATC
† In the SOIC package, the maximum RMS output power is thermally limited to 350 mW; 700 mW peaks can be driven, as long
as the RMS value is less than 350 mW.
‡ The D, DGN, and GQS packages are available taped and reeled. To order a taped and reeled part, add the suffix R to the
part number (e.g., TPA751DR).
Terminal Functions
TERMINAL
NO.
NAME
I/O
DESCRIPTION
I
BYPASS is the tap to the voltage divider for internal mid-supply bias. This terminal should be
connected to a 0.1-µF to 2.2-µF capacitor when used as an audio amplifier.
GQS
D, DGN
E3
2
GND
§
7
IN −
E5
4
I
IN − is the inverting input. IN − is typically used as the audio input terminal.
IN+
E4
3
I
IN + is the noninverting input. IN + is typically tied to the BYPASS terminal for SE input.
SHUTDOWN
E2
1
I
SHUTDOWN places the entire device in shutdown mode when held low (IDD = 1.5 nA).
VDD
VO+
A4
6
A5
5
O
VDD is the supply voltage terminal.
VO+ is the positive BTL output.
BYPASS
GND is the ground connection.
VO−
A2
8
O
VO− is the negative BTL output.
§ A1, A3, A5, B1−B5, C1−C5, D1−D5 are electrical and thermal connections to the ground plane.
absolute maximum ratings over operating free-air temperature range (unless otherwise noted)¶
Supply voltage, VDD . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 V
Input voltage, VI . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −0.3 V to VDD +0.3 V
Continuous total power dissipation . . . . . . . . . . . . . . . . . . . . . Internally limited (see Dissipation Rating Table)
Operating free-air temperature range, TA . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −40°C to 85°C
Operating junction temperature range, TJ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −40°C to 150°C
Storage temperature range, Tstg . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −65°C to 150°C
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 260°C
¶ Stresses beyond those listed under “absolute maximum ratings” may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated under “recommended operating conditions” is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
DISSIPATION RATING TABLE
PACKAGE
GQS||
TA = 25°C
1.66 W||
D
DGN
DERATING FACTOR
13.3 mW/°C
TA = 70°C
1.06 W
TA = 85°C
866 mW
725 mW
5.8 mW/°C
464 mW
377 mW
2.14 W#
17.1 mW/°C
1.37 W
1.11 W
# See the Texas Instruments document, PowerPAD Thermally Enhanced Package Application Report
(SLMA002), for more information on the PowerPAD package. The thermal data was measured on a PCB
layout based on the information in the section entitled Texas Instruments Recommended Board for PowerPAD
on page 33 of that document.
|| See the Texas Instruments document, MicroStar Junior Made Easy Application Brief (SSYA009A) for board
layout information on the MicroStar Junior package.
2
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
SLOS336C − DECEMBER 2000 − REVISED OCTOBER 2002
recommended operating conditions
ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ
ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ
ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ
ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ
ÁÁ
ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ
MIN
MAX
2.5
5.5
Supply voltage, VDD
High-level input voltage, VIH, (SHUTDOWN)
0.9VDD
V
V
Low-level input voltage, VIL, (SHUTDOWN)
Operating free-air temperature, TA
UNIT
0.1VDD
85
−40
V
°C
electrical characteristics at specified free-air temperature, VDD = 3.3 V, TA = 25°C (unless otherwise
noted)
ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ
ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ
ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ
ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ
ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ
ÁÁ
ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ
ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ
ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ
ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ
ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ
ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ
ÁÁÁÁÁ
ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ
ÁÁÁÁÁ
ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ
ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ
PARAMETER
TEST CONDITIONS
|VOS|
Output offset voltage (measured differentially)
SHUTDOWN = VDD, RL = 8 Ω, RF = 10 kΩ
PSRR
Power supply rejection ratio
IDD
IDD(SD)
Supply current
VDD = 3.2 V to 3.4 V
SHUTDOWN = VDD, RF = 10 kΩ
Supply current, shutdown mode (see Figure 4)
SHUTDOWN = 0 V, RF = 10 kΩ
MIN
TYP
MAX
UNIT
20
mV
1.25
2.5
mA
1.5
1000
nA
85
dB
|IIH|
SHUTDOWN, VDD = 3.3 V, Vi = VDD
1
µA
|IIL|
SHUTDOWN, VDD = 3.3 V, Vi = 0 V
1
µA
operating characteristics, VDD = 3.3 V, TA = 25°C, RL = 8 Ω
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
Output power, See Note 1
THD = 0.2%,
See Figure 9
Total harmonic distortion plus noise
f = 200 Hz to 4 kHz,
See Figure 7
BOM
B1
Maximum output power bandwidth
PO = 250 mW,
AV = −2 V/V,
THD = 2%,
See Figure 7
Unity-gain bandwidth
Open loop,
See Figure 15
Supply ripple rejection ratio
f = 1 kHz,
CB = 1 µF,
See Figure 2
79
dB
Noise output voltage
AV = −1V/V,
CB = 0.1 µF,
See Figure 19
17
µV(rms)
Vn
250
UNIT
PO
THD + N
mW
0.55%
20
kHz
1.4
MHz
NOTE 1: Output power is measured at the output terminals of the device at f = 1 kHz.
electrical characteristics at specified free-air temperature, VDD = 5 V, TA = 25°C (unless otherwise
noted)
ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ
ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ
ÁÁÁÁÁÁÁÁÁ
ÁÁÁ
ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ
ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ
ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ
ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ
PARAMETER
TEST CONDITIONS
|VOS|
Output offset voltage (measured differentially)
SHUTDOWN = VDD, RL = 8 Ω, RF = 10 kΩ
PSRR
Power supply rejection ratio
IDD
IDD(SD)
Supply current
VDD = 4.9 V to 5.1 V
SHUTDOWN = VDD, RF = 10 kΩ
Supply current, shutdown mode (see Figure 4)
SHUTDOWN = 0 V, RF = 10 kΩ
MIN
TYP
MAX
UNIT
20
mV
1.45
2.5
mA
5
78
dB
1500
nA
|IIH|
SHUTDOWN, VDD = 5.5 V, Vi = VDD
1
µA
|IIL|
SHUTDOWN, VDD = 5.5 V, Vi = 0 V
1
µA
operating characteristics, VDD = 5 V, TA = 25°C, RL = 8 Ω
ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ
ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ
ÁÁÁÁÁÁÁÁÁ
ÁÁÁ
ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ
ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ
ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ
ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ
PARAMETER
TEST CONDITIONS
MIN
PO
THD + N
Output power
THD = 0.5%,
See Figure 13
Total harmonic distortion plus noise
f = 200 Hz to 4 kHz,
See Figure 11
BOM
B1
Maximum output power bandwidth
PO = 250 mW,
AV = −2 V/V,
THD = 2%,
See Figure 11
Unity-gain bandwidth
Open loop,
See Figure 16
TYP
700†
MAX
UNIT
mW
0.5%
20
kHz
1.4
MHz
Supply ripple rejection ratio
f = 1 kHz,
CB = 1 µF,
See Figure 2
80
dB
Vn
Noise output voltage
AV = −1 V/V,
CB = 0.1 µF,
See Figure 20
17
µV(rms)
† The GQS and DGN packages, properly mounted, can conduct 700 mW RMS power continuously. The D package, can only conduct 350 mW
RMS power continuously, with peaks to 700 mW.
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
3
SLOS336C − DECEMBER 2000 − REVISED OCTOBER 2002
PARAMETER MEASUREMENT INFORMATION
VDD 6
RF
VDD/2
Audio
Input
RI
CI
VDD
CS
4
IN −
3
IN+
2
BYPASS
VO+ 5
−
+
RL = 8 Ω
CB
−
VO− 8
+
7
GND
VDD
1
SHUTDOWN
Bias
Control
Figure 1. BTL Mode Test Circuit
TYPICAL CHARACTERISTICS
Table of Graphs
FIGURE
kSVR
Supply ripple rejection ratio
vs Frequency
IDD
Supply current
vs Supply voltage
3, 4
vs Supply voltage
5
PO
Output power
vs Load resistance
vs Frequency
THD + N
Vn
PD
4
Total harmonic distortion plus noise
vs Output power
2
6
7, 8, 11, 12
9, 10, 13, 14
Open loop gain and phase
vs Frequency
15, 16
Closed loop gain and phase
vs Frequency
17, 18
Output noise voltage
vs Frequency
19, 20
Power dissipation
vs Output power
21, 22
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
SLOS336C − DECEMBER 2000 − REVISED OCTOBER 2002
TYPICAL CHARACTERISTICS
SUPPLY RIPPLE REJECTION RATIO
vs
FREQUENCY
SUPPLY CURRENT
vs
SUPPLY VOLTAGE
1.8
RL = 8 Ω
CB = 1 µF
Inputs Floating
−20
SHUTDOWN = VDD
RF = 10 kΩ
1.6
I DD − Supply Current − mA
−10
−30
−40
−50
−60
−70
VDD = 3.3 V
−80
−100
20
100
1.4
1.2
1
0.8
VDD = 5 V
−90
10k
1k
0.6
2.5
20k
3.5
3
f − Frequency − Hz
4
4.5
5
5.5
VDD − Supply Voltage − V
Figure 2
Figure 3
SUPPLY CURRENT
vs
SUPPLY VOLTAGE
10
9
SHUTDOWN = 0 V
RF = 10 kΩ
8
I DD − Supply Current − nA
k SVR − Supply Ripple Rejection Ratio − dB
0
7
6
5
4
3
2
1
0
2.5
3
3.5
4
4.5
5
5.5
VDD − Supply Voltage − V
Figure 4
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
5
SLOS336C − DECEMBER 2000 − REVISED OCTOBER 2002
TYPICAL CHARACTERISTICS
OUTPUT POWER
vs
SUPPLY VOLTAGE
1000
THD+N 1%
f = 1 kHz
PO − Output Power − mW
800
600
RL = 8 Ω
RL = 32 Ω
400
200
0
2.5
3
3.5
4
4.5
5
5.5
VDD − Supply Voltage − V
Figure 5
OUTPUT POWER
vs
LOAD RESISTANCE
800
THD+N = 1%
f = 1 kHz
PO − Output Power − mW
700
600
VDD = 5 V
500
400
300
VDD = 3.3 V
200
100
0
8
16
24
32
40
48
56
RL − Load Resistance − Ω
Figure 6
6
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
64
SLOS336C − DECEMBER 2000 − REVISED OCTOBER 2002
TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
10
THD+N −Total Harmonic Distortion + Noise − %
THD+N −Total Harmonic Distortion + Noise − %
10
VDD = 3.3 V
PO = 250 mW
RL = 8 Ω
AV = −20 V/V
1
AV =− 10 V/V
AV = −2 V/V
0.1
0.01
20
100
1k
10k
VDD = 3.3 V
RL = 8 Ω
AV = −2 V/V
PO = 50 mW
1
0.1
PO = 125 mW
PO = 250 mW
0.01
20
20k
100
1k
f − Frequency − Hz
Figure 7
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
10
VDD = 3.3 V
f = 1 kHz
AV = −2 V/V
THD+N −Total Harmonic Distortion + Noise − %
THD+N −Total Harmonic Distortion + Noise − %
10
1
RL = 8 Ω
0.1
0.01
0.05
0.1
20k
Figure 8
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
0
10k
f − Frequency − Hz
0.15
0.2
0.25
0.3
0.35
0.4
f = 20 kHz
1
f = 10 kHz
f = 1 kHz
0.1
f = 20 Hz
0.01
0.01
PO − Output Power − W
VDD = 3.3 V
RL = 8 Ω
CB = 1 µF
AV = −2 V/V
0.1
1
PO − Output Power − W
Figure 9
Figure 10
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
7
SLOS336C − DECEMBER 2000 − REVISED OCTOBER 2002
TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
10
VDD = 5 V
PO = 700 mW
RL = 8 Ω
THD+N −Total Harmonic Distortion + Noise − %
THD+N −Total Harmonic Distortion + Noise − %
10
AV = −20 V/V
1
AV = −10 V/V
AV = −2 V/V
0.1
0.01
20
100
1k
10k
20k
VDD = 5 V
RL = 8 Ω
AV = −2 V/V
1
PO = 700 mW
0.1
PO = 350 mW
0.01
20
100
f − Frequency − Hz
20k
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
10
10
THD+N −Total Harmonic Distortion + Noise − %
THD+N −Total Harmonic Distortion + Noise − %
10k
Figure 12
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
VDD = 5 V
f = 1 kHz
AV = −2 V/V
1
RL = 8 Ω
0.1
0.2
0.3
0.4
0.5
0.6
0.7
0.8
0.9
1
f = 20 kHz
1
f = 10 kHz
f = 1 kHz
f = 20 Hz
0.1
VDD = 5 V
RL = 8 Ω
CB = 1 µF
AV = −2 V/V
0.01
0.01
PO − Output Power − W
0.1
PO − Output Power − W
Figure 13
8
1k
f − Frequency − Hz
Figure 11
0.01
0.1
PO = 50 mW
Figure 14
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
1
SLOS336C − DECEMBER 2000 − REVISED OCTOBER 2002
TYPICAL CHARACTERISTICS
OPEN-LOOP GAIN AND PHASE
vs
FREQUENCY
80
180°
VDD = 3.3 V
RL = Open
70
140°
Phase
100°
50
60°
40
20°
30
Gain
20
−20°
10
Phase
Open-Loop Gain − dB
60
−60°
0
−100°
−10
−140°
−20
−30
1
101
102
103
104
−180°
f − Frequency − kHz
Figure 15
OPEN-LOOP GAIN AND PHASE
vs
FREQUENCY
80
180°
VDD = 5 V
RL = Open
70
140°
60
100°
60°
40
20°
30
Gain
20
−20°
10
Phase
Open-Loop Gain − dB
Phase
50
−60°
0
−100°
−10
−140°
−20
−30
1
101
102
103
104
−180°
f − Frequency − kHz
Figure 16
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
9
SLOS336C − DECEMBER 2000 − REVISED OCTOBER 2002
TYPICAL CHARACTERISTICS
CLOSED-LOOP GAIN AND PHASE
vs
FREQUENCY
1
180°
Phase
0.75
170°
0.25
0
160°
Gain
−0.25
150°
−0.5
−0.75
140°
−1
−1.25
−1.5
Phase
Closed-Loop Gain − dB
0.5
VDD = 3.3 V
RL = 8 Ω
PO = 250 mW
130°
−1.75
−2
101
102
103
104
105
106
120°
f − Frequency − Hz
Figure 17
CLOSED-LOOP GAIN AND PHASE
vs
FREQUENCY
1
180°
Phase
0.75
170°
0.25
0
160°
Gain
−0.25
150°
−0.5
−0.75
140°
−1
−1.25
−1.5
VDD = 5 V
RL = 8 Ω
PO = 700 m W
130°
−1.75
−2
101
102
103
104
105
f − Frequency − Hz
Figure 18
10
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
120°
106
Phase
Closed-Loop Gain − dB
0.5
SLOS336C − DECEMBER 2000 − REVISED OCTOBER 2002
TYPICAL CHARACTERISTICS
OUTPUT NOISE VOLTAGE
vs
FREQUENCY
100
VDD = 3.3 V
BW = 22 Hz to 22 kHz
RL = 8 Ω or 32 Ω
AV = −1 V/V
Vn − Output Noise Voltage − µV(rms)
Vn − Output Noise Voltage − µV(rms)
100
OUTPUT NOISE VOLTAGE
vs
FREQUENCY
VO BTL
VO+
10
1
20
100
1k
10k
VDD = 5 V
BW = 22 Hz to 22 kHz
RL = 8 Ω or 32 Ω
AV = −1 V/V
VO BTL
VO+
10
1
20
20k
100
f − Frequency − Hz
Figure 19
20k
POWER DISSIPATION
vs
OUTPUT POWER
350
800
VDD = 3.3 V
VDD = 5 V
RL = 8 Ω
RL = 8 Ω
700
PD − Power Dissipation − mW
300
PD − Power Dissipation − mW
10k
Figure 20
POWER DISSIPATION
vs
OUTPUT POWER
250
200
150
100
1k
f − Frequency − Hz
RL = 32 Ω
50
600
500
400
300
200
RL = 32 Ω
100
0
0
200
400
600
0
0
PD − Output Power − mW
200
400
600
800
1000
PD − Output Power − mW
Figure 21
Figure 22
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
11
SLOS336C − DECEMBER 2000 − REVISED OCTOBER 2002
APPLICATION INFORMATION
bridged-tied load
Figure 23 shows a linear audio power amplifier (APA) in a BTL configuration. The TPA751 BTL amplifier consists
of two linear amplifiers driving both ends of the load. There are several potential benefits to this differential drive
configuration, but initially consider power to the load. The differential drive to the speaker means that as one
side is slewing up, the other side is slewing down, and vice versa. This, in effect, doubles the voltage swing on
the load as compared to a ground referenced load. Plugging 2 × VO(PP) into the power equation, where voltage
is squared, yields 4× the output power from the same supply rail and load impedance (see equation 1).
V
V
(rms)
+
V
Power +
O(PP)
2 Ǹ2
2
(1)
(rms)
R
L
VDD
VO(PP)
RL
2x VO(PP)
VDD
−VO(PP)
Figure 23. Bridge-Tied Load Configuration
In a typical portable handheld equipment sound channel operating at 3.3 V, bridging raises the power into an
8-Ω speaker from a singled-ended (SE, ground reference) limit of 62.5 mW to 250 mW. In sound power that is
a 6-dB improvement, which is loudness that can be heard. In addition to increased power, there are frequency
response concerns. Consider the single-supply SE configuration shown in Figure 24. A coupling capacitor is
required to block the dc offset voltage from reaching the load. These capacitors can be quite large
(approximately 33 µF to 1000 µF), so they tend to be expensive, heavy, occupy valuable PCB area, and have
the additional drawback of limiting low-frequency performance of the system. This frequency-limiting effect, due
to the high pass filter network created with the speaker impedance and the coupling capacitance, is calculated
with equation 2.
fc +
12
1
2p R C
L C
(2)
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
SLOS336C − DECEMBER 2000 − REVISED OCTOBER 2002
APPLICATION INFORMATION
bridged-tied load (continued)
For example, a 68-µF capacitor with an 8-Ω speaker would attenuate low frequencies below 293 Hz. The BTL
configuration cancels the dc offsets, which eliminates the need for the blocking capacitors. Low-frequency
performance is then limited only by the input network and speaker response. Cost and PCB space are also
minimized by eliminating the bulky coupling capacitor.
VDD
−3 dB
VO(PP)
CC
RL
VO(PP)
fc
Figure 24. Single-Ended Configuration and Frequency Response
Increasing power to the load does carry a penalty of increased internal power dissipation. The increased
dissipation is understandable considering that the BTL configuration produces 4× the output power of a SE
configuration. Internal dissipation versus output power is discussed further in the thermal considerations
section.
BTL amplifier efficiency
The primary cause of linear amplifier inefficiencies is voltage drop across the output stage transistors. There
are two components of the internal voltage drop. One is the headroom or dc voltage drop that varies inversely
to output power. The second component is due to the sinewave nature of the output. The total voltage drop, can
be calculated by subtracting the RMS value of the output voltage from VDD. The internal voltage drop multiplied
by the RMS value of the supply current, IDDrms, determines the internal power dissipation of the amplifier.
An easy-to-use equation to calculate efficiency starts out being equal to the ratio of power from the power supply
to the power delivered to the load. To accurately calculate the RMS values of power in the load and in the
amplifier, the current and voltage waveform shapes must first be understood (see Figure 25).
IDD
VO
IDD(RMS)
V(LRMS)
Figure 25. Voltage and Current Waveforms for BTL Amplifiers
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
13
SLOS336C − DECEMBER 2000 − REVISED OCTOBER 2002
APPLICATION INFORMATION
BTL amplifier efficiency (continued)
Although the voltages and currents for SE and BTL are sinusoidal in the load, currents from the supply are very
different between SE and BTL configurations. In an SE application, the current waveform is a half-wave rectified
shape, whereas in BTL it is a full-wave rectified waveform. This means RMS conversion factors are different.
Keep in mind that for most of the waveform both the push and pull transistors are not on at the same time, which
supports the fact that each amplifier in the BTL device only draws current from the supply for half the waveform.
The following equations are the basis for calculating amplifier efficiency.
Efficiency of a BTL amplifier +
P
P
L
(3)
SUP
where
2
V rms 2
V
V
P + L
, and V
+ P , therefore, P + P
L
L
LRMS
Ǹ2
R
2R
L
L
1
and P SUP + V DD I DDavg and I DDavg + p
ŕ
p V
P sin(t) dt + 1
p
R
0
L
2V
P
P [cos(t)] p +
0
pR
R
L
L
V
therefore,
V
DD P
SUP
pR
L
substituting PL and PSUP into equation 7,
+
P
2V
2
Efficiency of a BTL amplifier +
where
V
P
+
VP
2 RL
2 V DD V P
p RL
+
p VP
4 V DD
Ǹ2 PL RL
PL = Power delivered to load
PSUP = Power drawn from power supply
VLRMS = RMS voltage on BTL load
RL = Load resistance
VP = Peak voltage on BTL load
IDDavg = Average current drawn from the
power supply
VDD = Power supply voltage
ηBTL = Efficiency of a BTL amplifier
therefore,
h BTL +
14
p
Ǹ2 PL RL
4V
(4)
DD
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
SLOS336C − DECEMBER 2000 − REVISED OCTOBER 2002
APPLICATION INFORMATION
application schematics
Figure 26 is a schematic diagram of a typical handheld audio application circuit, configured for a gain of
−10 V/V.
VDD 6
RF
50 kΩ
Audio
Input
RI
10 kΩ
CI
4
IN −
3
IN+
2
BYPASS
VDD
CS
1 µF
VDD/2
−
VO+ 5
+
CB
2.2 µF
−
VO− 8
+
700 mW
7
GND
1
From System Control
SHUTDOWN
Bias
Control
Figure 26. TPA751 Application Circuit
Figure 27 is a schematic diagram of a typical handheld audio application circuit, configured for a gain of
−10 V/V with a differential input.
VDD 6
RF
50 kΩ
Audio
Input−
RI
10 kΩ
CI
RI
10 kΩ
Audio
Input+
VDD/2
4
IN −
3
IN+
−
VO+ 5
+
RF
50 kΩ
2
CI
VDD
CS
1 µF
BYPASS
CB
2.2 µF
−
VO− 8
+
700 mW
7
GND
From System Control
1
SHUTDOWN
Bias
Control
Figure 27. TPA751 Application Circuit With Differential Input
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
15
SLOS336C − DECEMBER 2000 − REVISED OCTOBER 2002
APPLICATION INFORMATION
application schematics (continued)
It is important to note that using the additional RF resistor connected between IN+ and BYPASS causes VDD/2
to shift slightly, which could influence the THD+N performance of the amplifier. Although an additional external
operational amplifier could be used to buffer BYPASS from RF, tests in the lab have shown that the THD+N
performance is only minimally affected by operating in the fully differential mode as shown in Figure 27. The
following sections discuss the selection of the components used in Figures 26 and 27.
component selection
gain setting resistors, RF and RI
The gain for each audio input of the TPA751 is set by resistors RF and RI according to equation 5 for BTL mode.
ǒ Ǔ
BTL gain + * 2
R
F
R
I
(5)
BTL mode operation brings about the factor 2 in the gain equation due to the inverting amplifier mirroring the
voltage swing across the load. Given that the TPA751 is a MOS amplifier, the input impedance is very high;
consequently input leakage currents are not generally a concern, although noise in the circuit increases as the
value of RF increases. In addition, a certain range of RF values is required for proper start-up operation of the
amplifier. Taken together it is recommended that the effective impedance seen by the inverting node of the
amplifier be set between 5 kΩ and 20 kΩ. The effective impedance is calculated in equation 6.
Effective impedance +
R R
F I
R )R
F
I
(6)
As an example, consider an input resistance of 10 kΩ and a feedback resistor of 50 kΩ. The BTL gain of the
amplifier would be −10 V/V and the effective impedance at the inverting terminal would be 8.3 kΩ, which is well
within the recommended range.
For high performance applications, metal film resistors are recommended because they tend to have lower
noise levels than carbon resistors. For values of RF above 50 kΩ, the amplifier tends to become unstable due
to a pole formed from RF and the inherent input capacitance of the MOS input structure. For this reason, a small
compensation capacitor of approximately 5 pF should be placed in parallel with RF when RF is greater than
50 kΩ. This, in effect, creates a low-pass filter network with the cutoff frequency defined in equation 7.
−3 dB
fc +
1
2p R C
F F
fc
For example, if RF is 100 kΩ and CF is 5 pF, then fc is 318 kHz, which is well outside of the audio range.
16
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
(7)
SLOS336C − DECEMBER 2000 − REVISED OCTOBER 2002
APPLICATION INFORMATION
input capacitor, CI
In the typical application an input capacitor, CI, is required to allow the amplifier to bias the input signal to the
proper dc level for optimum operation. In this case, CI and RI form a high-pass filter with the corner frequency
determined in equation 8.
−3 dB
fc +
1
2p R C
I I
(8)
fc
The value of CI is important to consider, as it directly affects the bass (low frequency) performance of the circuit.
Consider the example where RI is 10 kΩ and the specification calls for a flat bass response down to 40 Hz.
Equation 8 is reconfigured as equation 9.
1
C +
I
2p R f c
I
(9)
In this example, CI is 0.40 µF, so one would likely choose a value in the range of 0.47 µF to 1 µF. A further
consideration for this capacitor is the leakage path from the input source through the input network (RI, CI) and
the feedback resistor (RF) to the load. This leakage current creates a dc offset voltage at the input to the amplifier
that reduces useful headroom, especially in high gain applications. For this reason a low-leakage tantalum or
ceramic capacitor is the best choice. When polarized capacitors are used, the positive side of the capacitor
should face the amplifier input in most applications, as the dc level there is held at VDD/2, which is likely higher
than the source dc level. It is important to confirm the capacitor polarity in the application.
power supply decoupling, CS
The TPA751 is a high-performance CMOS audio amplifier that requires adequate power supply decoupling to
ensure the output total harmonic distortion (THD) is as low as possible. Power supply decoupling also prevents
oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling is achieved
by using two capacitors of different types that target different types of noise on the power supply leads. For
higher frequency transients, spikes, or digital hash on the line, a good low equivalent-series-resistance (ESR)
ceramic capacitor, typically 0.1 µF, placed as close as possible to the device VDD lead, works best. For filtering
lower-frequency noise signals, a larger aluminum electrolytic capacitor of 10 µF or greater placed near the audio
power amplifier is recommended.
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
17
SLOS336C − DECEMBER 2000 − REVISED OCTOBER 2002
APPLICATION INFORMATION
midrail bypass capacitor, CB
The midrail bypass capacitor, CB, is the most critical capacitor and serves several important functions. During
start-up or recovery from shutdown mode, CB determines the rate at which the amplifier starts up. The second
function is to reduce noise produced by the power supply caused by coupling into the output drive signal. This
noise is from the midrail generation circuit internal to the amplifier, which appears as degraded PSRR and
THD + N. The capacitor is fed from a 250-kΩ source inside the amplifier. To keep the start-up pop as low as
possible, the relationship shown in equation 10 should be maintained. This insures the input capacitor is fully
charged before the bypass capacitor is fully charged and the amplifier starts up.
ǒCB
10
250 kΩ
v
1
Ǔ ǒRF ) RIǓ CI
(10)
As an example, consider a circuit where CB is 2.2 µF, CI is 0.47 µF, RF is 50 kΩ, and RI is 10 kΩ. Inserting these
values into the equation 10 we get:
18.2 v 35.5
which satisfies the rule. Bypass capacitor, CB, values of 0.1 µF to 2.2 µF ceramic or tantalum low-ESR capacitors
are recommended for the best THD and noise performance.
using low-ESR capacitors
Low-ESR capacitors are recommended throughout this applications section. A real (as opposed to ideal)
capacitor can be modeled simply as a resistor in series with an ideal capacitor. The voltage drop across this
resistor minimizes the beneficial effects of the capacitor in the circuit. The lower the equivalent value of this
resistance, the more the real capacitor behaves like an ideal capacitor.
5-V versus 3.3-V operation
The TPA751 operates over a supply range of 2.5 V to 5.5 V. This data sheet provides full specifications for 5-V
and 3.3-V operation, as these are considered to be the two most common standard voltages. There are no
special considerations for 3.3-V versus 5-V operation with respect to supply bypassing, gain setting, or stability.
The most important consideration is that of output power. Each amplifier in TPA751 can produce a maximum
voltage swing of VDD − 1 V. This means, for 3.3-V operation, clipping starts to occur when VO(PP) = 2.3 V as
opposed to VO(PP) = 4 V at 5 V. The reduced voltage swing subsequently reduces maximum output power into
an 8-Ω load before distortion becomes significant.
Operation from 3.3-V supplies, as can be shown from the efficiency formula in equation 4, consumes
approximately two-thirds the supply power of operation from 5-V supplies for a given output-power level.
18
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
SLOS336C − DECEMBER 2000 − REVISED OCTOBER 2002
APPLICATION INFORMATION
headroom and thermal considerations
Linear power amplifiers dissipate a significant amount of heat in the package under normal operating conditions.
A typical music CD requires 12 dB to 15 dB of dynamic headroom to pass the loudest portions without distortion
as compared with the average power output. From the TPA751 data sheet, one can see that when the TPA751
is operating from a 5-V supply into an 8-Ω speaker that 700 mW peaks are available. Converting watts to dB:
P
dB
+ 10 Log
P
P
W + 10Log 700 mW + –1.5 dB
1W
ref
Subtracting the headroom restriction to obtain the average listening level without distortion yields:
−1.5 dB − 15 dB = −16.5 (15 dB headroom)
−1.5 dB − 12 dB = −13.5 (12 dB headroom)
−1.5 dB − 9 dB = −10.5 (9 dB headroom)
−1.5 dB − 6 dB = −7.5 (6 dB headroom)
−1.5 dB − 3 dB = −4.5 (3 dB headroom)
Converting dB back into watts:
P W + 10 PdBń10 x P
ref
= 22 mW (15 dB headroom)
= 44 mW (12 dB headroom)
= 88 mW (9 dB headroom)
= 175 mW (6 dB headroom)
= 350 mW (3 dB headroom)
This is valuable information to consider when attempting to estimate the heat dissipation requirements for the
amplifier system. Comparing the absolute worst case, which is 700 mW of continuous power output with 0 dB
of headroom, against 12 dB and 15 dB applications drastically affects maximum ambient temperature ratings
for the system. Using the power dissipation curves for a 5-V, 8-Ω system, the internal dissipation in the TPA751
and maximum ambient temperatures is shown in Table 1.
Table 1. TPA751 Power Rating, 5-V, 8-Ω, BTL
D PACKAGE
(SOIC)
DGN PACKAGE
(MSOP)
GQS PACKAGE
(MicroStar Junior)
MAXIMUM AMBIENT
TEMPERATURE
MAXIMUM AMBIENT
TEMPERATURE
MAXIMUM AMBIENT
TEMPERATURE
675
34°C
110°C
99°C
595
47°C
115°C
105°C
176 mW (6 dB)
475
68°C
122°C
114°C
700
88 mW (9 dB)
350
89°C
125°C
123°C
700
44 mW (12 dB)
225
111°C
125°C
125°C
PEAK OUTPUT
POWER
(mW)
AVERAGE
OUTPUT POWER
POWER
DISSIPATION
(mW)
700
700 mW
700
350 mW (3 dB)
700
Table 1 shows that the TPA751 can be used to its full 700-mW rating without any heat sinking in still air up to
110°C, 34°C, and 99°C for the DGN package (MSOP), D package (SOIC), and GQS (MicroStar Junior)
package, respectively.
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
19
PACKAGE OPTION ADDENDUM
www.ti.com
10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
TPA751D
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
TPA751
TPA751DGN
ACTIVE
HVSSOP
DGN
8
80
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
ATC
TPA751DGNR
ACTIVE
HVSSOP
DGN
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
ATC
TPA751DR
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
TPA751
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
工商网监
湘ICP备2023018690号
