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TPS65270
SLVSAX7E – AUGUST 2011 – REVISED AUGUST 2016
TPS65270 4.5-V to 18-V Input Voltage, 2-A or 3-A Output Current, Dual Synchronous StepDown Regulator With Integrated MOSFET
1 Features
3 Description
•
•
•
The TPS65270 is a monolithic, dual synchronous
buck regulator with a wide operating input voltage
that can operate in 5-V, 9-V, 12-V, or 15-V bus
voltages and battery chemistries. The converters are
designed to simplify its application while giving the
designer the option to optimize their usage according
to the target application.
1
•
•
•
•
•
•
•
•
Wide Input Supply Voltage Range: 4.5 V to 18 V
0.8 V, ±1% Accuracy Reference
Up to 2-A (Buck 1) and 3-A (Buck 2) Maximum
Continuous Output Loading Current
Low-Power Pulse Skipping Mode to Achieve High
Light Load Efficiency
Adjustable Switching Frequency
300 kHz to 1.4 MHz Set by External Resistor
Startup With a Prebiased Output Voltage
Dedicated Enable and Soft Start for Each Buck
Peak Current-Mode Control With Simple
Compensation Circuit
Cycle-by-Cycle Overcurrent Protection
180° Out-of-Phase Operation to Reduce Input
Capacitance and Power Supply Induced Noise
Available in 24-Lead Thermally Enhanced
HTSSOP (PWP) and VQFN 4-mm × 4-mm (RGE)
Packages
2 Applications
•
•
•
•
•
•
•
DTV
DSL Modems
Cable Modems
Set-Top Boxes
Car DVD Players
Home Gateway and Access Point Networks
Wireless Routers
The TPS65270 features a precision 0.8-V reference
and can produce output voltages up to 15 V. Each
converter features an enable pin that allows
dedicated control of each channel that provides
flexibility for power sequencing. Soft-start time in
each channel can be adjusted by choosing different
external capacitors. TPS65270 is also able to start up
with a prebiased output. The converter begins
switching when output voltage reaches the prebiased
voltage.
Constant frequency peak current-mode control
simplifies the compensation and provides fast
transient response. Cycle-by-cycle overcurrent
protection and hiccup mode operation limits MOSFET
power dissipation in short-circuit or overloading fault
conditions. Low-side reverse current protection also
prevents excessive sinking current from damaging
the converter.
Device Information(1)
PART NUMBER
TPS65270
1
C2
10nF
FB1
VIN1
SS1
LX1
4
R3
383K
R2
10K
C6
10nF
80
VIN
C10
22uF
R4a
40.2K
C11
82pF
20
LOW_P
GND
VCC
GND
6
L1
4.7uH
19
TPS65270
18
AGND
GND
ROSC
GND
COMP2
LX2
SS2
LX2
8
17
L2
4.7uH
FB2
60
50
40
30
16
1.2V/3A
C14
22uF
15
10
R5a
40.2K
C15
82pF
11
R5b
80.6K
14
FB2
VIN2
12
EN2
BST2
VIN = 4.5 V
VIN = 12 V
VIN = 15 V
FB2
C13
10uF
25V
13
20
10
VIN
ENABLE Buck 2
70
1.8V/2A
LX1
9
C5
2.2nF
FB1
21
7
C16
100pF
(optional)
R4b
32.4K
22
5
C4
10uF
90
23
COMP1
C17
100pF
(optional)
4.00 mm × 4.00 mm
Efficiency vs Output Load
C9
10uF
25V
BST1
3
R1
10K
VQFN (24)
100
C8
47nF
Efficiency
FB1
C3
2.2nF
24
EN1
2
BODY SIZE (NOM)
7.80 mm × 4.40 mm
(1) For all available packages, see the orderable addendum at
the end of the data sheet.
Typical Application
ENABLE Buck 1
PACKAGE
HTSSOP (24)
C12
47nF
0
0
0.5
1
1.5
2
Output Current (A)
2.5
3
Copyright © 2016, Texas Instruments Incorporated
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
TPS65270
SLVSAX7E – AUGUST 2011 – REVISED AUGUST 2016
www.ti.com
Table of Contents
1
2
3
4
5
6
7
8
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Description (continued).........................................
Pin Configuration and Functions .........................
Specifications.........................................................
1
1
1
2
3
3
4
7.1
7.2
7.3
7.4
7.5
7.6
4
4
4
5
5
7
Absolute Maximum Ratings ......................................
ESD Ratings..............................................................
Recommended Operating Conditions.......................
Thermal Information ..................................................
Electrical Characteristics...........................................
Typical Characteristics ..............................................
Detailed Description .............................................. 8
8.1 Overview ................................................................... 8
8.2 Functional Block Diagram ......................................... 9
8.3 Feature Description................................................... 9
8.4 Device Functional Modes........................................ 11
9
Application and Implementation ........................ 13
9.1 Application Information............................................ 13
9.2 Typical Application .................................................. 13
10 Power Supply Recommendations ..................... 18
11 Layout................................................................... 18
11.1 Layout Guidelines ................................................. 18
11.2 Layout Example .................................................... 19
11.3 Power Dissipation ................................................. 19
12 Device and Documentation Support ................. 21
12.1
12.2
12.3
12.4
12.5
12.6
Documentation Support ........................................
Receiving Notification of Documentation Updates
Community Resources..........................................
Trademarks ...........................................................
Electrostatic Discharge Caution ............................
Glossary ................................................................
21
21
21
21
21
21
13 Mechanical, Packaging, and Orderable
Information ........................................................... 21
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision D (April 2013) to Revision E
Page
•
Added ESD Ratings table, Feature Description section, Device Functional Modes, Application and Implementation
section, Power Supply Recommendations section, Layout section, Device and Documentation Support section, and
Mechanical, Packaging, and Orderable Information section. ................................................................................................. 1
•
Deleted Ordering Information table; see POA at the end of the data sheet........................................................................... 1
•
Changed maximum value of Voltage on VIN1, VIN2, LX1, LX2 from 18 to 20...................................................................... 4
•
Changed maximum value of Voltage at LX1, LX2 (maximum withstand voltage transient < 10 ns) from 18 to 23 ............... 4
•
Changed maximum value of Operating virtual junction temperature, TJ from 125 to 150 ..................................................... 4
•
Changed Ambient temperature (TA) to Junction temperature (TJ) in Recommended Operating Conditions table and
maximum value from 85 to 125 .............................................................................................................................................. 4
•
Updated values in the Thermal Information table to align with JEDEC standards................................................................. 5
•
Changed EN1 and EN2 pin threshold (falling) typical value to maximum value in Electrical Characteristics table............... 5
•
Changed PSM low power mode threshold (falling) typical value to maximum value in Electrical Characteristics table........ 5
2
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SLVSAX7E – AUGUST 2011 – REVISED AUGUST 2016
5 Description (continued)
The switching frequency of the converters can be set from 300 KHz to 1.4 MHz with an external resistor. Two
converters have 180° out-of-phase clock signals to minimize the input filter requirements and alleviate EMI and
input capacitor requirements.
TPS65270 also features a light-load pulse skipping mode (PSM). The PSM mode allows a power loss reduction
on the input power supplied to the system at light loading in order to achieve light-load high efficiency.
The TPS65270 is available in a 24-pin, thermally enhanced HTSSOP (PWP) package and 24-pin VQFN (RGE)
package.
6 Pin Configuration and Functions
PWP Package
24-Pin HTSSOP
Top View
LX1
COMP1
4
21
LX1
LOW_P
5
20
GND
VCC
6
19
GND
18
COMP1
22
19
3
LOW_P
SS1
20
VIN1
VCC
23
21
2
AGND
FB1
22
BST1
ROSC
24
23
1
COMP2
EN1
24
RGE Package
24-Pin VQFN
Top View
SS2
1
18
SS1
FB2
2
17
FB1
GND
EN2
3
16
EN1
15
BST1
Thermal
AGND
7
ROSC
8
17
GND
BST2
4
COMP2
9
16
LX2
VIN2
5
14
VIN1
SS2
10
15
LX2
LX2
6
13
LX1
FB2
11
14
VIN2
EN2
12
13
BST2
8
9
10
11
12
GND
GND
GND
GND
LX1
Pad
7
Not to scale
Thermal
LX2
Pad
Not to scale
Pin Functions
PIN
I/O
DESCRIPTION
NAME
HTSSOP
VQFN
AGND
7
22
BST1
24
15
O
Bootstrapped power supply to high side floating gate driver in Buck 1. Connect a 47-nF ceramic
capacitor from this pin to the switching node pin LX1.
BST2
13
4
O
Bootstrapped power supply to high side floating gate driver in Buck 2. Connect a 47-nF ceramic
capacitor from this pin to the switching node pin LX2.
COMP1
4
19
O
Loop compensation pin for Buck 1. Connect a series RC circuit to this pin to compensate the
control loop of this converter.
COMP2
9
24
O
Loop compensation pin for Buck 2. Connect a series RC circuit to this pin to compensate the
control loop of this converter.
EN1
1
16
I
Enable for Buck 1. Logic high enables the Buck 1; Logic low disables Buck 1. If pin is left open a
weak internal pullup to internal V5V allows for automatic enable; For a delayed start-up add a
small ceramic capacitor from this pin to ground.
EN2
12
3
I
Enable for Buck 2. Logic high enables the Buck 2. Logic low disables Buck 2. If pin is left open a
weak internal pullup to internal V5V allows for automatic enable; For a delayed start-up add a
small ceramic capacitor from this pin to ground.
FB1
2
17
I
Feedback voltage for Buck 1. Connect a resistor divider to set 0.8 V from the output of the
converter to ground.
Power Analog ground. Connect all GND pins and power pad together.
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Pin Functions (continued)
PIN
NAME
I/O
DESCRIPTION
HTSSOP
VQFN
FB2
11
2
GND
17, 18,
19, 20
8, 9, 10,
11
5
20
I
Low power operation mode. With active high, Buck 1 and Buck 2 operate at pulse skipping mode
at light load; active low forces both Buck 1 and Buck 2 to PWM mode; this pin cannot be left
open.
LX1
21, 22
12, 13
O
Switching node connecting to inductor for Buck 1.
LX2
15, 16
6, 7
O
Switching node connecting to inductor for Buck 2.
ROSC
8
23
O
Oscillator frequency setup. Connect a resistor to ground to set the frequency of internal oscillator
clock.
SS1
3
18
O
Soft start input for Buck 1. An internal 5-µA charging current is sourcing to this pin. Connect a
small ceramic capacitor to this pin to set the Buck 1 soft-start time.
SS2
10
1
O
Soft start input for Buck 2. An internal 5-µA charging current is sourcing to this pin. Connect a
small ceramic capacitor to this pin to set the Buck 1 soft-start time.
VCC
6
21
O
Internal 6.5-V power supply bias. Connect a 10-µF ceramic capacitor from this pin to ground.
VIN1
23
14
Power Input supply for Buck 1. Conne ct a 10-µF ceramic capacitor close to this pin.
VIN2
14
5
Power Input supply for Buck 2. Connect a 10-µF ceramic capacitor close to this pin.
Thermal
Pad
—
—
LOW_P
Feedback voltage for Buck 2. Connect a resistor divider to set 0.8 V from the output of the
converter to ground.
I
Power Power ground for Buck 1 and Buck 2.
Must be soldered to PCB for optimal thermal performance. Have thermal vias on the PCB to
enhance power dissipation.
—
7 Specifications
7.1 Absolute Maximum Ratings
over operating free-air temperature range (unless otherwise noted) (1)
MIN
MAX
UNIT
–0.3
20
V
–1
23
V
Voltage at BST1, BST2, referenced to LX1, LX2 pin
–0.3
7
V
Voltage at VCC, EN1, EN2, COMP1, COMP2, LOW_P
–0.3
7
V
Voltage at SS1, SS2, FB1, FB2, ROSC
–0.3
3.6
V
Voltage at AGND, GND
–0.3
0.3
V
Operating virtual junction temperature, TJ
–40
150
°C
Storage temperature, Tstg
–55
150
°C
Voltage at VIN1, VIN2, LX1, LX2
Voltage at LX1, LX2 (maximum withstand voltage transient < 10 ns)
(1)
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended
Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
7.2 ESD Ratings
VALUE
V(ESD)
(1)
(2)
Electrostatic discharge
Human-body model (HBM), per ANSI/ESDA/JEDEC JS-001 (1)
±2000
Charged-device model (CDM), per JEDEC specification JESD22-C101 (2)
±500
UNIT
V
JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process.
7.3 Recommended Operating Conditions
over operating free-air temperature range (unless otherwise noted)
MIN
MAX
UNIT
VIN
Input operating voltage
4.5
18
V
TJ
Junction temperature
–40
125
°C
4
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7.4 Thermal Information
TPS65270
THERMAL METRIC (1) (2)
PWP (HTSSOP)
RGE (VQFN)
24 PINS
24 PINS
UNIT
RθJA
Junction-to-ambient thermal resistance
41.3
33.8
°C/W
RθJC(top)
Junction-to-case (top) thermal resistance
23.9
36.2
°C/W
RθJB
Junction-to-board thermal resistance
22.6
12.1
°C/W
ψJT
Junction-to-top characterization parameter
0.8
0.4
°C/W
ψJB
Junction-to-board characterization parameter
22.4
12.2
°C/W
RθJC(bot)
Junction-to-case (bottom) thermal resistance
4
2.5
°C/W
(1)
(2)
For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report.
See PowerPAD™ Thermally Enhanced Package.
7.5 Electrical Characteristics
TA = –40°C to 125°C, VIN = 12 V, fSW = 625 kHz (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
INPUT SUPPLY
VIN
Input Voltage
VIN1 and VIN2
IDDSDN
Shutdown
EN1 = EN2 = 0 V
4.5
18
V
IDDQ_nsw
Nonswitching quiescent power supply
current
VFB1 = VFB2 = 900 mV, LOW_P = high
UVLO
VIN undervoltage lockout
Hysteresis
0.35
VCC
Internal biasing supply
VCC load current = 0 A, VIN = 12 V
6.25
V
VCC_drop
VCCLDO Dropout Voltage
VIN = 5 V, VCC load current = 20 mA
180
mV
IVCC
VCC current limit
4.5 V < VIN < 18 V
200
mA
10
µA
1
mA
Rising VIN
4
4.2
4.45
Falling VIN
3.65
3.85
4.1
V
FEEDBACK AND ERROR AMPLIFIER
VFB
Regulated feedback voltage
VLINEREG
Line regulation: DC
VIN = 12 V , VCOMP = 1.2 V, TJ = 25°C
–1%
0.8
1%
VIN = 12 V, VCOMP = 1.2 V, TJ = –40°C to 125°C
–2%
0.8
2%
V
VIN = 4.5 V to 18 V, IOUT = 1 A
0.5
%/V
VLOADREG Load regulation: DC
IOUT = 10% to 90%, IOUT,MAX
0.4
%/A
Gm_EA
Error amplifier transconductance
–2 µA < ICOMP < 2 µA
130
µs
Gm_SRC
COMP voltage to inductor current Gm
ILX = 0.5 A
10
A/V
ENABLE, PFM MODE AND SOFT START
VEN
EN1 and EN2 pin threshold
VPSM
PSM low power mode threshold
ISS
SS1 and SS2 soft-start charging current
Rising
1.55
Falling
Rising
0.4
1.55
Falling
0.4
5
V
V
µA
OSCILLATOR
FSW_BK
FSW
Switching frequency
Programmable frequency
Set by external resistor ROSC
1.4
MHz
ROSC = 250 kΩ
0.85
0.3
1
1.15
MHz
ROSC = 500 kΩ
425
500
575
kHz
PROTECTION
ILIMIT1
Buck 1 peak inductor current limit
4.5 V < VIN < 18 V
3.2
A
ILIMIT1_LS1 Buck 1 low side MOSFET current limit
4.5 V < VIN < 18 V
2
A
ILIMIT2
4.5 V < VIN < 18 V
4.1
A
4.5 V < VIN < 18 V
2
A
Buck 2 peak inductor current limit
ILIMIT1_LS2 Buck 2 low side MOSFET current limit
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TPS65270
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Electrical Characteristics (continued)
TA = –40°C to 125°C, VIN = 12 V, fSW = 625 kHz (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
MOSFET ON-RESISTANCES
Rdson_HS1 On resistance of high side FET on CH1
BST1 to LX1 = 6.25 V
120
mΩ
Rdson_LS1
VIN = 12 V
80
mΩ
Rdson_HS2 On resistance of high side FET on CH2
BST2 to LX2 = 6.25 V
95
mΩ
Rdson_LS2
On resistance of low side FET on CH2
VIN = 12 V
50
Ton_min
Minimum in time
On resistance of low side FET on CH1
80
mΩ
120
ns
THERMAL SHUTDOWN
TTRIP
Thermal protection trip point
THYST
Thermal protection hysteresis
6
Rising temperature
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160
°C
20
°C
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7.6 Typical Characteristics
TA = 25°C, VIN = 12 V, fSW = 625 kHz (unless otherwise noted)
1.25
1.85
1.24
1.83
1.23
PWM, Vout1, Vin = 12 V
PWM, Vout2, Vin = 12 V
1.22
1.21
1.81
1.2
PSM, Vout2, Vin = 12 V
PSM, Vout2, Vin = 12 V
1.19
1.79
1.18
`
1.17
1.77
1.16
1.15
1.75
0
0.5
1
Buck 1 = 1.8 V
1.5
2
2.5
0
3
0.5
1
1.5
Buck 1 = 1.2 V
1% resistors
2
2.5
3
4
3.5
1% resistors
Figure 2. Load Regulation
Figure 1. Load Regulation
100
95
90
90
VI = 4.5 V
80
85
VI = 15 V
VI = 12 V
70
80
VI = 4.5 V
60
VI = 15 V
VI = 12 V
75
50
70
40
65
30
60
20
55
10
50
0
0
0.5
1
1.5
2
2.5
0
3
0.5
1
1.5
2
2.5
3
3.5
Buck 2 = 1 V
Buck 1 = 3.3 V
Figure 4. Efficiency
Figure 3. Efficiency
100
100
Vin = 12 V Vout2 = 5 V auto PSM-PWM
90
Vin = 12 V Vout1 = 3.3 V auto PSM-PWM
90
80
80
70
70
60
60
Forced PWM
50
50
Forced PWM
40
40
30
30
20
20
10
10
0
0
0
0.05
0.1
0.15
Buck 1 = 3.3 V
0.2
0
0.05
0.1
0.15
0.2
Buck 2 = 5 V
Figure 5. Efficiency
Figure 6. Efficiency
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8 Detailed Description
8.1 Overview
TPS65270 is a power management IC with two step-down buck converters. Both high-side and low-side
MOSFETs are integrated to provide fully synchronous conversion with higher efficiency. TPS65270 can support
4.5-V to 18-V input supply, 2-A continuous current for Buck 1 and 3 A for Buck 2. The buck converters have an
automatic PSM mode, which can improve power dissipation during light loads. Alternatively, the device
implements a constant frequency mode by connecting the LOW_P pin to ground. The wide switching frequency
of 300 kHz to 1.4 MHz allows for efficiency and size optimization. The switching frequency is adjustable by
selecting a resistor to ground on the ROSC pin. Input ripple is reduced by 180° out-of-phase operation between
Buck 1 and Buck 2.
Both buck converters have peak current mode control which simplifies the loop compensation. A traditional type
II compensation network can stabilize the system and achieve fast transient response. Moreover, an optional
capacitor in parallel with the upper resistor of the feedback divider provides one more zero and makes the
crossover frequency over 100 kHz. Each buck converter has an individual cycle-by-cycle current limit and low
side reverse current limit.
The device has a built-in LDO regulator. During a standby mode, the 6.5-V LDO can be used to drive MCU and
other active loads. with this LDO, system is able to turn off the two buck converters so as to reduce the power
consumption and improve the standby efficiency. Each converter has its own programmable soft start that can
reduce the input inrush current. The individual Enable pins for each independent control of each output voltage
and power sequence.
8
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8.2 Functional Block Diagram
LOW_P
5
VCC
6
ROSC
8
Pre-Regulator, Voltage Reference,
Current Bias
Logic
Internal
LDO
OSC
23
VIN1
24
BST1
VIN1
o
180
ILIM
SLP
CS
S
SS1
3
0.8V
FB1
EN 1 Logic
EN2 Logic
FB1
2
COMP1
4
EN1
1
Q
Latch
COMP
R
21,22
LX1
19, 20
GND
VOUT BUCK1
Q
EN
EA
FB1
BUCK 1
12
EN2
17,18
9
COMP2
FB2
GND
FB2
11
FB2
10
SS2
same as Buck 1
15,16
VOUT BUCK2
LX2
13
BUCK 2
AGND
14
BST2
VIN2
VIN2
6
Note: Pin numbers in block diagram are for HTSSOP (PWP) 24-pin package.
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8.3 Feature Description
8.3.1 Adjustable Switching Frequency
To select the internal switching frequency connect a resistor from ROSC to ground. Figure 7 shows the required
resistance for a given switching frequency.
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Feature Description (continued)
900
800
700
Resistance - W
600
500
400
300
200
100
0
0
0.5
1
1.5
2
fsw - Switching Frequency - MHz
2.5
3
Figure 7. ROSC vs Switching Frequency
ROSC (kW) = 239.13 ´ ƒSW -1.149
(1)
For operation at 800 kHz, a 300-kΩ resistor is required.
8.3.2 Out-of-Phase Operation
To reduce input ripple current, Buck 1 and Buck 2 operate 180° out-of-phase. This enables the system having
less input ripple, then to lower component cost, save board space and reduce EMI.
8.3.3 Delayed Start-Up
If a delayed start-up is required on any of the buck converters fit a ceramic capacitor to the ENx pins. The delay
added is approximately 0.75 ms per nF connected to the pin. The EN pins have a weak 1-MΩ pullup to the 5-V
rail.
8.3.4 Soft-Start Time
The device has an internal pullup current source of 5 µA that charges an external slow start capacitor to
implement a slow start time. Equation 2 shows how to select a slow start capacitor based on an expected slow
start time. The voltage reference (VREF) is 0.8 V and the slow start charge current (Iss) is 5 µA. The soft-start
circuit requires 1 nF per 160 µs to be connected at the SS pin. An 800-µs soft-start time is implemented for all
converters fitting 4.7 nF to the relevant pins.
tSS (ms)
§ C (nF) ·
VREF (V) u ¨ SS
¸
© ISS (µA) ¹
(2)
8.3.5 Adjusting the Output Voltage
The output voltage is set with a resistor divider from the output node to the FB pin. TI recommends using divider
resistors of 1% tolerance or better. To improve efficiency at light load, start with 40.2 kΩ for the R1 resistor and
use the Equation 3 to calculate R2.
R2
10
§ 0.8 V
·
R1 u ¨
¸
9¹
© 9O ±
(3)
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Feature Description (continued)
Vo
TPS65270
R1
FB
R2
0.8V
+
Figure 8. Voltage Divider Circuit
8.3.6 Error Amplifier
The device has a transconductance error amplifier. The transconductance of the error amplifier is 130 µA/V
during normal operation. The frequency compensation network is connected between the COMP pin and ground.
8.3.7 Slope Compensation
The device has a built-in slope compensation ramp. The slope compensation can prevent subharmonic
oscillations in peak current mode control when duty cycle becomes too large.
8.3.8 Overcurrent Protection
The current through the internal high-side MOSFET is sampled and scaled through an internal pilot device during
the hig time. The sampled current is compared to overcurrent limit. If the peak inductor current exceeds the
overcurrent limit reference level, an internal overcurrent fault counter is set to 1 and an internal flag is set. The
internal power MOSFET is immediately turned off and is not turned on again until the next switching cycle. The
protection circuitry continues to monitor the current and turns off the internal MOSFET as described. If the
overcurrent condition persists for four sequential clock cycles, the overcurrent fault counter overflows indicating
an overcurrent fault condition exists. The regulator is shut down and power good goes low. If the overcurrent
condition clears before the counter reaches four consecutive cycles, the internal flag and counter are reset. The
protection circuitry attempts to recover from the overcurrent condition after waiting four soft-start cycles. The
internal overcurrent flag and counter are reset. A normal soft-start cycle is attempted and normal operation
continues if the fault condition has cleared. If the overcurrent fault counter overflows during soft start, the
converter shuts down and this hiccup mode operation repeats.
8.3.9 Thermal Shutdown
The device implements an internal thermal shutdown to protect itself if the junction temperature exceeds 160°C.
The thermal shutdown forces the device to stop switching when the junction temperature exceeds thermal trip
threshold. Once the die temperature decreases below 140°C, the device reinitiates the power up sequence. The
thermal shutdown hysteresis is 20°C.
8.3.10 Low Power Mode Operation
By pulling the Low_P pin high all converters operate in pulse-skipping mode, greatly reducing the overall power
consumption at light and no load conditions. When LOW_P is tied to low, all converters run in forced PWM
mode.
8.4 Device Functional Modes
8.4.1 Operation With Minimum VIN (VIN < 4.45 V)
The device will operate with input voltages above the 4.45-V UVLO maximum voltage. The typical UVLO voltage
is 4.2 V and the device may operate at input voltage above this point. The device may also operate with lower
input voltages; the minimum UVLO voltage is 4 V (rising) and 3.65 V (falling). The device will not operate with
input voltages below the UVLO minimum voltage.
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Device Functional Modes (continued)
8.4.2 Operation With EN Control
The enable rising edge threshold voltage is 1.55 V (minimum) and falling edge threshold voltage is 0.4 V
(maximum). With EN held below 0.4 V the device is disabled and switching is inhibited. The IC quiescent current
is reduced in this state. The device becomes active when input voltage is above the UVLO threshold and the EN
voltage is increased above 1.55 V. Switching is enabled and the internal soft-start sequence is initiated as shown
in Figure 13.
12
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9 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
9.1 Application Information
The device is a dual-synchronous, step-down DC-DC converter. It is typically used to convert a higher DC
voltage to lower DC voltages with continuous available output current of 2 A or 3 A.
9.2 Typical Application
R3
383K
C5
2.2nF
C16
100pF
(optional)
R2
10K
2
3
ENABLE Buck 1
C12
47nF
4
19
COMP1
20
LOW_P
21
VCC
22
AGND
23
SS2
SS1
FB2
FB1
EN2
EN1
TPS65270
BST2
BST1
17
16
15
ENABLE Buck 2
C8
47nF
C9
VIN 10uF
14
13
LX1
L1
4.7uH
1.8V/2A
12
GND
11
10
9
7
1.2V/3A
GND
LX1
GND
LX2
GND
VIN1
LX2
VIN2
8
6
L2
4.7uH
C15
82pF
C2
10nF
18
25V
5
R5a
40.2K
ROSC
24
1
C16
100pF
(optional)
R1
10K
COMP2
C6
10nF
C13
10uF VIN
25V
C3
2.2nF
C4
10uF
C14
22uF
C10
22uF
R5b
80.6K
C11
82pF
R4a
40.2K
R4b
32.4K
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Figure 9. Typical Application Schematic
9.2.1 Design Requirements
For this design example, use the parameters listed in Table 1 as the input parameters.
Table 1. Design Parameters
PARAMETER
EXAMPLE VALUE
VOUT1
1.8 V
IOUT1
2A
VOUT2
1.2 V
IOUT2
3A
Transient response (1-A load step)
±5%
Input voltage
12 V (typical), 4.5 V to 18 V
Output voltage ripple
±1%
Switching frequency
625 kHz
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9.2.2 Detailed Design Procedure
9.2.2.1 Output Inductor Selection
To calculate the value of the output inductor, use Equation 4. LIR is a coefficient that represents the amount of
inductor ripple current relative to the maximum output current. The inductor ripple current is filtered by the output
capacitor. Therefore, choosing high inductor ripple currents impact the selection of the output capacitor since the
output capacitor must have a ripple current rating equal to or greater than the inductor ripple current. In general,
the inductor ripple value is at the discretion of the designer; however, LIR is normally from 0.1 to 0.3 for the
majority of applications.
VOUT
V
- VOUT
L = IN max
´
IO ´ LIR
VIN max ´ ƒSW
(4)
For the output filter inductor, it is important that the RMS current and saturation current ratings not be exceeded.
The RMS and peak inductor current can be found from Equation 6 and Equation 7.
VOUT
V
- VOUT
Iripple = IN max
´
L
VIN max ´ ƒSW
(5)
æ VOUT ´ (VIN max - VOUT ) ö
çç
÷÷
VIN max ´ L ´ ƒSW
2
è
ø
I Lrms = IO +
12
I ripple
I Lpeak = I OUT +
2
2
(6)
(7)
The current flowing through the inductor is the inductor ripple current plus the output current. During power up,
faults or transient load conditions, the inductor current can increase above the calculated peak inductor current
level calculated above. In transient conditions, the inductor current can increase up to the switch current limit of
the device. For this reason, the most conservative approach is to specify an inductor with a saturation current
rating equal to or greater than the switch current limit rather than the peak inductor current.
9.2.2.2 Output Capacitor Selection
There are three primary considerations for selecting the value of the output capacitor. The output capacitor
determines the modulator pole, the output voltage ripple, and how the regulator responds to a large change in
load current. The output capacitance needs to be selected based on the most stringent of these three criteria.
The desired response to a large change in the load current is the first criteria. The output capacitor needs to
supply the load with current when the regulator cannot. This situation would occur if there are desired hold-up
times for the regulator where the output capacitor must hold the output voltage above a certain level for a
specified amount of time after the input power is removed. The regulator is also temporarily not able to supply
sufficient output current if there is a large, fast increase in the current needs of the load such as a transition from
no load to full load. The regulator usually needs two or more clock cycles for the control loop to see the change
in load current and output voltage and adjust the duty cycle to react to the change. The output capacitor must be
sized to supply the extra current to the load until the control loop responds to the load change. The output
capacitance must be large enough to supply the difference in current for 2 clock cycles while only allowing a
tolerable amount of droop in the output voltage. Equation 8 shows the minimum output capacitance necessary to
accomplish this.
2 ´ DI OUT
CO =
ƒSW ´ DVOUT
where
•
•
•
ΔIout is the change in output current
ƒsw is the regulators switching frequency
ΔVout is the allowable change in the output voltage.
(8)
Equation 9 calculates the minimum output capacitance needed to meet the output voltage ripple specification.
14
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CO >
1
1
´
V
8 ´ ƒSW
Oripple
IOripple
where
•
•
•
ƒsw is the switching frequency.
Vripple is the maximum allowable output voltage ripple.
Iripple is the inductor ripple current.
(9)
Equation 10 calculates the maximum ESR an output capacitor can have to meet the output voltage ripple
specification.
VOripple
Re sr <
IOripple
(10)
Additional capacitance deratings for aging, temperature and DC bias should be factored in which increases this
minimum value. Capacitors generally have limits to the amount of ripple current they can handle without failing or
producing excess heat. An output capacitor that can support the inductor ripple current must be specified. Some
capacitor data sheets specify the root mean square (RMS) value of the maximum ripple current. Equation 11 can
be used to calculate the RMS ripple current the output capacitor needs to support.
I Lrms =
VOUT ´ (VIN
12 ´ VIN
max
max
- VOUT )
´ L ´ ƒSW
(11)
9.2.2.3 Input Capacitor Selection
The TPS65265 requires a high quality ceramic, type X5R or X7R, input decoupling capacitor of at least 10 µF of
effective capacitance on the PVIN input voltage pins. These capacitors must be connected as close as physically
possible to the input pins of the converters. In some applications additional bulk capacitance may also be
required for the PVIN input. The effective capacitance includes any DC bias effects. The voltage rating of the
input capacitor must be greater than the maximum input voltage. The capacitor must also have a ripple current
rating greater than the maximum input current ripple of The TPS65265. The input ripple current can be calculated
using Equation 12.
IINrms = I OUT ´
´ VOUT )
(V
VOUT
´ IN min
VIN min
VIN min
(12)
The value of a ceramic capacitor varies significantly over temperature and the amount of DC bias applied to the
capacitor. The capacitance variations due to temperature can be minimized by selecting a dielectric material that
is stable over temperature. X5R and X7R ceramic dielectrics are usually selected for power regulator capacitors
because they have a high capacitance to volume ratio and are fairly stable over temperature. The output
capacitor must also be selected with the DC bias taken into account. The capacitance value of a capacitor
decreases as the DC bias across a capacitor increases. The input capacitance value determines the input ripple
voltage of the regulator. The input voltage ripple can be calculated using Equation 13.
I OUT max ´ 0.25
DVIN =
CIN ´ ƒSW
(13)
9.2.2.4 Bootstrap Capacitor Selection
The device has two integrated boot regulators and requires a small ceramic capacitor between BST and LX pins
to provide the gate drive voltage for the high side MOSFET. TI recommends a ceramic capacitor of 0.047 µF. A
ceramic capacitor with an X7R or X5R grade dielectric is desired because of the stable characteristics over
temperature and voltage.
9.2.2.5 Loop Compensation
TPS65270 is a current mode control DC-DC converter. The error amplifier has 130-µA/V transconductance.
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Vo
iL
Co
RL
Resr
gmps=10A/V
R1
Current Sense
I/V Gain
Cff
FBx
Rc
Vref=0.8V
Cc
CRoll
R2
gm=130µA/V
Figure 10. Loop Compensation
A typical compensation circuit could be type II (RC and CC) to have a phase margin from 60 to 90 degrees, or
type III (RC, CC and Cff) to improve the converter transient response. CRoll adds a high frequency pole to
attenuate high-frequency noise when required. It may also prevent noise coupling from other rails if there is
possibility of cross coupling between rails when layout is very compact.
To calculate the external compensation components follow the following steps:
TYPE II CIRCUIT
TYPE III CIRCUIT
—
Use type III circuit for switching
frequencies higher than 500 kHz.
Select switching frequency that is appropriate for application
depending on L, C sizes, output ripple, EMI concerns and etc.
Switching frequencies from 500 kHz to 1 MHz give best trade off
between performance and cost. When using smaller L and Cs,
switching frequency can be increased. To optimize efficiency,
switching frequency can be lowered.
Select cross over frequency (fC) to be less than 1/5 to 1/10 of
switching frequency.
Suggested fC = fS / 10
RC =
Set and calculate RC.
2p ´ ƒc ´ VO ´ CO
gM ´ Vref ´ gmps
Calculate CC by placing a compensation zero at or before the
converter dominant pole
¦S
Cc =
1
CO u RL u 2S
Add CRoll if required to remove large signal coupling to high
impedance COMP node. Make sure that
¦SRoll
1
2 u S u RC u CRoll
CRoll
Suggested fC = fS / 10
RC =
RL ´ Co
Rc
Resr u CO
RC
2p ´ ƒc ´ CO
gM ´ gmps
Cc =
CRoll
RL ´ Co
Rc
Resr u CO
RC
is at least twice the cross over frequency.
Calculate Cff compensation zero at low frequency to boost the phase
margin at the crossover frequency. Make sure that the zero
frequency (fzff is smaller than soft start equivalent frequency (1 /
Tss).
16
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—
C¦¦
1
u S u ¦] ¦¦ u 5
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9.2.3 Application Curves
TA = 25°C, VIN = 12 V, fSW = 625 kHz (unless otherwise noted).
IO1 = 0 A
IO2 = 0 A
IO1 = 2 A
Figure 11. Buck 1 and Buck 2 in Steady State
VO1 = 1.8 V
VO2 = 1.2 V
Figure 12. Buck 1 and Buck 2 in Steady State
VO1 = 3.3 V
Figure 13. Start-Up With EN
VO2 = 1 V
IO2 = 3 A
IO1 = 1 A to 2 A
Figure 14. Buck 1 Load Transient
IO1 = 1 A to 2 A
Figure 15. Buck 2 Load Transient
Figure 16. Buck 1 and Buck 2 in PSM Mode
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TA = 25°C, VIN = 12 V, fSW = 625 kHz (unless otherwise noted).
Figure 17. Buck 2 Hard Short and Recover
10 Power Supply Recommendations
The device is designed to operate with an input voltage supply from 4.5 V to 18 V. This input power supply must
be well regulated. If the input supply is placed more than a few inches from the TPS65270 converter, bulk
capacitance may be required in addition to the ceramic bypass capacitors. An electrolytic capacitor with a value
of 47 µF is a typical choice.
11 Layout
11.1 Layout Guidelines
Layout is a critical portion of PMIC designs.
• Place VOUT, and LX on the top layer and an inner power plane for VIN.
• Fit also on the top layer connections for the remaining pins of the PMIC and a large top side area filled with
ground.
• The top layer ground area sould be connected to the bottom ground layer(s) using vias at the input bypass
capacitor, the output filter cpacitor and directly under the TPS65270 device to provide a thermal path from the
Powerpad land to ground.
• The AGND pin must be tied directly to the power pad under the IC and the power pad.
• For operation at full rated load, the top side ground area together with the bottom ground plane, must provide
adequate heat dissipating area.
• There are several signals paths that conduct fast changing currents or voltages that can interact with stray
inductance or parasitic capacitance to generate noise or degrade the power supplies performance. To help
eliminate these problems, the VIN pin must be bypassed to ground with a low ESR ceramic bypass capacitor
with X5R or X7R dielectric. Minimize the loop area formed by the bypass capacitor connections, the VIN pins,
and the ground connections. Because the LX connection is the switching node, the output inductor must be
placed close to the LX pins, and the area of the PCB conductor minimized to prevent excessive capacitive
coupling.
• The output filter capacitor ground must use the same power ground trace as the VIN input bypass capacitor.
Try to minimize this conductor length while maintaining adequate width.
• The compensation must be as close as possible to the COMP pins. The COMP and OSC pins are sensitive
to noise so the components associated to these pins must be placed as close as possible to the IC and
routed with minimal lengths of trace.
18
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COMP1
LOW_P
VCC
AGND
EN1
BST2
BST1
VIN2
VIN1
LX2
LX1
VIN1
LX1
EN2
GND
FB1
GND
FB2
GND
SS1
GND
SS2
LX2
VIN2
ROSC
COMP2
11.2 Layout Example
VOUT2
VOUT1
0.010 in. Diameter
Thermal VIA to Ground Plane
VIA to Ground Plane
Figure 18. Example Layout for the TPS65270
11.3 Power Dissipation
The total power dissipation inside TPS65270 must not exceed the maximum allowable junction temperature of
125°C to maintain reliable operation. The maximum allowable power dissipation is a function of the thermal
resistance of the package (RθJA) and ambient temperature.
To
1.
2.
3.
calculate the temperature inside the device under continuous loading use the following procedure.
Define the set voltage for each converter.
Define the continuous loading on each converter. Make sure do not exceed the converter maximum loading.
Determine from the graphs below the expected losses in watts per converter inside the device. The losses
depend on the input supply, the selected switching frequency, the output voltage and the converter chosen.
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Power Dissipation (continued)
4. To calculate the maximum temperature inside the IC use the following formula:
THOT _ SPOT = TA + PDIS ´ R qJA
where
•
•
PDIS is the sum of losses in all converters
RθJA is the junction to ambient thermal impedance of the device and it is heavily dependant on board layout
(14)
1.7
1.6
1.5
1.4
1.3
1.2
1.1
1.0
0.9
0.8
0.7
0.6
0.5
0.4
0.3
0.2
1.7
1.6
1.5
1.4
1.3
1.2
1.1
1.0
0.9
0.8
0.7
0.6
0.5
0.4
0.3
0.2
1
1.2
1.4
1.6
1.8
2
2.2
2.4
2.6
2.8
3
1
1.2
1.4
1.6
1.8
I (A)
2.2
2.4
2.6
2.8
3
I (A)
VIN = 12 V
fSW = 500 kHz
VOUT (from top to bottom) = 5 V, 3.3 V, 2.5 V, 1.8 V, 1.2 V
VIN = 12 V
fSW = 1.1 MHz
VOUT (from top to bottom) = 5 V, 3.3 V, 2.5 V, 1.8 V, 1.2 V
Figure 19. Buck 1
Figure 20. Buck 1
1.6
1.6
1.4
1.4
1.2
1.2
1.0
1.0
0.8
0.8
0.6
0.6
0.4
0.4
0.2
0.2
1
1.2
1.4
1.6
1.8
2
2.2
2.4
2.6
2.8
3
3.2
3.4
3.6
1
1.2
1.4
1.6
1.8
2
2.2
2.4
2.6
2.8
3
3.2
3.4
3.6
I (A)
I (A)
VIN = 12 V
fSW = 500 kHz
VOUT (from top to bottom) = 5 V, 3.3 V, 2.5 V, 1.8 V, 1.2 V
VIN = 12 V
fSW = 1.1 MHz
VOUT (from top to bottom) = 5 V, 3.3 V, 2.5 V, 1.8 V, 1.2 V
Figure 22. Buck 2
Figure 21. Buck 2
20
2
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12 Device and Documentation Support
12.1 Documentation Support
12.1.1 Related Documentation
For related documentation see the following:
PowerPAD™ Thermally Enhanced Package (SLMA002)
12.2 Receiving Notification of Documentation Updates
To receive notification of documentation updates, navigate to the device product folder on ti.com. In the upper
right corner, click on Alert me to register and receive a weekly digest of any product information that has
changed. For change details, review the revision history included in any revised document.
12.3 Community Resources
The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective
contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of
Use.
TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration
among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help
solve problems with fellow engineers.
Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and
contact information for technical support.
12.4 Trademarks
PowerPAD, E2E are trademarks of Texas Instruments.
All other trademarks are the property of their respective owners.
12.5 Electrostatic Discharge Caution
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more
susceptible to damage because very small parametric changes could cause the device not to meet its published specifications.
12.6 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
13 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
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PACKAGE OPTION ADDENDUM
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10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
TPS65270PWPR
ACTIVE
HTSSOP
PWP
24
2000
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
TPS65270
TPS65270RGER
ACTIVE
VQFN
RGE
24
3000
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
TPS
65270
TPS65270RGET
ACTIVE
VQFN
RGE
24
250
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
TPS
65270
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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