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MAX25612/MAX25612B
General Description
The MAX25612/MAX25612B are single-channel highbrightness LED (HB LED) drivers for automotive frontlight applications such as high beam, low beam, daytime
running light (DRL), turn indicator, fog light and other LED
lights. They can take an input voltage from 5V to 48V and
can drive a string of LEDs with a maximum output voltage
of 60V. The MAX25612/MAX25612B are fully synchronous
and are suitable for boost, buck-boost, SEPIC, and highside buck applications that need synchronous rectification
providing efficiencies greater than 90%.
The MAX25612/MAX25612B sense output current at
the high side of the LED string. High-side current
sensing is required to protect against shorts from the
output to the ground or battery input. It is also the most
flexible scheme for driving LEDs, allowing boost, highside buck, or buck-boost mode configurations. The
PWM input provides LED dimming ratios of up to
5000:1, and the ICTRL input provides additional analog
dimming capability in the MAX25612/MAX25612B.
The MAX25612/MAX25612B also include a FLT flag
that indicates open string, shorted string, and thermal
shutdown. The MAX25612/MAX25612B have built-in
spread-spectrum modulation for improved electromagnetic
compatibility
performance. The MAX25612/
MAX25612B are available in a space-saving (4mm x
4mm), 20-pin side-wettable TQFN or a 20-pin TSSOP
package and are specified to operate over the -40°C to
+125°C automotive temperature range.
Applications
●● Automotive Exterior Lighting:
• High-Beam/Low-Beam/Signal/Position Lights
• Daytime Running Lights (DRLs)
• Fog Light and Adaptive Front-Light Assemblies
Automotive Synchronous
High Voltage LED Controller
●● Simple to Optimize for Efficiency, Board Space, and
Input Operating Range
• Synchronous MOSFET Driver Improves Efficiency
by up to 5% for High-Current Boost, Buck-Boost,
SEPIC, and High-Side Buck Applications
• Programmable Switching Frequency (200kHz to
2.2MHz)
• 20-Pin TSSOP Package with Exposed Pad
and Thermally Enhanced 4mm x 4mm, 20-Pin
Side-Wettable TQFN Packages
●● Protection Features Increase System Reliability
• Short Circuit, Overvoltage and Thermal Protection
• Fault Diagnosis through Fault Flag
●● Automotive Ready
• -40ºC to +125ºC Operating Temperature Range
• AEC-Q100 Qualified
Simplified Typical Operating Circuit
L1
VIN
ROVP1
CBST
IN
UVEN
VCC
BST
DH
LX
MAX25612/
MAX25612B
P1
COUT
N1
DL
CSP
RCS_LED
N2
RSC
ROVP2
RCS_FET
CSN
PGND
OVP
ISENSE+
FLT
ISENSEPWMDIM
DIMOUT
ICTRL
COMP
RT
SGND EP
RCOMP
CCOMP_HF
CCOMP
●● Commercial, Industrial, and Architectural Lighting
Benefits and Features
●● Integration Minimizes BOM and Cost
• +5.0V to +48V Wide Input Voltage Range with a
Maximum +65V Boost Output
• Integrated pMOS Dimming FET Driver
• ICTRL Input for Analog Dimming
• Integrated High-Side Current-Sense Amplifier
• 200Hz Ramp Generator Simplifies PWM Dimming
19-100543; Rev 4; 1/20
Ordering Information appears at end of data sheet.
MAX25612/MAX25612B
Automotive Synchronous
High Voltage LED Controller
Absolute Maximum Ratings
IN, UVEN to PGND................................................-0.3V to +52V
ISENSE+, ISENSE-, DIMOUT to PGND................-0.3V to +65V
ISENSE- to ISENSE+............................................-0.6V to +0.3V
BST, DH to PGND..................................................-0.3V to +70V
LX to PGND...........................................................-0.3V to +65V
BST to LX.................................................................-0.3V to +6V
DH to LX........................................................ -0.3V to VCC+0.3V
DL to PGND.................................................. -0.3V to VCC+0.3V
CSP, CSN to SGND...................................... -0.3V to VCC+0.3V
CSP-CSN..............................................................-0.3V to +0.3V
COMP, RT to SGND........................................-0.3V to Vcc+0.3V
VCC to SGND...........................................................-0.3V to +6V
SGND to PGND.....................................................-0.3V to +0.3V
OVP, FLT, ICTRL, PWMDIM to SGND.....................-0.3V to +6V
Continuous Current on IN.................................................100mA
Continuous Current on DL.................................................+50mA
Short Circuit Duration on VCC....................................Continuous
Continuous Power Dissipation (20-Pin TSSOP) (TA =
+70°C, derate 26mW/°C above +70°C.)....................2122mW
Continuous Power Dissipation (20-Pin TQFN SW) (TA =
+70°C, derate 25.6mW/°C above +70°C.).................2050mW
Operating Temperature Range.......................... -40°C to +125°C
Junction Temperature.......................................................+150°C
Storage Temperature Range............................. -65°C to +150°C
Lead Temperature (Soldering, 10s).................................. +300ºC
Soldering Temperature (Reflow).......................................+260°C
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these
or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect
device reliability.
Package Information
20-TSSOP
PACKAGE CODE
U20E+3C
Outline Number
21-100132
Land Pattern Number
90-100049
Thermal Resistance, Single-Layer Board:
Junction to Ambient (θJA)
46ºC/W
Junction to Case (θJC)
2ºC/W
Thermal Resistance, Four-Layer Board:
Junction to Ambient (θJA)
37ºC/W
Junction to Case (θJC)
2ºC/W
20-TQFN SW
PACKAGE CODE
T2044Y+3C
Outline Number
21-100068
Land Pattern Number
90-0037
Thermal Resistance, Single-Layer Board:
Junction to Ambient (θJA)
59ºC/W
Junction to Case (θJC)
6ºC/W
Thermal Resistance, Four-Layer Board:
Junction to Ambient (θJA)
39ºC/W
Junction to Case (θJC)
6ºC/W
For the latest package outline information and land patterns (footprints), go to www.maximintegrated.com/packages. Note that a “+”,
“#”, or “-” in the package code indicates RoHS status only. Package drawings may show a different suffix character, but the drawing
pertains to the package regardless of RoHS status.
Package thermal resistances were obtained using the method described in JEDEC specification JESD51-7, using a four-layer board.
For detailed information on package thermal considerations, refer to www.maximintegrated.com/thermal-tutorial.
www.maximintegrated.com
Maxim Integrated │ 2
MAX25612/MAX25612B
Automotive Synchronous
High Voltage LED Controller
Electrical Characteristics
(VIN = 12V, CIN = CVCC = 1μF, DL = COMP = DIMOUT = PWMDIM = FLT = unconnected, VOVP = VCS = VPGND = VSGND = 0V,
VISENSE+ = VISENSE- = 45V, VICTRL = 1.40V, TA = TJ = -40ºC to +125ºC, unless otherwise noted. Typical values are at TA = +25ºC
(Note 2))
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
Supply Voltage
Operational Supply Voltage
VIN
Supply Current
IINQ
5
VOVP = 1.5V, no switching
48
V
1.8
5
mA
1.24
1.37
V
Undervoltage Lockout
Undervoltage Lockout Rising
VUVEN_THUP
Undervoltage Lockout Hysteresis
Hys
VUVEN rising
1.12
106
mV
VCC Regulator
Regulator Output Voltage
Undervoltage Lockout
VCC
VCC_UVLOR
IVCC = 0.1mA to 50mA,
6V < VIN < 16V
4.875
rising
Undervoltage Lockout Hysteresis
5.0
5.125
V
4.0
V
0.4
V
Oscillator (RT)
Switching Frequency Range
fSW
Bias Voltage at RT
VRT
Minimum OFF time
VCOMP = HIGH, VCS = 0V
Oscillator Frequency Accuracy
Frequency Dither
200
(dither disabled)
fDITH
Dither enabled, fsw = 200kHz to 2.2MHz
ISLOPE
Peak current ramp added to CS input per
switching cycle
2200
kHz
1.25
V
85
ns
-10
+10
±6
%
%
Slope Compensation
Slope-Compensation Current
Ramp Height
42.5
50
57.5
μA
1.2
V
V
Analog Dimming
ICTRL Input Control Voltage
Range
ICTRL Zero Current Threshold
ICTRL Clamp Voltage
ICTRL Input Bias Current
ICTRLRNG
ICTRLZC_VTH
0.2
(VISENSE+ - VISENSE-) < 5mV
0.16
0.18
0.2
ICTRLCLMP
ICTRL sink = 1μA
1.25
1.30
1.35
V
ICTRLIIN
VICTRL < = 5.5V
-500
20
500
nA
-0.2
+60
V
0
225
mV
LED Current-Sense Amp
Common-Mode Input Range
Differential Signal Range
ISENSE+ Input Bias Current
IBISENSE+
(VISENSE+ - VISENSE-) = 200mV, VISENSE+ = 60V
ISENSE- Input Bias Current
IBISENSE-
(VISENSE+ - VISENSE-) = 200mV, VIBSENSE- = 60V
Voltage Gain
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(VISENSE+ - VISENSE-) = 200mV,
3V < [VISENSE+, VISENSE-] < 60V
4.9
350
550
μA
22
60
μA
5.0
5.1
V/V
Maxim Integrated │ 3
MAX25612/MAX25612B
Automotive Synchronous
High Voltage LED Controller
Electrical Characteristics (continued)
(VIN = 12V, CIN = CVCC = 1μF, DL = COMP = DIMOUT = PWMDIM = FLT = unconnected, VOVP = VCS = VPGND = VSGND = 0V,
VISENSE+ = VISENSE- = 45V, VICTRL = 1.40V, TA = TJ = -40ºC to +125ºC, unless otherwise noted. Typical values are at TA = +25ºC
(Note 2))
PARAMETER
LED Current-Sense Regulation
Voltage
SYMBOL
VSENSE
LED Current-Sense Regulation
Voltage (Low Range)
VSENSE
Common-Mode Input Range
Selector
RNGSEL
CONDITIONS
MIN
TYP
MAX
VICTRL = 1.3V,
3V < [VISENSE+, VISENSE-] < 60V
213.8
220
226.2
VICTRL = 1.2V,
3V < [VISENSE+, VISENSE-] < 60V
194
200
206
VICTRL = 0.4V,
3V < [VISENSE+, VISENSE-] < 60V
36
40
44
VICTRL = 1.2V,
0V < [VISENSE+, VISENSE-] < 3V
192
200
208
VICTRL = 0.4V,
0V < [VISENSE+, VISENSE-] < 3V
35
40
45
VISENSE+ rising
2.72
2.85
2.98
VISENSE+ falling
2.48
2.6
2.72
(VISENSE+ - VISENSE-) = 200mV
1170
1800
2430
UNITS
mV
mV
V
ERROR AMP
Transconductance
gM
μS
COMP Sink Current
COMPISINK
VCOMP = 5V
300
μA
COMP Source Current
COMPISRC
VCOMP = 0V
300
μA
1
V
PWM Comparator
Input Offset Voltage
CS Limit Comparator
Current-Limit Threshold
VCS_LIMIT
190
210
230
mV
Gate Drivers (DH and DL)
RDS(ON) Pullup pMOS
1.3
Ω
RDS(ON) Pulldown nMOS
0.9
Ω
PWM Dimming
Internal Ramp Frequency
fRAMP
160
External Sync Frequency
Range
fDIM
60
External Sync Low-Level Voltage
VLTH
External Sync High-Level Voltage
VHTH
2.0
DIM Comparator Offset Voltage
VDIMOFS
170
DIM Voltage for 100% Duty
Cycle
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3.3
200
240
Hz
2000
Hz
0.4
V
V
200
230
mV
V
Maxim Integrated │ 4
MAX25612/MAX25612B
Automotive Synchronous
High Voltage LED Controller
Electrical Characteristics (continued)
(VIN = 12V, CIN = CVCC = 1μF, DL = COMP = DIMOUT = PWMDIM = FLT = unconnected, VOVP = VCS = VPGND = VSGND = 0V,
VISENSE+ = VISENSE- = 45V, VICTRL = 1.40V, TA = TJ = -40ºC to +125ºC, unless otherwise noted. Typical values are at TA = +25ºC
(Note 2))
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
pMOS Gate Driver (DIMOUT)
Peak Pullup Current
IDIMOUTPU
VPWMDIM = 0V, (VISENSE+ - VDIMOUT)
= 7V
40
73
120
mA
Peak Pulldown Current
IDIMOUTPD
(VISENSE+ - VDIMOUT) = 0V
15
35
65
mA
-8.4
-7.4
-6.1
V
1.23
1.29
V
DIMOUT Low Voltage with
Respect to ISENSE+
Overvoltage Protection (OVP)
OVP Threshold Rising
VOVP
Output rising
1.17
IBOVP
VOVP = 1.235V
-500
(VISENSE+ - VISENSE-)
369
Hysteresis
Input Bias Current
70
mV
+500
nA
427
mV
Short-Circuit Hiccup Mode
Short-Circuit Threshold
Hiccup Time
VSHORT-HIC
THICCUP
398
Clock
Cycles
8192
Buck-Boost Short Detect
Buck-Boost Short Detect
Threshold (MAX25612 only)
VSHORT-VOUT (VISENSE+ - VIN) falling, VIN = 12V
1.15
1.55
1.95
V
VIN = 4.75V, VOVP = 2V, ISINK = 5mA
68.6
200
mV
Temperature rising
165
ºC
10
ºC
Open-Drain Fault (FLT)
Output Voltage Low
VOL-FLT
Thermal Shutdown
Thermal Shutdown Temperature
TSHDN
Thermal Shutdown Hystersis
Note 1: All devices are 100% tested at TA = +25ºC. Limits over temperature are guaranteed by design
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Maxim Integrated │ 5
MAX25612/MAX25612B
Automotive Synchronous
High Voltage LED Controller
Typical Operating Characteristics
(VIN = 13.5V, TA = 25ºC unless otherwise noted.)
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Maxim Integrated │ 6
MAX25612/MAX25612B
Automotive Synchronous
High Voltage LED Controller
Typical Operating Characteristics (continued)
(VIN = 13.5V, TA = 25ºC unless otherwise noted.)
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Maxim Integrated │ 7
MAX25612/MAX25612B
Automotive Synchronous
High Voltage LED Controller
ISENSE+
IN
BST
DH
LX
VCC
DL
PGND
CSP
CSN
Pin Configurations
20
19
18
17
16
15
14
13
12
11
TOP VIEW
MAX25612/
MAX25612B
EP
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9
SGND
10
COMP
8
ICTRL
OVP
7
TSSOP
PGND
PWMDIM
RT
UVEN
6
DL
DIMOUT
TOP VIEW
5
VCC
4
FLT
3
LX
2
DH
1
ISENSE-
+
15
14
13
12
11
BST
16
10
CSP
IN
17
9
CSN
ISENSE+
18
8
COMP
ISENSE-
19
7
SGND
DIMOUT
20
6
ICTRL
MAX25612/
MAX25612B
1
2
3
4
5
UVEN
PWMDIM
FLT
RT
OVP
+
TQFN
4mm × 4mm
Maxim Integrated │ 8
MAX25612/MAX25612B
Automotive Synchronous
High Voltage LED Controller
Pin Description
PIN
NAME
FUNCTION
19
ISENSE-
Negative LED Current-Sense Input. A 100Ω resistor is recommended to be placed in series
with ISENSE- input and the negative terminal of the LED current-sense resistor.
20
DIMOUT
External Dimming pMOS Gate Driver
TSSOP
TQFN
1
2
3
1
UVEN
Undervoltage-Lockout (UVLO) Threshold/Enable Input. UVEN is a dual-function adjustable UVLO threshold input with an enable feature. Connect UVEN to VIN through a resistive
voltage-divider to program the UVLO threshold. Observe the absolute maximum value for
this pin.
Dimming Control Input. Connect PWMDIM to an external PWM signal for PWM dimming.
For analog-voltage-controlled PWM dimming, connect PWMDIM to VCC through a resistive
voltage-divider. The dimming frequency is 200Hz under these conditions. Connect PWMDIM to SGND to turn off the LEDs.
4
2
PWMDIM
5
3
FLT
Active-Low, Open-Drain Fault Indicator Output. See the Fault Indicator (FLT) section.
6
4
RT
PWM Switching Frequency Programming. Connect a resistor (RRT) from RT to SGND to
set the internal clock frequency.
7
5
OVP
Overvoltage Protection Input. Connect a resistive divider between the converter output,
OVP, and ground. When the voltage on OVP exceeds 1.23V, a fast-acting comparator immediately stops PWM switching. This comparator has hysteresis of 70mV.
8
6
ICTRL
Analog Dimming Control Input. The voltage at ICTRL sets the LED current level when
VICTRL < 1.25V. This voltage reference can be set using a resistor-divider from VCC to
SGND. For VICTRL > 1.25V, the internal reference sets the LED current.
9
7
SGND
Signal Ground
10
8
COMP
Compensation Network Connection. For proper compensation connect a suitable RC network from COMP to SGND.
11
9
CSN
Current-Sense Amplifier Negative Input for the Switching Regulator
12
10
CSP
Current-Sense Amplifier Positive Input for the Switching Regulator. Add a series resistor
from CSP to the switching MOSFET current-sense resistor terminal for programming the
slope compensation.
13
11
PGND
14
12
DL
15
13
VCC
16
14
LX
Switch Node of the Converter
17
15
DH
High-Side nMOS Gate Driver Output
18
16
BST
Bootstrap Supply Input for the High-Side Driver
19
17
IN
20
18
ISENSE+
Positive LED Current-Sense Input. The voltage between ISENSE+ and ISENSE- is proportionally regulated to the lesser of (VICTRL, 1.23V).
-
-
EP
Exposed Pad. Connect EP to the ground plane for heatsinking. Do not use EP as the only
electrical connection to ground
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Power Ground
Low-Side nMOS Gate Driver Output
5V Low-Dropout Voltage Regulator Output. VCC supplies the bias current for the gate drive
and internal control logic. Bypass VCC to GND with a 4.7µF and a 0.1µF ceramic capacitor.
Positive Power-Supply Input. Bypass with a 1µF ceramic capacitor to PGND.
Maxim Integrated │ 9
MAX25612/MAX25612B
Automotive Synchronous
High Voltage LED Controller
Functional Diagrams
UVEN
MAX25612/
MAX25612B
1.24V
EN
5V REG
IN
THERMAL TSHDN
SHUTDOWN
VCC
VCC
UVLO
BG
VCC
S
0.21V
CSN
ICTRL
ISENSE+
ISENSE-
LX
BST
DH
LX
LX
Q
BLANKING
x2
PWM
COMP
VCC
VCC
R
ISLOPE
CSP
BST
R-DOM
RT
OSCILLATOR
RT
BST
MAX DUTY
CYCLE
DL
VCC
PGND
1.0V
VICTRLCLMP
MIN
OUT
LPF
gM
COMP
x5
0.2V
PWMDIM
VISENSE+
BUCK-BOOST
SHORT DETECTION
(MAX25612 ONLY)
DIMOUT
200Hz
VISENSE+ - 7V
0.32V
2.2V
S
8192 x TOSC
HICCUP TIMER
Q
R
FLT
OVP
TSHDN
SGND
1.23V
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Maxim Integrated │ 10
MAX25612/MAX25612B
Detailed Description
The MAX25612/MAX25612B are single-channel HBLED
drivers for automotive front-light applications such as high
beam, low beam, daytime running light (DRL), turn
indicator, fog light, and other LED lights. They can take
an input voltage from 5V to 48V and can drive a string
of LEDs with a maximum output voltage of 60V. The
MAX25612/MAX25612B feature both low- and high-side
nMOS drivers for synchronous rectification. Synchronous
rectification greatly improves efficiency compared to
asynchronous switching converters, especially in highcurrent applications. Reverse recovery losses of the
synchronous MOSFET will increase at higher output
voltages; therefore, the efficiency benefit may be reduced
when driving large numbers of LEDs. Refer to the Typical
Operating Characteristics section for comparisons of
synchronous and asynchronous switching efficiency with
different currents and voltages.
The MAX25612/MAX25612B sense output current at the
high side of the LED string. High-side current sensing is
required to protect against shorts from the output to the
ground or battery input. It is also the most flexible scheme
for driving LEDs, allowing boost, high-side buck, SEPIC,
or buck-boost mode configurations. The PWMDIM input
provides LED dimming ratios of up to 5000:1, and the ICTRL
input provides additional analog dimming capability in the
MAX25612/MAX25612B. The MAX25612/MAX25612B
also include a FLT flag that indicates open string, shorted
string and thermal shutdown. The MAX25612/MAX25612B
have built-in spread-spectrum modulation for improved
electromagnetic compatibility performance.
Functional Operation
The operation of the MAX25612/MAX25612B is best
understood by referring to the block diagram of the
device. The devices are enabled when the UVEN pin
goes above 1.24V. In addition to the UVEN input, the
5V regulator input also needs to be above its respective
UVLO limit before switching on DL and DH can start.
The MAX25612/MAX25612B are constant-frequency,
current-mode controllers with low-side and high-side
NMOS gate drivers for synchronous switching. Switching
is initiated when PWM goes high. The RT oscillator can
be programmed from 200kHz to 2.2MHz by the resistor
between RT and SGND. Spread-spectrum dithering is
added to the oscillator to alleviate EMI problems in the
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Automotive Synchronous
High Voltage LED Controller
LED driver. The RT oscillator is synchronized to the
positive edge of the PWM pulse. This means that the
DL pulse goes high at the same instant as the positive
pulse on PWMDIM. Synchronizing the RT oscillator to
the PWMDIM pulse also guarantees that the switching
frequency variation over a period of a PWMDIM pulse
is the same from one PWMDIM pulse to the next. This
prevents flicker during PWM dimming when spread
spectrum is added to the RT oscillator.
Once PWMDIM transitions high, the external low-side
switching MOSFET is turned on. A current flows through
the low-side MOSFET, and this current is sensed by
the voltage across the current-sense resistor from the
source of the external low-side MOSFET to PGND.
The MOSFET source is connected to the CSP input
of the MAX25612/MAX25612B through a slopecompensation resistor (RSC). See the Typical Application
Circuits section. The ground side of the current-sense
resistor is connected to the CSN input. The slopecompensation current flows out of CSP and through the
RSC resistor. The differential voltage across CSP and
CSN is the voltage across the current-sense resistor
(RCS_FET) + (slope-compensation current x RSC). Slope
compensation prevents sub-harmonic oscillation when the
duty cycle exceeds 50%. Current in the external inductor
increases steadily when the external low-side MOSFET
is on. The differential voltage across CSP and CSN is fed
to the input of the current-limit comparator. This currentlimit comparator is used to protect the external low-side
switch from overcurrent and will cause switching to stop
for that particular cycle if (VCSP - VCSN) exceeds 0.21V.
The differential current-sense voltage signal is amplified
by a gain factor of two. The output of the amplifier has
a 1.0V offset added before being applied to the positive
input of a PWM comparator. The negative input of this
comparator is a control voltage from the error amplifier
that regulates the LED current. When the positive input
of the PWM comparator exceeds the control voltage
from the error amplifier, the switching is stopped for that
particular cycle and the external low-side nMOS stays off
until the next switching cycle. The inductor current decays
when the low-side nMOS is turned off. The inductor
current starts ramping back up when the next switching
cycle starts and the external low-side MOSFET turns
back on. Through this repetitive action, the PWM control
Maxim Integrated │ 11
MAX25612/MAX25612B
algorithm establishes a switch duty cycle to regulate the
current in the LED load.
When PWMDIM transitions high, the external dimming
MOSFET that is driven by DIMOUT is also turned
on. This external dimming MOSFET is a p-channel
MOSFET and is connected on the high side of the LED
load. The source of this pMOS is connected to ISENSEand the gate is connected to DIMOUT. The drain of this
MOSFET is connected to the anode of the external LED
string. In certain applications it is not necessary to use this
dimming MOSFET, and in these cases the DIMOUT output is left open. The external pMOS is turned on when
PWMDIM is high and is turned off when PWMDIM is
low. During normal operation when PWMDIM is high, the
voltage across the resistor from ISENSE+ to ISENSE- is
regulated to a programmed voltage. This programmed
voltage is 0.2 x (VICTRL - 0.2). The external pMOS switch
is also used for fault protection. Once a fault condition is
detected, DIMOUT is pulled high to turn off the pMOS
switch. This isolates the LED string from the fault condition and prevents excessive voltage or current from damaging the LEDs.
Input Voltage (IN)
The input supply (IN) must be locally bypassed with a
minimum of 1μF capacitance close to the pin. All the input
current that is drawn by the MAX25612/MAX25612B goes
through this input.
UVLO
The MAX25612/MAX25612B feature an adjustable UVLO
using the undervoltage enable input (UVEN). Connect
UVEN to IN through a resistive divider to set the UVLO
threshold. The MAX25612/MAX25612B are enabled
when VUVEN exceeds the 1.24V (typ) threshold. UVEN
also functions as an enable/disable input to the device.
Drive UVEN low to disable the device. Drive UVEN
high to enable the device.
VCC Regulator
The VCC supply is the low-voltage analog supply for the
chip and derives power from the input voltage from IN
to PGND. An internal power-on reset (POR) monitors
the VCC voltage and the IN voltage. The input voltage to
the VCC regulator is disconnected when the voltage at
IN goes below the UVLO threshold. A POR is generated
when VCC goes below its UVLO threshold, causing the
www.maximintegrated.com
Automotive Synchronous
High Voltage LED Controller
IC to reset. The chip will come out of reset state once the
input voltage goes back up and the VCC regulator output
is back in regulation.
Dimming MOSFET Driver (DIMOUT)
The IC requires an external p-channel MOSFET for
PWM dimming. For normal operation, connect the gate
of the MOSFET to the output of the dimming driver
(DIMOUT). The dimming driver can sink up to 35mA or
source up to 73mA of peak current for fast charging and
discharging of the p-MOSFET gate. When the PWMDIM
signal is high, this driver pulls the p-MOSFET gate to 7V
below VISENSE+ to completely turn on the p-channel dimming MOSFET. The DIMOUT output inverts and level-shifts
the signal on PWMDIM to drive the gate of the external
PMOS. In some applications, the dimming FET is not used.
In this case, the DIMOUT output can be left open.
LED Current-Sense Inputs (ISENSE+/ISENSE-)
The differential voltage from ISENSE+ to ISENSE- is
fed to an internal current-sense amplifier. This amplified signal is then connected to the negative input of the
transconductance error amplifier. The voltage-gain factor
of this amplifier is 5. The offset voltage for this amplifier
is +1mV. A resistor is connected between ISENSE+ and
ISENSE- to program the maximum LED current. The
full-scale signal is 200mV. The ISENSE+ input should be
connected to the positive terminal of the current-sense
resistor and the ISENSE- input should be connected to
the negative terminal of the current-sense resistor (LED
string anode side).
Internal Oscillator (RT)
The internal oscillator of the MAX25612/MAX25612B are
programmable from 200kHz to 2.2MHz using a single
resistor at RT. Use the following formula to calculate the
switching frequency:
fOSC(kHz) = 34200/RRT(kΩ)
where RRT is the resistor from RT to SGND. This equation
is a linear approximation of the relationship between
fOSC and RRT. See Table 1 and the Typical Operating
Maxim Integrated │ 12
MAX25612/MAX25612B
Characteristics section for more data points showing the
relationship between RRT and fOSC. The MAX25612/
MAX25612B have built-in frequency dithering of ±6% of
the programmed frequency to alleviate EMI problems.
Spread Spectrum
The devices have an internal spread-spectrum option to
optimize EMI performance. The switching frequency is
varied ±6%, centered on the oscillator frequency (fOSC).
The modulation signal is a triangular wave with a period
of 418 clocks. Therefore, fOSC ramps down 6% and back
to the set frequency in 418 clocks, and also ramps up 6%
and back to the set frequency in another 418 clocks.
Synchronous MOSFET Switch Driver
(DH and DL)
The IC drives an external high-side and low-side n-channel switching MOSFET. DH and DL can sink/source 2A
of peak current, allowing the ICs to switch MOSFETs in
high-power applications. The average current demanded
from the supply to drive the external MOSFETs depends
on the total gate charge (QG) and the operating frequency
of the converter, fSW. Use the following equation to calculate the driver supply current IDRIVER required for the
switching MOSFET:
IDRIVER = QG x fSW
The low-side gate driver (DL) drives an external nMOS (N1)
with either VCC or VPGND to turn the MOSFET on or off,
respectively. The high-side gate driver (DH) drives an external nMOS (N2) with either VBST or VLX to turn the MOSFET
on or off, respectively. During normal operation, DH will be
driven high while the DL is driven low. Likewise, DH will be
driven low while DL is driven high, thereby achieving synchronous switching. There is a small break-before-make
delay between the transitions to prevent any shoot-through
current that would occur as a result of both low- and highside MOSFETs being turned on at the same time.
Boost Capacitor Node (BST)
The BST input is used to provide a drive voltage to the
high-side switching MOSFET that is higher than LX.
Connect a 0.1μF ceramic capacitor from BST to the LX
switch node. Connect a diode from VCC to BST. Place the
capacitor as close as possible to BST.
Switching MOSFET Current-Sense Input
(CSP and CSN)
CSP and CSN are part of the current-mode control loop.
The switching control uses the voltage across CSP and
CSN, set by RCS and RSC, to terminate the ON pulse
width of the switching cycle, thus achieving peak currentmode control. Internal leading-edge blanking of 50ns is
provided to prevent premature turn-off of the switching
MOSFET in each switching cycle. Resistor RCS is conwww.maximintegrated.com
Automotive Synchronous
High Voltage LED Controller
nected between the source of the n-channel switching
MOSFET and PGND. During switching, a current ramp
with a slope of 50μAxfSW is sourced from the CSP input.
This current ramp, along with resistor RSC, programs the
amount of slope compensation.
Overvoltage Protection Input (OVP)
OVP sets the overvoltage threshold limit across the LEDs.
Use a resistive divider from ISENSE+ to OVP to SGND
to set the overvoltage threshold limit. An internal overvoltage protection comparator senses the differential voltage across OVP and SGND. If the differential voltage is
greater than 1.23V, DL goes low, DH and DIMOUT go high,
and FLT asserts. When the differential voltage drops by
70mV, DL is enabled, DIMOUT goes low, and FLT deasserts.
Output Short-Circuit Protection
The MAX25612/MAX25612B feature output short-circuit
protection. This feature is most useful where the LEDs are
connected over long cables and there exists the possibility
of shorts occurring when connectors are exposed.
For the MAX25612, short circuit is detected when the following two conditions are met:
●● VISENSE+ is lower than VIN by the VSHORT_
VOUT threshold, 1.55V (typ)
●● The current-sense voltage across VISENSE+ - VISENSE- exceeds the VSHORT_HIC threshold, 398mV
(typ)
The MAX25612B has disabled the VSHORT_VOUT threshold flag for applications where (VISENSE+ - VIN) is expected to be less than 1.55V (typ) during normal operation. In
this case, the VSHORT_HIC threshold is the only criteria
for detecting a short circuit.
The MAX25612/MAX25612B respond by stopping DL and
DH switching and pulling DIMOUT high to VISENSE+ to
turn off the dimming FET, which disconnects the output
capacitors from the shorted output. The device waits
8192 clock cycles before attempting to drive the LEDs
again. The 8192-clock-cycle counter is only active while
PWMDIM is HIGH.
Internal Transconductance Error Amplifier
The IC has a built-in transconductance amplifier that
is used to amplify the error signal inside the feedback loop. When the dimming signal is low, COMP
is disconnected from the output of the error amplifier
and DIMOUT goes high. When the dimming signal is high,
the output of the error amplifier is connected to COMP
and DIMOUT goes low. This enables the compensation
capacitor to hold the charge when the dimming signal
has turned off the internal switching MOSFET gate drive.
To maintain the charge on the compensation capacitor
Maxim Integrated │ 13
MAX25612/MAX25612B
Automotive Synchronous
High Voltage LED Controller
CCOMP, the capacitor should be a low-leakage ceramic
type. When the internal dimming signal is enabled, the
voltage on the compensation capacitor forces the converter into steady state almost instantaneously. The transconductance of the amplifier is 1800μS.
Signal Ground (SGND)
Analog Dimming
Thermal Shutdown
The devices offer an analog dimming control input
(ICTRL). The voltage at ICTRL sets the LED current
level when VICTRL < 1.3V (typ). The LED current can be
linearly adjusted from zero with the voltage on ICTRL. For
VICTRL > 1.3V (typ), an internal reference sets the LED
current. The LED current is guaranteed to be at zero when
the ICTRL voltage is at or below ICTRLZC_VTH(MIN). The
LED current can be linearly adjusted from zero to full
scale for the ICTRL voltage in the range of 0.2V to 1.2V.
Pulse-Dimming Input
The PWMDIM input of the MAX25612/MAX25612B
functions with either analog or PWM control signals.
Once the internal pulse detector detects three successive
edges of a PWM signal with a frequency between 60Hz
and 2kHz, the MAX25612/MAX25612B synchronize to
the external signal and pulse-width modulates the LED
current at the external DIM input frequency with the same
duty cycle as the DIM input. PWM dimming outside this
frequency range is also possible, with the caveat that the
switching clock may not be synchronized to the PWM
rising edge. If an analog control signal is applied to DIM,
the MAX25612/MAX25612B compare the DC input to an
internally generated 200Hz ramp to pulse-width-modulate
the LED current (fDIM = 200Hz). The output-current duty
cycle is linearly adjustable from 0% to 100% (0.2V < VDIM
< 3.0V). Use the following formula to calculate the voltage,
VDIM, necessary for a given output-current duty cycle D
VDIM = (D x 2.8) + 0.2V
where VDIM is the voltage applied to DIM in volts.
Power Ground (PGND)
This is the analog ground pin for all of the control circuitry
of the LED driver. Connect the PGND (power ground) and
the SGND together at the negative terminal of the input
bypass capacitor.
The devices feature thermal protection. When the junction temperature exceeds +165°C, the external switching
MOSFET starts operating at the minimum pulse width
to reduce the power dissipation in the internal power
MOSFETs. The part returns to regulation mode once the
junction temperature goes below +155°C. This results
in a cycled output during continuous thermal-overload
conditions.
Fault Indicator (FLT)
The MAX25612/MAX25612B feature an active-low, opendrain fault indicator (FLT). FLT asserts when one of
the following conditions occur:
1) Overvoltage across the LED string
2) Short-circuit condition across the LED string
3) Overtemperature condition
When the output voltage drops below the overvoltage
set point minus the hysteresis, FLT deasserts. Similarly,
during overtemperature fault, the FLT signal remains
asserted until the junction temperature falls 10ºC below
the thermal-shutdown threshold.
Exposed Paddle
The MAX25612/MAX25612B package features an exposed
thermal pad on its underside that should be used as a heat
sink. This pad lowers the package’s thermal resistance by
providing a direct heat-conduction path from the die to the
PCB. Connect the exposed pad and GND to the system
ground using a large pad or ground plane, or multiple vias
to the ground plane layer.
The power ground (PGND) connection acts as the ground
reference for the switching power components. Connect
PGND as close as possible to the negative plate of the
VCC decoupling capacitor.
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Maxim Integrated │ 14
MAX25612/MAX25612B
Automotive Synchronous
High Voltage LED Controller
Applications Information
VCC Regulator
The internal 5V regulator is used to power the internal
control circuitry inside the MAX25612/MAX25612B, as
well as the low-side FET gate driver. This regulator can
provide a load of 10mA to external circuitry. The 5V
regulator requires an external ceramic capacitor for stable
operation. A 4.7µF ceramic capacitor is adequate for most
applications. Place the ceramic capacitor close to the IC
to minimize trace length to the internal VCC pin and also
to the IC ground. Choose a 10V rated low-ESR, X7R
ceramic capacitor for optimal performance.
Programming the UVLO Enable Threshold
The UVLO threshold is set by resistors RUVEN1 and
RUVEN2 (see the Typical Application Circuits section).
The MAX25612/MAX25612B turn on when the voltage
across RUVEN2 exceeds 1.24V, the UVLO threshold. Use
the following equation to set the desired UVLO enable
threshold:
VUVEN
=
1.24
×
(RUVEN1
+ RUVEN2
RUVEN2
)
where VUVEN is the rising undervoltage threshold in volts.
The UVEN input can also be used as a digital enable
by applying an external logic signal that can turn the
MAX25612/MAX25612B on and off.
Programming LED Current
Normal sensing of the LED current should be done on the
high side where the LED current-sense resistor is connected to the anode of the LED string. The LED current is
programmed using the resistor RCS_LED (see the Typical
Application Circuits section). When ICTRL is connected to
a voltage greater than 1.3V, the internal reference regulates the voltage across RCS_LED to 220mV. The current
is given by:
ILED
www.maximintegrated.com
=
0.22
RCS_LED
The LED current can also be programmed by adjusting
the voltage on ICTRL when VICTRL ≤ 1.2V (analog dimming). The current is given by:
ILED
=
(VICTRL − 0.2)
(5 × RCS_LED)
Programming the Switching Frequency
The internal oscillator of the MAX25612/MAX25612B
is programmable from 200kHz to 2.2MHz using a single resistor at RT. Use the following formula to calculate
the value of the resistor RRT:
R RT (kΩ) =
34200
f OSC
where fOSC is the desired switching frequency in kHz.
This equation is a linear approximation of the relationship between RRT and fOSC. See Table 1 and the Typical
Operating Characteristics section for more data points
showing the relationship between RRT and fOSC.
Additional ±6% spread spectrum is added internally to
the oscillator to improve EMI performance.
Setting the Overvoltage Threshold
The overvoltage threshold is set by resistors ROVP1 and
ROVP2 (see the Typical Application Circuits section). The
overvoltage circuit in the MAX25612/MAX25612B is
activated when the voltage on OVP with respect to GND
exceeds 1.23V. Use the following equation to set the
desired overvoltage threshold:
VOVP = 1.23 x (ROVP1 + ROVP2)/ROVP2
Table 1. Typical RRT Programming Values
R RT (kΩ)
fOSC (kHz)
188
200
34.2
1000
14.7
2200
Maxim Integrated │ 15
MAX25612/MAX25612B
Inductor Selection
Boost Configuration
In the boost converter, the average inductor current varies with the line voltage. The maximum average current
occurs at the lowest line voltage. For the boost converter,
the average inductor current is equal to the input current.
Calculate maximum duty cycle using the equation below:
DMAX = (VLED - VFET2 - VINMIN)/
(VLED + VFET2 - VFET1)
where VLED is the forward voltage of the LED string
in volts, VINMIN is the minimum input supply voltage in
volts, and VFET1 and VFET2 are the average drain-to
source voltages of the MOSFETs N1 and N2 in volts when
they are on. Use an approximate value of 0.2V initially to
calculate DMAX. A more accurate value of the maximum
duty cycle can be calculated once the power MOSFET
is selected based on the maximum inductor current. Use
the following equations to calculate the maximum average inductor current ILAVG, peak-to-peak inductor current
ripple ∆IL, and the peak inductor current ILP in amperes:
ILAVG = ILED/(1 - DMAX)
Allowing the peak-to-peak inductor ripple to be
∆IL, the peak inductor current is given by:
ILP = ILAVG + 0.5 x ∆IL
The inductance value (L) of inductor L1 in Henries (H)
is calculated as:
L = (VINMIN - VFET1) x DMAX/(fSW x ∆IL)
where fSW is the switching frequency in Hertz, VINMIN and
VFET1 are in volts, and ∆IL is in amperes. Choose an
inductor that has a minimum inductance greater than the
calculated value. The current rating of the inductor should
be higher than ILP at the operating temperature.
Buck-Boost Configuration
In the buck-boost LED driver, the average inductor current is equal to the input current plus the LED current.
Calculate the maximum duty cycle using the following
equation:
DMAX = (VLED + VFET2)/
(VLED + VFET2 + VINMIN - VFET1)
where VLED is the forward voltage of the LED string in
volts, VINMIN is the minimum input supply voltage in volts,
and VFET1 and VFET2 are the average drain-to-source
www.maximintegrated.com
Automotive Synchronous
High Voltage LED Controller
voltages of the MOSFETs N1 and N2 in volts when they
are on. Use an approximate value of 0.2V initially to calculate DMAX. A more accurate value of maximum duty cycle
can be calculated once the power MOSFET is selected
based on the maximum inductor current.
Use the equations below to calculate the maximum average inductor current ILAVG, peak-to-peak inductor current
ripple ∆IL, and the peak inductor current ILP in amperes:
ILAVG = ILED/(1 - DMAX)
Allowing the peak-to-peak inductor ripple to be ∆IL
ILP = ILAVG + 0.5 x ∆IL
where ILP is the peak inductor current.
The inductance value (L) of inductor L1 in Henries is calculated as:
L = (VINMIN - VFET1) x DMAX/(fSW x ∆IL)
where fSW is the switching frequency in Hertz, VINMIN and
VFET1 are in volts, and ∆IL is in amperes. Choose an
inductor that has a minimum inductance greater than the
calculated value.
High-Side Buck Configuration
In the high-side buck LED driver, the average inductor
current is the same as the LED current. The peak inductor
current occurs at the maximum input line voltage where
the duty cycle is at the minimum.
DMIN = (VLED + VFET2)/(VINMAX - VFET1)
where VLED is the forward voltage of the LED string in
volts, VINMAX is the maximum input supply voltage in
volts, and VFET1 and VFET2 are the average drain-tosource voltages of the MOSFETs N1 and N2 in volts when
they are on. Use an approximate value of 0.2V initially
to calculate DMIN. The maximum peak-to-peak inductor
ripple ∆IL occurs at the maximum input line. The peak
inductor current is given by
ILP = ILED + 0.5 x ∆IL
The inductance value (L) of inductor L1 in Henries
is calculated as:
L = (VINMAX - VFET1 - VLED) x DMIN/(fSW x ∆IL)
where fSW is the switching frequency in Hertz, VINMAX and
VFET1 are in volts, and ∆IL is in amperes. Choose an
inductor that has a minimum inductance greater than the
calculated value.
Maxim Integrated │ 16
MAX25612/MAX25612B
SEPIC Configuration
The SEPIC converter provides the option to use either a
coupled inductor or two separate inductors (see Typical
Application Circuits). The average L1 inductor current
is equal to the input current. The average L2 inductor
current is equal to the LED current. Neglecting voltage
drops across the FETs, the maximum duty cycle can be
calculated as follows:
D MAX =
VLED
(VINMIN + VLED )
Where VLED is the LED string voltage and VINMIN is the
minimum input voltage. The inductor value of L1 is given by:
L1 =
VINMIN × D MAX
f SW + ∆IL IN
Where ΔILIN is the desired maximum input current ripple.
The L1 peak inductor current, ILINPK, is given by:
=
IL INPK ILED
∆IL IN
D MAX
+
1 − D MAX
2
The average current in inductor L2 is the same as the
LED current. The desired maximum peak-to-peak output
current ripple is ΔILOUT. The value of the inductor L2 is
given by:
L2 =
VINMIN × D MAX
f SW × ∆IL OUT
The L2 peak inductor current, ILOUTPK, is given by:
∆IL OUT
IL OUTPK
= ILED +
2
To simplify further SEPIC calculations, use the following
values of L and ILAVG:
L=
L1× L2
L1 + L2
ILAVG = IL1AVG + IL2AVG
choose the value of CSEPIC such that the peak to peak
ripple on it is less than 2% of the minimum input supply
voltage. This ensures that the second‐order effects
created by the series resonant circuit comprising L1,
CSEPIC, and L2 does not affect the normal operation of
the converter. Use the following equation:
C SEPIC ≥
www.maximintegrated.com
ILED × D MAX
VINMIN × 0.02 × f SW
Automotive Synchronous
High Voltage LED Controller
Switching MOSFET (N1) Selection
The switching MOSFET (N1) should have a voltage rating sufficient to withstand the maximum output voltage
together with the voltage drop of synchronous high-side
nMOS (N2), and any possible overshoot due to ringing
caused by parasitic inductances and capacitances. Use a
MOSFET with a drain-to-source voltage rating higher than
that calculated by the following equations:
Boost configuration:
VDS_MAX = (VLED + VFET2 + VRCS_LED + VPFET) x 1.2
Buck-boost configuration:
VDS_MAX = (VLED +VINMAX+ VFET2 +
VRCS_LED + VPFET) x 1.2
The factor 1.2 provides 20% safety margin.
A resistor is also typically added in series with the gate
of the switching MOSFET (N1) to adjust the slew rate,
minimize ringing on the switch node, and improve EMI
performance.
Synchronous MOSFET (N2) Selection
The synchronous MOSFET (N2) should have a similar
voltage rating as N1, such that it can withstand the output voltage while N1 is on and the LX node is pulled to
ground, including any possible undershoot due to ringing.
Dimming MOSFET Selection
Select a dimming MOSFET (P1) with continuous current
rating at the operating temperature higher than the LED
current by 30%. The drain-to-source voltage rating of the
dimming MOSFET must be higher than VLED by 20%.
A resistor may also be added in series with the gate of the
dimming MOSFET to control the slew rate and help reduce
current spikes that can occur when the dimming FET turns
on and connects the switching converter output capacitor to any capacitors at the LED load. A capacitor may be
added across the gate and drain of the dimming FET to get
better control of the RC time constant that controls the slew
rate. Otherwise, the RC time constant is controlled by the
parasitic capacitance of the chosen pMOS.
Slope Compensation
Slope compensation should be added to converters
with peak current-mode-control operating in continuousconduction mode with more than 50% duty cycle to avoid
current-loop instability and subharmonic oscillations. The
minimum amount of slope compensation required for stability is given by the following equation:
VSLOPE(MIN) = 0.5 x (inductor current downslope inductor current upslope) x RCS_FET
Maxim Integrated │ 17
MAX25612/MAX25612B
Automotive Synchronous
High Voltage LED Controller
In the MAX25612/MAX25612B, the slope-compensating
ramp is added to the current-sense signal before it is fed
to the PWM comparator. Connect a resistor (RSC) from
CSP to the switch current-sense resistor terminal for programming the amount of slope compensation.
The devices generate a current ramp with a slope of 50μA/
tOSC for slope compensation. The current-ramp signal is
forced into an external resistor (RSC) connected between
CSP and the source of the external MOSFET, thereby
adding a programmable slope-compensating voltage
(VSLOPE) at the current-sense input CSP. Therefore:
dVSLOPE/dt = (RSLOPE x 50μA)/tOSC
The slope-compensation voltage that needs to be added
to the current signal at minimum line voltage, with margin
of 1.2x, is given by the following equation:
RCS_FET =
Buck-boost and SEPIC configuration:
VSLOPE = DMAX
(VLED − VINMIN) × RCS_FET × 1.2
(2 × L × fSW)
High-side buck configuration:
VSLOPE = DMAX
(2 × VLED − VINMIN) × RCS_FET × 1.2
(2 × L × fSW)
MOSFET Current-Sense Resistor
The minimum value of the peak current-limit comparator
is 0.19V. The current-sense resistor value is given by:
RCS_FET = (0.19 - DMAX x VSLOPE)/ILPK
where ILPK is the peak inductor current that occurs at low
line in the boost and buck-boost configurations.
For boost configuration:
www.maximintegrated.com
[
ILPK + 0.75DMAX
(VLED − 2VINMIN)
L × fSW
]
For buck-boost configuration:
RCS_FET =
[
0.19
ILPK + 0.75DMAX
(VLED − VINMIN)
L × fSW
]
For SEPIC configuration:
R CS
FET
Boost configuration:
(VLED − 2 × VINMIN) × RCS_FET × 1.2
VSLOPE = DMAX
(2 × L × fSW)
0.19
=
0.19
(VLED − VINMIN)
IL1PK + IL2 PK + 0.75D MAX
L × f SW
For high-side buck configuration:
R CS
FET
=
0.19
(2VLED − VINMIN)
IL1PK + 0.75D MAX
L × f SW
Input Capacitor Selection
The input-filter capacitor bypasses the ripple current drawn by the converter and reduces the amplitude
of high-frequency current conducted to the input supply.
The ESR, ESL, and bulk capacitance of the input capacitor contribute to the input ripple. Use a low-ESR input
capacitor that can handle the maximum input RMS ripple
current from the converter. For the boost configuration, the input current is the same as the inductor current. For buck-boost configuration, the input current is
the inductor current minus the LED current. However, for
both configurations, the ripple current that the input filter
capacitor has to supply is the same as the inductor ripple
current with the condition that the output filter capacitor should be connected to ground for buck-boost configuration. This reduces the size of the input capacitor, as
the inductor current is continuous with maximum ∆IL/2.
Neglecting the effect of LED current ripple, the calculation of the input capacitor for boost, as well as buck-boost
configurations is the same.
Maxim Integrated │ 18
MAX25612/MAX25612B
Automotive Synchronous
High Voltage LED Controller
Neglecting the effect of the ESL, the ESR, and the
bulk capacitance at the input contributes to the inputvoltage ripple. For simplicity, assume that the contribution
from the ESR and the bulk capacitance is equal. This
allows 50% of the ripple for the bulk capacitance. The
capacitance is given by:
CIN
≥
(4
∆ IL
×
∆ VIN × fSW
)
where ∆IL is in amperes, CIN is in Farads, fSW is in
Hertz, and ∆VIN is in volts. The remaining 50% of allowable ripple is for the ESR of the output capacitor.
Use X7R ceramic capacitors for optimal performance.
The selected capacitor should have the minimum required
capacitance at the operating voltage.
In buck mode, the input capacitor has large pulsed
currents due to the current flowing in the synchronous
MOSFET N2 when the switching MOSFET N1 is off. It is
very important to consider the ripple-current rating of the
input capacitor in this application.
Output Capacitor Selection
The function of the output capacitor is to reduce the output ripple to acceptable levels. The ESR, ESL, and bulk
capacitance of the output capacitor contribute to the output ripple. In most applications, the output ESR and ESL
effects can be dramatically reduced by using low-ESR
ceramic capacitors. To reduce the ESL and ESR effects,
connect multiple ceramic capacitors in parallel to achieve
the required bulk capacitance. To minimize audible noise
generated by the ceramic capacitors during PWM dimming, it may be necessary to minimize the number of
ceramic capacitors on the output. In these cases, an
additional electrolytic or tantalum capacitor provides most
of the bulk capacitance.
Boost and Buck-Boost Configurations
The calculation of the output capacitance is the same
for both boost and buck-boost configurations. The output
ripple is caused by the ESR and bulk capacitance of the
output capacitor if the ESL effect is considered negligible.
For simplicity, assume that the contributions from ESR
and bulk capacitance are equal, allowing 50% of the ripple for the bulk capacitance. The capacitance is given by:
ILED × 2 × DMAX
COUT = VOUT
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RIPPLE × fSW
The remaining 50% of allowable ripple is for the ESR of
the output capacitor.
Based on this, the ESR of the output capacitor is given by:
ESRCOUT =
VOUTRIPPLE
ILPK × 2
Feedback Compensation
The LED current-control loop comprising the switching
converter, LED current amplifier, and error amplifier should
be compensated for stable control of the LED current.
The switching converter small-signal transfer function
has a right half-plane (RHP) zero for boost, SEPIC, and
buck-boost configurations, as the inductor current is in
continuous-conduction mode. The RHP zero adds a 20dB/
decade gain together with a 90° phase lag, which is difficult
to compensate. The easiest way to avoid this zero is to roll
off the loop gain to 0dB at a frequency less than 1/5 of the
RHP zero frequency with a -20dB/decade slope.
The worst-case RHP zero frequency (fZRHP) is calculated
as follows:
Boost configuration:
fZRHP =
(
VLED × 1 − DMAX
2π × L × ILED
)2
Buck-boost configuration:
fZRHP =
(VLED + VINMIN) × (1 − DMAX)2
2π × L × ILED
SEPIC configuration:
f ZRHP =
VLED (1 − D MAX )
2
2π × L × ILED × D MAX
The switching converter small-signal transfer function
also has an output pole for both boost and buck-boost
configurations. The effective output impedance that determines the output pole frequency together with the output
filter capacitance is calculated as:
Boost configuration:
ROUT =
(RLED + RCS_LED) × VLED
(RLED + RCS_LED) × ILED + VLED
Maxim Integrated │ 19
MAX25612/MAX25612B
Buck-boost configuration:
ROUT =
(RLED + RCS_LED) × VLED
(RLED + RCS_LED) × ILED × DMAX + VLED
where RLED is the dynamic impedance of the LED string
at the operating current.
The output pole frequency for both boost and buck-boost
configurations is calculated as follows:
1
fP = 2πR
OUTCOUT
The feedback-loop compensation is done by connecting
a resistor (RCOMP) and capacitor (CCOMP) in series from
COMP to SGND. RCOMP is chosen to set the high-frequency integrator gain for fast transient response, while
CCOMP is chosen to set the integrator zero to maintain
loop stability. For optimum performance, choose the components using the following equations:
fC = 0.2× fZRHP
The value of RCOMP and CCOMP can be calculated as:
RCOMP
=
2 x fZRHP x RCS_FET
fC x (1 − DMAX) x RCS_LED x 5 x GM
CCOMP
=
25
π x fZRHP x RCOMP
the CCOMP_HF capacitor will add a higher frequency pole,
which helps to ensure good gain margin and stability. It is
typically chosen to cancel the zero from the output capaci-
Automotive Synchronous
High Voltage LED Controller
tor ESR, or such that the pole is at one half the switching
frequency, whichever is lower.
1
fP2 = 2πR
COMPCCOMP_HF
A large resistor, such as 499kΩ or 1MΩ, should be added
from COMP to SGND in applications where the ICTRL
input is supplied by a programmed voltage source, which
may be less than 0.2V. For applications in which the
ICTRL is connected to a fixed voltage from a resistordivider, the COMP pulldown resistor is not needed.
High-Side Buck Compensation
The high-side buck configuration does not have a right halfplane zero to avoid, so in most cases a single capacitor
from COMP to GND will suffice to compensate the loop.
Calculate CCOMP according to the following equation:
G M × A V × R CS
C COMP =
2π × f C × R CS
FET
Where CCOMP is the compensation capacitor value in
nF, GM is the GM amplifier transconductance in μA/V,
AV is the LED current-sense voltage gain, and fC is the
desired crossover frequency in kHz. Choose a crossover
frequency that is lower than fSW/15.
The output pole is set by the dynamic resistance of the
LED string and the COUT capacitor
f POUT =
1
2π × R DYN × C OUT
If the output pole is within a decade of the crossover frequency, then it can be compensated by adding a resistor,
RCOMP, in series with CCOMP.
R=
COMP
www.maximintegrated.com
LED
C OUT
× R DYN
C COMP
Maxim Integrated │ 20
MAX25612/MAX25612B
Automotive Synchronous
High Voltage LED Controller
Typical Application Circuits
Boost LED Driver Using MAX25612
L1
VIN
N2
CBST
VCC
BST
IN
LX
CIN
COUT
DL
RUVEN1
CSP
UVEN
PWMDIM
C1
ISENSE+
FLT
ISENSE-
RCOMP
CCOMP_HF
ICTRL
www.maximintegrated.com
100Ω
COMP
R1
RRT
RCS_LED
DIMOUT
VCC
R2
ROVP2
P1
DH
OVP
MAX25612
VCC
RSC
CSN
PGND
VCC
RUVEN2
N1
RCS_FET
PWMDIM
ROVP1
RT
SGND
EP
CCOMP
Maxim Integrated │ 21
MAX25612/MAX25612B
Automotive Synchronous
High Voltage LED Controller
Typical Application Circuits (continued)
Buck-Boost LED Driver Using MAX25612
L1
VIN
N2
CBST
VCC
BST
IN
RUVEN1
LX
CIN
COUT
DL
CSP
UVEN
PWMDIM
C1
ISENSE+
FLT
ISENSE-
100Ω
COMP
R1
RCOMP
CCOMP_HF
ICTRL
RRT
P1
DIMOUT
VCC
R2
RCS_LED
ROVP2
DH
OVP
MAX25612
VCC
RSC
CSN
PGND
VCC
RUVEN2
N1
RCS_FET
PWMDIM
ROVP1
RT
SGND
EP
CCOMP
VIN
www.maximintegrated.com
Maxim Integrated │ 22
MAX25612/MAX25612B
Automotive Synchronous
High Voltage LED Controller
Typical Application Circuits (continued)
High-Side Buck LED Driver Using MAX25612B
COUT
L1
CBST
VCC
VIN
BST
IN
LX
CIN
DL
R3
CSP
UVEN
PWMDIM
R4
ISENSE+
FLT
ISENSE-
P1
100Ω
DIMOUT
VCC
COMP
R1
ROVP1
RCOMP
CCOMP_HF
ICTRL
RRT
RCS_LED
DH
OVP
MAX25612B
R2
RSC
CSN
PGND
VCC
C1
N1
RCS_FET
PWMDIM
VCC
RT
VIN
N2
LED-
SGND
EP
CCOMP
ROVP2
P2
LED-
www.maximintegrated.com
Maxim Integrated │ 23
MAX25612/MAX25612B
Automotive Synchronous
High Voltage LED Controller
Typical Application Circuits (continued)
SEPIC01 LED Driver Using MAX25612B
L1
N2
VIN
CBST
BST
IN
RGN2
LX
CIN
DL
R3
CSP
UVEN
PWMDIM
R4
CSEPIC
RSC
RCS_FET
PWMDIM
CDH
ROVP2
RCS_LED
ISENSE+
FLT
ISENSE-
C1
P1
DH
OVP
MAX25612B
100Ω
DIMOUT
VCC
COMP
R1
RCOMP
CCOMP_HF
ICTRL
RRT
ROVP1
CSN
PGND
VCC
R2
N1
COUT
RT
www.maximintegrated.com
SGND
EP
CCOMP
Maxim Integrated │ 24
MAX25612/MAX25612B
Automotive Synchronous
High Voltage LED Controller
Typical Application Circuits (continued)
SEPIC02 LED Driver Using MAX25612B
L2
L1
VIN
N2
CBST
VCC
BST
IN
CSEPIC
RGN2
LX
CIN
DL
R3
CSP
UVEN
PWMDIM
ISENSE+
FLT
ISENSE-
RCS_LED
100Ω
COMP
R1
RCOMP
CCOMP_HF
ICTRL
RRT
ROVP2
DIMOUT
VCC
R2
CDH
P1
DH
OVP
MAX25612B
C1
ROVP1
CSN
PGND
VCC
VCC
RSC
RCS_FET
PWMDIM
R4
N1
COUT
SGND
RT
CCOMP
EP
Ordering Information
TEMPERATURE
RANGE
PIN-PACKAGE
MAX25612AUP/V+
-40°C to +125°C
20-TSSOP-EP*
MAX25612ATP/VY+
-40°C to +125°C
20-TQFN-EP*
MAX25612BAUP/V+
-40°C to +125°C
20-TSSOP-EP*
MAX25612BATP/VY+
-40°C to +125°C
20-TQFN-EP*
PART
+Denotes a lead (Pb)-free/RoHS-compliant package.
/V denotes an automotive qualified part.
Y = Side-wettable package.
*EP = Exposed pad.
www.maximintegrated.com
Maxim Integrated │ 25
MAX25612/MAX25612B
Automotive Synchronous
High Voltage LED Controller
Revision History
REVISION
NUMBER
REVISION
DATE
PAGES
CHANGED
DESCRIPTION
0
6/19
Initial release
—
1
6/19
Added future-product notation to MAX25612ATP/VY+** in Ordering Information
21
2
12/19
Updated title to add MAX25612B; updated General Description, Benefits and
Features, Electrical Characteristics, Functional Diagrams, Detailed Description,
Applications Information, Typical Application Circuits, and Ordering Information
1–21
3
12/19
Updated Absolute Maximum Ratings, Pin Configurations, and Applications
Information; removed future-product notation from MAX25612ATP/VY+ in Ordering
Information
4
1/20
Removed all remaining future-product notation in Ordering Information
2, 8, 15, 25
25
For pricing, delivery, and ordering information, please visit Maxim Integrated’s online storefront at https://www.maximintegrated.com/en/storefront/storefront.html.
Maxim Integrated cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim Integrated product. No circuit patent licenses
are implied. Maxim Integrated reserves the right to change the circuitry and specifications without notice at any time. The parametric values (min and max limits)
shown in the Electrical Characteristics table are guaranteed. Other parametric values quoted in this data sheet are provided for guidance.
Maxim Integrated and the Maxim Integrated logo are trademarks of Maxim Integrated Products, Inc.
© 2019 Maxim Integrated Products, Inc. │ 26