EVAL45W19VFLYBP7TOBO1

AN_201604_PL52_018 45 W adapter demo board Using the new 800 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM controller Authors: Jared Huntington Stefan Preimel Scope and purpose The demo board described in this application note provides a test platform for the new 800 V CoolMOS™ P7 series of high voltage MOSFETs. The adapter uses ICE2QS03G, a second generation current mode control quasiresonant flyback controller and an IPA80R450P7 800 V CoolMOS™ P7 series power MOSFET. This application note is intended for those that have experience with flyback converter designs and will not go in depth about the overall design process for flyback converters, but will cover specific design aspects for this controller and 800 V CoolMOS™ P7 in charger and adapter applications. It will also look at the overall benefits that the 800 V CoolMOS™ P7 presents for switch mode power supplies. For a detailed introduction on flyback converter design please read Design guide for QR Flyback converter [1]. Intended audience Power supply design engineers Table of contents 1 Description.........................................................................................................................................2 2 Quasi-resonant flyback overview.......................................................................................................3 3 ICE2QS03G functional overview.........................................................................................................4 4 4.1 800 V CoolMOS P7 overview .............................................................................................................5 FullPAK vs. DPAK thermal performance .................................................................................................7 5 5.1 5.2 Design considerations........................................................................................................................9 800 V MOSFET ..........................................................................................................................................9 UVLO circuit ...........................................................................................................................................10 6 6.1 6.2 6.3 6.4 6.5 6.6 6.7 Demo board overview ......................................................................................................................12 Demo board pictures.............................................................................................................................12 Demo board specifications ...................................................................................................................12 Demo board features ............................................................................................................................13 Schematic ..............................................................................................................................................14 BOM with Infineon components in bold...............................................................................................15 PCB layout .............................................................................................................................................16 Transformer construction.....................................................................................................................17 7 7.1 7.2 7.3 7.4 Measurements .................................................................................................................................19 Test measurements under different line and load conditions ............................................................19 Normal operation..................................................................................................................................20 Surge testing..........................................................................................................................................22 Thermal performance under typical operating conditions.................................................................23 8 Conclusion .......................................................................................................................................26 9 References .......................................................................................................................................27 ™ Application Note www.infineon.com/p7 Please read the Important Notice and Warnings at the end of this document Revision 1.0 2016-06-27 45 W adapter demo board Using the new 800 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM controller Description 1 Description This 45 W adapter demo board is intended to be a form, fit and function test platform for charger and adapter applications to show the operation of the 800 V CoolMOS™ P7 as well as the overall controller design. The demo board is designed around a quasi-resonant flyback topology for improved switching losses which allows higher power density designs and lower radiated and conducted emissions. A 45 W universal input isolated flyback demo board with a 19 V output based on the ICE2QS03G controller and the P7 MOSFET is described in this application note and test results are presented. Figure 1 45 W flyback demo board Application Note 2 Revision 1.0 2016-06-27 45 W adapter demo board Using the new 800 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM controller Quasi-resonant flyback overview 2 Quasi-resonant flyback overview The QR flyback offers improved efficiency and EMI performance over the traditional fixed frequency flyback converter by reducing switching losses. This is accomplished by controlling the turn on time of the primary MOSFET (Qpri in Figure 3). In a flyback operating in discontinuous conduction mode (DCM) the energy is first stored in the primary side when the primary MOSFET Qpri is turned on allowing the primary current to ramp up. The primary MOSFET (Qpri) turns off and then the energy stored in the transformer transfers into the secondary side capacitor. The energy that is left in the primary inductance (Lpri) after transferring the energy to the secondary then resonates with the combined output capacitance of the MOSFET(CDS_parasitic) consisting of the MOSFET output capacitance (COSS), stray drain source capacitance from the transformer and layout, and any additional added external drain source capacitance on this node. In a fixed frequency flyback the switch turn on happens regardless of the VDS voltage from the MOSFET drain to source. If switching occurs at a higher VDS (Figure 2) this corresponds to more switching losses (EOSS losses). The QR flyback waits to turn on Qpri until the VDS voltage reaches the minimum possible voltage shown in Figure 2 and then turns on the MOSFET. _ = 0.5 Since the turn on switching losses are a function of V2 (as shown above), this drastically reduces the overall system switching losses. This has the added benefit of lowering the amount of switched energy which helps reduce switching noise from the converter, resulting in lower radiated and conducted emissions. 800 V CoolMOS™ P7 technology helps improve performance in a QR flyback by allowing an increase in the reflected voltage. This increase in reflected voltage increases the energy stored in the magnetizing inductance during the DCM period which allows switching at an even lower VDS voltage allowing even lower switching losses. The 800 V CoolMOS™ P7 also has a lower gate charge (QG) and output capacitance (COSS) which help to further reduce the switching losses of the MOSFET. MOSFET Turn ON MOSFET Turn ON Figure 2 Fixed frequency flyback primary MOSFET drain source waveform (left) vs. a quasi-resonant flyback primary MOSFET drain source waveform (right). Figure 3 Simplified flyback schematic Application Note 3 Revision 1.0 2016-06-27 45 W adapter demo board Using the new 800 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM controller ICE2QS03G functional overview 3 ICE2QS03G functional overview The PWM controller ICE2QS03G is a second generation quasi-resonant flyback controller IC developed by Infineon Technologies. The typical applications include TV-sets, DVD-players, set-top boxes, netbook adapters, home audio, and printer applications. This controller implements switching at the lowest ringing voltage and also includes pulse skipping at light loads for maximum efficiency across a wide range of loads. Figure 4 ICE2QS03G pinout Table 1 ICE2QS03G pin description Pin Name Description 1 Zero Crossing (ZC) Detects the minimum trough (valley) voltage for turn on for the primary switch turn on time 2 Feedback (FB) Voltage feedback for output regulation 3 Current Sense (CS) Primary side current sense for short circuit protection and current mode control 4 Gate drive output (GATE) MOSFET gate driver pin 5 High Voltage (HV) Connects to the bus voltage for the initial startup through the high voltage startup cell 6 No Connect (NC) No connection 7 Power supply (VCC) Positive IC for the power supply 8 Ground (GND) Controller ground Application Note 4 Revision 1.0 2016-06-27 45 W adapter demo board Using the new 800 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM controller 800 V CoolMOS™ P7 overview ™ 4 800 V CoolMOS P7 overview The 800 V P7 family of MOSFETs provides several advantages for flyback converters. The 800 V CoolMOS™ P7 offers a cost reduction for the same RDS(on) device when compared to the C3 series with an improvement in performance. The switching losses of the devices are lowered due to reduced device parasitic elements such as COSS and QG. These improvements diminish as the MOSFET drain source voltage during turn on gets lower. The greatest reduction in switching losses is seen at higher drain source switching voltages and at low output powers due to the improved output parasitic elements (COSS). The reduction in overall switching losses of the device allows moving to a higher RDS(ON) to further reduce the BOM cost or allow increasing the power density of the power supply design. PSpice models of the P7 800 V MOSFETs are provided on the Infineon website. These models have been fitted with measurements of the devices and provide a high level of accuracy. Below, Figure 5 shows the difference between the Infineon 45 W adapter measured waveforms and the simulated waveforms. These models can be used to better understand the loss mechanisms that are responsible for power dissipation in the primary MOSFET of the flyback converter and help optimize designs. Figure 5 Simulated switching vs. measured switching at 230 VAC operation. Application Note 5 Revision 1.0 2016-06-27 45 W adapter demo board Using the new 800 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM controller 800 V CoolMOS™ P7 overview Using the P7 PSpice models for the 45 W adapter, we can look at the losses occurring in the MOSFET of the 45 W adapter flyback converter. The figure below shows the breakdown of the MOSFET turn on losses, turn off losses, and conduction losses. As shown in Figure 6, the switching losses of the MOSFET are a more significant loss contributor at high line. The figure below shows the breakdown of MOSFET turn on losses, turn off losses, and conduction losses at high line. At low line the conduction losses (RDS(ON)) dominate and the improvement in COSS does not make such a large improvement. The IPA80R1K4P7 MOSFET offers lower switching losses which give a total power savings of 15.6 mW at high line over the original C3 series SPA06N80C3 - with a large reduction in cost. 160 140 Power Loss (mW) 120 100 80 60 Gate 40 Turn-Off Conduction 20 Turn-On 0 SPA11N80C3 High line IPA80R1K4P7 High line MOSFET and Line Voltage Figure 6 Switching losses of SPA11N80C3 vs. IPA80R450P7 at 230 VAC input 0,1 0,0 -0,1 5 10 15 20 25 30 35 40 45 Efficiency [%] -0,2 -0,3 SPA11N80C3 Comp. 1 Comp. 2 IPA80R450P7 -0,4 -0,5 -0,6 -0,7 -0,8 -0,9 Figure 7 Pout [W] IPA80R450P7 set as the efficiency reference measured in the 45 W adapter in comparison with the SPA11N80C3 and a competitor’s equivalent component at 230 VAC. Application Note 6 Revision 1.0 2016-06-27 45 W adapter demo board Using the new 800 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM controller 800 V CoolMOS™ P7 overview It can be seen in Figure 7 that the IPA80R450P7 has improved performance when compared to the C3 series of MOSFETs and two of our competitors latest generation of MOSFETs. At light loads the switching losses are dominant and it can be seen that the P7 switching performance is much better. *Simulations and modeling done by Stefano De Filippis 4.1 FullPAK vs. DPAK thermal performance The DPAK MOSFET package is ideal for low cost applications such as charger and adapters. The thermal performance is slightly lower than the TO-220 FullPAK (TO-220FP), but it has a lower package cost allowing for overall BOM savings. The DPAK also has a smaller form factor allowing for higher power density designs and the SMD placement to be used. In the Infineon 45 W adapter allows a TO-220FP or a DPAK footprint. The two packages were tested on the same board under full load (45 W) at 120 VAC and 230 VAC in a 25°C ambient to show the thermal performance difference between the two packages. Table 2 FullPAK vs. DPAK thermal performance (25°C ambient) Test conditions IPD80R450P7 IPA80R450P7 DPAK case temp. rise(°C) FullPAK case temp. rise(°C) DPAK temp. increase from FullPAK(°C) 45 W, 120 VAC, 60 Hz, 56.8°C 27.7°C 29.1°C 45 W, 230 VAC, 50 Hz 51.8°C 25.9°C 25.9°C In the infrared thermal images below, the primary MOSFET Q1 is called out in the black boxes. It can be seen that the temperature of the DPAK is 29.1°C higher than the FullPAK at 120 VAC. Most of this temperature difference is due to the fact that the MOSFET (when placed on the bottom side of the printed circuit board) receives some heating from the surrounding components (the snubber and transformer). Figure 10 shows the DPAK footprint temperature rise while the power supply is operating using the FullPAK. This increases the package temperature in addition to the difference in package thermal resistance leading to a higher temperature. The hottest components on the board are the snubber network resistors, R22 and R23, shown below in Figure 8. Table 3 takes the DPAK thermal rise and removes the PCB temperature rise of the footprint with the FullPAK in place. The DPAK temperature is then overcorrected due to some heating of the PCB from the FullPAK causing a higher footprint temperature. Table 3 FullPAK vs. DPAK thermal performance normalized for PCB rise (25 °C ambient) IPD80R450P7 DPAK case temp. rise(°C) IPD80R450P7 DPAK footprint temp. rise(°C) DPAK case temp. increase from PCB temp. (°C) DPAK temp. increase from FullPAK (°C) 45 W, 120 VAC, 60 Hz 56.8°C 30.1°C 26.7°C -1.0°C 45 W, 230 VAC, 50 Hz 51.8°C 29.5°C 22.3°C -3.6°C Test conditions A 50°C ambient would push the total DPAK temperature up to 106.8°C in this specific design. Depending on the required ambient operating conditions the DPAK package in this application would require a larger copper area or lower output power in order to have enough thermal margins under worst case conditions. The DPAK package can be used to give space, cost, and assembly savings, but the additional heating of surrounding components and reduced thermal performance needs to be considered when switching from a FullPAK to a DPAK package. Application Note 7 Revision 1.0 2016-06-27 45 W adapter demo board Using the new 800 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM controller 800 V CoolMOS™ P7 overview Q1 Figure 8 45 W adapter bottom using DPAK at 45 W load and 120 VAC. Q1 shown above in the black box is the flyback converter primary MOSFET. Note the MOSFET Q1 is receiving some heating from the surrounding components which contributes to the higher DPAK temperature. Q1 Figure 9 45 W adapter top using FullPAK at 45 W load and 120 VAC. Q1 shown above in the black box is the flyback converter primary MOSFET. Q1 Figure 10 45 W adapter bottom using FullPAK at 45 W load and 100 VAC. The DPAK footprint is shown and the local PCB temperature rise can be seen which further increases the DPAK temperature. Application Note 8 Revision 1.0 2016-06-27 45 W adapter demo board Using the new 800 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM controller Design considerations 5 Design considerations 5.1 800 V MOSFET The 800 V CoolMOS™ P7 provides several benefits for charger and adapter applications. An 800 V breakdown voltage allows a higher combination of bus voltage, reflected voltage, and snubber voltage than can be achieved with a 600 V or 650 V device. By allowing a higher reflected voltage and snubber voltage the system power losses can be reduced while maintaining higher breakdown voltage margins. Figure 11 MOSFET VDS during turn off in the Infineon 45 W adapter In this specific design the reflected voltage was increased from the Infineon 35 W adaptor which used a 600 V device. This section will compare the Infineon 35 W adapter design using a 600 V MOSFET with the Infineon 45 W adapter using an 800 V MOSFET to show the difference in performance between the two designs. The reflected voltage determines the trough (valley) voltage during DCM ringing where the switch turns on in the QR flyback converter. By allowing a higher reflected voltage there is a resulting lower trough in the ringing waveform. This allows the converter to switch at a lower VDS voltage and reduce the system's switching losses especially at high line (265 VAC) operation. _ = 0.5 Table 4 = _ + Parameter Symbol 600 V design 800 V P7 design Transformer primary turns NP 66 turns 87 turns Transformer secondary turns NS 11 turns 8 turns Output voltage Voutput 19 V 19 V Diode forward voltage Vforward 0.55 V 0.4 V Transformer reflected voltage Vreflected 117 V 211 V Application Note 9 Revision 1.0 2016-06-27 45 W adapter demo board Using the new 800 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM controller Design considerations The primary side resistor, capacitor, and diode (RCD) snubber network resistor power dissipation was reduced allowing the snubber voltage to reach a higher level and lowering the amount of energy that is dissipated in the snubber resistor. This especially comes into effect at very light load operation. = Table 5 +2 = − Parameter Symbol 600 V design 800 V P7 design Leakage inductance Lleakage 25 µH 25 µH Peak primary current under load at high line Ipri 0.43 A 0.48 A Snubber resistor Rsnubber 54 kΩ 300 kΩ Switching period Ts 28.6 µs 28.6 µs Snubber voltage Vsnubber 40.1 V 127 V Increasing the reflected voltage and lowering the amount of energy that is dissipated in the snubber lowers the overall system losses and would not be possible with a 600 V MOSFET as shown in Table 5. Even with increasing the reflected voltage by 94 V and increasing the snubber voltage by 30.4 V we still have an increase in margin from the MOSFET breakdown voltage. In this new design the margin has increased from 12% to 15% even with increasing the VDS voltages. This allows for the design of flyback converters running from higher input bus voltages or those that need margin for abnormal conditions such as surge. Table 6 Parameter Symbol 600 V design 800 V P7 design Primary bus voltage @265 VAC Vbus 373 V 373 V Reflected voltage Vreflected 117 V 211 V Snubber voltage Vsnubber 40.1 V 70.5 V Drain source voltage maximum VDS_max 526 V 622 V Margin from breakdown voltage VDS_margin 12 % 15 % 5.2 UVLO circuit The Under Voltage Lock Out (UVLO) circuit provides a mechanism to shut down the power supply when the AC line input voltage is lower than the specified voltage range. The UVLO event is detected by sensing the voltage level at U2’s (TL431) REF pin (VREF_typ = 2.5 V) through the voltage divider resistors (R12, R13, R14, and R17 in Figure 12) from the bulk capacitor C1. Q2 acts as a switch to enter or leave UVLO mode by controlling the FB pin voltage. Q3, together with R17, acts as voltage hysteresis for the UVLO circuit and U2 (TL431) as a comparator. The system enters the UVLO mode by controlling the FB pin voltage of U1 to 0 V (when the voltage input level goes back to input voltage range), VREF increases to 2.5 V (then switches Q2 and Q3 off) and Vcc hits 18 V, the UVLO mode is released. The calculation for the UVLO circuit is shown below: VREF= 2.5 V R12 = 4.99 MΩ R13 = 4.99 MΩ R14 = 330 kΩ R17 = 681 kΩ _ Application Note = ( 12 + 13 + 14) 14 10 Revision 1.0 2016-06-27 45 W adapter demo board Using the new 800 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM controller Design considerations 14 17 + 12 + 13 14 + 17 14 17 14 + 17 _ = _ = 77.8 = 114.3 _ The 'enter UVLO' threshold is set at 77.8 to allow for the BUS capacitance voltage to droop under 90 VAC at full load operation with some margin to avoid false triggering. Figure 12 Power supply status vs. AC input voltage showing the hysteretic behavior of the UVLO circuit. Application Note 11 Revision 1.0 2016-06-27 45 W adapter demo board Using the new 800 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM controller Demo board overview 6 Demo board overview 6.1 Demo board pictures Q1 IPA80R450P7 Figure 13 Top side of 45 W IFX adapter with a TO220 FullPAK populated Q2, Q3 – 2N7002 IC1 – ICE2QS03G Figure 14 Bottom side of 45 W IFX adapter highlighting Infineon components. The Q1 DPAK is not populated on the bottom side since the board is populated with a FullPAK device. 6.2 Demo board specifications Table 7 Section Parameter Specification Input ratings Input voltage 90 VAC – 265 VAC Input frequency 47 Hz – 63 Hz Input current at 100 VAC, 45 W 0.82 A maximum Power factor 0.55 @100 VAC 0.37 @265 VAC Peak efficiency 230 VAC, 45 W Peak efficiency 120 VAC, 45 W 91.4% 89.3% Surge 2 kV IEC61000-4-5 Application Note 12 Revision 1.0 2016-06-27 45 W adapter demo board Using the new 800 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM controller Demo board overview Section Parameter Specification Output ratings Nominal output voltage 19.0 V Tolerance 2% Output current 2.4 A Output power 45 W Line regulation 0.5% Load regulation 0.5% Output ripple 100 mVPP Quiescent power draw 42 mW @100 VAC 94 mW @265 VAC Switching frequency 25 – 60 kHz Mechanical Dimensions Length: 10.0 cm (3.94 in.) Width: 3.7 cm (1.46 in.) Height: 2.6 cm (1.02 in.) Environmental Ambient operating temperature -25°C to 50°C 6.3 Demo board features • Fold back point protection - For a quasi-resonant flyback converter, the maximum possible output power is • • • • • increased when a constant current limit value is used across the entire mains input voltage range. This is usually not desired as this will increase the cost of the transformer and output diode in the case of output over power conditions. The internal fold back protection is implemented to adjust the VCS voltage limit according to the bus voltage. Here, the input line voltage is sensed using the current flowing out of the ZC pin, during the MOSFET on-time. As the result, the maximum current limit adjusts with the AC line voltage. VCC over voltage and under voltage protection - During normal operation, the Vcc voltage is continuously monitored. When the Vcc voltage increases to VVCC OVP or Vcc voltage falls below the under voltage lock out level VVCC off, the IC will enter into auto restart mode. Over load/open loop protection - In the case of an open control loop, the feedback voltage is pulled up with an internal block. After a fixed blanking time, the IC enters into auto restart mode. In case of a secondary short-circuit or overload, the regulation voltage VFB will also be pulled up, the same protection is applied and the IC will auto restart. Adjustable output overvoltage protection - During the off-time of the power switch, the voltage at the zerocrossing pin, ZC, is monitored for output overvoltage detection. If the voltage is higher than the preset threshold 3.7 V for a preset period of 100 μs, the IC is latched off. Auto restart for over temperature protection - The IC has a built-in over temperature protection function. When the controller’s temperature reaches 140 °C, the IC will shut down the switch and enters into auto restart. This can protect the power MOSFET from overheating. Short winding protection - The source current of the MOSFET is sensed via external resistors, R15 and R16. If the voltage at the current sensing pin is higher than the preset threshold VCSSW of 1.68 V during the on-time of the power switch, the IC is latched off. This constitutes a short winding protection. To avoid an accidental latch off, a spike blanking time of 190 ns is integrated in the output of internal comparator. Application Note 13 Revision 1.0 2016-06-27 45 W adapter demo board Using the new 800 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM controller Demo board overview 6.4 Figure 15 Schematic 45 W adapter schematic Application Note 14 Revision 1.0 2016-06-27 45 W adapter demo board Using the new 800 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM controller Demo board overview 6.5 BOM with Infineon components in bold Table 8 Reference Description Part number Manufacturer C1 Electrolytic capacitor, 82 uF, 20%, 400 V EKXG401ELL820MM25S United Chemi-Con C2 Electrolytic capacitor, 470 uF, 20%, 25 V EKZE250ELL471MJ16S United Chemi-Con C3 Electrolytic capacitor, 100 uF, 20%, 25 V EEU-FR1E101 Panasonic C4 Capacitor ceramic, 22 nF, X7R, 50 V, CAP0805W VJ0805Y223KNAAO Vishay C5, C20 Capacitor ceramic, 100 nF, X7R, 50 V, CAP0805W C2012X7R2A104K125AA TDK C6 C_ELKO, 47uF, 20%, 25V, C_Aluminium Elektrolyt 5 mm UPM1E470MED Nichicon C7 Foil capacitor, 330 nF X2, 20%, 310 VAC, C_Foil 15 mm - V2 R463I33305002K Kemet C10 Capacitor ceramic, 1nF, NP0, 50 V, CAP0805W CGA4C2C0G1H102J060AA TDK C11 Capacitor Y2, 2.2 nF, Y2, 300 V, CAP-DISC 7.5 mm AY2222M35Y5US63L7 Vishay C13 Capacitor ceramic, 4.7 nF, NPO, 630 V, CAP1206W C1206C472JBGACTU Kemet C15 Capacitor ceramic, 220 nF, X7R, 25 V, CAP0805W C2012X7R1H224K125AA TDK C16 Capacitor ceramic, 100pF, NP0, 100 V, CAP0805W CGA4C2C0G2A101J060AA TDK C17, C21, C22 Capacitor ceramic, 2.2 uF, X7R, 25 V, CAP1206W C3216X7R1E225K160AA TDK C18, C19 220pF/250 VAC, 220pF, 250 Vac, C075045X100 VY2221K29Y5SS63V0 Vishay C24 Capacitor ceramic, 100 pF, NPO, 630 V, CAP1206W CGA5C4C0G2J101J060AA TDK CON1 ST-04A, IEC C6 AC Connector, ST-A04 6160.0003 Schurter D1 Diode, US1K-E3/61T, 600V, SMA US1K-E3/61T Vishay D2 Diode, NTST30100SG, 100V, TO220_standing NTST30100SG OnSemi D3 2KBP06M, 2KBP06M, 600V, KBPM 2KBP06M-E4/51 Vishay D4 Diode, BAS21-03W, 200V, SOD323 BAS21HT1G OnSemi F1 T2, 2 A, 250 Vac, Fuse small 40012000440 Littelfuse H1 Heatsink, TO-220 Heatsink 577202B00000G Aavid thermalloy H2 Hardware, Screw, M3, 8 mm M38 PRSTMCZ100- DURATOOL H3 Hardware, Nut, A2, M3 M3- HFA2-S100- DURATOOL H4 Hardware, insulator, Insert, 0.15 mm, 19 x 13 mm SPK10-0.006-00-54 Bergquist H5 Hardware, insulator, washer, TO220 insulating washer 7721-7PPSG AAVID THERMALLOY Application Note 15 Revision 1.0 2016-06-27 45 W adapter demo board Using the new 800 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM controller Demo board overview Reference Description Part number Manufacturer H6 Cable assembly 172-4202 Memory Protection Devices, Inc. IC1 QR PWM controller ICE2QS03G Infineon IC12 VOL617A-2, VOL617A-2, LSOP 4pin VOL617A-2X001T Vishay L1 Choke, 1.0 uH, 20%, INDUCTOR 4 u7 4,2 A 7447462010 Wurth L2 Inductance, 10 mH, Inductor common mode small 744821110 Wurth Q1 NMOS, IPA80R450P7, 800V, TO220FP IPA80R450P7 Infineon Q2, Q3 NMOS, 2N7002, 60V, SOT23 2N7002 Infineon R1 Resistor, 0R, 1%, RES0805R CRCW08050000Z0EA Vishay R2 Resistor, 39k2, 1%, RES0805R ERJ6ENF3922V Panasonic R3 Resistor, 4k99, 1%, RES0805R CRCW08054K99FKEA Vishay R4 Resistor, 33k2, 1%, RES0805R CRCW080533K2FKEA Vishay R5 Resistor, 100k, 1%, RES0805R CRCW0805100KFKEA Vishay R6, R8, R11 Resistor, 10k, 1%, RES0805R CRCW080510K0FKEA Vishay R15, R7 Resistor, 1R, 1%, RES1206W CRCW12061R00FKEA Vishay R10 Resistor, 2k, 1%, RES0805R CRCW08052K00FKEA Vishay R12, R13 Resistor, 4.99M, 1%, RES1206W CRCW12064M99FKEB Vishay R14 Resistor, 330k, 1%, RES0805R CRCW0805330KFKEA Vishay R16 Resistor, 1R5, 1%, RES1206W CRCW12061R50JNEAIF Vishay R17 Resistor, 681k, 1%, RES0805R CRCW0805681KFKEA Vishay R18 Resistor, 51k1, 1%, RES0805R ERJ6ENF5112V Panasonic R19, R24 Resistor, 200k, 1%, RES0805R CRCW0805200KFKEA Vishay R22, R23 Resistor, 150k, 1%, RES1206W CRCW1206150KFKEA Vishay R25 Resistor, 10R, 1%, RES1206W CRCW120610R0FKEA Vishay R27 Resistor, 27R, 1%, RES1206W CRCW120627R0FKEA Vishay 6.6 Figure 16 PCB layout Board layout top Application Note 16 Revision 1.0 2016-06-27 45 W adapter demo board Using the new 800 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM controller Demo board overview Figure 17 Board layout bottom The PCB was designed using Altium Designer 16. Schematic and board files are available on request. 6.7 Transformer construction The transformer for the 45 W adapter was built by I.C.E. Transformers: http://www.icetransformers.com/ Table 9 Transformer specification Manufacturer I.C.E. Transformers Core size RM10 Core material 3C95 Bobbin 8 pin RM10 vertical Primary inductance 1500 µH measured from pin 1 to pin 3 @10 kHz Leakage inductance < 25 µH measured from pin 1 to pin 3 with all other pins shorted @10 kHz *100% of components are Hi-Pot tested to 4.2 kV primary to secondary for 1 minute Figure 18 Transformer windings stackup Application Note 17 Revision 1.0 2016-06-27 45 W adapter demo board Using the new 800 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM controller Demo board overview 1. 2. 3. 4. 5. 6. 7. S- in red tube, S+ in black tube S- length 25 mm, solder length 5 mm S+ length 30 mm, solder length 5 mm Cut pin 4, pin 2, core clip PCB mount pins, and secondary pins. Add a flux band of 8mm copper foil with 2 layers of tape and 3mm of cuffing on each side. Add around the core with the tape side facing out. Using ɸ0.35 mm solder to pin 5. Vacuum varnish the entire assembly. Cut off core clamp pins Table 10 Transformer windings stackup Name Start Stop Turns Wire Layer Method P1 1 2 58 1 x ɸ0.35 mm primary tight S1 S- S+ 13 2 x ɸ0.5 mm triple insulated secondary tight P2 2 3 29 1 x ɸ0.35 mm primary tight P3 5 6 10 1 x ɸ0.15 mm, with margin tape auxiliary evenly spaced 2 tape T1 Application Note 18 Revision 1.0 2016-06-27 45 W adapter demo board Using the new 800 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM controller Measurements 7 Measurements 7.1 Test measurements under different line and load conditions 92,0 89,9 90,0 89,0 87,0 91,3 91,3 91,2 91,0 91,1 90,6 89,6 89,2 88,7 88,4 88,0 Efficiency (%) 91,1 91,2 90,9 90,5 91,0 86,6 86,0 Infineon 35W Adapter, 230VAC 50Hz, IPD60R600P6 85,0 84,0 Infineon 45W Adapter, 230VAC 50Hz, IPD80R450P7 83,2 83,0 82,0 81,0 80,0 0 Figure 19 5 10 15 20 25 Output Power (W) 30 35 40 45 45 W adapter efficiency at 230 VAC using IPA80R450P7 when compared to Infineon 35 W adapter using IPD60R600P6 92,0 90,6 91,0 90,0 89,3 89,0 89,8 90,7 90,4 90,3 90,3 89,1 89,1 90,2 90,2 89,8 89,5 89,7 88,5 88,0 Efficiency (%) 90,8 87,0 Infineon 35W Adapter, 120VAC 60Hz, IPD60R600P6 86,0 85,0 Infineon 45W Adapter, 120VAC 60Hz, IPD80R450P7 84,0 83,0 82,0 81,0 80,7 80,0 0 Figure 20 5 10 15 20 25 Output Power (W) 30 35 40 45 45 W Adapter efficiency at 120 VAC using IPA80R450P7 when compared to Infineon 35 W adapter using IPD60R600P6 Application Note 19 Revision 1.0 2016-06-27 45 W adapter demo board Using the new 800 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM controller Measurements 7.2 Figure 21 Normal operation Low line (100 VAC), no load, The ICE2QS03G is operating in burst mode to minimize the idle power consumption. The burst mode pulse train shown above occurs every 33.8 ms with the main switch inactive in the period between pulse trains to lower light load power consumption. CH1 (Yellow): Q1 VDS CH2 (Cyan): Q1 IDS CH3 (Magenta): Q1 VGS Figure 22 High line (265 VAC), no load, The ICE2QS03G is operating in burst mode to minimize idle power consumption. The burst mode pulse train shown above occurs every 33.8 ms. CH1 (Yellow): Q1 VDS CH2 (Cyan): Q1 IDS CH3 (Magenta): Q1 VGS Application Note 20 Revision 1.0 2016-06-27 45 W adapter demo board Using the new 800 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM controller Measurements Figure 23 Low line (100VAC), Full load (45 W) showing normal full load operation of the adapter. This is the worst case peak current that the primary MOSFET Q1 will encounter during normal operation. CH1 (Yellow): Q1 VDS CH2 (Cyan): Q1 IDS CH3 (Magenta): Q1 VGS Figure 24 High line (265 VAC), Full load (45 W) showing normal full load operation of the adapter. This is the worst case peak drain source voltage that the MOSFET will see under normal operating conditions. CH1 (Yellow): Q1 VDS CH2 (Cyan): Q1 IDS CH3 (Magenta): Q1 VGS Application Note 21 Revision 1.0 2016-06-27 45 W adapter demo board Using the new 800 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM controller Measurements 7.3 Surge testing In order for the power supply to be robust enough for abnormal line conditions such as lightning strikes or failures of other electronics on the line, it needs to survive surge testing. The 45 W power supply was tested to the 2 kV EN61000 surge conditions and still had 96 V of margin under worst case conditions for the MOSFET VDS. Table 11 EN61000 surge requirements Level Surge voltage L-N (kV) Surge voltage L-PE, N-PE (kV) Class 1 protected environment 0.25 0.5 Class 2 electrical cables are separated 0.5 1.0 1.0 2.0 2.0 4.0 Class 3 electrical cables run in parallel Class 4 outdoor Figure 25 IEC61000 2 kV surge test was performed on the adapter while operating under full load (45 W). The highest voltage that was reached across the Q1 VDS was 704 V. The surge event can be seen on CH1 when the VBUS rapidly rises. The bus capacitor (C1) and line filter values are critical for determining the peak surge voltage. CH1 (Yellow): VC1, VBUS CH2 (Cyan): CH3 (Magenta): CH4 (Green): Application Note Q1 VDS Q1 VGS Q1 IDS 22 Revision 1.0 2016-06-27 45 W adapter demo board Using the new 800 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM controller Measurements 7.4 Thermal performance under typical operating conditions Q1 Figure 26 100 VAC input, full load, top side. The line filter and bridge rectifier are hottest at this point due to higher AC input currents. Figure 27 100 VAC input, full load, bottom side. Q1 Figure 28 120 VAC input, full load, top side. The line filter and bridge rectifier are hotter at this point due to the higher primary side current. Application Note 23 Revision 1.0 2016-06-27 45 W adapter demo board Using the new 800 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM controller Measurements Figure 29 120 VAC input, full load, bottom side. Q1 Figure 30 230 VAC input, full load, top side. The primary MOSFET (Q1) is cooler at 230 VAC because conduction losses become less dominant with lower primary side peak currents. Figure 31 230 VAC input, full load, bottom side. Application Note 24 Revision 1.0 2016-06-27 45 W adapter demo board Using the new 800 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM controller Measurements Q1 Figure 32 265 VAC input, full load, top side. The MOSFET is cooler at 230 VAC because conduction losses become less dominant with the lower primary peak currents. Figure 33 265 VAC input, full load, bottom side. Application Note 25 Revision 1.0 2016-06-27 45 W adapter demo board Using the new 800 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM controller Conclusion 8 Conclusion The 800 V P7 series of CoolMOS™ MOSFETs offer an improvement in switching loss performance over the 800 V C3 MOSFETs. By switching from a 600 V to an 800 V device the performance of the converter can be further improved in flyback topologies by allowing a higher reflected voltage and snubber voltage, thus further reducing the converter losses while still allowing for an increased MOSFET drain source voltage margin. This allow for designs that improve overall system efficiency while reducing overall BOM cost. In addition, the CoolMOS™ P7 offers a new best-in-class RDS(ON). In DPAK a RDS(ON) of 280 mΩ is available, over 50% lower than the nearest 800 V MOSFET competitor. This new benchmark enables higher power density designs, BOM savings, and lower assembly costs. Application Note 26 Revision 1.0 2016-06-27 45 W adapter demo board Using the new 800 V CoolMOS™ P7 and ICE2QS03G quasi-resonant PWM controller References 9 References [1] Design Guide for QR Flyback Converter [2] IPA80R450P7 data sheet, 800 V CoolMOS™ P7 Power Transistor [3] ICE2QS03G data sheet, Infineon Technologies AG [4] 2N7002 data sheet, Infineon Technologies AG [5] ICE2QS03G design guide. [ANPS0027] [6] Converter Design Using the Quasi-Resonant PWM Controller ICE2QS03, Infineon Technologies AG, 2006. [ANPS0003] [7] Design tips for flyback converters using the Quasi-Resonant PWM controller ICE2QS01, Infineon Technologies, 2006. [ANPS0005] [8] Determine the switching frequency of Quasi-Resonant Flyback converters designed with ICE2QS01, Infineon Technologies, 2006. [ANPS0004] [9] 36W Evaluation Board with Quasi-Resonant PWM Controller ICE2QS03G, 2011. [AN-PS0040] Revision history Major changes since the last revision Page or reference Application Note Description of change 27 Revision 1.0 2016-06-27 Trademarks of Infineon Technologies AG AURIX™, C166™, CanPAK™, CIPOS™, CoolGaN™, CoolMOS™, CoolSET™, CoolSiC™, CORECONTROL™, CROSSAVE™, DAVE™, DI-POL™, DrBlade™, EasyPIM™, EconoBRIDGE™, EconoDUAL™, EconoPACK™, EconoPIM™, EiceDRIVER™, eupec™, FCOS™, HITFET™, HybridPACK™, Infineon™, ISOFACE™, IsoPACK™, i-Wafer™, MIPAQ™, ModSTACK™, my-d™, NovalithIC™, OmniTune™, OPTIGA™, OptiMOS™, ORIGA™, POWERCODE™, PRIMARION™, PrimePACK™, PrimeSTACK™, PROFET™, PRO-SIL™, RASIC™, REAL3™, ReverSave™, SatRIC™, SIEGET™, SIPMOS™, SmartLEWIS™, SOLID FLASH™, SPOC™, TEMPFET™, thinQ!™, TRENCHSTOP™, TriCore™. Trademarks updated August 2015 Other Trademarks All referenced product or service names and trademarks are the property of their respective owners. Edition 2016-06-27 Published by Infineon Technologies AG 81726 München, Germany © 2016 Infineon Technologies AG. All Rights Reserved. Do you have a question about this document? Email: erratum@infineon.com Document reference ifx1 IMPORTANT NOTICE The information given in this document shall in no event be regarded as a guarantee of conditions or characteristics (“Beschaffenheitsgarantie”) . With respect to any examples, hints or any typical values stated herein and/or any information regarding the application of the product, Infineon Technologies hereby disclaims any and all warranties and liabilities of any kind, including without limitation warranties of non-infringement of intellectual property rights of any third party. In addition, any information given in this document is subject to customer’s compliance with its obligations stated in this document and any applicable legal requirements, norms and standards concerning customer’s products and any use of the product of Infineon Technologies in customer’s applications. The data contained in this document is exclusively intended for technically trained staff. It is the responsibility of customer’s technical departments to evaluate the suitability of the product for the intended application and the completeness of the product information given in this document with respect to such application. For further information on the product, technology, delivery terms and conditions and prices please contact your nearest Infineon Technologies office (www.infineon.com). WARNINGS Due to technical requirements products may contain dangerous substances. For information on the types in question please contact your nearest Infineon Technologies office. Except as otherwise explicitly approved by Infineon Technologies in a written document signed by authorized representatives of Infineon Technologies, Infineon Technologies’ products may not be used in any applications where a failure of the product or any consequences of the use thereof can reasonably be expected to result in personal injury.