LT3971EDDTRPBF

LT3971 38V, 1.2A, 2MHz Step-Down Regulator with 2.8µA Quiescent Current FEATURES n n n n n n n n n n n n n n n DESCRIPTION The LT®3971 is an adjustable frequency monolithic buck switching regulator that accepts a wide input voltage range up to 38V. Low quiescent current design consumes only 2.8μA of supply current while regulating with no load. Low ripple Burst Mode operation maintains high efficiency at low output currents while keeping the output ripple below 15mV in a typical application. An internally compensated current mode topology is used for fast transient response and good loop stability. A high efficiency 0.33Ω switch is included on the die along with a boost Schottky diode and the necessary oscillator, control and logic circuitry. An accurate 1V threshold enable pin can be used to shut down the LT3971, reducing the input supply current to 700nA. A capacitor on the SS pin provides a controlled inrush current (soft-start). A power good flag signals when VOUT reaches 91% of the programmed output voltage. The LT3971 is available in small 10-pin MSOP and 3mm × 3mm DFN packages with exposed pads for low thermal resistance. L, LT, LTC, LTM, Linear Technology, the Linear logo and Burst Mode are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. Ultralow Quiescent Current: 2.8μA IQ Regulating 12VIN to 3.3VOUT Low Ripple Burst Mode® Operation: Output Ripple < 15mVP-P Wide Input Voltage Range: 4.3V to 38V 1.2A Maximum Output Current Adjustable Switching Frequency: 200kHz to 2MHz Synchronizable Between 250kHz to 2MHz Fast Transient Response Accurate 1V Enable Pin Threshold Low Shutdown Current: IQ = 700nA Power Good Flag Soft-Start Capability Internal Compensation Saturating Switch Design: 0.33Ω On-Resistance Output Voltage: 1.19V to 30V Small Thermally Enhanced 10-Pin MSOP Package and (3mm × 3mm) DFN Packages APPLICATIONS n n n Automotive Battery Regulation Power for Portable Products Industrial Supplies TYPICAL APPLICATION 3.3V Step Down Converter VIN 4.5V TO 38V VIN OFF ON EN PG 4.7μF SS RT BD 10pF 1.78M 49.9k SYNC GND FB 1M 3480 TA01 No Load Supply Current 4.0 3.5 4.7μH VOUT 3.3V 1.2A INPUT CURRENT (μA) 0.47μF 3.0 2.5 2.0 1.5 22μF 1.0 0 10 20 30 INPUT VOLTAGE (V) 40 3971 TA01b BOOST SW LT3971 3971f 1 LT3971 ABSOLUTE MAXIMUM RATINGS (Note 1) VIN, EN Voltage .........................................................38V BOOST Pin Voltage ...................................................55V BOOST Pin Above SW Pin.........................................30V FB, RT, SYNC, SS Voltage ...........................................6V PG, BD Voltage .........................................................30V Boost Diode Current....................................................1A Operating Junction Temperature Range (Note 2) LT3971E ............................................. –40°C to 125°C LT3971I .............................................. –40°C to 125°C Storage Temperature Range .............. –65°C to 150°C Lead Temperature (Soldering, 10 sec) (MSE Only) ....................................................... 300°C PIN CONFIGURATION TOP VIEW BD BOOST SW VIN EN 1 2 3 4 5 11 GND 10 SYNC 9 PG 8 RT 7 SS 6 FB TOP VIEW BD BOOST SW VIN EN 1 2 3 4 5 11 GND 10 9 8 7 6 SYNC PG RT SS FB DD PACKAGE 10-LEAD (3mm 3mm) PLASTIC DFN θJA = 45°C, θJC = 10°C/W EXPOSED PAD (PIN 11) IS GND, MUST BE SOLDERED TO PCB MSE PACKAGE 10-LEAD PLASTIC MSOP θJA = 45°C, θJC = 10°C/W EXPOSED PAD (PIN 11) IS GND, MUST BE SOLDERED TO PCB ORDER INFORMATION LEAD FREE FINISH LT3971EDD#PBF LT3971IDD#PBF LT3971EMSE#PBF LT3971IMSE#PBF TAPE AND REEL LT3971EDD#TRPBF LT3971IDD#TRPBF LT3971EMSE#TRPBF LT3971IMSE#TRPBF PART MARKING* LFJF LFJF LTFJG LTFJG PACKAGE DESCRIPTION 10-Lead (3mm × 3mm) Plastic DFN 10-Lead (3mm × 3mm) Plastic DFN 10-Lead Plastic MSOP 10-Lead Plastic MSOP TEMPERATURE RANGE –40°C to 125°C –40°C to 125°C –40°C to 125°C –40°C to 125°C Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. Consult LTC Marketing for information on non-standard lead based finish parts. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ 3971f 2 LT3971 ELECTRICAL CHARACTERISTICS PARAMETER Minimum Input Voltage Quiescent Current from VIN VEN Low VEN High, VSYNC Low VEN High, VSYNC Low VFB = 1.19V The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 12V, VEN = 12V, VBD = 3.3V unless otherwise noted. (Note 2) CONDITIONS l MIN TYP 4 0.7 1.7 0.1 MAX 4.3 1.2 2.7 4.5 12 1.205 1.215 0.01 2.4 1.2 240 150 3 1 1 1.8 28 1.07 20 140 1 1.0 1.6 UNITS V μA μA μA nA V V %/V MHz MHz kHz ns ns A mV μA mV μA V mA V mV nA mV mV μA μA V nA μA l l l FB Pin Current Feedback Voltage FB Voltage Line Regulation Switching Frequency 1.175 1.165 1.6 0.8 160 1.19 1.19 0.0002 2 1 200 80 110 4.3V < VIN < 40V RT = 11k RT = 35.7k RT = 255k Minimum Switch On Time Minimum Switch Off Time Switch Current Limit Switch VCESAT Switch Leakage Current Boost Schottky Forward Voltage Boost Schottky Reverse Leakage Minimum Boost Voltage (Note 3) BOOST Pin Current EN Voltage Threshold EN Voltage Hysteresis EN Pin Current PG Threshold Offset from VFB PG Hysteresis PG Leakage PG Sink Current SYNC Threshold SYNC Pin Current SS Source Current VSS = 1V 0.6 VPG = 3V VPG = 0.4V l 1.8 ISW = 1A ISH = 100mA VREVERSE = 12V VIN = 5V ISW = 1A, VBOOST = 15V EN Rising l l 2.4 330 0.02 770 0.02 1.4 20 0.95 1.01 30 0.2 VFB Rising 60 100 20 0.02 300 0.6 570 0.8 0.1 1 Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The LT3971E is guaranteed to meet performance specifications from 0°C to 125°C junction temperature. Specifications over the –40°C to 125°C operating junction temperature range are assured by design, characterization, and correlation with statistical process controls. The LT3971I is guaranteed over the full –40°C to 125°C operating junction temperature range. High junction temperatures degrade operating lifetimes. Operating lifetime is derated at junction temperatures greater than 125°C. Note 3: This is the minimum voltage across the boost capacitor needed to guarantee full saturation of the switch. 3971f 3 LT3971 TYPICAL PERFORMANCE CHARACTERISTICS Efficiency, VOUT = 5V 100 90 80 EFFICIENCY (%) 70 60 50 40 FRONT PAGE APPLICATION = 5V V 30 OUT R1 = 1M R2 = 309k 20 0 0.2 0.4 0.6 0.8 LOAD CURRENT (A) VIN = 24V EFFICIENCY (%) VIN = 36V VIN = 12V 100 90 80 70 60 50 40 30 1 1.2 3971 G01 TA = 25°C, unless otherwise noted. Efficiency, VOUT = 3.3V 100 VIN = 12V EFFICIENCY (%) Efficiency, VOUT = 5V FRONT PAGE APPLICATION 90 VOUT = 5V R1 = 1M 80 R2 = 309k VIN = 12V 70 60 50 40 30 20 10 VIN = 24V VIN = 36V VIN = 36V VIN = 24V 20 FRONT PAGE APPLICATION 0 0.2 0.4 0.6 0.8 LOAD CURRENT (A) 1 1.2 3971 G02 0 0.01 0.1 1 10 100 LOAD CURRENT (mA) 1000 3971 G03 Efficiency, VOUT = 3.3V 90 80 70 INPUT CURRENT (μA) FRONT PAGE APPLICATION VIN = 12V 100 No Load Supply Current DIODES, INC. DFLS2100 4.0 3.5 INPUT CURRENT (μA) 3.0 2.5 2.0 1.5 1 –55 1.0 –25 5 35 65 95 TEMPERATURE (°C) 125 155 No Load Supply Current FRONT PAGE APPLICATION VOUT = 3.3V EFFICIENCY (%) 60 50 40 30 20 10 0 0.01 0.1 VIN = 24V VIN = 36V 10 1 10 100 LOAD CURRENT (mA) 1000 3971 G04 0 10 20 30 INPUT VOLTAGE (V) 40 3971 G06 3971 G05 Feedback Voltage 1.205 1.200 FEEDBACK VOLTAGE (V) LOAD CURRENT (A) 1.195 1.190 1.185 1.180 1.175 –55 3.0 2.5 2.0 Maximum Load Current FRONT PAGE APPLICATION VOUT = 3.3V LOAD CURRENT (A) TYPICAL MINIMUM 1.5 1.0 0.5 0 2.5 Maximum Load Current FRONT PAGE APPLICATION VOUT = 5V TYPICAL 1.5 MINIMUM 2.0 1.0 0.5 –25 5 35 65 95 TEMPERATURE (°C) 125 155 5 10 15 20 25 30 INPUT VOLTAGE (V) 35 40 3971 G08 0 5 10 25 30 15 20 INPUT VOLTAGE (V) 35 40 3971 G09 3971 G07 3971f 4 LT3971 TYPICAL PERFORMANCE CHARACTERISTICS Load Regulation 0.30 0.25 0.20 LOAD REGULATION (%) 0.15 FREQUENCY (kHz) 0.10 0.05 0 –0.05 –0.10 –0.15 –0.20 FRONT PAGE APPLICATION –0.25 REFERENCED FROM V OUT AT 0.5A LOAD –0.30 0 200 400 600 800 1000 LOAD CURRENT (mA) 1000 950 900 850 800 750 700 650 1200 600 –55 –25 5 35 65 95 TEMPERATURE (°C) 125 155 SWITCH CURRENT LIMIT (A) TA = 25°C, unless otherwise noted. Switching Frequency 3.0 2.5 2.0 1.5 1.0 0.5 0 Switch Current Limit 0 20 40 60 DUTY CYCLE (%) 80 100 3971 G12 3971 G10 3971 G11 Switch Current Limit 2.5 2.4 SWITCH CURRENT LIMIT (A) 600 500 400 VCESAT (mV) 300 200 100 0 Switch VCESAT 30 25 BOOST PIN CURRENT (mA) 0 250 500 750 1000 1250 SWITCH CURRENT (mA) 1500 20 15 10 5 0 Boost Pin Current 2.3 2.2 2.1 2.0 1.9 1.8 1.7 1.6 DUTY CYCLE = 30% 1.5 –55 –25 5 35 65 95 TEMPERATURE (°C) 125 155 0 250 500 750 1000 1250 SWITCH CURRENT (mA) 1500 3971 G13 3971 G14 3971 G15 Frequency Foldback 900 800 SWITCHING FREQUENCY (kHz) SWITCH ON/OFF TIME (ns) 700 600 500 400 300 200 100 0 0 0.2 0.4 0.8 0.6 FB PIN VOLTAGE (V) 1 1.2 3971 G16 Minimum Switch On-Time/ Switch Off-Time 400 350 300 250 200 150 100 MIN TON 50 0 –55 0 –25 35 95 5 65 TEMPERATURE (°C) 125 155 MIN TOFF 1A LOAD MIN TOFF 0.5A LOAD SWITCH CURRENT LIMIT (A) 2.0 2.5 Soft-Start 1.5 1.0 0.5 0 0.25 0.5 0.75 1 1.25 1.5 1.75 SS PIN VOLTAGE (V) 2 3971 G17 3971 G18 3971f 5 LT3971 TYPICAL PERFORMANCE CHARACTERISTICS Minimum Input Voltage 5.0 FRONT PAGE APPLICATION 4.8 VOUT = 3.3V 4.6 INPUT VOLTAGE (V) INPUT VOLTAGE (V) 4.4 4.2 4.0 3.8 3.6 3.4 3.2 3.0 0 200 400 800 1000 600 LOAD CURRENT (mA) 1200 5.2 5.0 TO RUN TO START 6.0 TO START 5.8 5.6 5.4 TO RUN 6.4 6.2 TA = 25°C, unless otherwise noted. Minimum Input Voltage FRONT PAGE APPLICATION VOUT = 5V THRESHOLD VOLTAGE (V) 1.05 1.04 1.03 1.02 1.01 1.00 0.99 0.98 0.97 0.96 0 200 400 800 1000 600 LOAD CURRENT (mA) 1200 EN Threshold RISING THRESHOLD FALLING THRESHOLD 0.95 –55 –25 5 35 65 95 TEMPERATURE (°C) 125 155 3971 G19 3971 G20 3971 G21 Boost Diode Forward Voltage 1.6 1.4 BOOST DIODE VF (V) 1.2 1.0 0.8 0.6 0.4 0.2 0 0 250 500 750 1000 1250 BOOST DIODE CURRENT (mA) 1500 THRESHOLD VOLTAGE (%) 95 94 93 92 91 90 89 88 87 86 Power Good Threshold Transient Load Response, Load Current Stepped from 25mA (Burst Mode Operation) to 525mA VOUT 100mV/DIV IL 500mA/DIV –25 5 35 65 95 TEMPERATURE (°C) 125 155 10μs/DIV FRONT PAGE APPLICATION VIN = 12V, VOUT = 3.3V COUT = 47μF 3971 G24 85 –55 3971 G22 3971 G23 Transient Load Response, Load Current Stepped from 0.5A to 1A Switching Waveforms; Burst Mode Operation Switching Waveforms; Full Frequency Continuous Operation VOUT 100mV/ DIV VSW 5V/DIV VSW 5V/DIV IL 500mA/DIV IL 500mA/ DIV VOUT 20mV/DIV 10μs/DIV FRONT PAGE APPLICATION VIN = 12V, VOUT = 3.3V COUT = 47μF 3971 G25 IL 500mA/DIV VOUT 20mV/DIV 5μs/DIV FRONT PAGE APPLICATION VIN = 12V, VOUT = 3.3V ILOAD = 10mA COUT = 22μF 3971 G26 1μs/DIV FRONT PAGE APPLICATION VIN = 12V, VOUT = 3.3V ILOAD = 1A COUT = 22μF 3971 G27 3971f 6 LT3971 PIN FUNCTIONS BD (Pin 1): This pin connects to the anode of the boost diode. The BD pin is normally connected to the output. BOOST (Pin 2): This pin is used to provide a drive voltage, higher than the input voltage, to the internal bipolar NPN power switch. SW (Pin 3): The SW pin is the output of an internal power switch. Connect this pin to the inductor, catch diode, and boost capacitor. VIN (Pin 4): The VIN pin supplies current to the LT3971’s internal circuitry and to the internal power switch. This pin must be locally bypassed. EN (Pin 5): The part is in shutdown when this pin is low and active when this pin is high. The hysteretic threshold voltage is 1.005V going up and 0.975V going down. The EN threshold is only accurate when VIN is above 4.3V. If VIN is lower than 4.2V, ground EN to place the part in shutdown. Tie to VIN if shutdown feature is not used. FB (Pin 6): The LT3971 regulates the FB pin to 1.19V. Connect the feedback resistor divider tap to this pin. Also, connect a phase lead capacitor between FB and VOUT. Typically this capacitor is 10pF . SS (Pin 7): A capacitor is tied between SS and ground to slowly ramp up the peak current limit of the LT3971 on start-up. The soft-start capacitor is only actively discharged when EN is low. The SS pin is released when the EN pin goes high. Float this pin to disable soft-start. For applications with input voltages above 25V, add a 100k resistor in series with the soft-start capacitor. RT (Pin 8): A resistor is tied between RT and ground to set the switching frequency. PG (Pin 9): The PG pin is the open-drain output of an internal comparator. PGOOD remains low until the FB pin is within 9% of the final regulation voltage. PGOOD is valid when the LT3971 is enabled and VIN is above 4.3V. SYNC (Pin 10): This is the external clock synchronization input. Ground this pin for low ripple Burst Mode operation at low output loads. Tie to a clock source for synchronization, which will include pulse-skipping at low output loads. When in pulse-skipping mode, quiescent current increases to 1.5mA. GND (Exposed Pad Pin 11): Ground. The exposed pad must be soldered to PCB. 3971f 7 LT3971 BLOCK DIAGRAM VIN C1 INTERNAL 1.19V REF 1V EN RT + – SHDN SLOPE COMP OSCILLATOR 200kHz TO 2MHz RT SYNC PG ERROR AMP + – 1.09V + – VC GND R2 FB R1 C5 8 + – SWITCH LATCH R Q S Burst Mode DETECT VC CLAMP 1μA SHDN VIN BD BOOST C3 L1 VOUT SW D1 C2 SS R3 C4 3991 BD 3971f LT3971 OPERATION The LT3971 is a constant frequency, current mode stepdown regulator. An oscillator, with frequency set by RT, sets an RS flip-flop, turning on the internal power switch. An amplifier and comparator monitor the current flowing between the VIN and SW pins, turning the switch off when this current reaches a level determined by the voltage at VC (see Block Diagram). An error amplifier measures the output voltage through an external resistor divider tied to the FB pin and servos the VC node. If the error amplifier’s output increases, more current is delivered to the output; if it decreases, less current is delivered. An active clamp on the VC node provides current limit. The VC node is also clamped by the voltage on the SS pin; soft-start is implemented by generating a voltage ramp at the SS pin using an external capacitor. If the EN pin is low, the LT3971 is shut down and draws 700nA from the input. When the EN pin exceeds 1V, the switching regulator will become active. The switch driver operates from either VIN or from the BOOST pin. An external capacitor is used to generate a voltage at the BOOST pin that is higher than the input supply. This allows the driver to fully saturate the internal bipolar NPN power switch for efficient operation. To further optimize efficiency, the LT3971 automatically switches to Burst Mode operation in light load situations. Between bursts, all circuitry associated with controlling the output switch is shut down, reducing the input supply current to 1.7μA. In a typical application, 2.8μA will be consumed from the supply when regulating with no load. The oscillator reduces the LT3971’s operating frequency when the voltage at the FB pin is low. This frequency foldback helps to control the output current during startup and overload. The LT3971 contains a power good comparator which trips when the FB pin is at 91% of its regulated value. The PG output is an open-drain transistor that is off when the output is in regulation, allowing an external resistor to pull the PG pin high. Power good is valid when the LT3971 is enabled and VIN is above 4.2V. APPLICATIONS INFORMATION Achieving Ultralow Quiescent Current To enhance efficiency at light loads, the LT3971 operates in low ripple Burst Mode, which keeps the output capacitor charged to the desired output voltage while minimizing the input quiescent current. In Burst Mode operation the LT3971 delivers single pulses of current to the output capacitor followed by sleep periods where the output power is supplied by the output capacitor. When in sleep mode the LT3971 consumes 1.7μA, but when it turns on all the circuitry to deliver a current pulse, the LT3971 consumes 1.5mA of input current in addition to the switch current. Therefore, the total quiescent current will be greater than 1.7μA when regulating. As the output load decreases, the frequency of single current pulses decreases (see Figure 1) and the percentage of time the LT3971 is in sleep mode increases, resulting in much higher light load efficiency. By maximizing the time between pulses, the converter quiescent current 1000 SWITCHING FREQUENCY (kHz) 800 FRONT PAGE APPLICATION VIN = 12V VOUT = 3.3V 600 400 200 0 0 20 40 60 80 LOAD CURRENT (mA) 100 120 3971 F01 Figure 1. Switching Frequency in Burst Mode Operation gets closer to the 1.7μA ideal. Therefore, to optimize the quiescent current performance at light loads, the current in the feedback resistor divider and the reverse current in the catch diode must be minimized, as these appear to the output as load currents. Use the largest possible 3971f 9 LT3971 APPLICATIONS INFORMATION feedback resistors and a low leakage Schottky catch diode in applications utilizing the ultralow quiescent current performance of the LT3971. The feedback resistors should preferably be on the order of MΩ and the Schottky catch diode should have less than 1μA of typical reverse leakage at room temperature. These two considerations are reiterated in the FB Resistor Network and Catch Diode Selection sections. It is important to note that another way to decrease the pulse frequency is to increase the magnitude of each single current pulse. However, this increases the output voltage ripple because each cycle delivers more power to the output capacitor. The magnitude of the current pulses was selected to ensure less than 15mV of output ripple in a typical application. See Figure 2. VSW 5V/DIV To ensure proper Burst Mode operation, the SYNC pin must be grounded. When synchronized with an external clock, the LT3971 will pulse skip at light loads. The quiescent current will significantly increase to 1.5mA in light load situations when synchronized with an external clock. Holding the SYNC pin high yields no advantages in terms of output ripple or minimum load to full frequency, so is not recommended. FB Resistor Network The output voltage is programmed with a resistor divider between the output and the FB pin. Choose the resistor values according to: ⎞ ⎛V R1= R2 ⎜ OUT − 1⎟ ⎝ 1.19 V ⎠ Reference designators refer to the Block Diagram. 1% resistors are recommended to maintain output voltage accuracy. The total resistance of the FB resistor divider should be selected to be as large as possible to enhance low current performance. The resistor divider generates a small load on the output, which should be minimized to optimize the low supply current at light loads. When using large FB resistors, a 10pF phase lead capacitor should be connected from VOUT to FB. Setting the Switching Frequency The LT3971 uses a constant frequency PWM architecture that can be programmed to switch from 200kHz to 2MHz by using a resistor tied from the RT pin to ground. A table showing the necessary RT value for a desired switching frequency is in Table 1. Table 1. Switching Frequency vs RT Value SWITCHING FREQUENCY (MHz) 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 RT VALUE (kΩ) 255 118 71.5 49.9 35.7 28.0 22.1 17.4 14.0 11.0 3971f IL 500mA/DIV VOUT 20mV/DIV 5μs/DIV FRONT PAGE APPLICATION VIN = 12V VOUT = 3.3V ILOAD = 10mA 3971 F02 Figure 2. Burst Mode Operation While in Burst Mode operation, the burst frequency and the charge delivered with each pulse will not change with output capacitance. Therefore, the output voltage ripple will be inversely proportional to the output capacitance. In a typical application with a 22μF output capacitor, the output ripple is about 10mV, and with a 47μF output capacitor the output ripple is about 5mV. The output voltage ripple can continue to be decreased by increasing the output capacitance. At higher output loads (above 92mA for the front page application) the LT3971 will be running at the frequency programmed by the RT resistor, and will be operating in standard PWM mode. The transition between PWM and low ripple Burst Mode operation will exhibit slight frequency jitter, but will not disturb the output voltage. 10 LT3971 APPLICATIONS INFORMATION Operating Frequency Tradeoffs Selection of the operating frequency is a tradeoff between efficiency, component size, minimum dropout voltage, and maximum input voltage. The advantage of high frequency operation is that smaller inductor and capacitor values may be used. The disadvantages are lower efficiency, lower maximum input voltage, and higher dropout voltage. The highest acceptable switching frequency (fSW(MAX)) for a given application can be calculated as follows: fSW(MAX ) = VOUT + VD tON(MIN)( VIN − VSW + VD ) Input Voltage Range The minimum input voltage is determined by either the LT3971’s minimum operating voltage of 4.3V or by its maximum duty cycle (see equation in Operating Frequency Tradeoffs section). The minimum input voltage due to duty cycle is: VIN(MIN) = VOUT + VD −V +V 1− fSW tOFF(MIN) D SW where VIN is the typical input voltage, VOUT is the output voltage, VD is the catch diode drop (~0.5V), and VSW is the internal switch drop (~0.5V at max load). This equation shows that slower switching frequency is necessary to safely accommodate high VIN/VOUT ratio. Also, as shown in the Input Voltage Range section, lower frequency allows a lower dropout voltage. The input voltage range depends on the switching frequency because the LT3971 switch has finite minimum on and off times. The minimum switch on and off times are strong functions of temperature. Use the typical minimum on and off curves to design for an application’s maximum temperature, while adding about 30% for part-to-part variation. The minimum and maximum duty cycles that can be achieved taking minimum on and off times into account are: DCMIN = fSW tON(MIN) DCMAX = 1− fSW tOFF(MIN) where fSW is the switching frequency, the tON(MIN) is the minimum switch on-time, and the tOFF(MIN) is the minimum switch off-time. These equations show that duty cycle range increases when switching frequency is decreased. A good choice of switching frequency should allow adequate input voltage range (see Input Voltage Range section) and keep the inductor and capacitor values small. where VIN(MIN) is the minimum input voltage, VOUT is the output voltage, VD is the catch diode drop (~0.5V), VSW is the internal switch drop (~0.5V at max load), fSW is the switching frequency (set by RT), and tOFF(MIN) is the minimum switch off-time. Note that higher switching frequency will increase the minimum input voltage. If a lower dropout voltage is desired, a lower switching frequency should be used. The maximum input voltage for LT3971 applications depends on switching frequency, the Absolute Maximum Ratings of the VIN and BOOST pins, and the operating mode. For a given application where the switching frequency and the output voltage are already selected, the maximum input voltage (VIN(OP-MAX)) that guarantees optimum output voltage ripple for that application can be found by applying the following equation: VIN(OP-MAX ) = VOUT + VD –V +V fSW • tON(MIN) D SW where tON(MIN) is the minimum switch on-time. Note that a higher switching frequency will decrease the maximum operating input voltage. Conversely, a lower switching frequency will be necessary to achieve normal operation at higher input voltages. The circuit will tolerate inputs above the maximum operating input voltage and up to the Absolute Maximum Ratings of the VIN and BOOST pins, regardless of chosen switching frequency. However, during such transients 3971f 11 LT3971 APPLICATIONS INFORMATION where VIN is higher than VIN(OP-MAX), the LT3971 will enter pulse-skipping operation where some switching pulses are skipped to maintain output regulation. The output voltage ripple and inductor current ripple will be higher than in typical operation. Do not overload when VIN is greater than VIN(OP-MAX). Inductor Selection and Maximum Output Current A good first choice for the inductor value is: L= VOUT + VD fSW Coilcraft Sumida www.coilcraft.com www.sumida.com Table 2. Inductor Vendors VENDOR Murata TDK Toko URL www.murata.com PART SERIES LQH55D TYPE Open Shielded Shielded Shielded Shielded Shielded Open Shielded Shielded Open Shielded Shielded Open www.componenttdk.com SLF7045 SLF10145 www.toko.com D62CB D63CB D73C D75F MSS7341 MSS1038 CR54 CDRH74 CDRH6D38 CR75 where fSW is the switching frequency in MHz, VOUT is the output voltage, VD is the catch diode drop (~0.5V) and L is the inductor value in μH. The inductor’s RMS current rating must be greater than the maximum load current and its saturation current should be about 30% higher. For robust operation in fault conditions (start-up or short-circuit) and high input voltage (>30V), the saturation current should be above 2.8A. To keep the efficiency high, the series resistance (DCR) should be less than 0.1Ω, and the core material should be intended for high frequency applications. Table 2 lists several vendors and suitable types. The inductor value must be sufficient to supply the desired maximum output current (IOUT(MAX)), which is a function of the switch current limit (ILIM) and the ripple current. IOUT(MAX ) = ILIM – ΔIL 2 When the switch is off, the potential across the inductor is the output voltage plus the catch diode drop. This gives the peak-to-peak ripple current in the inductor: ΔIL = (1− DC)•( VOUT + VD) L • fSW Where fSW is the switching frequency of the LT3971, DC is the duty cycle and L is the value of the inductor. Therefore, the maximum output current that the LT3971 will deliver depends on the switch current limit, the inductor value, and the input and output voltages. The inductor value may have to be increased if the inductor ripple current does not allow sufficient maximum output current (IOUT(MAX)) given the switching frequency, and maximum input voltage used in the desired application. The optimum inductor for a given application may differ from the one indicated by this simple design guide. A larger value inductor provides a higher maximum load current and reduces the output voltage ripple. If your load is lower than the maximum load current, than you can relax the value of the inductor and operate with higher ripple current. This allows you to use a physically smaller inductor, or one with a lower DCR resulting in higher efficiency. Be aware that if the inductance differs from the simple rule above, then the maximum load current will depend on the input voltage. In addition, low inductance may result in discontinuous mode operation, which further reduces maximum load current. The LT3971 limits its peak switch current in order to protect itself and the system from overload faults. The LT3971’s switch current limit (ILIM) is at least 2.4A at low duty cycles and decreases linearly to 1.75A at DC = 0.8. 3971f 12 LT3971 APPLICATIONS INFORMATION For details of maximum output current and discontinuous operation, see Linear Technology’s Application Note 44. Finally, for duty cycles greater than 50% (VOUT/VIN>0.5), a minimum inductance is required to avoid sub-harmonic oscillations. See Application Note 19. One approach to choosing the inductor is to start with the simple rule given above, look at the available inductors, and choose one to meet cost or space goals. Then use the equations above to check that the LT3971 will be able to deliver the required output current. Note again that these equations assume that the inductor current is continuous. Discontinuous operation occurs when IOUT is less than ΔIL/2. Input Capacitor Bypass the input of the LT3971 circuit with a ceramic capacitor of X7R or X5R type. Y5V types have poor performance over temperature and applied voltage, and should not be used. A 4.7μF to 10μF ceramic capacitor is adequate to bypass the LT3971 and will easily handle the ripple current. Note that larger input capacitance is required when a lower switching frequency is used (due to longer on-times). If the input power source has high impedance, or there is significant inductance due to long wires or cables, additional bulk capacitance may be necessary. This can be provided with a low performance electrolytic capacitor. Step-down regulators draw current from the input supply in pulses with very fast rise and fall times. The input capacitor is required to reduce the resulting voltage ripple at the LT3971 and to force this very high frequency switching current into a tight local loop, minimizing EMI. A 4.7μF capacitor is capable of this task, but only if it is placed close to the LT3971 (see the PCB Layout section). A second precaution regarding the ceramic input capacitor concerns the maximum input voltage rating of the LT3971. A ceramic input capacitor combined with trace or cable inductance forms a high quality (under damped) tank circuit. If the LT3971 circuit is plugged into a live supply, the input voltage can ring to twice its nominal value, possibly exceeding the LT3971’s voltage rating. This situation is easily avoided (see the Hot Plugging Safely section). Output Capacitor and Output Ripple The output capacitor has two essential functions. Along with the inductor, it filters the square wave generated by the LT3971 to produce the DC output. In this role it determines the output ripple, so low impedance (at the switching frequency) is important. The second function is to store energy in order to satisfy transient loads and stabilize the LT3971’s control loop. Ceramic capacitors have very low equivalent series resistance (ESR) and provide the best ripple performance. A good starting value is: 100 COUT = VOUT fSW where fSW is in MHz, and COUT is the recommended output capacitance in μF Use X5R or X7R types. This choice will . provide low output ripple and good transient response. Transient performance can be improved with a higher value capacitor. Increasing the output capacitance will also decrease the output voltage ripple. A lower value of output capacitor can be used to save space and cost but transient performance will suffer. When choosing a capacitor, look carefully through the data sheet to find out what the actual capacitance is under operating conditions (applied voltage and temperature). A physically larger capacitor or one with a higher voltage rating may be required. Table 3 lists several capacitor vendors. Table 3. Recommended Ceramic Capacitor Vendors MANUFACTURER AVX Murata Taiyo Yuden Vishay Siliconix TDK WEBSITE www.avxcorp.com www.murata.com www.t-yuden.com www.vishay.com www.tdk.com Catch Diode Selection The catch diode (D1 from Block Diagram) conducts current only during switch off time. Average forward current in normal operation can be calculated from: ID( AVG) = IOUT VIN – VOUT VIN where IOUT is the output load current. The only reason to consider a diode with a larger current rating than necessary 3971f 13 LT3971 APPLICATIONS INFORMATION for nominal operation is for the worst-case condition of shorted output. The diode current will then increase to the typical peak switch current. Peak reverse voltage is equal to the regulator input voltage. Use a diode with a reverse voltage rating greater than the input voltage. Table 4. Schottky Diodes. The Reverse Current Values Listed Are Estimates Based Off of Typical Curves for Reverse Current vs Reverse Voltage at 25°C. VR (V) 20 40 20 40 30 40 20 30 40 50 20 30 40 40 40 60 100 40 40 60 40 40 60 40 40 60 IAVE (A) 0.5 0.5 1 1 0.5 0.5 1 1 1 1 2 2 1 2 1.1 1 2 2 1 1 1 2 2 2 2 2 500 700 400 550 700 400 500 700 510 500 770 860 500 500 620 500 500 500 700 500 500 620 530 550 595 VF at 1A (mV) VF at 2A (mV) IR at VR = 20V 25°C (μA) 30 0.4 0.5 20 15 1 1.1 1.1 1.1 0.4 20 0.6 1 4 1 2.5 0.01 0.45 PART NUMBER On Semiconductor MBR0520L MBR0540 MBRM120E MBRM140 Diodes Inc. B0530W B0540W B120 B130 B140 B150 B220 B230 B140HB DFLS240L DFLS140 DFLS160 DFLS2100 B240 CMSH1 - 40M CMSH1 - 60M CMSH1 - 40ML CMSH2 - 40M CMSH2 - 60M CMSH2 - 40L CMSH2 - 40 CMSH2 - 60M An additional consideration is reverse leakage current. When the catch diode is reversed biased, any leakage current will appear as load current. When operating under light load conditions, the low supply current consumed by the LT3971 will be optimized by using a catch diode with minimum reverse leakage current. Low leakage Schottky diodes often have larger forward voltage drops at a given current, so a trade-off can exist between low load and high load efficiency. Often Schottky diodes with larger reverse bias ratings will have less leakage at a given output voltage than a diode with a smaller reverse bias rating. Therefore, superior leakage performance can be achieved at the expense of diode size. Table 4 lists several Schottky diodes and their manufacturers. Ceramic Capacitors Ceramic capacitors are small, robust and have very low ESR. However, ceramic capacitors can cause problems when used with the LT3971 due to their piezoelectric nature. When in Burst Mode operation, the LT3971’s switching frequency depends on the load current, and at very light loads the LT3971 can excite the ceramic capacitor at audio frequencies, generating audible noise. Since the LT3971 operates at a lower current limit during Burst Mode operation, the noise is typically very quiet to a casual ear. If this is unacceptable, use a high performance tantalum or electrolytic capacitor at the output. A final precaution regarding ceramic capacitors concerns the maximum input voltage rating of the LT3971. As previously mentioned, a ceramic input capacitor combined with trace or cable inductance forms a high quality (under damped) tank circuit. If the LT3971 circuit is plugged into a live supply, the input voltage can ring to twice its nominal value, possibly exceeding the LT3971’s rating. This situation is easily avoided (see the Hot Plugging Safely section). BOOST and BD Pin Considerations Capacitor C3 and the internal boost Schottky diode (see the Block Diagram) are used to generate a boost voltage that is higher than the input voltage. In most cases a 0.47μF capacitor will work well. Figure 3 shows three ways to arrange the boost circuit. The BOOST pin must be more than 2.3V above the SW pin for best efficiency. 3971f Central Semiconductor 14 LT3971 APPLICATIONS INFORMATION For outputs of 3V and above, the standard circuit (Figure 3a) is best. For outputs between 2.8V and 3V, use a 1μF boost capacitor. A 2.5V output presents a special case because it is marginally adequate to support the boosted drive stage while using the internal boost diode. For reliable BOOST pin operation with 2.5V outputs use a good external Schottky diode (such as the ON Semi MBR0540), and a 1μF boost capacitor (Figure 3b). For output voltages below 2.5V, the boost diode can be tied to the input (Figure 3c), or to another external supply greater than 2.8V. However, the circuit in Figure 3a is more efficient because the BOOST pin current comes from a lower voltage source. You must also be sure that the maximum voltage ratings of the BOOST and BD pins are not exceeded. BD VIN VIN BOOST LT3971 4.7μF SW C3 VOUT 5.0 4.8 4.6 INPUT VOLTAGE (V) 4.4 4.2 4.0 3.8 3.6 3.4 VOUT = 3.3V TA = 25°C 3.2 L = 4.7μH f = 800kHz 3.0 10 TO RUN TO START The minimum operating voltage of an LT3971 application is limited by the minimum input voltage (4.3V) and by the maximum duty cycle as outlined in the Input Voltage Range section. For proper start-up, the minimum input voltage is also limited by the boost circuit. If the input voltage is ramped slowly, the boost capacitor may not be fully charged. Because the boost capacitor is charged with the energy stored in the inductor, the circuit will rely on some minimum load current to get the boost circuit running properly. This minimum load will depend on input and output voltages, and on the arrangement of the boost circuit. The minimum load generally goes to zero once the circuit has started. Figure 4 shows a plot of minimum load to start and to run as a function of input voltage. In many cases the discharged output capacitor will present a load to the switcher, which will allow it to start. The plots show the worst-case situation where VIN is ramping very slowly. GND (3a) For VOUT > 2.8V BD VIN VIN BOOST LT3971 4.7μF SW C3 D2 GND VOUT 100 LOAD CURRENT (mA) 1000 6.4 6.2 INPUT VOLTAGE (V) 6.0 5.8 5.6 TO RUN 5.4 5.2 5.0 3971 FO3 (3b) For 2.5V < VOUT < 2.8V TO START BD VIN VIN BOOST LT3971 4.7μF SW C3 VOUT GND VOUT = 5V TA = 25°C L = 4.7μH f = 800kHz 10 100 LOAD CURRENT (mA) 1000 3971 F04 (3c) For VOUT < 2.5V; VIN(MAX) = 27V Figure 3. Three Circuits for Generating the Boost Voltage Figure 4. The Minimum Input Voltage Depends on Output Voltage, Load Current and Boost Circuit 3971f 15 LT3971 APPLICATIONS INFORMATION For lower start-up voltage, the boost diode can be tied to VIN; however, this restricts the input range to one-half of the absolute maximum rating of the BOOST pin. At light loads, the inductor current becomes discontinuous and this reduces the minimum input voltage to approximately 400mV above VOUT. At higher load currents, the inductor current is continuous and the duty cycle is limited by the maximum duty cycle of the LT3971, requiring a higher input voltage to maintain regulation. Enable Pin The LT3971 is in shutdown when the EN pin is low and active when the pin is high. The rising threshold of the EN comparator is 1.01V, with 30mV of hysteresis. The EN pin can be tied to VIN if the shutdown feature is not used. Adding a resistor divider from VIN to EN programs the LT3971 to regulate the output only when VIN is above a desired voltage (see Figure 5). Typically, this threshold, VIN(EN), is used in situations where the input supply is current limited, or has a relatively high source resistance. A switching regulator draws constant power from the source, so source current increases as source voltage drops. This looks like a negative resistance load to the source and can cause the source to current limit or latch low under low source voltage conditions. The VIN(EN) threshold prevents the regulator from operating at source voltages where the problems might occur. This threshold can be adjusted by setting the values R3 and R4 such that they satisfy the following equation: R3 VIN(EN) = +1 R4 where output regulation should not start until VIN is above VIN(EN). Due to the comparator’s hysteresis, switching will not stop until the input falls slightly below VIN(EN). VIN R3 EN R4 3971 F05 Be aware that when the input voltage is below 4.3V, the input current may rise to several hundred μA. And the part may be able to switch at cold or for VIN(EN) thresholds less than 7V. Figure 6 shows the magnitude of the increased input current in a typical application with different programmed VIN(EN). When operating in Burst Mode for light load currents, the current through the VIN(EN) resistor network can easily be greater than the supply current consumed by the LT3971. Therefore, the VIN(EN) resistors should be large to minimize their effect on efficiency at low loads. 12V VIN(EN) Input Current 500 400 INPUT CURRENT (μA) 300 200 100 0 0 1 2 3 4 5 6 7 8 9 10 11 12 INPUT VOLTAGE (V) VIN(EN) = 12V R3 = 11M R4 = 1M 6V VIN(EN) Input Current 500 400 INPUT CURRENT (μA) 300 200 100 LT3971 0 0 1 2 3 4 INPUT VOLTAGE (V) 5 6 3971 F06 1V + – SHDN VIN(EN) = 6V R3 = 5M R4 = 1M Figure 5. Programmed Enable Threshold Figure 6. Input Current vs Input Voltage for a Programmed VIN(EN) of 6V and 12V 3971f 16 LT3971 APPLICATIONS INFORMATION Soft-Start The SS pin can be used to soft-start the LT3971 by throttling the maximum input current during start-up. An internal 1μA current source charges an external capacitor generating a voltage ramp on the SS pin. The SS pin clamps the internal VC node, which slowly ramps up the current limit. Maximum current limit is reached when the SS pin is about 1.5V or higher. By selecting a large enough capacitor, the output can reach regulation without overshoot. For applications with input voltages above 25V, a 100k resistor in series with the soft-start capacitor is recommended. Figure 7 shows start-up waveforms for a typical application with a 10nF capacitor on SS for a 3.3Ω load when the EN pin is pulsed high for 13ms. The external SS capacitor is only actively discharged when EN is low. With EN low, the external SS cap is discharged through approximately 150Ω. The EN pin needs to be low long enough for the external cap to completely discharge through the 150Ω pull-down prior to start-up. The LT3971 will not enter Burst Mode operation at low output loads while synchronized to an external clock, but instead will pulse skip to maintain regulation. The LT3971 may be synchronized over a 250kHz to 2MHz range. The RT resistor should be chosen to set the LT3971 switching frequency 20% below the lowest synchronization input. For example, if the synchronization signal will be 250kHz and higher, the RT should be selected for 200kHz. To assure reliable and safe operation the LT3971 will only synchronize when the output voltage is near regulation as indicated by the PG flag. It is therefore necessary to choose a large enough inductor value to supply the required output current at the frequency set by the RT resistor (see the Inductor Selection section). The slope compensation is set by the RT value, while the minimum slope compensation required to avoid subharmonic oscillations is established by the inductor size, input voltage, and output voltage. Since the synchronization frequency will not change the slopes of the inductor current waveform, if the inductor is large enough to avoid subharmonic oscillations at the frequency set by RT, than the slope compensation will be sufficient for all synchronization frequencies. Shorted and Reversed Input Protection VOUT 2V/DIV VSS 1V/DIV IL 0.5A/DIV 2ms/DIV 3971 F07 Figure 7. Soft-Start Waveforms for Front-Page Application with 10nF Capacitor on SS. EN is Pulsed High for About 13ms with a 3.3Ω Load Resistor Synchronization To select low ripple Burst Mode operation, tie the SYNC pin below 0.6V (this can be ground or a logic low output). Synchronizing the LT3971 oscillator to an external frequency can be done by connecting a square wave (with 20% to 80% duty cycle) to the SYNC pin. The square wave amplitude should have valleys that are below 0.6V and peaks above 1.0V (up to 6V). If the inductor is chosen so that it won’t saturate excessively, a LT3971 buck regulator will tolerate a shorted output. There is another situation to consider in systems where the output will be held high when the input to the LT3971 is absent. This may occur in battery charging applications or in battery backup systems where a battery or some other supply is diode ORed with the LT3971’s output. If the VIN pin is allowed to float and the EN pin is held high (either by a logic signal or because it is tied to VIN), then the LT3971’s internal circuitry will pull its quiescent current through its SW pin. This is fine if your system can tolerate a few μA in this state. If you ground the EN pin, the SW pin current will drop to essentially zero. However, if the VIN pin is grounded while the output is held high, regardless of EN, parasitic diodes inside the LT3971 can pull current from the output through the SW pin and the VIN pin. Figure 8 shows a circuit that will run only when the input voltage is present and that protects against a shorted or reversed input. 3971f 17 LT3971 APPLICATIONS INFORMATION D4 MBRS140 VIN VIN EN LT3971 BD FB BOOST SW VOUT L1 VOUT C2 GND + BACKUP RPG C3 3971 F07 GND RT C4 Figure 8. Diode D4 Prevents a Shorted Input from Discharging a Backup Battery Tied to the Output. It Also Protects the Circuit from a Reversed Input. The LT3971 Runs Only When the Input is Present C5 D1 C1 GND R1 R2 PCB Layout For proper operation and minimum EMI, care must be taken during printed circuit board layout. Figure 9 shows the recommended component placement with trace, ground plane and via locations. Note that large, switched currents flow in the LT3971’s VIN and SW pins, the catch diode (D1), and the input capacitor (C1). The loop formed by these components should be as small as possible. These components, along with the inductor and output capacitor, should be placed on the same side of the circuit board, and their connections should be made on that layer. Place a local, unbroken ground plane below these components. The SW and BOOST nodes should be as small as possible. Finally, keep the FB and RT nodes small so that the ground traces will shield them from the SW and BOOST nodes. The Exposed Pad on the bottom of the package must be soldered to ground so that the pad acts as a heat sink. To keep thermal resistance low, extend the ground plane as much as possible, and add thermal vias under and near the LT3971 to additional ground planes within the circuit board and on the bottom side. Hot Plugging Safely The small size, robustness and low impedance of ceramic capacitors make them an attractive option for the input bypass capacitor of LT3971 circuits. However, these capacitors can cause problems if the LT3971 is plugged into a live supply. The low loss ceramic capacitor, combined with stray inductance in series with the power source, forms an under damped tank circuit, and the voltage at 3971 F09 VIAS TO LOCAL GROUND PLANE VIAS TO VOUT VIAS TO SYNC VIAS TO RUN/SS VIAS TO PG VIAS TO VIN OUTLINE OF LOCAL GROUND PLANE Figure 9. A Good PCB Layout Ensures Proper, Low EMI Operation the VIN pin of the LT3971 can ring to twice the nominal input voltage, possibly exceeding the LT3971’s rating and damaging the part. If the input supply is poorly controlled or the user will be plugging the LT3971 into an energized supply, the input network should be designed to prevent this overshoot. See Linear Technology Application Note 88 for a complete discussion. High Temperature Considerations For higher ambient temperatures, care should be taken in the layout of the PCB to ensure good heat sinking of the LT3971. The Exposed Pad on the bottom of the package must be soldered to a ground plane. This ground should be tied to large copper layers below with thermal vias; these layers will spread heat dissipated by the LT3971. Placing additional vias can reduce thermal resistance further. The maximum load current should be derated as the ambient temperature approaches the maximum junction rating. Power dissipation within the LT3971 can be estimated by calculating the total power loss from an efficiency measurement and subtracting the catch diode loss and inductor 3971f 18 LT3971 APPLICATIONS INFORMATION loss. The die temperature is calculated by multiplying the LT3971 power dissipation by the thermal resistance from junction to ambient. Also keep in mind that the leakage current of the power Schottky diode goes up exponentially with junction temperature. When the power switch is closed, the power Schottky diode is in parallel with the power converter’s output filter stage. As a result, an increase in a diode’s leakage current results in an effective increase in the load, and a corresponding increase in input power. Therefore, the catch Schottky diode must be selected with care to avoid excessive increase in light load supply current at high temperatures. Other Linear Technology Publications Application Notes 19, 35 and 44 contain more detailed descriptions and design information for buck regulators and other switching regulators. The LT1376 data sheet has a more extensive discussion of output ripple, loop compensation and stability testing. Design Note 318 shows how to generate a bipolar output supply using a buck regulator. TYPICAL APPLICATIONS 5V Step-Down Converter VIN 7V TO 38V VIN OFF ON EN PG 4.7μF SS RT BD 10pF 1M SYNC f = 800kHz GND FB 309k 22μF 3971 TA02 BOOST 0.47μF SW LT3971 4.7μH 49.9k VOUT 5V 1.2A 3.3V Step-Down Converter VIN 4.3V TO 38V VIN OFF ON EN PG 4.7μF SS RT BD 10pF 1M SYNC f = 600kHz GND FB 562k 22μF 3971 TA03 BOOST 0.47μF SW LT3971 4.7μH 71.5k VOUT 3.3V 1.2A 3971f 19 LT3971 TYPICAL APPLICATIONS 2.5V Step-Down Converter VIN 4.3V TO 38V VIN OFF ON EN PG 4.7μF SS RT BD 10pF 1M SYNC f = 400kHz GND FB 909k 47μF 3971 TA04 1.8V Step-Down Converter VIN 4.3V TO 27V VIN BD BOOST 0.47μF SW LT3971 4.7μH BOOST 1μF SW LT3971 4.7μH OFF ON EN PG SS 4.7μF RT 10pF 118k VOUT 2.5V 1.2A 118k SYNC f = 400kHz GND FB 1M 511k 100μF 3971 TA05 VOUT 1.8V 1.2A 12V Step-Down Converter VIN 15V TO 38V VIN OFF ON EN PG 10μF SS RT BD 10pF 1M 49.9k f = 800kHz SYNC GND FB 110k 10μF 3971 TA06 BOOST 0.47μF SW LT3971 10μH VOUT 12V 1.2A 3.3V Step-Down Converter with Undervoltage Lockout, Soft-Start, and Power Good VIN 6V TO 38V 5M EN SW 4.7μF 100k 1M 1nF 49.9k SYNC GND FB 562k 22μF 3971 TA07 VIN BOOST 0.47μF 4.7μH SS RT LT3971 PG BD 10pF 1M 150k PGOOD VOUT 3.3V 1.2A f = 800kHz 3971f 20 LT3971 TYPICAL APPLICATIONS 5V, 2MHz Step-Down Converter with Soft-Start VIN 9V TO 25V VIN OFF ON EN PG SS 2.2μF RT 1nF 11k SYNC f = 2MHz GND FB 309k 22μF 3971 TA08 BOOST 0.47μF SW LT3971 2.2μH BD 10pF 1M VOUT 5V 1.2A 4V Step-Down Converter with a High Impedance Input Source + 24V 11M VIN EN BOOST 0.47μF SW LT3971 4.7μH * AVERAGE OUTPUT POWER CANNOT EXCEED THAT WHICH CAN BE PROVIDED BY HIGH IMPEDANCE SOURCE. NAMELY, V2 • POUT(MAX) = 4R VOUT 4V 1.2A* 100μF 3971 TA09a – + CBULK 100μF 4.7μF 1M PG SS RT 1nF 49.9k SYNC f = 800kHz GND FB BD 10pF 1M 412k WHERE V IS VOLTAGE OF SOURCE, R IS INTERNAL SOURCE IMPEDANCE, AND N IS LT3971 EFFICIENCY. MAXIMUM OUTPUT CURRENT OF 1.2A CAN BE SUPPLIED FOR A SHORT TIME BASED ON THE ENERGY WHICH CAN BE SOURCED BY THE BULK INPUT CAPACITANCE. Sourcing a Maximum Load Pulse VOUT 200mV/DIV VIN 5V/DIV Start-Up from High Impedance Input Source VIN 1V/DIV VOUT 2V/DIV IL 1A/DIV 500μs/DIV 3971 TA09b IL 500mA/DIV 2ms/DIV 3971 TA09c 3971f 21 LT3971 PACKAGE DESCRIPTION DD Package 10-Lead Plastic DFN (3mm × 3mm) (Reference LTC DWG # 05-08-1699) R = 0.115 TYP 6 0.675 ±0.05 0.38 ± 0.10 10 3.50 ±0.05 1.65 ±0.05 2.15 ±0.05 (2 SIDES) PACKAGE OUTLINE 0.25 ± 0.05 0.50 BSC 2.38 ±0.05 (2 SIDES) RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS NOTE: 1. DRAWING TO BE MADE A JEDEC PACKAGE OUTLINE M0-229 VARIATION OF (WEED-2). CHECK THE LTC WEBSITE DATA SHEET FOR CURRENT STATUS OF VARIATION ASSIGNMENT 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE PIN 1 TOP MARK (SEE NOTE 6) 3.00 ±0.10 (4 SIDES) 1.65 ± 0.10 (2 SIDES) (DD) DFN 1103 5 0.200 REF 0.75 ±0.05 2.38 ±0.10 (2 SIDES) 1 0.25 ± 0.05 0.50 BSC 0.00 – 0.05 BOTTOM VIEW—EXPOSED PAD 3971f 22 LT3971 PACKAGE DESCRIPTION MSE Package 10-Lead Plastic MSOP Exposed Die Pad , (Reference LTC DWG # 05-08-1664 Rev C) BOTTOM VIEW OF EXPOSED PAD OPTION 2.794 (.110 0.102 .004) 0.889 (.035 0.127 .005) 1 2.06 0.102 (.081 .004) 1.83 0.102 (.072 .004) 0.29 REF 5.23 (.206) MIN 2.083 (.082 0.102 3.20 – 3.45 .004) (.126 – .136) 0.05 REF DETAIL “B” CORNER TAIL IS PART OF DETAIL “B” THE LEADFRAME FEATURE. FOR REFERENCE ONLY NO MEASUREMENT PURPOSE 0.497 0.076 (.0196 .003) REF 10 0.50 0.305 0.038 (.0197) (.0120 .0015) BSC TYP RECOMMENDED SOLDER PAD LAYOUT 3.00 0.102 (.118 .004) (NOTE 3) 10 9 8 7 6 4.90 0.152 (.193 .006) 0.254 (.010) GAUGE PLANE 0.53 0.152 (.021 .006) DETAIL “A” 0.18 (.007) SEATING PLANE 1.10 (.043) MAX DETAIL “A” 0 – 6 TYP 12345 3.00 0.102 (.118 .004) (NOTE 4) 0.86 (.034) REF NOTE: 1. DIMENSIONS IN MILLIMETER/(INCH) 2. DRAWING NOT TO SCALE 3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS. INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX 0.17 – 0.27 (.007 – .011) TYP 0.50 (.0197) BSC 0.1016 (.004 0.0508 .002) MSOP (MSE) 0908 REV C 3971f Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 23 LT3971 RELATED PARTS PART LT3980 LT3970 LT3695 LT3689 DESCRIPTION COMMENTS 58V, 80V Transient Protection, 2A, 2.4MHz High Efficiency Micropower VIN(MIN) = 3.6V, VIN(MAX) = 58V, Transient to 80V, VOUT(MIN) = 0.79V, IQ = 75μA, ISD