DATASHEET
ISL6754
FN6754
Rev 2.00
June 2, 2016
ZVS Full-Bridge PWM Controller with Adjustable Synchronous Rectifier Control
The ISL6754 is a high performance extension of the Intersil
family of Zero-Voltage Switching (ZVS) full-bridge PWM
controllers. Like the ISL6752, it achieves ZVS operation by
driving the upper bridge FETs at a fixed 50% duty cycle while
the lower bridge FETs are trailing-edge modulated with
adjustable resonant switching delays.
Features
Adding to the ISL6752’s feature set are average current
monitoring and soft-start. The average current signal may be
used for average current limiting, current sharing circuits and
average current mode control. Additionally, the ISL6754
supports both voltage- and current-mode control.
• 3% current limit threshold
The ISL6754 features complemented PWM outputs for
Synchronous Rectifier (SR) control. The complemented
outputs may be dynamically advanced or delayed relative to
the PWM outputs using an external control voltage.
• Supply UVLO
This advanced BiCMOS design features precision dead time
and resonant delay control, and an oscillator adjustable to
2MHz operating frequency. Additionally, multi-pulse
suppression ensures alternating output pulses at low duty
cycles where pulse skipping may occur.
• Buffered oscillator sawtooth output
Applications
• Multi-pulse suppression
• Adjustable resonant delay for ZVS operation
• Synchronous rectifier control outputs with adjustable
delay/advance
• Voltage- or current-mode control
• Adjustable average current limit
• Adjustable dead time control
• 175µA start-up current
• Adjustable oscillator frequency up to 2MHz
• Internal over-temperature protection
• Fast current sense to output delay
• Adjustable cycle-by-cycle peak current limit
• 70ns leading edge blanking
• Pb-free (RoHS compliant)
• ZVS full-bridge converters
• Telecom and datacom power
Related Literature
• Wireless base station power
• AN1603, “ISL6752/54EVAL1Z ZVS DC/DC Power Supply
with Synchronous Rectifiers User Guide”
• File server power
• AN1619, “Designing with ISL6752DBEVAL1Z and
ISL6754DBEVAL1Z Control Cards”
• Industrial power systems
Pin Configuration
ISL6754
(20 LD QSOP)
TOP VIEW
VREF 1
20 SS
VERR 2
19 VADJ
CTBUF 3
18 VDD
RTD 4
17 OUTLL
RESDEL 5
16 OUTLR
CT 6
15 OUTUL
FB 7
14 OUTUR
RAMP 8
13 OUTLLN
CS 9
12 OUTLRN
IOUT 10
FN6754 Rev 2.00
June 2, 2016
11 GND
Page 1 of 20
ISL6754
Ordering Information
PART NUMBER
(Notes 1, 2, 3)
PART MARKING
TEMP. RANGE (°C)
ISL6754AAZA
6754 AAZ
ISL6752/54EVAL1Z
Evaluation Board
ISL6754DBEVAL1Z
ISL6754 Evaluation Daughter Board
-40 to +105
PACKAGE
(RoHS Compliant)
20 Ld QSOP
PKG.
DWG. #
M20.15
NOTES:
1. Add -T suffix for 2.5k unit tape and reel option. Please refer to TB347 for details on reel specifications.
2. These Intersil Pb-free plastic packaged products employ special Pb-free material sets, molding compounds/die attach materials, and 100% matte
tin plate plus anneal (e3 termination finish, which is RoHS compliant and compatible with both SnPb and Pb-free soldering operations). Intersil
Pb-free products are MSL classified at Pb-free peak reflow temperatures that meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020.
3. For Moisture Sensitivity Level (MSL), please see product information page for ISL6754. For more information on MSL, please see tech brief TB363.
Pin Descriptions
PIN
NUMBER
PIN
NAME
1
VREF
The 5.00V reference voltage output having 3% tolerance over line, load and operating temperature. Bypass to GND with a
0.1µF to 2.2µF low ESR capacitor.
2
VERR
The control voltage input to the inverting input of the PWM comparator. The output of an external Error Amplifier (EA) is applied
to this input, either directly or through an opto-coupler, for closed loop regulation. VERR has a nominal 1mA pull-up current
source.
When VERR is driven by an opto-coupler or other current source device, a pull-up resistor from VREF is required to linearize the
gain. Generally, a pull-up resistor on the order of 5kΩ is acceptable.
3
CTBUF
CTBUF is the buffered output of the sawtooth oscillator waveform present on CT and is capable of sourcing 2mA. It is offset
from ground by 0.40V and has a nominal valley-to-peak gain of 2. It may be used for slope compensation.
4
RTD
This is the oscillator timing capacitor discharge current control pin. The current flowing in a resistor connected between this
pin and GND determines the magnitude of the current that discharges CT. The CT discharge current is nominally 20x the
resistor current. The PWM dead time is determined by the timing capacitor discharge duration. The voltage at RTD is nominally
2.00V.
5
RESDEL
Sets the resonant delay period between the toggle of the upper FETs and the turn on of either of the lower FETs. The voltage
applied to RESDEL determines when the upper FETs switch relative to a lower FET turning on. Varying the control voltage from
0V to 2.00V increases the resonant delay duration from 0 to 100% of the dead time. The control voltage divided by 2
represents the percent of the dead time equal to the resonant delay. In practice the maximum resonant delay must be set
lower than 2.00V to ensure that the lower FETs, at maximum duty cycle, are OFF prior to the switching of the upper FETs.
6
CT
The oscillator timing capacitor is connected between this pin and GND. It is charged through an internal 200µA current source
and discharged with a user adjustable current source controlled by RTD.
7
FB
FB is the inverting inputs to the Error Amplifier (EA). The amplifier may be used as the error amplifier for voltage feedback or
used as the average current limit amplifier (IEA). If the amplifier is not used, FB should be grounded.
8
RAMP
This is the input for the sawtooth waveform for the PWM comparator. The RAMP pin is shorted to GND at the termination of
the PWM signal. A sawtooth voltage waveform is required at this input. For current-mode control this pin is connected to CS
and the current loop feedback signal is applied to both inputs. For voltage-mode control, the oscillator sawtooth waveform
may be buffered and used to generate an appropriate signal, RAMP may be connected to the input voltage through a RC
network for voltage feed forward control, or RAMP may be connected to VREF through a RC network to produce the desired
sawtooth waveform.
9
CS
This is the input to the overcurrent comparator. The overcurrent comparator threshold is set at 1.00V nominal. The CS pin is
shorted to GND at the termination of either PWM output.
Depending on the current sensing source impedance, a series input resistor may be required due to the delay between the
internal clock and the external power switch. This delay may result in CS being discharged prior to the power switching device
being turned off.
10
IOUT
Output of the 4X buffer amplifier of the sample and hold circuitry that captures and averages the CS signal.
11
GND
Signal and power ground connections for this device. Due to high peak currents and high frequency operation, a low
impedance layout is necessary. Ground planes and short traces are highly recommended.
FN6754 Rev 2.00
June 2, 2016
DESCRIPTION
Page 2 of 20
ISL6754
Pin Descriptions (Continued)
PIN
NUMBER
PIN
NAME
12
OUTLRN
13
OUTLLN
14
OUTUR
15
OUTUL
16
OUTLR
17
OUTLL
18
VDD
VDD is the power connection for the IC. To optimize noise immunity, bypass VDD to GND with a ceramic capacitor as close to
the VDD and GND pins as possible. VDD is monitored for supply voltage Undervoltage Lockout (UVLO). The start and stop
thresholds track each other resulting in relatively constant hysteresis.
19
VADJ
A 0V to 5V control voltage applied to this input sets the relative delay or advance between OUTLL/OUTLR and
OUTLLN/OUTLRN. The phase relationship between OUTUL/OUTUR and OUTLL/OUTLR is maintained regardless of the phase
adjustment between OUTLL/OUTLR and OUTLLN/OUTLRN.
Voltages below 2.425V result in OUTLLN/OUTLRN being advanced relative to OUTLL/OUTLR. Voltages above 2.575V result in
OUTLLN/OUTLRN being delayed relative to OUTLL/OUTLR. A voltage of 2.50V ±75mV results in zero phase difference. A weak
internal 50% divider from VREF results in no phase delay if this input is left floating.
The range of phase delay/advance is either zero or 40 to 300ns with the phase differential increasing as the voltage deviation
from 2.5V increases. The relationship between the control voltage and phase differential is non-linear. The gain (t/V) is low
for control voltages near 2.5V and rapidly increases as the voltage approaches the extremes of the control range. This behavior
provides the user increased accuracy when selecting a shorter delay/advance duration.
When the PWM outputs are delayed relative to the SR outputs (VADJ < 2.425V), the delay time should not exceed 90% of the
dead time as determined by RTD and CT.
20
SS
Connect the soft-start timing capacitor between this pin and GND to control the duration of soft-start. The value of the
capacitor and the internal current source determine the rate of increase of the duty cycle during start-up.
SS may also be used to inhibit the outputs by grounding through a small transistor in an open collector/drain configuration.
FN6754 Rev 2.00
June 2, 2016
DESCRIPTION
These outputs are the complements of the PWM (lower) bridge FETs. OUTLLN is the complement of OUTLL and OUTLRN is the
complement of OUTLR. These outputs are suitable for control of synchronous rectifiers. The phase relationship between each
output and its complement is controlled by the voltage applied to VADJ.
These outputs control the upper bridge FETs and operate at a fixed 50% duty cycle in alternate sequence. OUTUL controls the
upper left FET and OUTUR controls the upper right FET. The left and right designation may be switched as long as they are
switched in conjunction with the lower FET outputs, OUTLL and OUTLR.
These outputs control the lower bridge FETs, are pulse width modulated, and operate in alternate sequence. OUTLL controls
the lower left FET and OUTLR controls the lower right FET. The left and right designation may be switched as long as they are
switched in conjunction with the upper FET outputs, OUTUL and OUTUR.
Page 3 of 20
ISL6754
FN6754 Rev 2.00
June 2, 2016
Functional Block Diagram
VDD
OUTUL
50%
VDD
VREF
OUTUR
DELAY/
ADVANCE
TIMING
CONTROL
PWM
STEERING
LOGIC
UVLO
OVERTEMPERATURE
PROTECTION
OUTLL
OUTLR
PWM
OUTLLN
OUTLRN
GND
SAMPLE
AND
HOLD
VREF
VADJ
RESDEL
IOUT
CS
+
-
4X
+70 nS
LEADING
EDGE
BLANKING
1.00V
OVERCURRENT
COMPARATOR
CT
RTD
OSCILLATOR
VREF
RAMP
PWM
COMPARATOR
CTBUF
VREF
80mV
1 mA
+
-
SS
0.33
Page 4 of 20
SOFT-START
CONTROL
FIGURE 1. FUNCTIONAL BLOCK DIAGRAM
VERR
+
-
0.6V
FB
ISL6754
FN6754 Rev 2.00
June 2, 2016
Typical Application - High Voltage Input Primary Side Control ZVS Full-Bridge Converter
VIN+
CR2
CR3
T3
Q2
Q8A
R15
R16
Q8B
Q1
C3
Q5A
Q5B
C2
+
T1
C1
R18
400 VDC
Q10A
Q9A
C16
C15
Q9B
Q10B
+ VOUT
L1
Q12
C4
+
C14
Q13
R17
RETURN
Q4
Q6A
Q7A
Q6B
Q7B
C13
Q3
VIN-
R20
R19
R13
R7
T2
R6
CR1
1 VREF
SS 20
2 VERR
VADJ 19
4 RTD
ISL6754
3 CTBUF
R8
5 RESDEL
R5
R1
R4
VDD 18
C12
OUTLL 17
EL7212
OUTLR 16
6 CT
OUTUL 15
7 FB
OUTUR 14
8 RAMP
OUTLLN 13
9 CS
U1
10 IOUT
OUTLRN 12
R11
R23
C17
EL7212
T4
CR4
R21
U4
U5
R22
C18
GND 11
R24
U1
R12
BIAS
C9
VDD
C5
R2
C6
C7
U3
TL431
C10
C8
C11
U2
R14
R3
R9
R10
FIGURE 2. HIGH VOLTAGE INPUT PRIMARY SIDE CONTROL ZVS FULL-BRIDGE CONVERTER
R25
Page 5 of 20
VIN+
T3
1:1:1
Q1
Q2
Q6
Q5
CR2
R17
CR3
T1
Np:Ns:Ns=9:2:2
R18
R20
Ns
Np
+ Vout
C12
Q10A
L1
Q16
Ns
Q9A
C14
Q9B
Q10B
C15
+
+
400 VDC
C1
R19
Q15
T4
1:1:1
Q4
Q3
CR5
CR4
Q7A
C13
Q8A
R15
Q7B
R16
Q8B
RETURN
C11
C10
C9
Q11A
Q12A
Q12B
Q11B
Q13A
VIN-
Q13B
VREF
U1
R11
T2
R12
1 VREF
SS 20
2 VERR
VADJ 19
3 CTBUF
R10
4 RTD
ISL6754
CR1
5 RESDEL
R4
OUTUL 15
7 FB
OUTUR 14
9 CS
R5
10 IOUT
R9
OUTLL 17
OUTLR 16
6 CT
8 RAMP
R1
R23
VDD 18
C16
OUTLLN 13
Q14A
VREF
OUTLRN 12
Q14B
GND 11
C17
R24
C6
R14
R8
CR6
Page 6 of 20
SECONDARY
BIAS SUPPLY
R22
U3
+
C2
R21
R2
R3
C3
C4
C5
R6
R7
R13
C7
R25
C8
FIGURE 3. HIGH VOLTAGE INPUT SECONDARY SIDE CONTROL ZVS FULL-BRIDGE CONVERTER(20 LD QSOP)
C18
ISL6754
FN6754 Rev 2.00
June 2, 2016
Typical Application - High Voltage Input Secondary Side Control ZVS Full-Bridge Converter
(20 ld QSOP)
ISL6754
Absolute Maximum Ratings
Thermal Information
Supply Voltage, VDD . . . . . . . . . . . . . . . . . . . . . . . . . . (GND - 0.3V) to +22.0V
OUTxxx . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . (GND - 0.3V) to VDD
Signal Pins. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . (GND - 0.3V) to VREF + 0.3V
VREF . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . (GND - 0.3V) to 6.0V
Peak GATE Current . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .0.1A
Latch-Up (Note 6) . . . . . . . . . . . . . . . . . . . . . . . . . Class II, Level B at +85°C
Thermal Resistance Junction to Ambient (Typical)
JA (°C/W)
88
20 Lead QSOP (Note 4) . . . . . . . . . . . . . . . . . . . . . . . . . .
Maximum Junction Temperature . . . . . . . . . . . . . . . . . . . .-55°C to +150°C
Maximum Storage Temperature Range . . . . . . . . . . . . . .-65°C to +150°C
Pb-Free Reflow Profile . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . see TB493
Operating Conditions
Temperature Range
ISL6754AAxx . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .-40°C to +105°C
Supply Voltage Range (Typical). . . . . . . . . . . . . . . . . . . . . . . 9VDC to 16VDC
CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact product
reliability and result in failures not covered by warranty.
NOTES:
4. JA is measured with the component mounted on a high effective thermal conductivity test board in free air. See Tech Brief TB379 for details.
5. All voltages are with respect to GND.
6. Jedec Class II pulse conditions and failure criterion used. Level B exceptions are using a pulse limited to 50mA.
Electrical Specifications Recommended operating conditions unless otherwise noted. Refer to “Functional Block Diagram” on page 4
and “Typical Application schematics” beginning on page 5. 9V < VDD < 20V, RTD = 10.0kΩ, CT = 470pF, TA = -40°C to +105°C, Typical values are at
TA = +25°C; Parameters with MIN and/or MAX limits are 100% tested at +25°C, unless otherwise specified. Temperature limits established by
characterization and are not production tested.
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
-
-
20
V
SUPPLY VOLTAGE
Supply Voltage
Start-Up Current, IDD
VDD = 5.0V
-
175
400
µA
Operating Current, IDD
RLOAD, COUT = 0
-
11.0
15.5
mA
UVLO START Threshold
8.00
8.75
9.00
V
UVLO STOP Threshold
6.50
7.00
7.50
V
-
1.75
-
V
4.850
5.000
5.150
V
-
3
-
mV
-10
-
-
mA
5
-
-
mA
VREF = 4.85V
-15
-
-100
mA
Current Limit Threshold
VERR = VREF
0.97
1.00
1.03
V
CS to OUT Delay
Excl. LEB
-
35
-
ns
-
70
-
ns
Hysteresis
REFERENCE VOLTAGE
Overall Accuracy
IVREF = 0mA to -10mA
Long Term Stability
TA = +125°C, 1000 hours (Note 7)
Operational Current (source)
Operational Current (sink)
Current Limit
CURRENT SENSE
Leading Edge Blanking (LEB) Duration
CS to OUT Delay + LEB
TA = +25°C
-
-
150
ns
CS Sink Current Device Impedance
VCS = 1.1V
-
-
20
Ω
Input Bias Current
VCS = 0.3V
-1.0
-
1.0
µA
IOUT Sample and Hold Buffer Amplifier Gain
TA = +25°C
3.85
4.00
4.15
V/V
IOUT Sample and Hold VOH
VCS = max, ILOAD = -300µA
3.9
-
-
V
IOUT Sample and Hold VOL
VCS = 0.00V, ILOAD = 10µA
-
-
0.3
V
FN6754 Rev 2.00
June 2, 2016
Page 7 of 20
ISL6754
Electrical Specifications Recommended operating conditions unless otherwise noted. Refer to “Functional Block Diagram” on page 4
and “Typical Application schematics” beginning on page 5. 9V < VDD < 20V, RTD = 10.0kΩ, CT = 470pF, TA = -40°C to +105°C, Typical values are at
TA = +25°C; Parameters with MIN and/or MAX limits are 100% tested at +25°C, unless otherwise specified. Temperature limits established by
characterization and are not production tested. (Continued)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
-
-
20
W
RAMP
RAMP Sink Current Device Impedance
VRAMP = 1.1V
RAMP to PWM Comparator Offset
TA = +25°C
65
80
95
mV
Bias Current
VRAMP = 0.3V
-5.0
-
-2.0
µA
Minimum Duty Cycle
VERR < 0.6V
-
-
0
%
Maximum Duty Cycle (per half-cycle)
VERR = 4.20V, VCS = 0V (Note 8)
-
94
-
%
RTD = 2.00kΩ, CT = 220pF
-
97
-
%
RTD = 2.00kΩ, CT = 470pF
-
99
-
%
0.85
-
1.20
V
0.7
0.8
0.9
V
0.31
0.33
0.35
V/V
(Note 7)
0
-
4.45
V
Input Common-Mode (CM) Range
(Note 7)
0
-
VREF
V
GBWP
(Note 7)
5
-
-
MHz
VERR VOL
ILOAD = 2mA
-
-
0.4
V
VERR VOH
ILOAD = 0mA
4.20
-
-
V
VERR Pull-Up Current Source
VERR = 2.5V
0.8
1.0
1.3
mA
EA Reference
TA = +25°C
0.594
0.600
0.606
V
0.590
0.600
0.612
V
165
183
201
kHz
-10
-
10
%
PULSE WIDTH MODULATOR
Zero Duty Cycle VERR Voltage
VERR to PWM Comparator Input Offset
TA = +25°C
VERR to PWM Comparator Input Gain
Common-Mode (CM) Input Range
ERROR AMPLIFIER
EA Reference + EA Input Offset Voltage
OSCILLATOR
Frequency Accuracy, Overall
(Note 7)
Frequency Variation with VDD
TA = +25°C, (F20V- - F10V)/F10V
-
0.3
1.7
%
Temperature Stability
VDD = 10V, |F-40°C - F0°C|/F0°C
-
4.5
-
%
|F0°C - F105°C|/F25°C (Note 7)
-
1.5
-
%
-193
-200
-207
µA
19
20
23
µA/µA
0.75
0.80
0.88
V
Charge Current
TA = +25°C
Discharge Current Gain
CT Valley Voltage
Static Threshold
CT Peak Voltage
Static Threshold
2.75
2.80
2.88
V
CT Peak-to-Peak Voltage
Static Value
1.92
2.00
2.05
V
1.97
2.00
2.03
V
0
-
2.00
V
RTD Voltage
RESDEL Voltage Range
CTBUF Gain (VCTBUFP-P/VCTP-P)
VCT = 0.8V, 2.6V
1.95
2.00
2.05
V/V
CTBUF Offset from GND
VCT = 0.8V
0.34
0.40
0.44
V
CTBUF VOH
V(ILOAD = 0mA, ILOAD = -2mA), VCT = 2.6V
-
-
0.10
V
CTBUF VOL
V(ILOAD = 2mA, ILOAD = 0mA), VCT = 0.8V
-
-
0.10
V
FN6754 Rev 2.00
June 2, 2016
Page 8 of 20
ISL6754
Electrical Specifications Recommended operating conditions unless otherwise noted. Refer to “Functional Block Diagram” on page 4
and “Typical Application schematics” beginning on page 5. 9V < VDD < 20V, RTD = 10.0kΩ, CT = 470pF, TA = -40°C to +105°C, Typical values are at
TA = +25°C; Parameters with MIN and/or MAX limits are 100% tested at +25°C, unless otherwise specified. Temperature limits established by
characterization and are not production tested. (Continued)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
-60
-70
-80
µA
4.410
4.500
4.590
V
SOFT-START
Charging Current
SS = 3V
SS Clamp Voltage
SS Discharge Current
SS = 2V
Reset Threshold Voltage
TA = +25°C
10
-
-
mA
0.23
0.27
0.33
V
OUTPUT
High Level Output Voltage (VOH)
IOUT = -10mA, VDD - VOH
-
0.5
1.0
V
Low Level Output Voltage (VOL)
IOUT = 10mA, VOL - GND
-
0.5
1.0
V
Rise Time
COUT = 220pF, VDD = 15V (Note 7)
-
110
200
ns
Fall Time
COUT = 220pF, VDD = 15V (Note 7)
-
90
150
ns
UVLO Output Voltage Clamp
VDD = 7V, ILOAD = 1mA (Note 9)
-
-
1.25
V
Output Delay/Advance Range
OUTLLN/OUTLRN relative to OUTLL/OUTLR
VADJ = 2.50V
-
2
-
ns
VADJ < 2.425V
-40
-
-300
ns
VADJ > 2.575V
40
-
300
ns
2.575
-
5.000
V
0
-
2.425
V
VADJ = 0
-
300
-
ns
VADJ = 0.5V
-
105
-
ns
VADJ = 1.0V
-
70
-
ns
VADJ = 1.5V
-
55
-
ns
VADJ = 2.0V
-
50
-
ns
VADJ = VREF
-
300
-
ns
Delay/Advance Control Voltage Range
OUTLLN/OUTLRN relative to OUTLL/OUTLR
OUTLxN Delayed
VADJ Delay Time
TA = +25°C (OUTLx Delayed) (Note 10)
OUTLxN Advanced
TA = +25°C (OUTLxN Delayed)
VADJ = VREF - 0.5V
-
100
-
ns
VADJ = VREF - 1.0V
-
68
-
ns
VADJ = VREF - 1.5V
-
55
-
ns
VADJ = VREF - 2.0V
-
48
-
ns
(Note 7)
-
140
-
°C
Thermal Shutdown Clear
(Note 7)
-
125
-
°C
Hysteresis, Internal Protection
(Note 7)
-
15
-
°C
THERMAL PROTECTION
Thermal Shutdown
NOTES:
7. Limits established by characterization and are not production tested.
8. This is the maximum duty cycle achievable using the specified values of RTD and CT. Larger or smaller maximum duty cycles may be obtained using
other values for these components. See Equations 1 through 3.
9. Adjust VDD below the UVLO stop threshold prior to setting at 7V.
10. When OUTLx is delayed relative to OUTLxN (VADJ < 2.425V), the delay duration as set by VADJ should not exceed 90% of the CT discharge time (dead
time) as determined by CT and RTD.
FN6754 Rev 2.00
June 2, 2016
Page 9 of 20
ISL6754
Typical Performance Curves
1.01
1
0.99
0.98
-40
25
CT DISCHARGE CURRENT GAIN
NORMALIZED VREF
1.02
-25
-10
5
20
35
50
65
80
95
24
23
22
21
20
19
18
110
0
200
FIGURE 4. REFERENCE VOLTAGE vs TEMPERATURE
600
800
1000
FIGURE 5. CT DISCHARGE CURRENT GAIN vs RTD CURRENT
1-103
CT = 680pF
CT = 1000pF
FREQUENCY (kHz)
1-104
CT = 470pF
DEAD TIME (ns)
400
RTD CURRENT (µA)
TEMPERATURE (°C)
1-103
CT = 100pF
CT = 220pF
CT = 330pF
100
RTD = 10kΩ
100
RTD = 50kΩ
RTD = 100kΩ
10
0
10
20
30
40 50 60
RTD (kΩ)
70
80
90
FIGURE 6. DEAD TIME (DT) vs CAPACITANCE
FN6754 Rev 2.00
June 2, 2016
100
10
0.1
1
CT (nF)
10
FIGURE 7. CAPACITANCE vs FREQUENCY
Page 10 of 20
ISL6754
Functional Description
Overcurrent Operation
Features
The ISL6754 PWM is an excellent choice for low cost ZVS fullbridge applications requiring adjustable synchronous rectifier
drive. With its many protection and control features, a highly
flexible design with minimal external components is possible.
Among its many features are a very accurate overcurrent limit
threshold, thermal protection, a buffered sawtooth oscillator
output suitable for slope compensation, synchronous rectifier
outputs with variable delay/advance timing, and adjustable
frequency.
If synchronous rectification is not required, please consider the
ISL6755 controller.
Oscillator
The ISL6754 has an oscillator with a programmable frequency
range to 2MHz, which can be programmed with a resistor and
capacitor.
The switching period is the sum of the timing capacitor charge
and discharge durations. The charge duration is determined by
CT and a fixed 200µA internal current source. The discharge
duration is determined by RTD and CT.
3
t C 11.5 10 CT
S
(EQ. 1)
t D 0.06 RTD CT + 50 10
1
t SW = t C + t D = ---------f SW
S
–9
S
(EQ. 2)
(EQ. 3)
Where tC and tD are the charge and discharge times,
respectively, CT is the timing capacitor in Farads, RTD is the
discharge programming resistance in ohms, tSW is the oscillator
period, and fSW is the oscillator frequency. One output switching
cycle requires two oscillator cycles. The actual times will be
slightly longer than calculated due to internal propagation delays
of approximately 10ns/transition. This delay adds directly to the
switching duration, but also causes overshoot of the timing
capacitor peak and valley voltage thresholds, effectively
increasing the peak-to-peak voltage on the timing capacitor.
Additionally, if very small discharge currents are used, there will
be increased error due to the input impedance at the CT pin. The
maximum recommended current through RTD is 1mA, which
produces a CT discharge current of 20mA.
Two overcurrent protection mechanisms are available to the
power supply designer. The first method is cycle-by-cycle peak
overcurrent protection which provides fast response. The
cycle-by-cycle peak current limit results in pulse-by-pulse duty cycle
reduction when the current feedback signal exceeds 1.0V. When
the peak current exceeds the threshold, the active output pulse is
immediately terminated. This results in a decrease in output
voltage as the load current increases beyond the current limit
threshold. The ISL6754 operates continuously in an overcurrent
condition without shutdown.
The second method is a slower, averaging method which
produces constant or “brick-wall” current limit behavior. If
voltage-mode control is used, the average overcurrent protection
also maintains flux balance in the transformer by maintaining
duty cycle symmetry between half-cycles. If voltage-mode control
is used in a bridge topology, it should be noted that peak current
limit results in inherently unstable operation. The DC blocking
capacitors used in voltage-mode bridge topologies become
unbalanced, as does the flux in the transformer core. Average
current limit will prevent the instability and allow continuous
operation in current limit provided the control loop is designed
with adequate bandwidth.
The propagation delay from CS exceeding the current limit
threshold to the termination of the output pulse is increased by
the Leading Edge Blanking (LEB) interval. The effective delay is
the sum of the two delays and is nominally 105ns.
The current sense signal applied to the CS pin connects to the
peak current comparator and a sample and hold averaging
circuit. After a 70ns Leading Edge Blanking (LEB) delay, the
current sense signal is actively sampled during the on time, the
average current for the cycle is determined, and the result is
amplified by 4x and output on the IOUT pin. If an RC filter is
placed on the CS input, its time constant should not exceed
~50ns or significant error may be introduced on IOUT.
The maximum duty cycle, D, and percent Dead Time, DT, can be
calculated from:
tC
D = ---------t SW
(EQ. 4)
DT = 1 – D
(EQ. 5)
CHANNEL 1 (YELLOW): OUTLL
CHANNEL 3 (BLUE): CS
CHANNEL 2 (RED): OUTLR
CHANNEL 4 (GREEN): IOUT
FIGURE 8. CS INPUT vs IOUT
Figure 8 on page 11 shows the relationship between the CS
signal and IOUT under steady state conditions. IOUT is 4x the
FN6754 Rev 2.00
June 2, 2016
Page 11 of 20
ISL6754
average of CS. Figure 9 shows the dynamic behavior of the
current averaging circuitry when CS is modulated by an external
sine wave. Notice IOUT is updated by the sample and hold
circuitry at the termination of the active output pulse.
EA as described above. An external op-amp may be used as
either the current or voltage EA providing the circuit is not
allowed to source current into VERR. The external EA must only
sink current, which may be accomplished by adding a diode in
series with its output.
The 4x gain of the sample and hold buffer allows a range of 150
to 1000mV peak on the CS signal, depending on the resistor
divider placed on IOUT. The overall bandwidth of the average
current loop is determined by the integrating current EA
compensation and the divider on IOUT.
1
20 VREF
2 VERR
19 SS
3
18 VDD
4
C10
CHANNEL 1 (YELLOW): OUTLL
CHANNEL 3 (BLUE): CS
CHANNEL 2 (RED): OUTLR
CHANNEL 4 (GREEN): IOUT
150 - 1000 mV
If average overcurrent limit is desired, IOUT may be used with the
error amplifier of the ISL6754. Typically IOUT is divided down and
filtered as required to achieve the desired amplitude. The
resulting signal is input to the current error amplifier (IEA). The
IEA is similar to the voltage EA found in most PWM controllers,
except it cannot source current. Instead, VERR has a separate
internal 1mA pull-up current source.
Configure the IEA as an integrating (Type I) amplifier using the
internal 0.6V reference. The voltage applied at FB is integrated
against the 0.6V reference. The resulting signal, VERR, is applied
to the PWM comparator where it is compared to the sawtooth
voltage on RAMP. If FB is less than 0.6V, the IEA will be open loop
(can’t source current), VERR will be at a level determined by the
voltage loop, and the duty cycle is unaffected. As the output load
increases, IOUT will increase, and the voltage applied to FB will
increase until it reaches 0.6V. At this point the IEA will reduce
VERR as required to maintain the output current at the level that
corresponds to the 0.6V reference. When the output current
again drops below the average current limit threshold, the IEA
returns to an open loop condition, and the duty cycle is again
controlled by the voltage loop.
The average current control loop behaves much the same as the
voltage control loop found in typical power supplies except it
regulates current rather than voltage.
The EA available on the ISL6754 may also be used as the voltage
EA for the voltage feedback control loop rather than the current
FN6754 Rev 2.00
June 2, 2016
16 OUTLR
6
15 OUTUL
7 FB
0.6V +
8
14 OUTUR
10 IOUT
R6
17 OUTLL
5
9 CS
FIGURE 9. DYNAMIC BEHAVIOR OF CS vs IOUT
The average current signal on IOUT remains accurate provided
the output inductor current remains continuous (CCM operation).
Once the inductor current becomes discontinuous (DCM
operation), IOUT represents 1/2 the peak inductor current rather
than the average current. This occurs because the sample and
hold circuitry is active only during the on time of the switching
cycle. It is unable to detect when the inductor current reaches
zero during the off time.
ISL6754
S&H
4x
13 N/C
12 GND
11 GND
R5
R4
FIGURE 10. AVERAGE OVERCURRENT IMPLEMENTATION
The current EA cross-over frequency, assuming R6 >> (R4||R5),
is:
1
f CO = --------------------------------2 R 6 C 10
Hz
(EQ. 6)
Where fCO is the cross-over frequency. A capacitor in parallel with
R4 may be used to provide a double-pole roll-off.
The average current loop bandwidth is normally set to be much
less than the switching frequency, typically less than 5kHz and
often as slow as a few hundred hertz or less. This is especially
useful if the application experiences large surges. The average
current loop can be set to the steady state overcurrent threshold
and have a time response that is longer than the required
transient. The peak current limit can be set higher than the
expected transient so that it does not interfere with the transient,
but still protects for short-term larger faults. In essence a 2-stage
overcurrent response is possible.
The peak overcurrent behavior is similar to most other PWM
controllers. If the peak current exceeds 1.0V, the active output
pulse is terminated immediately.
If voltage-mode control is used in a bridge topology, it should be
noted that peak current limit results in inherently unstable
operation. DC blocking capacitors used in voltage-mode bridge
topologies become unbalanced, as does the flux in the
transformer core. The average overcurrent circuitry prevents this
behavior by maintaining symmetric duty cycles for each
Page 12 of 20
ISL6754
half-cycle. If the average current limit circuitry is not used, a
latching overcurrent shutdown method using external
components is recommended.
VIN
The CS to output propagation delay is increased by the Leading
Edge Blanking (LEB) interval. The effective delay is the sum of
the two delays and is 130ns maximum.
20
2
19
3
18
4
R3
Voltage Feed Forward Operation
Voltage feed forward is a technique used to regulate the output
voltage for changes in input voltage without the intervention of
the control loop. Voltage feed forward is implemented in
voltage-mode control loops, but is redundant and unnecessary in
peak current-mode control loops.
Voltage feed forward operates by modulating the sawtooth ramp
in direct proportion to the input voltage. Figure 11 demonstrates
the concept.
1
C7
ISL6754
17
5
16
6
15
7
14
8 RAMP
13
9
12
10
GND 11
FIGURE 12. VOLTAGE FEED FORWARD CONTROL
The charging time of the ramp capacitor is:
VIN
ERROR VOLTAGE
RAMP
CT
OUTLL, LR
V RAMP PEAK
t = – R 3 C 7 ln 1 – ----------------------------------------
V IN MIN
(EQ. 7)
For optimum performance, the maximum value of the capacitor
should be limited to 10nF. The maximum DC current through the
resistor should be limited to 2mA maximum. For example, if the
oscillator frequency is 400kHz, the minimum input voltage is
300V, and a 4.7nF ramp capacitor is selected, the value of the
resistor can be determined by rearranging Equation 7.
–6
–t
– 2.5 10
R 3 = ------------------------------------------------------------------------- = -----------------------------------------------------------–9
1
V
RAMP PEAK
4.7 10 ln 1 – ----------
C 7 ln 1 – ----------------------------------------
300
V
IN MIN
FIGURE 11. VOLTAGE FEED FORWARD BEHAVIOR
Input voltage feed forward may be implemented using the RAMP
input. An RC network connected between the input voltage and
ground, as shown in Figure 12, generates a voltage ramp whose
charging rate varies with the amplitude of the source voltage. At
the termination of the active output pulse, RAMP is discharged to
ground so that a repetitive sawtooth waveform is created. The
RAMP waveform is compared to the VERR voltage to determine
duty cycle. The selection of the RC components depends upon
the desired input voltage operating range and the frequency of
the oscillator. In typical applications, the RC components are
selected so that the ramp amplitude reaches 1.0V at minimum
input voltage within the duration of one half-cycle.
S
= 159
k
(EQ. 8)
Where t is equal to the oscillator period minus the dead time. If
the dead time is short relative to the oscillator period, it can be
ignored for this calculation.
If feed forward operation is not desired, the RC network may be
connected to VREF rather than the input voltage. Alternatively, a
resistor divider from CTBUF may be used as the sawtooth signal.
Regardless, a sawtooth waveform must be generated on RAMP
as it is required for proper PWM operation.
Gate Drive
The ISL6754 outputs are capable of sourcing and sinking 10mA
(at rated VOH, VOL) and are intended to be used in conjunction
with integrated FET drivers or discrete bipolar totem pole drivers.
The typical on resistance of the outputs is 50Ω.
FN6754 Rev 2.00
June 2, 2016
Page 13 of 20
ISL6754
Slope Compensation
Peak current-mode control requires slope compensation to
improve noise immunity, particularly at lighter loads, and to
prevent current loop instability, particularly for duty cycles
greater than 50%. Slope compensation may be accomplished by
summing an external ramp with the current feedback signal or by
subtracting the external ramp from the voltage feedback error
signal. Adding the external ramp to the current feedback signal is
the more popular method.
From the small signal current-mode model [1] it can be shown
that the naturally-sampled modulator gain, Fm, without slope
compensation, is:
1
Fm = ------------------Snt SW
(EQ. 9)
Where Sn is the slope of the sawtooth signal and tSW is the
duration of the half-cycle. When an external ramp is added, the
modulator gain becomes:
1
1
Fm = -------------------------------------- = --------------------------m c Snt SW
Sn + Se t SW
(EQ. 10)
N S R CS
D t SW
NS
V CS = ------------------------ I O + ------------------- V IN -------- – V O
N P N CT
2L O
NP
V
(EQ. 16)
Where VCS is the voltage across the current sense resistor and IO
is the output current at current limit.
Since the peak current limit threshold is 1.00V, the total current
feedback signal plus the external ramp voltage must sum to this
value.
V e + V CS = 1
(EQ. 17)
Substituting Equations 15 and 16 into Equation 17 and solving
for RCS yields:
(EQ. 18)
O
Se
m c = 1 + ------Sn
(EQ. 11)
The criteria for determining the correct amount of external ramp
can be determined by appropriately setting the damping factor of
the double-pole located at half the oscillator frequency. The
double-pole will be critically damped if the Q-factor is set to 1,
and over-damped for Q > 1, and under-damped for Q < 1. An
under-damped condition can result in current loop instability.
1
Q = ------------------------------------------------ m c 1 – D – 0.5
(EQ. 12)
Where D is the percent of on time during a half cycle. Setting
Q = 1 and solving for Se yields:
1
1
S e = S n --- + 0.5 ------------- – 1
1 –D
(EQ. 13)
Since Sn and Se are the on time slopes of the current ramp and
the external ramp, respectively, they can be multiplied by tON to
obtain the voltage change that occurs during tON.
1
1
V e = V n --- + 0.5 ------------- – 1
1 –D
(EQ. 14)
Where Vn is the change in the current feedback signal during the
on time and Ve is the voltage that must be added by the external
ramp.
Vn can be solved for in terms of input voltage, current transducer
components, and output inductance yielding:
V
(EQ. 15)
Where RCS is the current sense burden resistor, NCT is the
current transformer turns ratio, LO is the output inductance, VO is
FN6754 Rev 2.00
June 2, 2016
The inductor current, when reflected through the isolation
transformer and the current sense transformer to obtain the
current feedback signal at the sense resistor yields:
N P N CT
1
R CS = ------------------------ ---------------------------------------------------NS
VO
1 D
I O + -------- t SW --- + ----
2
L
Where Se is slope of the external ramp and:
t SW V R CS N
O
S 1
V e = ---------------------------------------- -------- --- + D – 0.5
NP
N CT L O
the output voltage, and NS and NP are the secondary and
primary turns, respectively.
For simplicity, idealized components have been used for this
discussion, but the effect of magnetizing inductance must be
considered when determining the amount of external ramp to
add. Magnetizing inductance provides a degree of slope
compensation to the current feedback signal and reduces the
amount of external ramp required. The magnetizing inductance
adds primary current in excess of what is reflected from the
inductor current in the secondary.
V IN Dt SW
I P = ----------------------------Lm
(EQ. 19)
A
Where VIN is the input voltage that corresponds to the duty cycle
D and Lm is the primary magnetizing inductance. The effect of
the magnetizing current at the current sense resistor, RCS, is:
I P R CS
V CS = -------------------------N CT
(EQ. 20)
V
If VCS is greater than or equal to Ve, then no additional slope
compensation is needed and RCS becomes:
N CT
R CS = -----------------------------------------------------------------------------------------------------------------------------------NS
DT SW
NS
V IN Dt SW
-------- I O + ---------------- V ------- – V O + ----------------------------Lm
NP
2L O IN N P
(EQ. 21)
If VCS is less than Ve, then Equation 16 is still valid for the value
of RCS, but the amount of slope compensation added by the
external ramp must be reduced by VCS.
Adding slope compensation may be accomplished in the
ISL6754 using the CTBUF signal. CTBUF is an amplified
representation of the sawtooth signal that appears on the CT pin.
It is offset from ground by 0.4V and is 2x the peak-to-peak
Page 14 of 20
ISL6754
amplitude of CT (0.4V to 4.4V). A typical application sums this
signal with the current sense feedback and applies the result to
the CS pin as shown in Figure 13.
1
20
2
19
3 CTBUF
18
4
17
16
5
R9
6
15
7
14
8 RAMP
13
9 CS
12
GND 11
10
R6
RCS
ISL6754
C4
Determine the amount of voltage, Ve, that must be added to the
current feedback signal using Equation 15.
Ve = 153mV
Next, determine the effect of the magnetizing current from
Equation 20.
VCS = 91mV
Using Equation 23, solve for the summing resistor, R9, from
CTBUF to CS.
R9 = 30.1kΩ
Determine the new value of RCS, R’CS, using Equation 24.
R’CS = 15.4Ω
The above discussion determines the minimum external ramp
that is required. Additional slope compensation may be
considered for design margin.
FIGURE 13. ADDING SLOPE COMPENSATION
Assuming the designer has selected values for the RC filter
placed on the CS pin, the value of R9 required to add the
appropriate external ramp can be found by superposition.
D V CTBUF – 0.4 + 0.4 R 6
V e – V CS = -----------------------------------------------------------------------------R6 + R9
RCS = 15.1Ω.
If the application requires dead time less than about 500ns, the
CTBUF signal may not perform adequately for slope
compensation. CTBUF lags the CT sawtooth waveform by 300ns
to 400ns. This behavior results in a non-zero value of CTBUF
when the next half-cycle begins when the dead time is short.
Under these situations, slope compensation may be added by
externally buffering the CT signal as shown in Figure 14.
(EQ. 22)
V
Rearranging to solve for R9 yields:
D V CTBUF – 0.4 – V e + V CS + 0.4 R 6
R 9 = -----------------------------------------------------------------------------------------------------------------V e – V CS
20
2
19
ISL6754
3
(EQ. 23)
R9
The value of RCS determined in Equation 18 or 21 must be
rescaled so that the current sense signal presented at the CS pin
is that predicted by Equation 16. The divider created by R6 and
R9 makes this necessary.
R6 + R9
R CS = --------------------- R CS
R9
1 VREF
18
4
17
5 CT
16
6
15
7
14
8 RAMP
13
12
9 CS
R6
10
GND 11
(EQ. 24)
RCS
CT
C4
Example:
VIN = 280V
VO = 12V
LO = 2.0µH
Np/Ns = 20
Lm = 2mH
IO = 55A
FIGURE 14. ADDING SLOPE COMPENSATION USING CT
Using CT to provide slope compensation instead of CTBUF
requires the same calculations, except that Equations 22 and 23
require modification. Equation 22 becomes:
Oscillator Frequency, fSW = 400kHz
2D R 6
V e – V CS = --------------------R6 + R9
Duty Cycle, D = 85.7%
and Equation 23 becomes:
NCT = 50
2D – V e + V CS R 6
R 9 = -----------------------------------------------------------V e – V CS
R6 = 499Ω
V
(EQ. 25)
(EQ. 26)
Solve for the current sense resistor, RCS, using Equation 18.
FN6754 Rev 2.00
June 2, 2016
Page 15 of 20
ISL6754
The buffer transistor used to create the external ramp from CT
should have a sufficiently high gain (>200) so as to minimize the
required base current. Whatever base current is required reduces
the charging current into CT and will reduce the oscillator
frequency.
UR and LL off. The direction of the primary and secondary
currents are indicated by IP and IS, respectively.
VIN+
UL
D1
The ISL6754 is a full-bridge Zero-Voltage Switching (ZVS) PWM
controller that behaves much like a traditional hard-switched
topology controller. Rather than drive the diagonal bridge
switches simultaneously, the upper switches (OUTUL, OUTUR) are
driven at a fixed 50% duty cycle and the lower switches (OUTLL,
OUTLR) are pulse width modulated on the trailing edge.
DEADTIME
PWM
PWM
OUTLR
VOUT+
IP
RTN
LL
LR
D2
VIN-
FIGURE 17. UL - LR POWER TRANSFER CYCLE
The UL - LR power transfer period terminates when switch LR
turns off as determined by the PWM. The current flowing in the
primary cannot be interrupted instantaneously, so it must find an
alternate path. The current flows into the parasitic switch
capacitance of LR and UR which charges the node to VIN and
then forward biases the body diode of upper switch UR.
CT
PWM
IS
LL
ZVS Full-Bridge Operation
OUTLL
UR
PWM
VIN+
UL
UR
D1
IS
VOUT+
LL
OUTUR
RESONANT
DELAY
IP
RTN
OUTUL
LL
RESDEL
WINDOW
D2
FIGURE 15. BRIDGE DRIVE SIGNAL TIMING
VIN-
To understand how the ZVS method operates one must include
the parasitic elements of the circuit and examine a full switching
cycle.
VIN+
UL
UR
D1
VOUT+
LL
RTN
LL
LR
D2
VIN-
FIGURE 16. IDEALIZED FULL-BRIDGE
In Figure 16, the power semiconductor switches have been
replaced by ideal switch elements with parallel diodes and
capacitance, the output rectifiers are ideal, and the transformer
leakage inductance has been included as a discrete element.
The parasitic capacitance has been lumped together as switch
capacitance, but represents all parasitic capacitance in the
circuit including winding capacitance. Each switch is designated
by its position, Upper Left (UL), Upper Right (UR), Lower Left (LL),
and Lower Right (LR). The beginning of the cycle, shown in
Figure 17, is arbitrarily set as having switches UL and LR on and
FN6754 Rev 2.00
June 2, 2016
LR
FIGURE 18. UL - UR FREE-WHEELING PERIOD
The primary leakage inductance, LL, maintains the current which
now circulates around the path of switch UL, the transformer
primary, and switch UR. When switch LR opens, the output
inductor current free-wheels through both output diodes, D1 and
D2. During the switch transition, the output inductor current
assists the leakage inductance in charging the upper and lower
bridge FET capacitance.
The current flow from the previous power transfer cycle tends to
be maintained during the free-wheeling period because the
transformer primary winding is essentially shorted. Diode D1
may conduct very little or none of the free-wheeling current,
depending on circuit parasitics. This behavior is quite different
than occurs in a conventional hard-switched full-bridge topology
where the free-wheeling current splits nearly evenly between the
output diodes, and flows not at all in the primary.
This condition persists through the remainder of the half-cycle.
During the period when CT discharges, also referred to as the
dead time, the upper switches toggle. Switch UL turns off and
switch UR turns on. The actual timing of the upper switch toggle
is dependent on RESDEL, which sets the resonant delay. The
voltage applied to RESDEL determines how far in advance the
toggle occurs prior to a lower switch turning on. The ZVS
transition occurs after the upper switches toggle and before the
diagonal lower switch turns on. The required resonant delay is
1/4 of the period of the LC resonant frequency of the circuit
Page 16 of 20
ISL6754
formed by the leakage inductance and the parasitic capacitance.
The resonant transition may be estimated from Equation 27.
1
= --- ----------------------------------2
2
R
1
--------------- – ---------2
LL CP
4L L
(EQ. 27)
Where is the resonant transition time, LL is the leakage
inductance, CP is the parasitic capacitance, and R is the
equivalent resistance in series with LL and CP.
The resonant delay is always less than or equal to the dead time
and may be calculated using Equation 28.
V resdel
resdel = -------------------- DT
2
(EQ. 28)
S
The UR - LL power transfer period terminates when switch LL
turns off as determined by the PWM. The current flowing in the
primary must find an alternate path. The current flows into the
parasitic switch capacitance which charges the node to VIN and
then forward biases the body diode of upper switch UL. As before,
the output inductor current assists in this transition. The primary
leakage inductance, LL, maintains the current, which now
circulates around the path of switch UR, the transformer primary,
and switch UL. When switch LL opens, the output inductor
current free-wheels predominantly through diode D1. Diode D2
may actually conduct very little or none of the free-wheeling
current, depending on circuit parasitics. This condition persists
through the remainder of the half-cycle.
VIN+
UL
UR
D1
Where resdel is the desired resonant delay, Vresdel is a voltage
between 0V and 2V applied to the RESDEL pin, and DT is the
dead time (see Equations 1 through 5).
IS
VOUT+
LL
IP
RTN
When the upper switches toggle, the primary current that was
flowing through UL must find an alternate path. It
charges/discharges the parasitic capacitance of switches UL and
LL until the body diode of LL is forward biased. If RESDEL is set
properly, switch LL will be turned on at this time. The output
inductor does not assist this transition. It is purely a resonant
transition driven by the leakage inductance.
VIN+
UL
UR
D1
IS
VOUT+
LL
IP
LL
LR
D2
VIN-
FIGURE 21. UR - UL FREE-WHEELING PERIOD
When the upper switches toggle, the primary current that was
flowing through UR must find an alternate path. It
charges/discharges the parasitic capacitance of switches UR
and LR until the body diode of LR is forward biased. If RESDEL is
set properly, switch LR will be turned on at this time.
VIN+
UL
UR
D1
RTN
LL
VOUT+
LL
LR
D2
IS
IP
VIN-
RTN
LL
FIGURE 19. UPPER SWITCH TOGGLE AND RESONANT TRANSITION
LR
D2
The second power transfer period commences when switch LL
closes. With switches UR and LL on, the primary and secondary
currents flow as indicated in Figure 20.
VIN+
UL
UR
D1
VOUT+
LL
RTN
LL
LR
D2
VIN-
FIGURE 22. UPPER SWITCH TOGGLE AND RESONANT TRANSITION
The first power transfer period commences when switch LR
closes and the cycle repeats. The ZVS transition requires that the
leakage inductance has sufficient energy stored to fully charge
the parasitic capacitances. Since the energy stored is
proportional to the square of the current (1/2 LLIP2), the ZVS
resonant transition is load dependent. If the leakage inductance
is not able to store sufficient energy for ZVS, a discrete inductor
may be added in series with the transformer primary.
VIN-
FIGURE 20. UR - LL POWER TRANSFER CYCLE
FN6754 Rev 2.00
June 2, 2016
Page 17 of 20
ISL6754
Synchronous Rectifier Outputs and Control
The ISL6754 provides double-ended PWM outputs, OUTLL and
OUTLR, and Synchronous Rectifier (SR) outputs, OUTLLN and
OUTLRN. The SR outputs are the complements of the PWM
outputs. It should be noted that the complemented outputs are
used in conjunction with the opposite PWM output, i.e., OUTLL
and OUTLRN are paired together and OUTLR and OUTLLN are
paired together.
CT
OUTLL
OUTLR
OUTLLN
(SR1)
CT
OUTLRN
(SR2)
OUTLL
OUTLR
FIGURE 25. WAVEFORM TIMING WITH SR OUTPUTS DELAYED,
2.575V < VADJ < 5.00V
OUTLLN
(SR1)
Setting VADJ to VREF/2 results in no delay on any output. The no
delay voltage has a ±75mV tolerance window. Control voltages
below the VREF/2 zero delay threshold cause the PWM outputs,
OUTLL/LR, to be delayed. Control voltages greater than the
VREF/2 zero delay threshold cause the SR outputs, OUTLLN/LRN,
to be delayed. It should be noted that when the PWM outputs,
OUTLL/LR, are delayed, the CS to output propagation delay is
increased by the amount of the added delay.
OUTLRN
(SR2)
FIGURE 23. BASIC WAVEFORM TIMING
Referring to Figure 23, the SRs alternate between being both on
during the free-wheeling portion of the cycle (OUTLL/LR off), and
one or the other being off when OUTLL or OUTLR is on. If OUTLL is
on, its corresponding SR must also be on, indicating that OUTLRN
is the correct SR control signal. Likewise, if OUTLR is on, its
corresponding SR must also be on, indicating that OUTLLN is the
correct SR control signal.
A useful feature of the ISL6754 is the ability to vary the phase
relationship between the PWM outputs (OUTLL, OUT LR) and the
their complements (OUTLLN, OUTLRN) by ±300ns. This feature
allows the designer to compensate for differences in the
propagation times between the PWM FETs and the SR FETs. A
voltage applied to VADJ controls the phase relationship.
CT
OUTLL
The delay feature is provided to compensate for mismatched
propagation delays between the PWM and SR outputs as may be
experienced when one set of signals crosses the
primary-secondary isolation boundary. If required, individual
output pulses may be stretched or compressed as required using
external resistors, capacitors, and diodes.
When the PWM outputs are delayed, the 50% upper outputs are
equally delayed, so the resonant delay setting is unaffected.
On/Off Control
The ISL6754 does not have a separate enable/disable control pin.
The PWM outputs, OUTLL/OUTLR, may be disabled by pulling VERR
to ground. Doing so reduces the duty cycle to zero, but the upper
50% duty cycle outputs, OUTUL/OUTUR, will continue operation.
Likewise, the SR outputs OUTLLN/OUTLRN will be active high.
Pulling soft-start to ground will disable all outputs and set them
to a low condition
Fault Conditions
OUTLR
OUTLLN
(SR1)
OUTLRN
(SR2)
A fault condition occurs if VREF or VDD fall below their Undervoltage
Lockout (UVLO) thresholds or if the thermal protection is triggered.
When a fault is detected the outputs are disabled low. When the
fault condition clears the outputs are re-enabled.
An overcurrent condition is not considered a fault and does not
result in a shutdown.
Thermal Protection
FIGURE 24. WAVEFORM TIMING WITH PWM OUTPUTS DELAYED,
0V < VADJ < 2.425V
FN6754 Rev 2.00
June 2, 2016
Internal die over-temperature protection is provided. An
integrated temperature sensor protects the device should the
junction temperature exceed +140°C. There is approximately
+15°C of hysteresis.
Page 18 of 20
ISL6754
Ground Plane Requirements
Careful layout is essential for satisfactory operation of the device.
A good ground plane must be employed. VDD and VREF should
be bypassed directly to GND with good high frequency
capacitance.
References
[1] Ridley, R., “A New Continuous-Time Model for Current Mode
Control”, IEEE Transactions on Power Electronics, Vol. 6, No.
2, April 1991.
Revision History
The revision history provided is for informational purposes only and is believed to be accurate, but not warranted.
Please go to the web to make sure that you have the latest revision.
DATE
REVISION
CHANGE
June 2, 2016
FN6754.2
Updated entire datasheet applying Intersil’s new standards.
Added Related Literature, Revision History and About Intersil sections.
Updated Note 1.
Added Note 3.
Added evaluation boards to the ordering information table on page 2.
In the “Electrical Specifications” on page 7 under the “REFERENCE VOLTAGE” section updated the test
conditions from “IVREF = 0mA to 10mA” to “IVREF = 0mA to -10mA”.
Updated POD M20.15 to the latest revision changes are as follows:
- Note 2 changed from “Dimensioning and tolerancing per ANSI Y14.5M-1982.” to “Dimensioning and
tolerancing conform to AMSE Y14.5m-1994.”
-Changed title from “20 LEAD SHRINK SMALL OUTLINE PLASTIC PACKAGE” to “20 LEAD QUARTER SIZE
OUTLINE PLASTIC PACKAGE (QSOP)”
- Update to new POD format by removing table with dimensions and placing dimensions on drawing
instead. Added land pattern.
About Intersil
Intersil Corporation is a leading provider of innovative power management and precision analog solutions. The company's products
address some of the largest markets within the industrial and infrastructure, mobile computing and high-end consumer markets.
For the most updated datasheet, application notes, related documentation and related parts, please see the respective product
information page found at www.intersil.com.
You may report errors or suggestions for improving this datasheet by visiting www.intersil.com/ask.
Reliability reports are also available from our website at www.intersil.com/support.
© Copyright Intersil Americas LLC 2008-2016. All Rights Reserved.
All trademarks and registered trademarks are the property of their respective owners.
For additional products, see www.intersil.com/en/products.html
Intersil products are manufactured, assembled and tested utilizing ISO9001 quality systems as noted
in the quality certifications found at www.intersil.com/en/support/qualandreliability.html
Intersil products are sold by description only. Intersil may modify the circuit design and/or specifications of products at any time without notice, provided that such
modification does not, in Intersil's sole judgment, affect the form, fit or function of the product. Accordingly, the reader is cautioned to verify that datasheets are
current before placing orders. Information furnished by Intersil is believed to be accurate and reliable. However, no responsibility is assumed by Intersil or its
subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Intersil or its subsidiaries.
For information regarding Intersil Corporation and its products, see www.intersil.com
FN6754 Rev 2.00
June 2, 2016
Page 19 of 20
ISL6754
Package Outline Drawing
M20.15
20 LEAD QUARTER SIZE OUTLINE PLASTIC PACKAGE (QSOP)
Rev 2, 1/11
20
INDEX
AREA
1
2
0.244 (6.19)
0.157 (3.98) 0.228 (5.80)
0.150 (3.81)
4
3
GAUGE
PLANE
TOP VIEW
6
0.050 (1.27)
0.25
0.010
SEATING PLANE
3
0.069 (1.75)
0.053 (1.35)
0.344 (8.74)
0.337 (8.56)
0.016 (0.41)
0.0196 (0.49)
5
0.0099 (0.26)
8°
0°
0.012 (0.30)
0.008 (0.20)
0.025
(0.635 BSC)
8
0.010 (0.25)
0.004 (0.10)
0.061 MAX (1.54 MIL)
SIDE VIEW
0.010 (0.25)
0.007 (0.18)
DETAIL "X"
NOTES:
0.015 (0.38) x 20
0.025 (0.64) x 18
1. Symbols are defined in the “MO Series Symbol List” in Section 2.2 of
Publication Number 95.
2. Dimensioning and tolerancing conform to AMSE Y14.5m-1994.
20
0.060 (1.52) x 20
3. Dimension does not include mold flash, protrusions or gate burrs.
Mold flash, protrusion and gate burrs shall not exceed 0.15mm
(0.006 inch) per side.
4. Dimension does not include interlead flash or protrusions. Interlead
flash and protrusions shall not exceed 0.25mm (0.010 inch) per
side.
0.220(5.59)
5. The chamfer on the body is optional. If it is not present, a visual index
feature must be located within the crosshatched area.
6. Length of terminal for soldering to a substrate.
7. Terminal numbers are shown for reference only.
1
2
3
TYPICAL RECOMMENDED LAND PATTERN
8. Dimension does not include dambar protrusion. Allowable
dambar protrusion shall be 0.10mm (0.004 inch) total in excess of
dimension at maximum material condition.
9. Controlling dimension: INCHES. Converted millimeter dimensions
are not necessarily exact.
FN6754 Rev 2.00
June 2, 2016
Page 20 of 20
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