PM6670AS

PM6670AS Complete DDR2/3 memory power supply controller Features ■ ■ Switching section (VDDQ) – 4.5 V to 36 V input voltage range – 0.9 V, ±1% voltage reference – 1.8 V (DDR2) or 1.5 V (DDR3) fixed output voltages – 0.9 V to 2.6 V adjustable output voltage – 1.237 V ±1% reference voltage available – Very fast load transient response using constant-on-time control loop – No RSENSE current sensing using low side MOSFETs' RDS(ON) – Negative current limit – Latched OVP and UVP – Soft-start internally fixed at 3 ms – Selectable pulse skipping at light load – selectable no-audible (33 kHz) pulse skip mode – Ceramic output capacitors supported – Output voltage ripple compensation VTT LDO and VTTREF – 2 Apk LDO with foldback for VTT – Remote VTT sensing – High-Z VTT output in S3 – Ceramic output capacitors supported – ±15 mA Low noise buffered reference Applications ■ DDR2/3 memory supply ■ Digital TV system ■ SSTL18, SSTL15 and HSTL bus termination Table 1. Description The device PM6670AS is a complete DDR2/3 power supply regulator designed to meet JEDEC specifications. It integrates a constant on-time (COT) buck controller, a 2 Apk sink/source low drop out regulator and a 15 mA low noise buffered reference. The COT architecture assures fast transient response supporting both electrolytic and ceramic output capacitors. An embedded integrator control loop compensates the DC voltage error due to the output ripple. The 2 Apk sink/source linear regulator provides the memory termination voltage with fast load transient response. The device is full compliant with system sleep states S3 and S4/S5, providing LDO output high impedance in Suspend-To-RAM and Tracking Discharge of all outputs in Suspend-To-Disk. Device summary Order codes Package PM6670AS Packaging Tube VFQFPN-24 4x4 (Exposed pad) PM6670ASTR February 2010 Tape and reel Doc ID 14436 Rev 2 1/53 www.st.com 53 Contents PM6670AS Contents 1 Typical application circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 2 Pin settings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 3 2.1 Connections . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 2.2 Pin description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 Electrical data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 3.1 Maximum rating . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 3.2 Thermal data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 3.3 Recommended operating conditions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 4 Electrical characteristics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9 5 Typical operating characteristics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13 6 Block diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18 7 Device description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19 7.1 7.2 2/53 VDDQ section - constant on-time PWM controller . . . . . . . . . . . . . . . . . . 20 7.1.1 Constant-on-time architecture . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22 7.1.2 Output ripple compensation and loop stability . . . . . . . . . . . . . . . . . . . . 23 7.1.3 Pulse-skip and no-audible pulse-skip modes . . . . . . . . . . . . . . . . . . . . . 28 7.1.4 Mode-of-operation selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29 7.1.5 Current sensing and current limit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30 7.1.6 POR, UVLO and soft-start . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31 7.1.7 Power-Good signal . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33 7.1.8 VDDQ output discharge . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33 7.1.9 Gate drivers . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34 7.1.10 Reference voltage and bandgap . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34 7.1.11 Over voltage and under voltage protections . . . . . . . . . . . . . . . . . . . . . 35 7.1.12 Device thermal protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35 VTTREF buffered reference and VTT LDO section . . . . . . . . . . . . . . . . . 36 7.2.1 VTT and VTTREF soft-start . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 36 7.2.2 VTTREF and VTT outputs discharge . . . . . . . . . . . . . . . . . . . . . . . . . . . 36 Doc ID 14436 Rev 2 PM6670AS Contents 7.3 8 S3 and S5 power management pins . . . . . . . . . . . . . . . . . . . . . . . . . . . . 37 Application information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 38 8.1 External components selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 38 8.1.1 Inductor selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 40 8.1.2 Input capacitor selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 41 8.1.3 Output capacitor selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 42 8.1.4 MOSFETs selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 43 8.1.5 Diode selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 45 8.1.6 VDDQ current limit setting . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 46 8.1.7 All ceramic capacitors application . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 47 9 Package mechanical data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 50 10 Revision history . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 52 Doc ID 14436 Rev 2 3/53 Typical application circuit PM6670AS 1 Typical application circuit Figure 1. Application circuit +5V R LP VIN C C IN2 IN3 R1 C IN R2 BOOT VOSC VCC AVCC DSCG 11 MODE 23 LDOIN PHASE 4 VTTREF LGATE 2 VTTSNS 1 PM6670AS PM6670A HGATE CSNS 21 16 VTTGND VSNS 9 14 13 7 C OUT2 4/53 C OUT R LIM C 10 C VDDQ 19 PGND 15 L 17 VTT 5 C BOOT 20 COMP 24 VTT 22 VREF///. OUT3 8 S5///’// C 18 S3/////////’ VTTREF 6 PGO/ IN4 12 SGND_- C SEL 3 VDDQ (LDO input) BYP Doc ID 14436 Rev 2 INT PM6670AS Pin settings 2 Pin settings 2.1 Connections 24 CSNS PHASE HGATE BOOT LDOIN VTT Pin connection (through top view) 19 1 18 VTTGND VCC VTTSNS LGATE DDRSEL PGND PM6670A PM6675S PM6670AS VTTREF PG SGND S3 6 Doc ID 14436 Rev 2 COMP MODE VSNS 7 S5 DSCG 13 12 VOSC AVCC VREF Figure 2. 5/53 Pin settings 2.2 PM6670AS Pin description Table 2. N° Pin 1 VTTGND LDO power ground. Connect to negative terminal of VTT output capacitor. 2 VTTSNS LDO remote sensing. Connect as close as possible to the load via a low noise PCB trace. 3 DDRSEL DDR voltage selector (if MODE is tied to VCC) or pulse-skip/no-audible pulse-skip selector in adjustable mode (MODE voltage lower than 3 V). See Mode of Operation Selection section for details. 4 VTTREF Low noise buffered DDR Reference Voltage. A 22 nF (minimum) ceramic bypass capacitor is required in order to achieve stability. 5 SGND Ground reference for analog circuitry, control logic and VTTREF buffer. Connect together with the thermal pad and VTTGND to a low impedance ground plane. See the Application Note for details. 6 AVCC +5 V supply for internal logic. Connect to +5 V rail through a simple RC filtering network. 7 VREF High accuracy output voltage reference (1.237 V) for multilevel pins setting. It can deliver up to 50 μA. Connect a 100 nF capacitor between VREF and SGND in order to enhance noise rejection. 8 VOSC Frequency selection. Connect to the central tap of a resistor divider to set the desired switching frequency. The pin cannot be left floating. See Device Description section for details. VSNS VDDQ output remote sensing. Discharge path for VDDQ in non-tracking discharge. Input for internal resistor divider that provides VDDQ/2 to VTTREF and VTT. Connect as close as possible to the load via a low noise PCB trace. 10 MODE Mode of operation selector. If MODE pin voltage is higher than 4 V, the fixed output mode is selected. If MODE pin voltage is lower than 4 V, it is used as negative input of the error amplifier. See Mode of Operation Selection section for details. 11 COMP DC voltage error compensation input for the switching section. Refer to Mode of Operation Selection section for more details. 12 DSCG Discharge mode selection. Refer to output discharge selection section for tracking/non-tracking discharge or no-discharge options. 13 S5 Switching controller enable. Connect to S5 system status signal to meet S0S5 power management states compliance. See Power Management Pins section for details. S5 pin can't be left floating. 14 S3 Linear regulator enable. Connect to S3 system status signal to meet S0-S5 power management states compliance. See Power Management Pins section for details. S3 pin can't be left floating. 15 PG Power-Good signal (open drain output). High when VDDQ output voltage is within ±10% of nominal value. 16 PGND Power ground for the switching section. 17 LGATE Low-side gate driver output. 9 6/53 Pin functions Function Doc ID 14436 Rev 2 PM6670AS Pin settings Table 2. Pin functions (continued) N° Pin Function 18 VCC +5 V low-side gate driver supply. Bypass with a 100 nF capacitor to PGND. 19 CSNS Current Sense Input for the switching section. This pin must be connected through a resistor to the drain of the synchronous rectifier (RDSon sensing) 20 PHASE Switch node connection and return path for the high-side gate driver. 21 HGATE High-side gate driver output 22 BOOT Bootstrap capacitor connection. Positive supply input of the high-side gate driver. 23 LDOIN Linear regulator input. Connect to VDDQ in normal configuration or to a lower supply to reduce the power dissipation. A 10 μF bypass ceramic capacitor is suggested for noise rejection enhancement. See Device Description section for more details. 24 VTT LDO linear regulator output. Bypass with a 20 μF (2x10 μF MLCC) filter capacitor. Doc ID 14436 Rev 2 7/53 Electrical data PM6670AS 3 Electrical data 3.1 Maximum rating Table 3. Absolute maximum ratings (1) Symbol Parameter Value VAVCC AVCC to SGND -0.3 to 6 VVCC VCC to SGND -0.3 to 6 PGND, VTTGND to SGND VPHASE PTOT Unit -0.3 to 0.3 HGATE and BOOT to PHASE -0.3 to 6 HGATE and BOOT to PGND -0.3 to 44 PHASE to SGND -0.3 to 38 LGATE to PGND -0.3 to VVCC +0.3 V CSNS, PG, S3, S5, DSCG, COMP, VSNS, VOSC, VREF, MODE, DDRSEL to GND -0.3 to VAVCC + 0.3 VTTREF, VREF, VTT, VTTSNS to SGND -0.3 to VAVCC + 0.3 LDOIN, VTT, VTTREF, LDOIN to VTTGND -0.3 to VAVCC + 0.3 Power dissipation @TA = 25 °C 2.3 W 1. Free air operating conditions unless otherwise specified. Stresses beyond those listed under “absolute maximum ratings” may cause permanent damage to the device. Exposure to absolute maximum rated conditions for extended periods may affect device reliability. 3.2 Thermal data Table 4. Thermal data Symbol 3.3 Parameter Unit 42 °C/W RthJA Thermal resistance junction to ambient TSTG Storage temperature range - 50 to 150 °C TA Operating ambient temperature range - 40 to 125 °C TJ Junction operating temperature range - 40 to 125 °C Recommended operating conditions Table 5. Recommended operating conditions Symbol Min Typ Max Input voltage range 4.5 - 36 VAVCC IC supply voltage 4.5 - 5.5 VVCC IC supply voltage 4.5 - 5.5 VIN 8/53 Value Parameter Doc ID 14436 Rev 2 Unit V PM6670AS 4 Electrical characteristics Electrical characteristics TA = -25 °C to 85 °C, VCC = AVCC = +5 V and LDOIN connected to VDDQ output if not otherwise specified (a) Table 6. Electrical characteristics Symbol Parameter Test condition Min Typ Max 0.8 2 Unit Supply section Operating current S3, S5, MODE and DDRSEL connected to AVCC, No Load on VTT and VTTREF outputs. VCC connected to AVCC ISTR Operating current in STR S5, MODE and DDRSEL connected to AVCC, S3 tied to SGND, No Load on VTTREF. VCC connected to AVCC ISH Operating current in shutdown S3 and S5 tied to SGND. Discharge mode active. VCC connected to AVCC IIN AVCC under voltage lockout upper threshold mA 4.1 0.6 1 1 10 4.25 4.4 μA V UVLO AVCC under voltage lockout lower threshold 3.85 UVLO hysteresis 4.0 4.1 70 mV ON-time (SMPS) tON On-time duration MODE and DDRSEL high, VVSNS = 2 V VOSC = 300 mV 650 750 850 VOSC = 500 mV 390 450 510 300 350 ns 1.237 1.249 V 4 mV ns OFF-time (SMPS) tOFFMIN Minimum off time Voltage reference Voltage accuracy 4.5 V < VIN < 36 V Load regulation -50 μA< IVREF < 50 μA Undervoltage lockout fault threshold 1.224 -4 800 VDDQ output a. Specifications referred to TJ = TA. All the parameters at operating temperatures extremes are guaranteed by design and statistical correlation (not production tested) Doc ID 14436 Rev 2 9/53 Electrical characteristics Table 6. PM6670AS Electrical characteristics (continued) Symbol Parameter Test condition DDR3 VDDQ output voltage MODE connected to AVCC, DDRSEL tied to SGND, no load DDR2 VDDQ output voltage MODE and DDRSEL connected to AVCC, no load Feedback accuracy MODE and DDRSEL connected to AVCC, no load Min Typ Max Unit 1.5 V VVDDQ 1.8 -1.5 1.5 % 130 μA 6 mV Current limit and zero crossing comparator ICSNS CSNS input bias current 110 Comparator offset Positive current limit threshold -6 Rsense = 1 kΩ VPGND - VCSNS Fixed negative current limit threshold VZC,OFFS 120 Zero crossing comparator offset -11 120 mV 110 mV -5 1 HGATE high state (pull-up) 2.0 3 HGATE low state (pull-down) 1.8 2.7 LGATE high state (pull-up) 1.4 2.1 LGATE low state (pull-down) 0.6 0.9 mV High and low side gate drivers HGATE driver on-resistance Ω LGATE driver on-resistance UVP/OVP protections and PGOOD SIGNAL (SMPS only) OVP Over voltage threshold 112 115 118 UVP Under voltage threshold 67 70 73 Power-good upper threshold 107 110 113 Power-good lower threshold 86 90 93 % PGOOD IPG,LEAK PG leakeage current PG forced to 5 V VPG,LOW PG low-level voltage IPG,SINK = 4 mA 1 μA 150 250 mV 3 4 ms Soft-start section (SMPS) Soft-start ramp time (4 steps current limit) 1.5 Soft-start current limit step μA 30 Soft-end section 10/53 VDDQ discharge resistance in non-tracking discharge mode 15 VTT discharge resistance in non-tracking discharge mode 15 25 35 Ω Doc ID 14436 Rev 2 25 35 PM6670AS Table 6. Electrical characteristics Electrical characteristics (continued) Symbol Parameter Test condition VTTREF discharge resistance in non-tracking discharge mode VDDQ Output threshold synchronous for final tracking to non-tracking discharge transition Min Typ Max Unit 1 1.5 2 kΩ 0.2 0.4 0.6 V 1 10 VTT LDO section ILDOIN,ON ILDOIN, STR ILDOIN, STD IVTTSNS, LDO input bias current in full- S3 = S5 = +5 V, on state No Load on VTT LDO input bias current in suspend-to-RAM state S3 = 0 V, S5 = +5 V, No Load on VTT 10 LDO input bias current in suspend-to-disk state S3 = S5 = 0 V, No Load on VTT 1 VTTSNS bias current S3 = +5 V, S5 = +5 V, VVTTSNS = VVSNS /2 1 VTTSNS leakage current S3 = 0V, S5 = +5 V, VVTTSNS = VVSNS /2 1 BIAS IVTTSNS, LEAK IVTT,LEAK VTT leakage current S3 = 0V, S5 = +5 V, VVTT = VVSNS /2 LDO linear regulator output voltage (DDR2) S3 = S5= +5 V, IVTT = 0 A, MODE = DDRSEL = +5 V LDO linear regulator output voltage (DDR3) S3 = S5= +5 V, IVTT = 0 A, MODE = +5 V, DDRSEL = 0 V -10 0.9 V 0.75 -20 20 LDO output accuracy respect S3 = S5 = MODE = +5 V, to VTTREF -1 A < IVTT < 1 A -25 25 S3 = S5 = MODE = +5 V, -2 A < IVTT < 2 A -35 35 LDO source current limit IVTT,CL LDO sink current limit μA 10 S3 = S5 = MODE = + 5 V, -1 mA < IVTT < 1 mA VVTT μA VVTT < 1.10*(VVSNS /2) 2 2.3 3 VVTT > 1.10*(VVSNS /2) 1 1.15 1.4 VVTT > 0.90*(VVSNS /2) -3 -2.3 -2 VVTT < 0.90*(VVSNS /2) -1.4 -1.15 -1 mV A VTTREF section VTTREF output voltage VVTTREF VTTREF output voltage accuracy respect to VSNS/2 IVTTREF VTTREF current limit IVTTREF = 0 A, VVSNS = 1.8 V -15 mA < IVTTREF < 15 mA, VVSNS = 1.8 V VTTREF = 0 or VSNS Doc ID 14436 Rev 2 0.9 -2 V 2 ±40 % mA 11/53 Electrical characteristics Table 6. PM6670AS Electrical characteristics (continued) Symbol Parameter Test condition Min Typ Max Unit Power management section Turn OFF level 0.4 S3,S5 Turn ON level 1.6 MODE pin high level threshold VMODE VAVCC 0.7 VAVCC 1.3 MODE pin low level threshold DDRSEL pin high level threshold VDDRSEL VAVCC 0.8 DDRSEL pin middle level window VAVCC 1.5 1.0 DDRSEL pin low level threshold 0.5 DSCG pin high level threshold VDSCG V VAVCC 0.8 DSCG pin middle level window 2.0 1.0 DSCG pin low level threshold 0.5 IIN,LEAK Logic inputs leakage current S3, S5 = 5 V 10 μA IIN3,LEAK Multilevel inputs leakage current MODE, DDRSEL and DSCG = 5 V 10 μA VOSC input leakage current VOSC = 500 mV 1 μA IOSC, LEAK Thermal shutdown TSHDN Shutdown temperature (1) 150 1. Guaranteed by design. Not production tested 12/53 Doc ID 14436 Rev 2 °C PM6670AS Typical operating characteristics Typical operating characteristics Figure 3. Efficiency vs load - 1.5 V and 1.8 V, Figure 4. Vin = 12 V 100 90 80 Efficiency (%) 70 60 50 40 DDR2 - Forced PWM DDR2 - No-Audible P-S DDR2 - Pulse-Skip DDR3 - Forced PWM DDR3 - No-Audible P-S DDR3 - Pulse-Skip 30 20 10 0 0.001 0.01 0.1 1 Swiching frequency (kHz 5 10 Output current (A) 500 450 400 350 300 250 200 150 100 50 0 0.001 Forced PW M No-Audible P-S Pulse-Skip 0.01 0.1 Switching frequency vs input voltage, 1.8 V Figure 6. 500 500 450 450 400 350 300 250 200 10 Switching frequency vs input voltage, 1.5 V 400 350 300 250 200 150 150 0.0 5.0 10.0 15.0 20.0 25.0 30.0 0.0 5.0 10.0 Input voltage (V) Figure 7. 15.0 20.0 25.0 30.0 Input voltage (V) VDDQ line regulation, 1.8 V, 7 A Figure 8. 1.8000 VDDQ line regulation, 1.5 V, 7 A 1.4980 1.4975 1.7990 Output voltage (V) 1.7980 Forced PWM 1.4970 Forced PWM Output voltage (V) 1 Output curre nt (A) Switching frequency (kHz) Switching frequency (kHz) Figure 5. Switching frequency vs load - 1.8 V, Vin = 12 V No-Audible P-S Pulse-Skip 1.7970 1.7960 No-Audible P-S 1.4965 Pulse-Skip 1.4960 1.4955 1.4950 1.7950 1.4945 1.7940 1.4940 0.0 5.0 10.0 15.0 20.0 25.0 30.0 0.0 Input voltage (V) 5.0 10.0 15.0 20.0 25.0 30.0 Input voltage (V) Doc ID 14436 Rev 2 13/53 Typical operating characteristics Figure 9. PM6670AS VDDQ load regulation, 1.8 V, Vin = 12 V Figure 10. VDDQ load regulation, 1.5 V, Vin = 12 V 1.530 Forced PWM No-Audible P-S Pulse-Skip 1.850 1.840 Output voltage (V) Output voltage (V) 1.860 1.830 1.820 1.810 1.800 0.001 0.01 0.1 1 1.510 1.500 1.490 1.480 1.470 0.001 10 Forced PWM No-Audible P-S Pulse-Skip 1.520 0.01 Output current (A) 0.940 0.790 0.930 0.780 0.920 0.910 0.900 0.890 10 0.770 0.760 0.750 0.740 -1.5 -0.5 0.5 1.5 2.5 0.730 -2.5 Output current (A) Figure 13. VTTREF load regulation, 0.9 V, VSNS = 1.8 V 14/53 1 Figure 12. VTT load regulation, 0.75 V, LDOIN = 1.5 V Output voltage (V) Output voltage (V) Figure 11. VTT load regulation, 0.9 V, LDOIN = 1.8 V 0.880 -2.5 0.1 Output current (A) -1.5 -0.5 0.5 1.5 2.5 Output current (A) Figure 14. No-audible pulse-skip waveforms Doc ID 14436 Rev 2 PM6670AS Typical operating characteristics Figure 15. Power-up sequence - AVCC above UVLO Figure 16. VDDQ soft-start, 1.8 V, heavy load Figure 17. - 1.8 A to 1.8 A VTT load transient, 0.9 V Figure 18. 0 mA to 9 mA VTTREF load transient, 0.9 V Doc ID 14436 Rev 2 15/53 Typical operating characteristics PM6670AS Figure 19. Non-tracking (soft) discharge Figure 20. Tracking (fast) discharge, LDOIN = VDDQ Figure 21. 0 A to 10 A VDDQ load transient, PWM Figure 22. 10 A to 0 A VDDQ load transient, PWM 16/53 Doc ID 14436 Rev 2 PM6670AS Figure 23. Over-voltage protection, VDDQ = 1.8 V Typical operating characteristics Figure 24. Under-voltage protection, VDDQ = 1.8 V Doc ID 14436 Rev 2 17/53 Block diagram 6 PM6670AS Block diagram Figure 25. Functional and block diagram VOSC VREF Vr = 0.9V 1.236V Bandgap BOOT VTTSNS Level shifter Ton HGATE 1-shot LDOIN PHASE Ton min 1-shot VTT Anti Cross Conduction VCC Toff min 1-shot TD LGATE NTD PGND HIZ Zero Crossing & Current Limit VTTGND _ BEN R VTTREF VREF COMP + SWEN Vr +10% R NTD CSNS gm - + - Vr + PG + UVP/OVP SGND AVCC Vr Vr -10% UVLO SWEN TD NTD BEN HIZ VSNS DDR3 DDRSEL CONTROL LOGIC DSCG adj S3 Table 7. 18/53 NTD Thermal Shutdown S5 MODE Legend SWEN Switching controller enable TD Tracking discharge enable NTD Non-tracking discharge enable BEN VTTREF buffer enable HIZ LDO high impedance mode enable Doc ID 14436 Rev 2 fix PM6670AS 7 Device description Device description The PM6670AS is designed to satisfy DDR2-3 power supply requirements combining a synchronous buck controller, a 15 mA buffered reference and a high-current low-drop out (LDO) linear regulator capable of sourcing and sinking up to 2 Apk. The switching controller section is a high-performance, pseudo-fixed frequency, constant-on-time (COT) based regulator specifically designed for handling fast load transient over a wide range of input voltages. The DDR2-3 supply voltage VDDQ can be easily set to 1.8 V (DDR2) or 1.5 V (DDR3) without additional components. The output voltage can also be adjusted in the 0.9 V to 2.6 V range using an external resistor divider. The switching mode power supply (SMPS) can handle different modes of operation in order to minimize noise or power consumption, depending on the application needs. A lossless current sensing scheme, based on the low-side MOSFET’s on resistance avoids the need for an external current sense resistor. The output of the linear regulator (VTT) tracks the memory’s reference voltage VTTREF within ±30 mV over the full operating load conditions. The input of the LDO can be either VDDQ or a lower voltage rail in order to reduce the total power dissipation. Linear regulator stability is achieved by filtering its output with a ceramic capacitor (20 μF or greater). The reference voltage (VTTREF) section provides a voltage equal to one half of VSNS with an accuracy of 1%. This regulator can source and sink up to ±15 mA. A 10 nF to 100 nF bypass capacitor is required between VTTREF and SGND for stability. According to DDR2/3 JEDEC specifications, when the system enters the Suspend-To-RAM state the LDO output is left in high impedance while VTTREF and VDDQ are still alive. When the suspend-to-disk state (S3 and S5 tied to ground) is entered, all outputs are actively discharged when either tracking or non-tracking discharge is selected. Doc ID 14436 Rev 2 19/53 Device description 7.1 PM6670AS VDDQ section - constant on-time PWM controller The PM6670AS uses a pseudo-fixed frequency, constant on-time (COT) controller as the core of the switching section. It is well known that the COT controller uses a relatively simple algorithm and uses the ripple voltage derived across the output capacitor’s ESR to trigger the on-time one-shot generator. In this way, the output capacitor’s ESR acts as a current sense resistor providing the appropriate ramp signal to the PWM comparator. Nearly constant switching frequency is achieved by the system’s loop in steady-state operating conditions by varying the on-time duration, avoiding thus the need for a clock generator. The on-time one shot duration is directly proportional to the output voltage, sensed at VSNS pin, and inversely proportional to the input voltage, sensed at the VOSC pin, as follows: Equation 1 TON = K OSC VSNS +τ VOSC where KOSC is a constant value (130ns typ.) and τ is the internal propagation delay (40 ns typ.). The one-shot generator directly drives the high-side MOSFET at the beginning of each switching cycle allowing the inductor current to increase; after the on-time has expired, an off-time phase, in which the low-side MOSFET is turned on, follows. The off-time duration is solely determined by the output voltage: when lower than the set value (i.e. the voltage at VSNS pin is lower than the internal reference VR = 0.9 V), the synchronous rectifier is turned off and a new cycle begins (Figure 26). Figure 26. Inductor current and output voltage in steady state conditions Inductor current Output voltage Vreg Ton 20/53 Toff Doc ID 14436 Rev 2 t PM6670AS Device description The duty-cycle of the buck converter is, in steady-state conditions, given by Equation 2 V OUT D = -------------V IN The switching frequency is thus calculated as Equation 3 fSW VOUT α VIN D 1 = = = OSC ⋅ V TON α OUT K OSC K OSC SNS VOSC where Equation 4a V OSC α OSC = -------------V IN Equation 4b V SNS α OUT = -------------V OUT Referring to the typical application schematic (figures on cover page and Figure 27), the final expression is then: Equation 5 fSW = α OSC R2 1 = ⋅ K OSC R1 + R 2 K OSC Even if the switching frequency is theoretically independent from battery and output voltages, parasitic parameters involved in power path (like MOSFETs' on-resistance and inductor's DCR) introduce voltage drops responsible for slight dependence on load current. In addition, the internal delay is due to a small dependence on input voltage. The PM6670AS switching frequency can be set by an external divider connected to the VOSC pin. Doc ID 14436 Rev 2 21/53 Device description PM6670AS Figure 27. Switching frequency selection and VOSC pin VIN PM6670AS PM6670 R1 VOSC R2 The suggested voltage range for VOSC pin is 0.3V to 2V, for better switching frequency programmability. 7.1.1 Constant-on-time architecture Figure 28 shows the simplified block diagram of the constant-on-time controller. The switching regulator of the PM6670AS owns a one-shot generator that ignites the highside MOSFET when the following conditions are simultaneously satisfied: the PWM comparator is high (i.e. output voltage is lower than Vr = 0.9 V), the synchronous rectifier current is below the current limit threshold and the minimum off-time has expired. A minimum off-time constraint (300 ns typ.) is introduced to assure the boot capacitor charge and allow inductor valley current sensing on low-side MOSFET. A minimum on-time is also introduced to assure the start-up switching sequence. Once the on-time has timed out, the high side switch is turned off, while the synchronous rectifier is ignited according to the anti-cross conduction management circuitry. When the output voltage reaches the valley limit (determined by internal reference Vr = 0.9 V), the low-side MOSFET is turned off according to the anti-cross conduction logic once again, and a new cycle begins. 22/53 Doc ID 14436 Rev 2 PM6670AS Device description Figure 28. Switching section simplified block diagram 7.1.2 Output ripple compensation and loop stability The loop is closed connecting the center tap of the output divider (internally, when the fixed output voltage is chosen, or externally, using the MODE pin in the adjustable output voltage mode). The feedback node is the negative input of the error comparator, while the positive input is internally connected to the reference voltage (Vr = 0.9 V). When the feedback voltage becomes lower than the reference voltage, the PWM comparator goes to high and sets the control logic, turning on the high-side MOSFET. After the On-Time (calculated as previously described) the system releases the high-side MOSFET and turns on the synchronous rectifier. The voltage drop along ground and supply PCB paths, used to connect the output capacitor to the load, is a source of DC error. Furthermore the system regulates the output voltage valley, not the average, as shown in Figure 26. Thus, the voltage ripple on the output capacitor is an additional source of DC error. To compensate this error, an integrative network is introduced in the control loop, by connecting the output voltage to the COMP pin through a capacitor (CINT) as shown in Figure 29. Doc ID 14436 Rev 2 23/53 Device description PM6670AS Figure 29. Circuitry for output ripple compensation COMP PIN VOLTAGE ΔV Vr VREF t COMP OUTPUT VOLTAGE I=gm(V1-Vr) + - PWM Comparator CFILT gm ΔV CINT RINT t + VCINT Vr RFb1 V1 RFb2 ESR VSNS COUT The additional capacitor is used to reduce the voltage on the COMP pin when higher than 300 mVpp and is unnecessary for most of applications. The trans conductance amplifier (gm) generates a current, proportional to the DC error, used to charge the CINT capacitor. The voltage across the CINT capacitor feeds the negative input of the PWM comparator, forcing the loop to compensate the total static error. An internal voltage clamp forces the COMP pin voltage range to ±150 mV with respect to VREF. This is useful to avoid or smooth output voltage overshoot during a load transient. When the pulse-skip mode is entered, the clamping range is automatically reduced to 60mV in order to enhance the recovering capability. In the ripple amplitude is larger than 150 mV, an additional capacitor CFILT can be connected between the COMP pin and ground to reduce ripple amplitude, otherwise the integrator will operate out of its linearity range. This capacitor is unnecessary for most of applications and can be omitted. The design of the external feedback network depends on the output voltage ripple. If the ripple is higher than approximately 20mV, the correct CINT capacitor is usually enough to keep the loop stable. The stability of the system depends firstly on the output capacitor zero frequency. The following condition must be satisfied: Equation 6 fSW > k ⋅ fZout = k 2π ⋅ C out ⋅ ESR where k is a fixed design parameter (k > 3). It determines the minimum integrator capacitor value: 24/53 Doc ID 14436 Rev 2 PM6670AS Device description Equation 7 CINT > gm Vr ⎛f ⎞ Vout 2π ⋅ ⎜ SW − fZout ⎟ k ⎝ ⎠ ⋅ where gm = 50 μs is the integrator trans conductance. In order to ensure stability it must be also verified that: Equation 8 CINT > gm Vr ⋅ 2π ⋅ fZout VOUT If the ripple on the COMP pin is greater than the integrator 150 mV, the auxiliary capacitor CFILT can be added. If q is the desired attenuation factor of the output ripple, CFILT is given by: Equation 9 CFILT = CINT ⋅ (1 − q) q In order to reduce the noise on the COMP pin, it is possible to add a resistor RINT that, together with CINT and CFILT, becomes a low pass filter. The cutoff frequency fCUT must be much greater (10 or more times) than the switching frequency: Equation 10 RINT = 1 CINT ⋅ CFILT 2π ⋅ fCUT ⋅ CINT + CFILT If the ripple is very small (lower than approximately 20 mV), a different compensation network, called “Virtual-ESR” Network, is needed. This additional circuit generates a triangular ripple that is added to the output voltage ripple at the input of the integrator. The complete control scheme is shown in Figure 30. Doc ID 14436 Rev 2 25/53 Device description PM6670AS Figure 30. “Virtual-ESR” Network COMP PIN VOLTAGE T NODE VOLTAGE ΔV2 VREF ΔV1 t VREF t COMP I=gm(V1-Vr) - PWM Comparator RINT CINT CFILT gm R1 Vr C VSNS OUTPUT VOLTAGE ΔV + - T R + RFb1 V1 RFb2 ESR COUT t The ripple on the COMP pin is the sum of the output voltage ripple and the triangular ripple generated by the Virtual-ESR Network. In fact the Virtual-ESR Network behaves like a another equivalent series resistor RVESR. A good trade-off is to design the network in order to achieve an RVESR given by: Equation 11 R VESR = VRIPPLE − ESR ΔIL where ΔIL is the inductor current ripple and VRIPPLE is the total ripple at the T node, chosen greater than approximately 20 mV. The new closed-loop gain depends on CINT. In order to ensure stability it must be verified that: Equation 12 CINT > gm Vr ⋅ 2π ⋅ fZ Vout where: Equation 13 fZ = 1 2π ⋅ C out ⋅ R TOT and: 26/53 Doc ID 14436 Rev 2 PM6670AS Device description Equation 14 RTOT = ESR + RVESR Moreover, the CINT capacitor must meet the following condition: Equation 15 fSW > k ⋅ fZ = k 2π ⋅ C out ⋅ R TOT where RTOT is the sum of the ESR of the output capacitor and the equivalent ESR given by the Virtual-ESR Network (RVESR). The k parameter must be greater than unity (k > 3) and determines the minimum integrator capacitor value CINT: Equation 16 CINT > gm Vr ⋅ ⎛ fSW ⎞ Vout 2π ⋅ ⎜ − fZ ⎟ ⎝ k ⎠ The capacitor of the Virtual-ESR Network, C, is chosen as follows: Equation 17 C > 5 ⋅ CINT and R is calculated to provide the desired triangular ripple voltage: Equation 18 R= L R VESR ⋅ C Finally the R1 resistor is calculated according to expression 19: Equation 19 ⎛ 1 ⎞ ⎟ R ⋅ ⎜⎜ ⋅ π ⋅ fZ ⎟⎠ C ⎝ R1 = 1 R− C ⋅ π ⋅ fZ Doc ID 14436 Rev 2 27/53 Device description 7.1.3 PM6670AS Pulse-skip and no-audible pulse-skip modes High efficiency at light load conditions is achieved by PM6670AS entering the pulse-skip mode (if enabled). When one of the two fixed output voltages is set, pulse-skip power saving is a default feature. At light load conditions the zero-crossing comparator truncates the lowside switch on-time as soon as the inductor current becomes negative; in this way the comparator determines the on-time duration instead of the output ripple (see Figure 31). Figure 31. Inductor current and output voltage at light load with pulse-skip Inductor current VDDQ Output Vreg TON TOFF t TIDLE As a consequence, the output capacitor is left floating and its discharge depends solely on the current drained from the load. When the output ripple on the pin COMP falls under the reference, a new shot is triggered and the next cycle begins. The pulse-skip mode is naturally obtained enabling the zero-crossing comparator and automatically takes part in the COT algorithm when the inductor current is about half the ripple current amount, i.e. migrating from continuous conduction mode (C.C.M.) to discontinuous conduction mode (D.C.M.). The output current threshold related to the transition between PWM mode and pulse-skip mode can be approximately calculated as: Equation 20 ILOAD (PWM2Skip) = VIN − VOUT ⋅ TON 2 ⋅L At higher loads, the inductor current never crosses zero and the device works in pure PWM mode with a switching frequency around the nominal value. A physiological consequence of pulse-skip mode is a more noisy and asynchronous (than normal conditions) output, mainly due to very low load. If the pulse-skip is not compatible with the application, the PM6670AS, when set in adjustable mode-of-operation, allows the user to choose between forced-PWM and no-audible pulse-skip alternative modes (see 4.1.4 for details). No-audible pulse-skip mode Some audio-noise sensitive applications cannot accept the switching frequency to enter the audible range as is possible in pulse-skip mode with very light loads. For this reason, the PM6670AS implements an additional feature to maintain a minimum switching frequency of 33 kHz despite a slight efficiency loss. At very light load conditions, if any switching cycle 28/53 Doc ID 14436 Rev 2 PM6670AS Device description has taken place within 30 μs (typ.) since the last one (because the output voltage is still higher than the reference), a no-audible pulse-skip cycle begins. The low-side MOSFET is turned on and the output is driven to fall until the reference has been crossed. Then, the high-side switch is turned on for a TON period and, once it has expired, the synchronous rectifier is enabled until the inductor current reaches the zero-crossing threshold (see Figure 32). Figure 32. Inductor current and output voltage at light load with non-audible pulse-skip Inductor current VDDQ Output Vreg TMAX TON TOFF t TIDLE For frequencies higher than 33 kHz (due to heavier loads) the device works in the same way as in Pulse-Skip mode. It is important to notice that in both Pulse-Skip and No-Audible Pulse-Skip modes the switching frequency changes not only with the load but also with the input voltage. 7.1.4 Mode-of-operation selection Figure 33. MODE and DDRSEL multifunction pin configurations VDDQ VDDQ +5V +5V PM6670 PM6670AS PM6670 PM6670AS R9 R9 MODE MODE R8 R8 DDRSEL (a) VREF DDRSEL (b) The PM6670AS has been designed to satisfy the widest range of applications involving DDR2/3 memories, SSTL15-18 buses termination and I/O supplies for CPU/Chipset. The device is provided with multilevel pins which allow the user to choose the appropriate configuration. The MODE pin is used to firstly decide between fixed preset or adjustable (user defined) output voltages. Doc ID 14436 Rev 2 29/53 Device description PM6670AS When the MODE pin is connected to +5 V, the PM6670AS allows setting the VDDQ voltage to 1.8 V or 1.5 V just forcing the DDRSEL multilevel pin to +5 V or to ground respectively (see Figure 33a). In this condition the pulse-skip feature is enabled. This device configuration is suitable for standard DDR2/3 memory supply applications avoiding the need for an external, high accuracy, divider for output voltage setting. Applications requiring different output voltages can be managed by PM6670AS simply setting the adjustable mode. If MODE pin voltage is higher than 4 V, the fixed output mode is selected. Connecting an external divider to the MODE pin (Figure 33b), it is used as negative input of the error amplifier and the output voltage is given by expression (21). Equation 21 VDDQ ADJ = 0.9 ⋅ R8 + R9 R8 VDDQ output voltage can be set in the range from 0.9 V to 2.6 V. Adjustable mode automatically switches DDRSEL pin to become the power saving algorithm selector: if tied to +5 V, the forced-PWM (fixed frequency) control is performed. If grounded or connected to VREF pin (1.237 V reference voltage), the pulse-skip or non-audible pulse-skip modes are respectively selected. Table 8. Mode-of-operation settings summary Mode VMODE > 4.3 V DDRSEL VDDQ VDDRSEL > 4.2 V 1.8 V 1 V < VDDRSEL < 3.5 V Operating mode Pulse-Skip 1.5 V 4.2 V VMODE < 3.7 V 1V < VDDRSEL < 3.5 V VDDRSEL < 0.5 V 7.1.5 Forced-PWM ADJ Non-Audible Pulse-skip Pulse-Skip Current sensing and current limit The PM6670AS switching controller uses a valley current sensing algorithm to properly handle the current limit protection and the inductor current zero-crossing information. The current is sensed during the conduction time of the low-side MOSFET. The current sensing element is the on-resistance of the low-side switch. The sensing scheme is visible in Figure 34. 30/53 Doc ID 14436 Rev 2 PM6670AS Device description Figure 34. Current sensing scheme VIN PM6670AS PM6670 HGATE VOUT PHASE 100µA·RILIM CSNS IVALLEY·RDSon LGATE PGND An internal 120 μA current source is connected to CSNS pin that is also the non-inverting input of the positive current limit comparator. When the voltage drop developed across the sensing parameter equals the voltage drop across the programming resistor RILIM, the controller skips subsequent cycles until the overcurrent condition is detected or the output UV protection latches off the device (see Section 7.1.11: Over voltage and under voltage protections on page 35). Referring to Figure 34, the RDSon sensing technique allows high efficiency performance without the need for an external sensing resistor. The on-resistance of the MOSFET is affected by temperature drift and nominal value spread of the parameter itself; this must be considered during the RILIM setting resistor design. It must be taken into account that the current limit circuit actually regulates the inductor valley current. This means that RILIM must be calculated to set a limit threshold given by the maximum DC output current plus half of the inductor ripple current: Equation 22 ICL = 120μA ⋅ RILIM R DSon The PM6670AS provides also a fixed negative current limit to prevent excessive reverse inductor current when the switching section sinks current from the load in forced-PWM (3rd quadrant working conditions). This negative current limit threshold is measured between PHASE and PGND pins, comparing the drop magnitude on PHASE pin with an internal 120 mV fixed threshold. 7.1.6 POR, UVLO and soft-start The PM6670AS automatically performs an internal startup sequence during the rising phase of the analog supply of the device (AVCC). The switching controller remains in a stand-by state until AVCC crosses the upper UVLO threshold (4.2 V typ.), keeping active the internal discharge MOSFETs (only if AVCC > 1 V). Doc ID 14436 Rev 2 31/53 Device description PM6670AS The soft-start allows a gradual increase of the internal current limit threshold during start-up reducing the input/output surge currents. At the beginning of start-up, the PM6670AS current limit is set to 25% of nominal value and the under voltage protection is disabled. Then, the current limit threshold is sequentially brought to 100% in four steps of approximately 750 μs (Figure 35). Figure 35. Soft-start waveforms Switching output Current limit threshold Time S5 After a fixed 3 ms total time, the soft-start finishes and UVP is released: if the output voltage doesn't reach the Power-Good lower threshold within soft-start duration, the UVP condition is detected and the device performs a soft end and latches off. Depending on the load conditions, the inductor current may or may not reach the nominal value of the current limit during the soft-start (Figure 36 shows two examples). Figure 36. Soft-start at heavy load (a) and short-circuit (b) conditions, pulse-skip enabled (a) 32/53 (b) Doc ID 14436 Rev 2 PM6670AS 7.1.7 Device description Power-Good signal The PG pin is an open drain output used to monitor output voltage through VSNS (in fixed output voltage mode) or MODE (in adjustable output voltage mode) pins and is enabled after the soft-start timer has expired. PG signal is held low if the VDDQ output voltage drops 10% below or rises 10% above the nominal regulated value. The PG output can sink current up to 4 mA. 7.1.8 VDDQ output discharge Active discharge of VDDQ output occurs when PM6670AS enters the suspend-to-disk system state (S3 and S5 tied to GND) and DSCG pin has been properly set. Figure 37. DSCG pin connection for discharge mode selection +5V PM6670AS PM6670 VREF DSCG The PM6670AS allows the user to choose between fast discharge (tracking discharge), soft discharge (non-tracking discharge) or no discharge modes. Voltage on DSCG multilevel pin determines discharge mode as shown in Table 2 on page 6. Table 9. Discharge mode selection DSCG voltage Soft-end type Description VDSCG >4.2V No discharge All outputs left floating. 1V< VDSCG +150 °C LGATE pin is forced high after the soft-end, then the device latches off. Exit by a power-on reset or toggling S5 after 15°C temperature drop. Doc ID 14436 Rev 2 35/53 Device description 7.2 PM6670AS VTTREF buffered reference and VTT LDO section The PM6670AS provides the required DDR2/3 reference voltage on the VTTREF pin. The internal buffer tracks half the voltage on the VSNS pin and has a sink and source capability up to 15 mA. Higher currents rapidly deteriorate the output accuracy. A 10 nF to 100 nF (33 nF typ.) bypass capacitor to SGND is required for stability. The VTT low-drop-out linear regulator has been designed to sink and source up to 2 A peak current and 1 A continuously. The VTT voltage tracks VTTREF within ±35 mV. A remote voltage sensing pin (VTTSNS) is provided to recovery voltage drops due to parasitic resistance. In DDR2/3 applications, the linear regulator input LDOIN is typically connected to VDDQ output; connecting LDOIN pin to a lower voltage, if available in the system, reduces the power dissipation of the LDO. A minimum output capacitance of 20 μF (2x10 μF MLCC capacitors) is enough to assure stability and fast load transient response. 7.2.1 VTT and VTTREF soft-start Soft-start on VTT and VTTREF outputs is achieved by current clamping. The LDO linear regulator is provided with a current foldback protection: when the output voltage exits the internal ±10% VTT-Good window, the output current is clamped at ±1 A. Re-entering VTT-Good window releases the current limit clamping. The foldback mechanism naturally implements a two steps soft-start charging the output capacitors with a 1 A constant current. Something similar occurs at VTTREF pin, where the output capacitor is smoothly charged at a fixed 40 mA current limit. 7.2.2 VTTREF and VTT outputs discharge The tracking discharge mechanism involves the VTT linear regulator. When the suspend-todisk state is entered, the switching regulator is turned OFF. At the same time the LDO drains a 1A constant current from LDOIN and keeps VTT in track with VTTREF that, in turn, is half the voltage at the VSNS pin. When the VDDQ output reaches 400mV, the PM6670AS switches on the internal discharge MOSFETs to complete the process (see Section 7.1.8: VDDQ output discharge on page 33). In Soft Discharge (i.e. non-tracking discharge) the PM6670AS disables the internal regulators and suddenly turns on the discharge MOSFETs on each output. 36/53 Doc ID 14436 Rev 2 PM6670AS 7.3 Device description S3 and S5 power management pins According to DDR2/3 memories supply requirements, the PM6670AS can manage all S0 to S5 system states by connecting S3-S5 pins to their respective sleep-mode signals in the notebook's motherboard. Keeping S3 and S5 high, the S0 (Full-On) state is decoded and the outputs are alive. In S3 state (S5 = 1, S3 = 0), the PM6670AS maintains VDDQ and VTTREF outputs active and VTT output in high-impedance as needed. In S4/S5 states (S5 = S3 = 0) all outputs are turned off and, according to DSCG pin voltage, the proper soft-end is performed. Table 11. S3 and S5 Sleep-states decoding S3 S5 System State VDDQ VTTREF VTT 1 1 S0 (Full-On) On On On 0 1 S3 (Suspend-To-RAM) On On Off (Hi-Z) 0 0 S4/S5 (Suspend-To-Disk) Off (Discharge) Off (Discharge) Off (Discharge) Doc ID 14436 Rev 2 37/53 Application information 8 PM6670AS Application information The purpose of this chapter is to show the design procedure of the switching section. The design starts from three main specifications: ● The input voltage range, provided by the battery or the AC adapter. The two extreme values (VINMAX and VINmin) are important for the design. ● The maximum load current, indicated by ILOAD,MAX. ● The maximum allowed output voltage ripple VRIPPLE,MAX. It's also possible that specific designs should involve other specifications. The following paragraphs will guide the user into a step-by-step design. 8.1 External components selection The PM6670AS uses a pseudo-fixed frequency, constant on-time (COT) controller as the core of the switching section. The switching frequency can be set by connecting an external divider to the VOSC pin. The voltage seen at this pin must be greater than 0.8 V and lower than 2 V in order to ensure system's linearity. Nearly constant switching frequency is achieved by the system's loop in steady-state operating conditions by varying the on-time duration, avoiding thus the need for a clock generator. The on-time one shot duration is directly proportional to the output voltage, sensed at VSNS pin, and inversely proportional to the input voltage, sensed at the VOSC pin, as follows: Equation 24 TON = K OSC VSNS +τ VOSC where KOSC is a constant value (130 ns typ.) and τ is the internal propagation delay (40 ns typ.). The duty cycle of the buck converter is, under steady state conditions, given by Equation 25 D= VOUT VIN The switching frequency is thus calculated as Equation 26 fSW 38/53 VOUT α VIN 1 D = = = OSC ⋅ VSNS TON α OUT K OSC K OSC ⋅ VOSC Doc ID 14436 Rev 2 PM6670AS Application information where Equation 27a α OSC = VOSC VIN α OUT = VSNS VOUT Equation 27b Referring to the typical application schematic (figure in cover page and Figure 27), the final expression is then: Equation 28 fSW = α OSC R2 1 = ⋅ K OSC R1 + R 2 K OSC The switching frequency directly affects two parameters: ● Inductor size: greater frequencies mean smaller inductances. In most of the applications, real estate solutions (i.e. low-profile power inductors) are mandatory also with high saturation and r.m.s. currents. ● Efficiency: switching losses are proportional to the frequency. Generally, higher frequencies imply lower efficiency. Even if the switching frequency is theoretically independent from input and output voltages, parasitic parameters involved in power path (like MOSFETs' on-resistance and inductor's DCR) introduce voltage drops responsible for a slight dependence on load current. In addition, the internal delay is due to a light dependence on input voltage. Table 12. Typical values for switching frequency selection R1 (kΩ) R2 (kΩ) Approx switching frequency (kHz) 330 11 250 330 13 300 330 15 350 330 18 400 330 20 450 330 22 500 Doc ID 14436 Rev 2 39/53 Application information 8.1.1 PM6670AS Inductor selection Once the switching frequency has been defined, the inductance value depends on the desired inductor ripple current. Low inductance value means great ripple current that brings poor efficiency and great output noise. On the other hand a great current ripple is desirable for fast transient response when a load step is applied. High inductance brings higher efficiency, but the transient response is critical, especially if VINmin - VOUT is small. Moreover a minimum output ripple voltage is necessary to assure system stability and jitter-free operations (see Output capacitor selection paragraph). The product of the output capacitor's ESR multiplied by the inductor ripple current must be taken into consideration. A good trade-off between the transient response time, the efficiency, the cost and the size is choosing the inductance value in order to maintain the inductor ripple current between 20% and 50% (usually 40%) of the maximum output current. The maximum inductor ripple current, ΔIL,MAX, occurs at the maximum input voltage. Given these considerations, the inductance value can be calculated with the following expression: Equation 29 L= VIN − VOUT VOUT ⋅ fsw ⋅ ΔIL VIN where fSW is the switching frequency, VIN is the input voltage, VOUT is the output voltage and ΔIL is the inductor ripple current. Once the inductor value is determined, the inductor ripple current is then recalculated: Equation 30 ΔIL,MAX = VIN,MAX − VOUT fsw ⋅ L ⋅ VOUT VIN,MAX The next step is the calculation of the maximum r.m.s. inductor current: Equation 31 IL,RMS = (ILOAD,MAX )2 + (ΔIL,MAX )2 12 The inductor must have an r.m.s. current greater than IL,RMS in order to assure thermal stability. Then the calculation of the maximum inductor peak current follows: Equation 32 IL,PEAK = ILOAD,MAX + ΔIL,MAX 2 IL,PEAK is important in inductor selection in term of its saturation current. 40/53 Doc ID 14436 Rev 2 PM6670AS Application information The saturation current of the inductor should be greater than IL,PEAK not only in case of hard saturation core inductors. Using soft-ferrite cores it is possible (but not advisable) to push the inductor working near its saturation current. In Table 13 some inductors are listed. Table 13. Evaluated inductors (@fsw = 400 kHz) Manufacturer Series Inductance (µH) +40 °C rms current (A) -30% saturation current (A) COILCRAFT MLC1538-102 1 13.4 21.0 COILCRAFT MLC1240-901 0.9 12.4 24.5 COILCRAFT MVR1261C-112 1.1 20 20 WURTH 7443552100 1 16 20 COILTRONICS HC8-1R2 1.2 16.0 25.4 In pulse-skip mode, low inductance values produce a better efficiency versus load curve, while higher values result in higher full-load efficiency because of the smaller current ripple. 8.1.2 Input capacitor selection In a buck topology converter the current that flows through the input capacitor is pulsed and with zero average value. The RMS input current can be calculated as follows: Equation 33 2 ICinRMS = ILOAD ⋅ D ⋅ (1 − D) + 1 D ⋅ (ΔIL )2 12 Neglecting the second term, the equation 10 is reduced to: Equation 34 ICinRMS = ILOAD D ⋅ (1 − D) The losses due to the input capacitor are thus maximized when the duty-cycle is 0.5: Equation 35 Ploss = ESR Cin ⋅ ICinRMS (max)2 = ESR Cin ⋅ (0.5 ⋅ ILOAD (max))2 The input capacitor should be selected with a RMS rated current higher than ICINRMS(max). Tantalum capacitors are good in terms of low ESR and small size, but they occasionally can burn out if subjected to very high current during operation. Multi-layers-ceramic-capacitors (MLCC) have usually a higher RMS current rating with smaller size and they remain the best choice. The drawback is their quite high cost. Doc ID 14436 Rev 2 41/53 Application information PM6670AS It must be taken into account that in some MLCC the capacitance decreases when the operating voltage is near the rated voltage. In Table 14 some MLCC suitable for most of applications are listed. Table 14. Evaluated MLCC for input filtering Manufacturer 8.1.3 Capacitance (μF) Rated voltage (V) Series Maximum Irms @100 kHz (A) TAIYO YUDEN UMK325BJ106KM-T 10 50 2 TAIYO YUDEN GMK316F106ZL-T 10 35 2.2 TAIYO YUDEN GMK325F106ZH-T 10 35 2.2 TAIYO YUDEN GMK325BJ106KN 10 35 2.5 TDK C3225X5R1E106M 10 25 Output capacitor selection Using tantalum or electrolytic capacitors, the selection is made referring to ESR and voltage rating rather than by a specific capacitance value. The output capacitor has to satisfy the output voltage ripple requirements. At a given switching frequency, small inductor values are useful to reduce the size of the choke but increase the inductor current ripple. Thus, to reduce the output voltage ripple a low ESR capacitor is required. To reduce jitter noise between different switching regulators in the system, it is preferable to work with an output voltage ripple greater than 25 mV. As far as it concerning the load transient requirements, the Equivalent Series Resistance (ESR) of the output capacitor must satisfy the following relationship: Equation 36 ESR ≤ VRIPPLE,MAX ΔIL,MAX where VRIPPLE is the maximum tolerable ripple voltage. In addition, the ESR must be high enough high to meet stability requirements. The output capacitor zero must be lower than the switching frequency: Equation 37 fSW > fZ = 42/53 1 2π ⋅ ESR ⋅ C out Doc ID 14436 Rev 2 PM6670AS Application information If ceramic capacitors are used, the output voltage ripple due to inductor current ripple is negligible. Then the inductance could be smaller, reducing the size of the choke. In this case it is important that output capacitor can adsorb the inductor energy without generating an over-voltage condition when the system changes from a full load to a no load condition. The minimum output capacitance can be chosen by the following equation: Equation 38 C OUT ,min = L ⋅ ILOAD ,MAX 2 Vf 2 − Vi 2 where Vf is the output capacitor voltage after the load transient, while VI is the output capacitor voltage before the load transient. In Table 15 are listed some tested polymer capacitors are listed. Table 15. Evaluated output capacitors Series Capacitance (μF) Rated voltage (V) ESR max @100 kHz (mΩ) 4TPE220MF 220 4V 15 to 25 4TPE150MI 220 4V 18 4TPC220M 220 4V 40 TNCB OE227MTRYF 220 2.5 V 25 Manufacturer SANYO HITACHI 8.1.4 MOSFETs selection In most of the applications, power management efficiency is a high level requirement. Power dissipation on the power switches becomes an important factor in the selection of switches. Losses of high-side and low-side MOSFETs depend on their working condition. Considering the high-side MOSFET, the power dissipation is calculated as: Equation 39 PDHighSide = Pconduction + Pswitching Maximum conduction losses are approximately given by: Equation 40 Pconduction = RDSon ⋅ VOUT 2 ⋅ ILOAD,MAX VIN. min Doc ID 14436 Rev 2 43/53 Application information PM6670AS where RDSon is the drain-source on-resistance of the control MOSFET. Switching losses are approximately given by: Equation 41 Pswitching = VIN ⋅ (ILOAD (max) − 2 ΔIL ΔI ) ⋅ t on ⋅ fsw VIN ⋅ (ILOAD (max) + L ) ⋅ t off ⋅ fsw 2 2 + 2 where tON and tOFF are the turn-on and turn-off times of the MOSFET and depend on the gate-driver current capability and the gate charge Qgate. A greater efficiency is achieved with low RDSon. Unfortunately low RDSon MOSFETs have a great gate charge. As general rule, the RDSon x Qgate product should be minimized to find out the suitable MOSFET. Logic-level MOSFETs are recommended, as long as low-side and high-side gate drivers are powered by VVCC = +5 V. The breakdown voltage of the MOSFETs (VBRDSS) must be greater than the maximum input voltage.VINmax. Table 16 lists tested high-side MOSFETs. Table 16. Evaluated high-side MOSFETs Manufacturer Type RDSon (mΩ) Gate charge (nC) Rated reverse voltage (V) ST STS12NH3LL 10.5 12 30 IR IRF7811 9 18 30 In buck converters the power dissipation of the synchronous MOSFET is mainly due to conduction losses: Equation 42 PDLowSide ≅ Pconduction Maximum conduction losses occur at the maximum input voltage: Equation 43 ⎛ V Pconduction = RDSon ⋅ ⎜1 − OUT ⎜ V IN,MAX ⎝ ⎞ ⎟ ⋅ ILOAD,MAX 2 ⎟ ⎠ The synchronous rectifier should have the lowest RDSon as possible. When the high-side MOSFET turns on, high dV/dt of the phase node can bring up even the low-side gate through its gate-drain capacitance CRRS, causing a cross-conduction problem. Once again, the choice of the low-side MOSFET is a trade-off between on resistance and gate charge; a good selection should minimize the ratio CRSS / CGS where Equation 44 CGS = CISS − CRSS 44/53 Doc ID 14436 Rev 2 PM6670AS Application information Tested low-side MOSFETs are listed in Table 17. Table 17. Evaluated low-side MOSFETs Manufacturer Type RDSon (mΩ) CGD \ CGS Rated reverse voltage (V) ST STS12NH3LL 13.5 0.069 30 ST STS25NH3LL 4.0 0.011 30 IR IRF7811 24 0.054 30 Dual N-MOS can be used in applications with lower output current. Table 18 shows some suitable dual MOSFETs for applications requiring about 3 A. Table 18. 8.1.5 Suitable dual MOSFETs Manufacturer Type RDSon (mΩ) Gate charge (nC) Rated reverse voltage (V) ST STS8DNH3LL 25 10 30 IR IRF7313 46 33 30 Diode selection A rectifier across the synchronous switch is recommended. The rectifier works as a voltage clamp across the synchronous rectifier and reduces the negative inductor swing during the dead time between turning the high-side MOSFET off and the synchronous rectifier on. Moreover it increases the efficiency of the system. Choose a schottky diode as long as its forward voltage drop is very little (0.3 V). The reverse voltage should be greater than the maximum input voltage VINmax and a minimum recovery reverse charge is preferable. Table 19 shows some evaluated diodes. Table 19. Evaluated recirculation rectifiers Manufacturer Type Forward voltage (V) Rated reverse voltage (V) Reverse current (μA) ST STPS1L30M 0.34 30 0.00039 ST STPS1L30A 0.34 30 0.00039 Doc ID 14436 Rev 2 45/53 Application information 8.1.6 PM6670AS VDDQ current limit setting The valley current limit is set by RCSNS and must be chosen to support the maximum load current. The valley of the inductor current ILvalley is: Equation 45 ILvalley = ILOAD (max) − ΔIL 2 The output current limit depends on the current ripple as shown I Figure 39: Figure 39. Valley current limit waveforms Inductor current Current Inductor current MAX LOAD 2 MAX LOAD 1 Valley current limit Time As the valley threshold is fixed, the greater the current ripple, the greater the DC output current will be. If an output current limit greater than ILOAD(max) over all the input voltage range is required, the minimum current ripple must be considered in the previous formula. Then the resistor RCSNS is: Equation 46 RCSNS = RDSon ⋅ ILvalley 100uA where RDSon is the drain-source on-resistance of the low-side switch. Consider the temperature effect and the worst case value in RDSon calculation (typically +0.4%/°C). The accuracy of the valley current also depends on the offset of the internal comparator (±5 mV). The negative valley-current limit (if the device works in forced-PWM mode) is given by: Equation 47 INEG = 46/53 120mV RDSon Doc ID 14436 Rev 2 PM6670AS 8.1.7 Application information All ceramic capacitors application Design of external feedback network depends on the output voltage ripple across the output capacitors' ESR. If the ripple is great enough (at least 20 mV), the compensation network simply consists of a CINT capacitor. Figure 40. Integrative compensation Ton One-shot generator VSNS VDDQ + PWM Comparator - VREF + COMP gm Integrator CFILT RINT - Vr=0.9 CINT The stability of the system depends firstly on the output capacitor zero frequency. It must be verified that: Equation 48 fSW > k ⋅ fZout = k 2π ⋅ R out C out where k is a free design parameter greater than unity (k > 3). It determines the minimum integrator capacitor value CINT: Equation 49 CINT > gm Vref ⎛f ⎞ Vo 2π ⋅ ⎜ SW − fZout ⎟ k ⎝ ⎠ Doc ID 14436 Rev 2 ⋅ 47/53 Application information PM6670AS If the ripple on pin COMP is greater than the integrator output dynamic (150 mV), an additional capacitor Cfilt could be added in order to reduce its amplitude. If q is the desired attenuation factor of the output ripple, select: Equation 50 C filt = CINT ⋅ (1 − q) q In order to reduce noise on pin COMP, it's possible to introduce a resistor RINT that, together with CINT and Cfilt, becomes a low psas filter. The cutoff frequency fCUT must be much greater (10 or more times) than the switching frequency of the section: Equation 51 RINT = 2π ⋅ fCUT 1 CINT ⋅ CFILT CINT + CFILT For most of applications both RINT and Cfilt are unnecessary. If the ripple is very small (e.g. such as with ceramic capacitors), an additional compensation network, called “Virtual ESR” network, is needed. This additional part generates a triangular ripple that substitutes the ESR output voltage ripple. The complete compensation scheme is represented in Figure 41. Figure 41. Virtual ESR network L R R1 CINT VDDQ C RINT PWM Comparator Ton Generation Block 48/53 + gm + - VREF CFILT Integrator Doc ID 14436 Rev 2 0.9V PM6670AS Application information Select C as shown: Equation 52 C > 5 ⋅ CINT Then calculate R in order to have enough ripple voltage on the integrator input: Equation 53 R= L R VESR ⋅ C Where RVESR is the new virtual output capacitor ESR. A good trade-off is to consider an equivalent ESR of 30-50 mΩ, even though the choice depends on inductor current ripple. Then choose R1 as follows: Equation 54 ⎛ 1 ⎞ ⎟ R ⋅ ⎜⎜ CπfZ ⎟⎠ ⎝ R1 = 1 R− CπfZ Doc ID 14436 Rev 2 49/53 Package mechanical data 9 PM6670AS Package mechanical data In order to meet environmental requirements, ST offers these devices in different grades of ECOPACK® packages, depending on their level of environmental compliance. ECOPACK® specifications, grade definitions and product status are available at: www.st.com. ECOPACK® is an ST trademark. Table 20. VFQFPN-24 4mm x 4mm mechanical data mm. Dim. Min Typ Max 0.80 0.90 1.00 A1 0.0 0.05 A2 0.65 0.80 D 4.00 D1 3.75 E 4.00 E1 3.75 A θ P 50/53 12° 0.24 0.42 e 0.50 N 24.00 Nd 6.00 Ne 6.00 L 0.30 b 0.18 0.30 D2 2.40 2.70 E2 2.40 2.70 Doc ID 14436 Rev 2 0.40 0.60 0.50 PM6670AS Package mechanical data Figure 42. Package dimensions Doc ID 14436 Rev 2 51/53 Revision history 10 PM6670AS Revision history Table 21. 52/53 Document revision history Date Revision Changes 14-Feb-2008 1 Initial release. 17-Feb-2010 2 Updated: Coverpage, Table 2, Table 6, Section 7.1, Figure 28 and Table 8 Doc ID 14436 Rev 2 PM6670AS Please Read Carefully: Information in this document is provided solely in connection with ST products. STMicroelectronics NV and its subsidiaries (“ST”) reserve the right to make changes, corrections, modifications or improvements, to this document, and the products and services described herein at any time, without notice. All ST products are sold pursuant to ST’s terms and conditions of sale. Purchasers are solely responsible for the choice, selection and use of the ST products and services described herein, and ST assumes no liability whatsoever relating to the choice, selection or use of the ST products and services described herein. No license, express or implied, by estoppel or otherwise, to any intellectual property rights is granted under this document. If any part of this document refers to any third party products or services it shall not be deemed a license grant by ST for the use of such third party products or services, or any intellectual property contained therein or considered as a warranty covering the use in any manner whatsoever of such third party products or services or any intellectual property contained therein. UNLESS OTHERWISE SET FORTH IN ST’S TERMS AND CONDITIONS OF SALE ST DISCLAIMS ANY EXPRESS OR IMPLIED WARRANTY WITH RESPECT TO THE USE AND/OR SALE OF ST PRODUCTS INCLUDING WITHOUT LIMITATION IMPLIED WARRANTIES OF MERCHANTABILITY, FITNESS FOR A PARTICULAR PURPOSE (AND THEIR EQUIVALENTS UNDER THE LAWS OF ANY JURISDICTION), OR INFRINGEMENT OF ANY PATENT, COPYRIGHT OR OTHER INTELLECTUAL PROPERTY RIGHT. UNLESS EXPRESSLY APPROVED IN WRITING BY AN AUTHORIZED ST REPRESENTATIVE, ST PRODUCTS ARE NOT RECOMMENDED, AUTHORIZED OR WARRANTED FOR USE IN MILITARY, AIR CRAFT, SPACE, LIFE SAVING, OR LIFE SUSTAINING APPLICATIONS, NOR IN PRODUCTS OR SYSTEMS WHERE FAILURE OR MALFUNCTION MAY RESULT IN PERSONAL INJURY, DEATH, OR SEVERE PROPERTY OR ENVIRONMENTAL DAMAGE. ST PRODUCTS WHICH ARE NOT SPECIFIED AS "AUTOMOTIVE GRADE" MAY ONLY BE USED IN AUTOMOTIVE APPLICATIONS AT USER’S OWN RISK. Resale of ST products with provisions different from the statements and/or technical features set forth in this document shall immediately void any warranty granted by ST for the ST product or service described herein and shall not create or extend in any manner whatsoever, any liability of ST. ST and the ST logo are trademarks or registered trademarks of ST in various countries. Information in this document supersedes and replaces all information previously supplied. The ST logo is a registered trademark of STMicroelectronics. All other names are the property of their respective owners. © 2010 STMicroelectronics - All rights reserved STMicroelectronics group of companies Australia - Belgium - Brazil - Canada - China - Czech Republic - Finland - France - Germany - Hong Kong - India - Israel - Italy - Japan Malaysia - Malta - Morocco - Philippines - Singapore - Spain - Sweden - Switzerland - United Kingdom - United States of America www.st.com Doc ID 14436 Rev 2 53/53