MIC28510
75V/4A Hyper Speed Control
Synchronous DC/DC Buck Regulator
SuperSwitcher II
General Description
Features
The Micrel MIC28510 is an adjustable–frequency,
synchronous buck regulator featuring unique adaptive on–
time control architecture. The MIC28510 operates over an
input supply range of 4.5V to 75V and provides a regulated
output of up to 4A of output current. The output voltage is
adjustable down to 0.8V with a guaranteed accuracy of ±1%.
Micrel’s Hyper Speed Control architecture allows for ultra–
fast transient response while reducing the output capacitance
and also makes (High VIN) / (Low VOUT) operation possible.
This adaptive tON ripple control architecture combines the
advantages of fixed–frequency operation and fast transient
response in a single device.
The MIC28510 offers a full suite of protection features to
ensure protection of the IC during fault conditions. These
include undervoltage lockout to ensure proper operation
under power–sag conditions, internal soft–start to reduce
inrush current, foldback current limit, “hiccup” mode shortcircuit protection, and thermal shutdown.
All support documentation can be found on Micrel’s web
site at: www.micrel.com.
Hyper Speed Control architecture enables:
High Delta V operation (VIN = 75V and VOUT = 0.8V)
Small output capacitance
4.5V to 75V voltage input
4A output current capability, up to 95% efficiency
Adjustable output voltage form 0.8V to 24V
±1% FB accuracy
Any Capacitor stable:
Zero-ESR to high–ESR output capacitors
100kHz to 500kHz switching frequency
Internal compensation
Foldback current–limit and “hiccup” mode short-circuit
protection
Thermal shutdown
Supports safe startup into a pre–biased load
–40C to +125C junction temperature range
28-pin 5mm 6mm MLF® package
Applications
Distributed power systems
Communications/networking infrastructure
Industrial power
Solar energy
___________________________________________________________________________________________________________
Typical Application
Efficiency (VIN = 48V)
vs. Output Current
100
90
5.0V
3.3V
2.5V
1.8V
1.5V
1.2V
1.0V
0.8V
EFFICIENCY (%)
80
70
60
50
40
fSW = 250kHz
30
20
10
0
1
2
3
4
5
6
OUTPUT CURRENT (A)
Hyper Speed Control, SuperSwitcher II and Any Capacitor are trademarks of Micrel, Inc.
MLF and MicroLeadFrame are registered trademarks of Amkor Technology, Inc.
Micrel Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel +1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com
March 2012
M9999-030912-A
Micrel, Inc.
MIC28510
Ordering Information
Part Number
Junction Temperature Range
Package
Lead Finish
MIC28510YJL
40C to 125C
28-pin 5mm 6mm MLF®
Pb-Free
Pin Configuration
28-Pin 5mm 6mm MLF® (JL)
Pin Description
Pin Number
Pin Name
Pin Function
1, 3
NC
No Connect.
Power Ground. PGND is the ground path for the MIC28510 buck converter power stage. The
PGND pin connects to the sources of low-side N-Channel internal MOSFETs, the negative
terminals of input capacitors, and the negative terminals of output capacitors. The loop for the
power ground should be as small as possible and separate from the signal ground (SGND) loop.
2, 5, 6,
7, 8, 21
PGND
4, 9, 10,
11, 12
SW
13, 14, 15, 16,
17, 18, 19
PVIN
High-Side Internal N-Channel MOSFET Drain Connection (Input).The PVIN operating voltage
range is from 4.5V to 75V. Input capacitors between the PVIN pins and the power ground (PGND)
are required and keep the connection short.
20
BST
Boost (Output). Bootstrapped voltage to the high-side N-channel internal MOSFET driver. A
Schottky diode is connected between the VDD pin and the BST pin. A boost capacitor of 0.1μF is
connected between the BST pin and the SW pin.
CS
Current Sense (Input). High current output driver return. The CS pin connects directly to the switch
node. Due to the high-speed switching on this pin, the CS pin should be routed away from
sensitive nodes. CS pin also senses the current by monitoring the voltage across the low-side
internal MOSFET during OFF-time.
22
March 2012
Switch Node (Output). Internal connection for the high-side MOSFET source and low-side
MOSFET drain.
2
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Micrel, Inc.
MIC28510
Pin Description (Continued)
Pin Number
Pin Name
23
FS
Frequency Setting Pin.
24
EN
Enable (Input). A logic level control of the output. The EN pin is CMOS-compatible. Logic high =
enable, logic low = shutdown. In the off state, the VDD supply current of the device is reduced
(typically 0.7mA). Do not pull the EN pin above the VDD supply. This pin has 100k pull-up resistor to
VDD.
25
FB
Feedback (Input). Input to the transconductance amplifier of the control loop. The FB pin is
regulated to 0.8V. A resistor divider connecting the feedback to the output is used to adjust the
desired output voltage.
26
SGND
Signal Ground. SGND must be connected directly to the ground planes. Do not route the SGND pin
to the PGND Pad on the top layer; see PCB layout guidelines for details.
27
VDD
VDD Bias (Input). Power to the internal reference and control sections of the MIC28510. The VDD
operating voltage range is from 4.5V to 5.5V. A 2.2µF ceramic capacitor from the VDD pin to the
PGND pin must be placed next to the IC. VDD must be powered up at the same time or after PVIN
to make the soft-start function correctly.
28
PVDD
March 2012
Pin Function
Power Supply for Gate Driver of Bottom MOSFET.
3
M9999-030912-A
Micrel, Inc.
MIC28510
Absolute Maximum Ratings(1)
Operating Ratings(3)
PVIN to PGND................................................ 0.3V to +76V
FS to PGND ....................................................0.3V to PVIN
PVDD, VDD to PGND ......................................... 0.3V to +6V
VSW, VCS to PGND .............................. 0.3V to (PVIN +0.3V)
VBST to VSW ........................................................ 0.3V to 6V
VBST to PGND .................................................. 0.3V to 82V
VEN to PGND ...................................... 0.3V to (VDD + 0.3V)
VFB to PGND....................................... 0.3V to (VDD + 0.3V)
PGND to SGND ........................................... 0.3V to +0.3V
Junction Temperature (TJ) ....................................... +150°C
Storage Temperature (TS).........................65C to +150C
Lead Temperature (soldering, 10s)............................ 260°C
ESD Rating(2) ..............................................................1000V
Supply Voltage (PVIN) ....................................... 4.5V to 75V
Bias Voltage (PVDD, VDD).................................. 4.5V to 5.5V
Enable Input (VEN) ................................................. 0V to VDD
Junction Temperature (TJ) ........................ 40C to +125C
Maximum Power Dissipation......................................Note 4
Package Thermal Resistance(4)
5mm x 6mm MLF® (JA) ....................................36C/W
Electrical Characteristics(5)
PVIN = VFS = 48V, VDD = 5V; VBST – VSW = 5V; TA = 25°C, unless noted. Bold values indicate 40°C ≤ TJ ≤ +125°C.
Parameter
Condition
Min.
Typ.
Max.
Units
4.5
75
V
2
75
V
Power Supply Input
Input Voltage Range ( PVIN)
FS Voltage Range
VDD Bias Voltage
Operating Bias Voltage (VDD)
Undervoltage Lockout Trip Level
VDD Rising
4.5
5
5.5
V
3.2
3.85
4.45
V
UVLO Hysteresis
Quiescent Supply Current (IVDD)
Shutdown Supply Current (IVDD)
380
VFB = 1.5V
VDD = VBST = 5.5V, VIN = 48V
SW = unconnected, VEN = 0V
mV
1.4
3
mA
0.7
2
mA
Reference
Feedback Reference Voltage
0°C ≤ TJ ≤ 85°C (±1.0%)
0.792
0.8
0.808
40°C ≤ TJ ≤ 125°C (±1.5%)
0.788
0.8
0.812
V
Load Regulation
IOUT = 0A to 4A
0.04
%
Line Regulation
PVIN = 4.5 to 75V
0.1
%
FB Bias Current
VFB = 0.8V
-0.5
EN Logic Level High
4.5V < VDD < 5.5V
1.2
EN Logic Level Low
4.5V < VDD < 5.5V
EN Bias Current
VEN = 0V
0.005
0.5
µA
0.4
V
100
µA
Enable Control
V
50
Notes:
1.
Exceeding the absolute maximum rating may damage the device.
2.
Devices are ESD sensitive. Handling precautions recommended. Human body model, 1.5kΩ in series with 100pF.
3.
The device is not guaranteed to function outside operating range.
4.
PD(MAX) = (TJ(MAX) – TA)/ JA, where JA depends upon the printed circuit layout. See “Applications Information.”
5.
Specification for packaged product only.
March 2012
4
M9999-030912-A
Micrel, Inc.
MIC28510
Electrical Characteristics(5) (Continued)
PVIN = VFS = 48V, VDD = 5V; VBST – VSW = 5V; TA = 25°C, unless noted. Bold values indicate 40°C ≤ TJ ≤ +125°C.
Parameter
Condition
Min.
Typ.
Max.
Units
VFS = PVIN
375
500
625
kHz
Oscillator
Switching Frequency
Maximum Duty Cycle
(6)
Minimum Duty Cycle
VFB = 0V, VFS=PVIN
80
%
VFB > 0.8V
0
%
360
ns
6
ms
Minimum Off-time
Soft-Start
Soft-Start time
Short-Circuit Protection
Current–Limit Threshold
Short–Circuit Current
VFB = 0.8V, TJ = 25°C
4.8
VFB = 0.8V, TJ = 125°C
4
VFB = 0V
2
7
10
10
4.3
5.7
A
A
Internal FETs
Top–MOSFET RDS (ON)
ISW = 1A
31
mΩ
Bottom–MOSFET RDS (ON)
ISW = 1A
31
mΩ
SW Leakage Current
PVIN = 36V, VSW = 36V, VEN = 0V, VBST = 41V
55
µA
PVIN Leakage Current
PVIN = 36V, VSW = 0V, VEN = 0V, VBST = 41V
55
µA
Thermal Protection
Over–Temperature Shutdown
TJ Rising
Over–Temperature Shutdown
Hysteresis
160
°C
2
°C
Note:
6.
The maximum duty–cycle is limited by the fixed mandatory off-time (tOFF ) of typically 360ns.
March 2012
5
M9999-030912-A
Micrel, Inc.
MIC28510
Typical Characteristics
400
SHUTDOWN CURRENT (uA)
SUPPLY CURRENT (mA)
20
VOUT = 3.3V
IOUT = 0A
16
VDD = 5V
fSW = 250kHz
12
8
4
0
5
15
25
35
45
55
65
VIN Shutdown Current
vs. Input Voltage
300
200
V DD = 5V
V EN = 0V
100
0
75
VDD Operating Supply Current
vs. Input Voltage
10
VDD SUPPLY CURRENT (mA)
VIN Operating Supply Current
vs. Input Voltage
8
6
VOUT = 3.3V
VDD = 5V
4
fSW = 250kHz
2
0
5
15
25
35
45
55
65
75
5
15
25
35
45
55
65
75
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
Feedback Voltage
vs. Input Voltage
Total Regulation
vs. Input Voltage
Output Peak Current Limit
vs. Input Voltage
1.0%
TOTAL REGULATION (%)
VOUT = 3.3V
VDD = 5V
0.804
15
0.8%
IOUT = 0A
fSW = 250kHz
0.800
0.796
VOUT = 3.3V
VOUT = 3.3V
0.6%
VDD = 5V
0.4%
IOUT = 0A to 4A
CURRENT LIMIT (A)
0.808
FEEDBACK VOLTAGE (V)
INPUT VOLTAGE (V)
fSW = 250kHz
0.2%
0.0%
-0.2%
-0.4%
-0.6%
VDD = 5V
12
fSW = 250kHz
9
6
3
-0.8%
0.792
-1.0%
5
15
25
35
45
55
65
75
0
5
15
INPUT VOLTAGE (V)
35
45
55
65
5
75
VOUT = 3.3V
VDD = 5V
IOUT = 1A
R18 =100k
R19 =100k
10
8
VIN = 48V
6
VOUT = 3.3V
VDD = 5V
4
IOUT = 0A
fSW = 250kHz
2
0
100
5
15
25
35
45
55
65
75
45
55
65
75
1.0
SHUTDOWN CURRENT (mA)
SUPPLY CURRENT (mA)
250
35
VDD Shutdown Current
vs. Temperature
12
150
25
INPUT VOLTAGE (V)
VDD Operating Supply Current
vs. Temperature
300
200
15
INPUT VOLTAGE (V)
Switching Frequency
vs. Input Voltage
SWITCHING FREQUENCY (kHz)
25
0.8
0.6
0.4
VIN = 48V
0.2
VEN = 0V
IOUT = 0A
VDD = 5V
0.0
-50
-25
0
25
50
75
TEMPERATURE (°C)
100
125
-50
-25
0
25
50
75
100
125
TEMPERATURE (°C)
INPUT VOLTAGE (V)
March 2012
6
M9999-030912-A
Micrel, Inc.
MIC28510
Typical Characteristics (Continued)
20
VIN = 48V
IOUT = 0A
4.0
3.9
3.8
RISING
3.7
3.6
3.5
FALLING
3.4
400
VIN = 48V
SUPPLY CURRENT (mA)
4.1
VIN Shutdown Current
vs. Temperature
SHUTDOWN CURRENT (uA)
4.2
VDD THRESHOLD (V)
VIN Operating Supply Current
vs. Temperature
VDD UVLO Threshold
vs. Temperature
VOUT = 3.3V
16
VDD = 5V
IOUT = 0A
fSW = 250kHz
12
8
4
320
240
VIN = 48V
VEN = 0V
-25
0
25
50
75
100
125
0
0
-50
-25
TEMPERATURE (°C)
0
50
75
100
-50
125
0
V DD = 5V
fSW = 250kHz
9
6
VOUT = 3.3V
0.804
50
75
100
125
100
125
0.4%
VIN = 48V
LOAD REGULATION (%)
FEEBACK VOLTAGE (V)
V OUT = 3.3V
25
Load Regulation
vs. Temperature
0.808
V IN = 48V
VDD = 5V
IOUT = 0A
0.800
0.796
VIN = 48V
0.3%
VOUT = 3.3V
VDD = 5V
0.2%
IOUT = 0A to 4A
fSW = 250kHz
0.1%
0.0%
-0.1%
-0.2%
3
0.792
-50
0
-50
-25
0
25
50
75
100
-25
0
25
50
75
100
-0.3%
125
-50
Line Regulation
vs. Temperature
0.4%
IOUT = 0A
0.1%
0.0%
-0.1%
-0.2%
100
VIN = 48V
250
25°C
125°C
200
VIN = 48V
VOUT = 3.3V
150
VDD = 5V
R18 = 100k
R19 =100k
100
-0.3%
0
25
50
75
TEMPERATURE (°C)
March 2012
100
75
EN Bias Current
vs. Temperature
EN BIAS CURRENT (µA)
SWITCHING FREQUENCY
(kHz)
V DD = 5V
-25
50
Switching Frequency
vs. Output Current
-40°C
V OUT = 3.3V
-50
25
TEMPERATURE (°C)
300
0.2%
0
TEMPERATURE (°C)
V IN = 5V to 75V
0.3%
-25
125
TEMPERATURE (°C)
LINE REGULATION (%)
-25
TEMPERATURE (°C)
Feedback Voltage
vs. Temperature
15
CURRENT LIMIT (A)
25
TEMPERATURE (°C)
Output Peak Current Limit
vs. Temperature
12
IOUT = 0A
80
3.3
-50
VDD = 5V
160
125
VDD = 5V
80
VEN = 0V
60
40
20
0
0
1
2
3
OUTPUT CURRENT (A)
7
4
-50
-25
0
25
50
75
100
125
TEMPERATURE (°C)
M9999-030912-A
Micrel, Inc.
MIC28510
Typical Characteristics (Continued)
Enable Threshold
vs. Temperature
100
EFFICIENCY (%)
RISING
0.7
FALLING
0.6
85
80
75
65
VIN = 48V
60
VDD = 5V
55
0.5
-50
-25
0
25
50
75
100
48VIN
70
75VIN
0
1
100
0.2%
90
EFFICIENCY (%)
LINE REGULATION (%)
0.804
fSW = 250kHz
0.800
0.796
2
3
0.792
4
0
1
0.1%
VIN = 5V to 75V
VOUT = 3.3V
VDD = 5V
2
3
4
OUTPUT CURRENT (A)
Efficiency (VIN = 5V)
vs. Output Current
0.3%
Efficiency (VIN = 48V)
vs. Output Current
100
90
3.3V
2.5V
1.8V
1.5V
1.2V
1.0V
0.9V
0.8V
80
70
fsw = 250kHz
60
5.0V
3.3V
2.5V
1.8V
1.5V
1.2V
1.0V
0.9V
0.8V
80
50
fSW = 250kHz
-0.2%
VOUT = 3.3V
VDD = 5V
OUTPUT CURRENT (A)
Line Regulation
vs. Output Current
-0.1%
VIN = 48V
fSW = 250kHz
50
125
TEMPERATURE (°C)
0.0%
VOUT = 3.3V
VDD = 5V
EFFICIENCY (%)
ENABLE THRESHOLD (V)
6VIN
90
0.9
0.808
4.5VIN
95
FEEDBACK VOLTAGE (V)
1.0
0.8
Feedback Voltage
vs. Output Current
Efficiency
vs. Output Current
70
60
50
40
fSW = 250kHz
30
40
20
-0.3%
30
1
2
3
4
0
OUTPUT CURRENT (A)
EFFICIENCY (%)
DIE TEMPERATURE (°C)
90
5.0V
3.3V
2.5V
1.8V
1.5V
1.2V
1.0V
0.9V
60
50
40
30
0.8V
fSW = 250kHz
3
4
5
3
4
OUTPUT CURRENT (A)
March 2012
2
3
5
6
4
5
6
Die Temperature* (VIN = 48V)
vs. Output Current
60
40
VIN = 5.0V
VOUT = 3.3V
20
VDD = 5V
60
40
`
V IN = 48V
V OUT = 3.3V
20
V DD = 5V
fSW = 250kHz
10
2
1
OUTPUT CURRENT (A)
fSW = 250kHz
0
1
0
80
20
0
10
6
Die Temperature* (VIN = 5.0V)
vs. Output Current
80
100
70
2
OUTPUT CURRENT (A)
Efficiency (VIN = 75V)
vs. Output Current
80
1
DIE TEMPERATURE (°C)
0
0
0
1
2
3
OUTPUT CURRENT (A)
8
4
0
1
2
3
4
OUTPUT CURRENT (A)
M9999-030912-A
Micrel, Inc.
MIC28510
Typical Characteristics (Continued)
Efficiency (VIN =12V)
vs. Output Current
Die Temperature* (VIN = 75V)
vs. Output Current
100
100
5.0V
3.3V
2.5V
1.8V
1.2V
0.8V
90
EFFICIENCY (%)
80
60
40
V IN = 75V
80
70
60
50
fSW = 250kHz
V OUT = 3.3V
20
V DD = 5V
5.0V
3.3V
2.5V
1.8V
1.2V
0.8V
90
EFFICIENCY (%)
100
DIE TEMPERATURE (°C)
Efficiency (VIN = 18V)
vs. Output Current
40
80
70
60
50
fSW = 250kHz
40
fSW = 250kHz
30
0
0
1
2
3
0
4
1
2
3
4
5
30
6
0
1
OUTPUT CURRENT (A)
2
3
4
5
6
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
100
4
4
VOUT = 5V
5.0V
3.3V
2.5V
1.8V
1.2V
0.8V
80
70
60
50
V OUT = 0.8V
LOAD CURRENT (A)
LOAD CURRENT (A)
90
EFFICIENCY (%)
Thermal Derating
Thermal Derating
Efficiency (VIN = 24V)
vs. Output Current
VIN = 48V
3
fSW = 250kHz
L = 10µH
Tj_MAX =125°C
VOUT = 3.3V
2
VOUT =2.5V
1
3
V OUT = 1.2V
2
VIN = 48V
fSW = 250kHz
L = 10µH
Tj_MAX =125°C
1
fSW = 250kHz
40
0
0
30
25
0
1
2
3
4
5
6
40
55
70
85
25
100
MAXIMUM AMBIENT TEMPERATURE
(°C)
OUTPUT CURRENT (A)
Thermal Derating
40
70
85
100
Thermal Derating
Thermal Derating
4
55
MAXIMUM AMBIENT TEMPERATURE
(°C)
4
4
VOUT =3.3V
2
V OUT = 5V
V IN = 12V
fSW = 250kHz
L = 10µH
Tj_MAX = 125°C
1
VOUT = 2.5V
3
LOAD CURRENT (A)
3
LOAD CURRENT (A)
LOAD CURRENT (A)
V OUT = 2.5V
VOUT =3.3V
2
VOUT = 5V
VIN = 18V
1
fSW = 250kHz
L = 10µH
Tj_MAX = 125°C
VOUT = 2.5V
25
40
55
70
85
MAXIMUM AMBIENT TEMPERATURE
(°C)
March 2012
100
3
2
V OUT = 5V
V IN = 24V
fSW = 250kHz
L = 10µH
Tj_MAX =125°C
1
0
0
0
V OUT =3.3V
25
40
55
70
85
MAXIMUM AMBIENT TEMPERATURE
(°C)
9
100
25
40
55
70
85
100
MAXIMUM AMBIENT TEMPERATURE
(°C)
M9999-030912-A
Micrel, Inc.
MIC28510
Typical Characteristics (Continued)
Thermal Derating
Thermal Derating
VOUT = 1.8V
2
VOUT = 0.8V
V IN = 12V
fSW = 250kHz
L = 10µH
Tj_MAX = 125°C
1
LOAD CURRENT (A)
V OUT = 1.2V
3
LOAD CURRENT (A)
LOAD CURRENT (A)
Thermal Derating
4
4
4
VOUT =1.8V
3
VOUT = 1.2V
2
V IN = 18V
1
fSW = 250kHz
L = 10µH
Tj_MAX = 125°C
VOUT = 0.8V
25
40
55
70
85
100
25
40
55
70
85
MAXIMUM AMBIENT TEMPERATURE
(°C)
MAXIMUM AMBIENT TEMPERATURE
(°C)
VOUT = 1.8V
2
VOUT = 1.2V
V IN = 24V
1
fSW = 250kHz
L = 10µH
Tj_MAX = 125°C
V OUT = 0.8V
0
0
0
3
100
25
40
55
70
85
100
MAXIMUM AMBIENT TEMPERATURE
(°C)
Thermal Derating
LOAD CURRENT (A)
4
3
24VIN
2
48VIN
VOUT = 12V
1
fSW = 250kHz
L = 27µH
Tj_MAX = 125°C
0
25
40
55
70
85
100
MAXIMUM AMBIENT TEMPERATURE
(°C)
Die Temperature*: The temperature measurement was taken at the hottest point on the MIC28510 case mounted on a five–square inch, four–layer,
0.62”, FR–4 PCB with 2 oz. finish copper weight–pre–layer (see Thermal Measurement section). Actual results will depend upon the size of the PCB,
ambient temperature and proximity to other heat–emitting components.
March 2012
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Micrel, Inc.
MIC28510
Functional Characteristics
March 2012
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Micrel, Inc.
MIC28510
Functional Characteristics (Continued)
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MIC28510
Functional Characteristics (Continued)
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MIC28510
Functional Diagram
Figure 1. MIC28510 Block Diagram
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Micrel, Inc.
MIC28510
The maximum duty cycle is obtained from the 360ns
tOFF(MIN):
Functional Description
The MIC28510 is an adaptive ON-time synchronous
step-down DC/DC regulator. It is designed to operate
over a wide input voltage range from, 4.5V to 75V, and
provides a regulated output voltage at up to 4A of output
current. A digitally-modified adaptive ON-time control
scheme is employed in order to obtain a constantswitching frequency and to simplify the control
compensation. Over current protection is implemented
without the use of an external sense resistor. The device
includes an internal soft-start function which reduces the
power supply input surge current at start-up by
controlling the output voltage rise time.
D MAX
VOUT
VIN fSW
360ns
tS
Eq. 2
Eq. 1
where VOUT is the output voltage and VIN is the power
stage input voltage and fSW is the switching frequency.
At the end of the ON–time period, the internal high–side
driver turns off the high–side MOSFET and the low–side
driver turns on the low–side MOSFET. The OFF–time
period length depends upon the feedback voltage in
most cases. When the feedback voltage decreases and
the output of the gm amplifier is below 0.8V, the ON-time
period is triggered and the OFF–time period ends. If the
OFF–time period determined by the feedback voltage is
less than the minimum OFF–time tOFF(MIN), which is about
360ns, then the MIC28510 control logic will apply the
tOFF(MIN) instead. The minimum tOFF(MIN) period is required
to maintain enough energy in the boost capacitor (CBST)
to drive the high–side MOSFET.
March 2012
tS
1
where tS = 1/fSW. It is not recommended to use
MIC28510 with a OFF-time close to tOFF(MIN) during
steady-state operation.
The actual ON–time and resulting switching frequency
will vary with the part–to–part variation in the rise and fall
times of the internal MOSFETs, the output load current,
and variations in the VDD voltage. Also, the minimum tON
results in a lower switching frequency in high VIN to VOUT
applications, such as 75V to 1.0V.
Figure 2 shows the MIC28510 control loop timing during
steady-state operation. During steady-state, the gm
amplifier senses the feedback voltage ripple, which is
proportional to the output voltage ripple and the inductor
current ripple, to trigger the ON–time period. The ON–
time is predetermined by the tON estimator. The
termination of the OFF–time is controlled by the
feedback voltage. At the valley of the feedback voltage
ripple, which occurs when VFB falls below VREF, the OFF
period ends and the next ON–time period is triggered
through the control logic circuitry.
Theory of Operation
Figure 1 illustrates the block diagram for the control loop
of the MIC28510. The output voltage is sensed by the
MIC28510 feedback pin FB via the voltage divider R1
and R2, and compared to a 0.8V reference voltage VREF
at the error comparator through a low–gain
transconductance (gm) amplifier. If the feedback voltage
decreases and the output of the gm amplifier is below
0.8V, then the error comparator will trigger the control
logic and generate an ON–time period. The ON–time
period length is predetermined by the “FIXED tON
ESTIMATION” circuitry:
t ON(estimated)
t S t OFF(MIN)
Figure 2. MIC28510 Control Loop Timing
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Micrel, Inc.
MIC28510
Figure 3 shows the operation of the MIC28510 during
load transient. The output voltage drops due to the
sudden load increase, which causes the VFB to be less
than VREF. This will cause the error comparator to trigger
an ON–time period. At the end of the ON–time period, a
minimum OFF–time tOFF(min) is generated to charge CBST
since the feedback voltage is still below VREF. Then, the
next ON–time period is triggered due to the low
feedback voltage. Therefore, the switching frequency
changes during the load transient, but returns to the
nominal fixed frequency once the output has stabilized at
the new load current level. With the varying duty cycle
and switching frequency, the output recovery time is fast
and the output voltage deviation is small in MIC28510
converter.
Soft–Start
Soft–start reduces the power supply input surge current
at startup by controlling the output voltage rise time. The
input surge appears while the output capacitor is
charged up. A slower output rise time will draw a lower
input surge current.
The MIC28510 implements an internal digital soft–start
by making the 0.8V reference voltage VREF ramp from 0
to 100% in approximately 6ms with 9.7mV steps.
Therefore, the output voltage is controlled to increase
slowly with a stair–case VFB ramp. Once the soft–start
cycle ends, the related circuitry is disabled to reduce
current consumption. VDD must be powered up at the
same time or after VIN to allow the soft–start function
correctly.
Current Limit
The MIC28510 uses the RDS(ON) of the internal low–side
power MOSFET to sense over–current conditions. This
method will avoid adding cost, use of additional board
space and power losses taken by a discrete current
sense resistor.
In each switching cycle of the MIC28510 converter, the
inductor current is sensed by monitoring the low–side
MOSFET in the OFF period. If the peak inductor current
is greater than 7A, then the MIC28510 turns off the
high–side MOSFET and a soft–start sequence is
triggered. This mode of operation is called “hiccup
mode” and its purpose is to protect the downstream load
in case of a hard short. The current–limit threshold has a
foldback characteristic related to the feedback voltage,
as shown in Figure 4.
Figure 3. MIC28510 Load Transient Response
Unlike true current–mode control, the MIC28510 uses
the output voltage ripple to trigger an ON–time period.
The output voltage ripple is proportional to the inductor
current ripple if the ESR of the output capacitor is large
enough.
In order to meet the stability requirements, the
MIC28510 feedback voltage ripple should be in phase
with the inductor current ripple and large enough to be
sensed by the gm amplifier and the error comparator.
The recommended feedback voltage ripple is
20mV~100mV. If a low–ESR output capacitor is
selected, then the feedback voltage ripple may be too
small to be sensed by the gm amplifier and the error
comparator. Also, the output voltage ripple and the
feedback voltage ripple are not necessarily in phase with
the inductor current ripple if the ESR of the output
capacitor is very low. In these cases, ripple injection is
required to ensure proper operation. Please refer to
“Ripple Injection” subsection in Application Information of
this datatsheet for more details regarding the ripple
injection technique.
March 2012
Current-Limit Threshold
vs. Feedback Voltage
CURRENT LIMIT THRESHOLD (A)
12
VIN = 48V
10
8
6
4
2
0
0.0
0.2
0.4
0.6
0.8
1.0
FEEDBACK VOLTAGE (V)
Figure 4. MIC28510 Current–Limit Foldback Characteristic
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M9999-030912-A
Micrel, Inc.
MIC28510
Internal MOSFET Gate Drive
Figure 1 (the block diagram) shows a bootstrap circuit,
consisting of D1 (a Schottky diode is recommended) and
a capacitor connected from the SW pin to the BST pin
(CBST). This circuit supplies energy to the high–side drive
circuit. Capacitor CBST is charged, while the low–side
MOSFET is on, and the voltage on the SW pin is
approximately 0V. When the high–side MOSFET driver
is turned on, energy from CBST is used to turn the
MOSFET on. As the high–side MOSFET turns on, the
voltage on the SW pin increases to approximately VIN.
Diode D1 is reverse biased and CBST floats high while
continuing to keep the high–side MOSFET on. The bias
current of the high–side driver is less than 10mA so a
0.1μF to 1μF is sufficient to hold the gate voltage with
minimal droop for the power stroke (high–side switching)
cycle, i.e. ΔBST = 10mA x 4μs/0.1μF = 400mV. When
the low–side MOSFET is turned back on, CBST is
recharged through D1. A small resistor in series with
CBST, can be used to slow down the turn–on time of the
high–side N–channel MOSFET.
The drive voltage is derived from the PVDD supply
voltage. The nominal low–side gate drive voltage is PVDD
and the nominal high–side gate drive voltage is
approximately PVDD – VDIODE, where VDIODE is the voltage
drop across D1. An approximate 30ns delay between the
high–side and low–side driver transitions is used to
prevent current from simultaneously flowing unimpeded
through both MOSFETs.
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MIC28510
Application Information
Inductor Selection
Values for inductance, peak, and RMS currents are
required to select the output inductor. The input and
output voltages and the inductance value determine the
peak–to–peak inductor ripple current. Generally, higher
inductance values are used with higher input voltages.
Larger peak–to–peak ripple currents will increase the
power dissipation in the inductor and MOSFETs. Larger
output ripple currents will also require more output
capacitance to smooth out the larger ripple current.
Smaller peak–to–peak ripple currents require a larger
inductance value and therefore a larger and more
expensive inductor. A good compromise between size,
loss and cost is to set the inductor ripple current to be
equal to 20% of the maximum output current. The
inductance value is calculated by Equation 4:
Setting the Switching Frequency
The MIC28510 is an adjustable–frequency, synchronous
buck regulator featuring a unique digitally-modified,
adaptive on–time control architecture. The switching
frequency can be adjusted between 100kHz and 500kHz
by changing the resistor divider network consisting of
R18 and R19.
L
The following formula gives the estimated switching
frequency:
fSW _ ADJ
VIN(max) fsw 20% IOUT(max)
Eq. 4
where:
fSW = Switching frequency
20% = Ratio of AC ripple current to DC output current
VIN(MAX) = Maximum power stage input voltage
Figure 5. Switching Frequency Adjustment
R19
fO
R18 R19
VOUT (VIN(max) VOUT )
The peak–to–peak inductor current ripple is:
Eq. 3
IL(pp)
where fO = switching frequency when R18 is 100k and
R19 being open, fO is typically 450kHz. For more precise
setting, it is recommended to use the graph illustrated
in Figure 6:
VOUT (VIN(max) VOUT )
VIN(max) fsw L
Eq. 5
The peak inductor current is equal to the average output
current plus one half of the peak–to–peak inductor
current ripple.
IL(pk) =IOUT(max) + 0.5 ΔIL(pp)
Eq. 6
The RMS inductor current is used to calculate the I2R
losses in the inductor:
2
IL(RMS) IOUT(max)
ΔIL(PP)
12
2
Eq. 7
Figure 6. Switching Frequency vs. R19
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Micrel, Inc.
MIC28510
The maximum value of ESR is calculated:
Maximizing efficiency requires the proper selection of
core material while minimizing the winding resistance.
The high frequency operation of the MIC28510 requires
the use of ferrite materials for all but the most cost
sensitive applications. Lower cost iron powder cores
may be used but the increase in core loss will reduce the
efficiency of the power supply. This is especially
noticeable at low output power. The winding resistance
decreases efficiency at the higher output current levels.
The winding resistance must be minimized although this
usually comes at the expense of a larger inductor. The
power dissipated in the inductor is equal to the sum of
the core and copper losses. At higher output loads, the
core losses are usually insignificant and can be ignored.
At lower output currents, the core losses can be a
significant contributor. Core loss information is usually
available from the magnetics vendor. Copper loss in the
inductor is calculated by Equation 8:
PINDUCTOR(Cu) = IL(RMS)2 × RWINDING
ESR COUT
ΔVOUT(pp)
Eq. 10
ΔIL(PP)
where:
ΔVOUT(pp) = peak–to–peak output voltage ripple
ΔIL(PP) = peak–to–peak inductor current ripple
The total output ripple is a combination of the ESR and
output capacitance. The total ripple is calculated in
Equation 10:
2
ΔVOUT(pp)
ΔIL(PP)
ΔIL(PP) ESR C
OUT
C
f
8
OUT
SW
2
Eq. 11
Eq. 8
where:
COUT = Output capacitance value
fSW = Switching frequency
The resistance of the copper wire, RWINDING, increases
with the temperature. The value of the winding
resistance used should be at the operating temperature:
PWINDING(Ht) = RWINDING(20°C) × (1 + 0.0042 × (TH – T20°C))
Eq. 9
As described in the “Theory of Operation” subsection in
Functional Description, the MIC28510 requires at least
20mV peak–to–peak ripple at the FB pin to make the gm
amplifier and the error comparator behave properly. Also,
the output voltage ripple should be in phase with the
inductor current. Therefore, the output voltage ripple
caused by the output capacitors value should be much
smaller than the ripple caused by the output capacitor
ESR. If low–ESR capacitors, such as ceramic capacitors,
are selected as the output capacitors, a ripple injection
method should be applied to provide the enough
feedback voltage ripple. Please refer to the “Ripple
Injection” subsection for more details.
The voltage rating of the capacitor should be 20%
greater for aluminum electrolytic or OS–CON. The
output capacitor RMS current is calculated in Equation
12:
where:
TH = Temperature of wire under full load
T20°C = Ambient temperature
RWINDING(20°C) = Room temperature winding resistance
(usually specified by the manufacturer)
Output Capacitor Selection
The type of the output capacitor is usually determined by
its equivalent series resistance (ESR). Voltage and RMS
current capability are two other important factors for
selecting the output capacitor. Recommended capacitor
types are ceramic, low–ESR aluminum electrolytic, OS–
CON and POSCAP. The output capacitor’s ESR is
usually the main cause of the output ripple. The output
capacitor ESR also affects the control loop from a
stability point of view.
ICOUT (RMS)
ΔIL(PP)
Eq. 12
12
The power dissipated in the output capacitor is:
2
PDISS(COUT ) ICOUT (RMS) ESR COUT
March 2012
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Eq. 13
M9999-030912-A
Micrel, Inc.
MIC28510
The applications are divided into three situations
according to the amount of the feedback voltage ripple:
1) Enough ripple at the feedback voltage due to the large
ESR of the output capacitors.
As shown in Figure 7a, the converter is stable without
any ripple injection. The feedback voltage ripple is:
Input Capacitor Selection
The input capacitor for the power stage input VIN should
be selected for ripple current rating and voltage rating.
Tantalum input capacitors may fail when subjected to
high inrush currents, caused by turning the input supply
on. A tantalum input capacitor’s voltage rating should be
at least two times the maximum input voltage to
maximize reliability. Aluminum electrolytic, OS–CON,
and multilayer polymer film capacitors can handle the
higher inrush currents without voltage de–rating. The
input voltage ripple will primarily depend on the input
capacitor’s ESR. The peak input current is equal to the
peak inductor current, so:
ΔVIN = IL(pk) × CESR
ΔVFB(pp)
Eq. 14
Eq. 15
ΔVFB(pp) ESR ΔIL (pp)
The power dissipated in the input capacitor is:
PDISS(CIN) = ICIN(RMS)2 × CESR
Eq. 17
where ΔIL(pp) is the peak–to–peak value of the inductor
current ripple.
2) Inadequate ripple at the feedback voltage due to the
small ESR of the output capacitors.
The output voltage ripple is fed into the FB pin through a
feedforward capacitor CFF in this situation, as shown in
Figure 7b. The typical CFF value is between 1nF and
22nF.
With the feedforward capacitor, the feedback voltage
ripple is very close to the output voltage ripple:
The input capacitor must be rated for the input current
ripple. The RMS value of input capacitor current is
determined at the maximum output current. Assuming
the peak–to–peak inductor current ripple is low:
ICIN(RMS) IOUT(MAX ) D (1 D)
R2
ESR COUT ΔIL (pp)
R1 R2
Eq. 18
3) Virtually no ripple at the FB pin voltage due to the very
low ESR of the output capacitors.
Eq. 16
Ripple Injection
The VFB ripple required for proper operation of the
MIC28510 gm amplifier and error comparator is 20mV to
100mV. However, the output voltage ripple is generally
designed as 1% to 2% of the output voltage. For a low
output voltage, such as a 1V, the output voltage ripple is
only 10mV to 20mV, and the feedback voltage ripple is
less than 20mV. If the feedback voltage ripple is so small
that the gm amplifier and error comparator can’t sense it,
then the MIC28510 will lose control and the output
voltage is not regulated. In order to have some amount
of VFB ripple, a ripple injection method is applied for low
output voltage ripple applications.
Figure 7a. Enough Ripple at FB
Figure 7b. Inadequate Ripple at FB
March 2012
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M9999-030912-A
Micrel, Inc.
MIC28510
The process of sizing the ripple injection resistor and
capacitors is:
Step 1. Select CFF to feed all output ripples into the
feedback pin and make sure the large time constant
assumption is satisfied. Typical choice of CFF is 1nF to
22nF if R1 and R2 are in kΩ range.
Step 2. Select RINJ according to the expected feedback
voltage ripple using Equation 22:
K DIV
Figure 7c. Invisible Ripple at FB
In this situation, the output voltage ripple is less than
20mV. Therefore, additional ripple is injected into the FB
pin from the switching node SW via a resistor RINJ and a
capacitor CINJ, as shown in Figure 7c.
The injected ripple is:
ΔVFB(PP) VIN K DIV D (1- D)
K DIV
R1//R2
R INJ R1//R2
1
f SW τ
ΔVFB(PP)
VIN
f SW τ
D (1 D)
Eq. 22
Then the value of Rinj is obtained as:
R INJ (R1//R2) (
1
K DIV
1)
Eq. 23
Step 3. Select CINJ as 100nF, which could be considered
as short for a wide range of the frequencies.
Setting Output Voltage
The MIC28510 requires two resistors to set the output
voltage as shown in Figure 8.
Eq. 19
Eq. 20
where:
VIN = Power stage input voltage
D = Duty cycle
fSW = Switching frequency
τ = (R1//R2//RINJ) × CFF
In Equations 19 and 20, it is assumed that the time
constant associated with CFF must be much greater than
the switching period:
Figure 8. Voltage–Divider Configuration
1
fSW
T
1
Eq. 21
If the voltage divider resistors R1 and R2 are in the kΩ
range, a CFF of 1nF to 22nF can easily satisfy the large
time constant requirements. Also, a 100nF injection
capacitor CINJ is used in order to be considered as short
for a wide range of the frequencies.
March 2012
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M9999-030912-A
Micrel, Inc.
MIC28510
The output voltage is determined by Equation 24:
R1
VO VFB 1
R2
Eq. 24
where, VFB = 0.8V. A typical value of R1 can be between
3kΩ and 10kΩ. If R1 is too large, it may allow noise to be
introduced into the voltage feedback loop. If R1 is too
small, it will decrease the efficiency of the power supply,
especially at light loads. Once R1 is selected, R2 can be
calculated using:
R2
VFB R1
VOUT VFB
VIN (V)
L
COUT
R3
(RINJ)
C3
(CINJ)
0.8V to
3.3V
5V to
75V
10H
100F
16.5k
0.1F
5V
5.88V to
75V
10H
100F
10k
0.1F
12V
15V to
75V
27H
2x47F
45.3k
0.1F
24V
30V to
75V
47H
10F+220F
133k
0.1F
Table 1. Recommended Component values
Eq. 25
`
The inverting input voltage VINJ is clamped to 1.2V. As
the injected ripple increases, the swing of VINJ will be
clamped. The clamped VINJ reduces the line regulation
because it is reflected back as a DC error on the FB
terminal.
The Figure 9 shows the typical input output relationship.
The typical operating point should fall under the nongrey part of the graph shown in Figure 9. This should be
used in conjunction with the recommended component
values indicated in Table 1 and Bill of Matrials Table.
Thermal Measurements
Measuring the IC’s case temperature is recommended to
ensure it is within its operating limits. Although this might
seem like a very elementary task, it is easy to get
erroneous results. The most common mistake is to use
the standard thermal couple that comes with a thermal
meter. This thermal couple wire gauge is large, typically
22 gauge, and behaves like a heatsink, resulting in a
lower case measurement.
Two methods of temperature measurement are using a
smaller thermal couple wire or an infrared thermometer.
If a thermal couple wire is used, it must be constructed
of 36 gauge wire or higher (smaller wire size) to
minimize the wire heat–sinking effect. In addition, the
thermal couple tip must be covered in either thermal
grease or thermal glue to make sure that the thermal
couple junction is making good contact with the case of
the IC. Omega brand thermal couple (5SC–TT–K–36–
36) is adequate for most applications.
Wherever possible, an infrared thermometer is
recommended. The measurement spot size of most
infrared thermometers is too large for an accurate
reading on a small form factor ICs. However, an IR
thermometer from Optris has a 1mm spot size, which
makes it a good choice for measuring the hottest point
on the case. An optional stand makes it easy to hold the
beam on the IC for long periods of time.
Figure 9. Output Voltage vs Input Voltage
March 2012
VOUT (V)
22
M9999-030912-A
Micrel, Inc.
MIC28510
In “Hot–Plug” applications, a Tantalum or Electrolytic
bypass capacitor must be used to limit the over–
voltage spike seen on the input supply with power is
suddenly applied.
Inductor
PCB Layout Guidelines
Warning!!! To minimize EMI and output noise, follow
these layout recommendations.
PCB Layout is critical to achieve reliable, stable and
efficient performance. A ground plane is required to
control EMI and minimize the inductance in power,
signal and return paths. Thickness of the copper planes
is also important in terms of dissipating heat. The 2 oz
copper thickness is adequate from thermal point of view
and also thick copper plain helps in terms of noise
immunity. Keep in mind thinner planes can be easily
penetrated by noise
The following guidelines should be followed to insure
proper operation of the MIC28510 converter.
IC
The 2.2µF ceramic capacitor, which is connected to
the VDD pin, must be located right at the IC. The
VDD pin is very noise sensitive and placement of the
capacitor is very critical. Use wide traces to connect
to the VDD and PGND pins.
The signal ground pin (SGND) must be connected
directly to the ground planes. The SGND and PGND
connection should be done at a single point near the
IC. Do not route the SGND pin to the PGND Pad on
the top layer.
Place the IC close to the point–of–load (POL).
Use fat traces to route the input and output power
lines.
Signal and power grounds should be kept separate
and connected at only one location.
Place the input capacitor next to the power pins.
Place the input capacitors on the same side of the
board and as close to the IC as possible.
Keep both the PVIN pin and PGND connections
short.
Place several vias to the ground plane close to the
input capacitor ground terminal.
Use either X7R or X5R dielectric input capacitors.
Do not use Y5V or Z5U type capacitors.
Do not replace the ceramic input capacitor with any
other type of capacitor. Any type of capacitor can be
placed in parallel with the input capacitor.
If a Tantalum input capacitor is placed in parallel
with the input capacitor, it must be recommended for
switching regulator applications and the operating
voltage must be derated by 50%.
March 2012
Keep the inductor connection to the switch node
(SW) short.
Do not route any digital lines underneath or close to
the inductor.
Keep the switch node (SW) away from the feedback
(FB) pin.
The CS pin should be connected directly to the SW
pin to accurate sense the voltage across the low–
side MOSFET.
To minimize noise, place a ground plane underneath
the inductor.
The inductor can be placed on the opposite side of
the PCB with respect to the IC. It does not matter
whether the IC or inductor is on the top or bottom as
long as there is enough air flow to keep the power
components within their temperature limits. The
input and output capacitors must be placed on the
same side of the board as the IC.
Output Capacitor
Use a wide trace to connect the output capacitor
ground terminal to the input capacitor ground
terminal.
Phase margin will change as the output capacitor
value and ESR changes. Contact the factory if the
output capacitor is different from what is shown in
the BOM.
The feedback trace should be separate from the
power trace and connected as close as possible to
the output capacitor. Sensing a long high current
load trace can degrade the DC load regulation.
RC Snubber
Input Capacitor
23
Place the RC snubber on either side of the board
and as close to the SW pin as possible.
M9999-030912-A
Micrel, Inc.
MIC28510
Evaluation Board Schematic
Figure 10. Schematic of MIC28510 Evaluation Board
(J9, J10, J11, R13, R15 are for testing purposes)
March 2012
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M9999-030912-A
Micrel, Inc.
MIC28510
Bill of Materials
Item
C1
C2, C3
C13
C6
Part Number
EEU–FC2A101B
GRM32ER72A225KA35L
C3225X7R2A225KT5
GRM32ER60J107ME20L
C8, C9
GRM188R72A104KA35D
D1
D2
L1
Q1
R1
R2, R16
R3
R4
R5
R6
March 2012
Murata(2)
TDK
Murata(2)
0805ZC225MAT2A
AVX(4)
GRM21BR71A225KA01L
Murata(2)
TDK
06031C102KAT2A
(4)
C2012X7R1A105K
1
2.2µF Ceramic Capacitor, X7R, Size 1210, 100V
2
100µF Ceramic Capacitor, X5R, Size 1210, 6.3V
1
0.1µF Ceramic Capacitor, X7R, Size 0603, 50V
1
0.1µF Ceramic Capacitor, X7R, Size 0603, 100V
1
2.2µF Ceramic Capacitor, X7R, Size 0805, 10V
2
1nF Ceramic Capacitor, X7R, Size 0603, 100V
1
4.7nF Ceramic Capacitor, X7R, Size 0603, 50V
1
1µF Ceramic Capacitor, X7R, Size 0805, 10V
1
Small Signal Schottky Diode
1
Murata(2)
TDK(3)
GRM21BR71A105KA01L
100µF Aluminum Capacitor, SMD, 100V
(3)
C1608X7R2A102K
C1608X7R2A472K
Qty.
(3)
TDK(3)
06035C472KAT2A
C4, C5, C7,
C14, C15
Murata(2)
C1608X7S2A104K
GRM188R71H472KA01D
C16
TDK(3)
AVX(4)
GRM188R72A102KA01D
C12
Murata
06035C104KAT2A
GRM188R71H104KA93D
Description
(1)
(2)
AVX(4)
C2012X7R1A225K
C11
Panasonic
12106D107MAT2A
C1608X7R1H104K
C10
Manufacturer
AVX
(2)
Murata
(3)
TDK
(4)
AVX
Murata(2)
TDK(3)
Open
BAT46W–TP
MCC(5)
BAT46W–7–F
Diodes Inc.(6)
MMXZ5232B–TP
CMDZ5L6
DR125–100–R
FCX493
CRCW06034R75FKEA
CRCW08051R21FKEA
CRCW060316K5FKEA
CRCW060310K0FKEA
CRCW060380K6FKEA
CRCW060340K2FKEA
MCC(5)
5.6V Zener Diode
(7)
Central Semi
Cooper Bussmann(8) 10µH Inductor, 5.35A RMS, 7A Saturation Current
(6)
Diodes Inc/ZETEX
1
1
100V NPN Transistor
1
(9)
4.75Ω Resistor, Size 0603, 1%
1
(9)
1.21Ω Resistor, Size 0805, 1%
2
(9)
16.5kΩ Resistor, Size 0603, 1%
1
(9)
10kΩ Resistor, Size 0603, 1%
1
(9)
80.6kΩ Resistor, Size 0603, 1%
1
(9)
40.2kΩ Resistor, Size 0603, 1%
1
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale
25
M9999-030912-A
Micrel, Inc.
MIC28510
Bill of Materials (Continued)
Item
R7
Part Number
CRCW060320K0FKEA
Manufacturer
Description
Qty.
(9)
20kΩ Resistor, Size 0603, 1%
1
(9)
Vishay Dale
R8
CRCW060311K5FKEA
Vishay Dale
11.5kΩ Resistor, Size 0603, 1%
1
R9
CRCW06038K06FKEA
Vishay Dale(9)
8.06kΩ Resistor, Size 0603, 1%
1
CRCW06034K75FKEA
(9)
4.75kΩ Resistor, Size 0603, 1%
1
(9)
3.24kΩ Resistor, Size 0603, 1%
1
(9)
1.91kΩ Resistor, Size 0603, 1%
1
(9)
0Ω Resistor, Size 0603
1
(9)
10kΩ Resistor, Size 0805, 1%
1
(9)
49.9Ω Resistor, Size 0603, 1%
1
(9)
715Ω Resistor, Size 0603, 1%
(9)
100kΩ Resistor, Size 0603, 1%
2
(9)
1
R10
R11
R12
R13
R14
R15
CRCW06033K24FKEA
CRCW06031K91FKEA
CRCW06030000Z0EAHP
CRCW080510K0JNEA
CRCW060349R9FKEA
R17 (OPEN) CRCW0603715RFKEA
R18, R19
R20
CRCW0603100KFKEAHP
CRCW06032R00FKEA
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale
2Ω Resistor, Size 0603, 1%
R21 (OPEN) CRCW0603348RFKEA
Vishay Dale(9)
348Ω Resistor, Size 0603, 1%
U1
Micrel. Inc.(10)
75V/4A Synchronous Buck DC/DC Regulator
MIC28510YJL
1
Notes:
1.
Panasonic: www.panasonic.com.
2.
Murata: www.murata.com.
3.
TDK: www.tdk.com.
4.
AVX: www.avx.com.
5.
MCC: www.mccsemi.com.
6.
Diodes Inc.: www.diodes.com.
7.
Central Semi: www.centralsemi.com.
8.
Cooper: www.cooperbussman.com.
9.
Vishay Dale : www.vishay.com.
10. Micrel, Inc.: www.micrel.com.
March 2012
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M9999-030912-A
Micrel, Inc.
MIC28510
PCB Layout Recommendations
Figure 11. MIC28510 Evaluation Board Top Layer
Figure 12. MIC28510 Evaluation Board Mid–Layer 1 (Ground Plane)
March 2012
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M9999-030912-A
Micrel, Inc.
MIC28510
PCB Layout Recommendations (Continued)
Figure 13. MIC28510 Evaluation Board Mid–Layer 2
Figure 14. MIC28510 Evaluation Board Bottom Layer
March 2012
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M9999-030912-A
Micrel, Inc.
MIC28510
Recommended Land Pattern
Red circle indicates Thermal Via. Size and must be connected to GND plane for maximum thermal performance.
Green rectangle (with shaded area) indicates Solder Stencil Opening on exposed pad area.
Blue and Magenta colored pads indicate different potential. DO NOT connect to GND plane.
Thermal Via
Via Size/Pitch
Solder Stencil Opening/Pitch
X
0.300 0.35mm/0.80mm
1.551.20mm/1.75mm
Blue Circle/Black Pad
X
0.300 0.35mm/0.80mm
0.801.11mm/1.31mm
Magenta Circle/Black Pad
X
0.300 0.35mm/0.80mm
0.501.11mm/1.31mm
Red Circle/Black Pad
March 2012
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M9999-030912-A
Micrel, Inc.
MIC28510
Package Information
28–Pin 5mm 6mm MLF® (JL)
MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA
TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com
Micrel makes no representations or warranties with respect to the accuracy or completeness of the information furnished in this data sheet. This
information is not intended as a warranty and Micrel does not assume responsibility for its use. Micrel reserves the right to change circuitry,
specifications and descriptions at any time without notice. No license, whether express, implied, arising by estoppel or otherwise, to any intellectual
property rights is granted by this document. Except as provided in Micrel’s terms and conditions of sale for such products, Micrel assumes no liability
whatsoever, and Micrel disclaims any express or implied warranty relating to the sale and/or use of Micrel products including liability or warranties
relating to fitness for a particular purpose, merchantability, or infringement of any patent, copyright or other intellectual property right.
Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product
can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant
into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A
Purchaser’s use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser’s own risk and Purchaser agrees to fully
indemnify Micrel for any damages resulting from such use or sale.
© 2011 Micrel, Incorporated.
March 2012
30
M9999-030912-A
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