LTC1735
High Efficiency
Synchronous Step-Down
Switching Regulator
DESCRIPTIO
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FEATURES
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The LTC®1735 is a synchronous step-down switching
regulator controller that drives external N-channel power
MOSFETs using a fixed frequency architecture. Burst
ModeTM operation provides high efficiency at low load
currents. The precision 0.8V reference is compatible with
future generation microprocessors. OPTI-LOOP compensation allows the transient response to be optimized over
a wide range of output capacitance and ESR values.
Synchronizable/Programmable Fixed Frequency
OPTI-LOOPTM Compensation Minimizes COUT
±1% Output Voltage Accuracy
Dual N-Channel MOSFET Synchronous Drive
Wide VIN Range: 4V to 36V Operation
VOUT Range: 0.8V to 6V
Internal Current Foldback
Output Overvoltage Crowbar Protection
Latched Short-Circuit Shutdown Timer
with Defeat Option
Very Low Dropout Operation: 99% Duty Cycle
Forced Continuous Control Pin
Optional Programmable Soft-Start
Remote Output Voltage Sense
Power Good Output (LTC1735F Only)
Logic Controlled Micropower Shutdown: IQ < 25µA
LTC1435 Pin Compatible with
Minor Component Changes
Available in 16-Lead Narrow SSOP, SO Packages and
20-Lead TSSOP Package (LTC1735F Only)
The operating frequency (synchronizable up to 500kHz) is
set by an external capacitor allowing maximum flexibility
in optimizing efficiency. A forced continuous control pin
reduces noise and RF interference and can assist secondary winding regulation by disabling Burst Mode operation
when the main output is lightly loaded.
Protection features include internal foldback current limiting, output overvoltage crowbar and optional shortcircuit shutdown. Soft-start is provided by an external
capacitor that can be used to properly sequence supplies.
The operating current level is user-programmable via an
external current sense resistor. Wide input supply range
allows operation from 4V to 30V (36V maximum).
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APPLICATIO S
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, LTC and LT are registered trademarks of Linear Technology Corporation.
Burst Mode and OPTI-LOOP are trademarks of Linear Technology Corporation.
Notebook and Palmtop Computers, PDAs
Cellular Telephones and Wireless Modems
DC Power Distribution Systems
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TYPICAL APPLICATIO
CSS
0.1µF
COSC
COSC
47pF
COUT: PANASONIC EEFUEOG181R
CIN: MARCON THCR70E1H226ZT
L1: PANASONIC ETQP6F2R0HFA
RSENSE: IRC LRF2010-01-R005J
SW
CC2
100pF
LTC1735
SGND
R2
20k
1%
CB
0.22µF
BOOST
ITH
CC
330pF
RC
33k
TG
RUN/SS
L1
2µH
DB
CMDSH-3
VIN
VIN
5V TO 24V
CIN
22µF
50V
M1
FDS6680A
RSENSE
0.005Ω
VOUT
1.6V
9A
100pF
VOSENSE
R1
20k
1%
INTVCC
BG
SENSE +
PGND
SENSE
+
4.7µF
–
M2
FDS6680A
D1
MBRS340T3
+
1000pF
COUT
180µF
4V
×4
SP
1735 F01
Figure 1. High Efficiency Step-Down Converter
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LTC1735
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ABSOLUTE
AXI U RATI GS
(Note 1)
Input Supply Voltage (VIN).........................36V to – 0.3V
Topside Driver Supply Voltage (BOOST)....42V to – 0.3V
Switch Voltage (SW) ....................................36V to – 5V
EXTVCC Voltage ...........................................7V to – 0.3V
Boosted Driver Voltage (BOOST – SW) .......7V to – 0.3V
SENSE +, SENSE – Voltages .......... 1.1 (INTVCC) to – 0.3V
FCB Voltage ............................(INTVCC + 0.3V) to – 0.3V
ITH, VOSENSE Voltages ...............................2.7V to – 0.3V
RUN/SS, PGOOD (LTC1735F Only)
Voltages .......................................................7V to – 0.3V
Peak Driver Output Current 7.5%) as well as other more serious
conditions that may overvoltage the output. In this case,
the top MOSFET is turned off and the bottom MOSFET is
turned on until the overvoltage condition is cleared.
Foldback current limiting for an output shorted to ground
is provided by amplifier A. As VOSENSE drops below 0.6V,
the buffered ITH input to the current comparator is gradually pulled down to a 0.86V clamp. This reduces peak
inductor current to about 1/4 of its maximum value.
Low Current Operation
The LTC1735 has three low current modes controlled by
the FCB pin. Burst Mode operation is selected when the
FCB pin is above 0.8V (typically tied to INTVCC). In Burst
Mode operation, if the error amplifier drives the ITH voltage
below 0.86V, the buffered ITH input to the current comparator will be clamped at 0.86V. The inductor current
peak is then held at approximately 20mV/RSENSE (about
1/4 of maximum output current). If ITH then drops below
0.5V, the Burst Mode comparator B will turn off both
MOSFETs to maximize efficiency. The load current will be
supplied solely by the output capacitor until ITH rises
above the 60mV hysteresis of the comparator and switching is resumed. Burst Mode operation is disabled by
comparator F when the FCB pin is brought below 0.8V.
This forces continuous operation and can assist secondary winding regulation.
When the FCB pin is driven by an external oscillator, a low
noise cycle-skipping mode is invoked and the internal
oscillator is synchronized to the external clock by comparator C. In this mode the 25% minimum inductor
current clamp is removed, providing constant frequency
discontinuous operation over the widest possible output
current range. This constant frequency operation is not
quite as efficient as Burst Mode operation, but provides a
lower noise, constant frequency spectrum.
Tying the FCB pin to ground enables forced continuous
operation. This is the least efficient mode, but is desirable
in certain applications. The output can source or sink
current in this mode. When sinking current while in forced
continuous operation, current will be forced back into the
main power supply potentially boosting the input supply to
dangerous voltage levels—BEWARE.
Foldback Current, Short-Circuit Detection and
Short-Circuit Latchoff
The RUN/SS capacitor, CSS, is used initially to limit the
inrush current of the switching regulator. After the controller has been started and been given adequate time to
charge up the output capacitors and provide full load current, CSS is used as a short-circuit time-out circuit. If the
output voltage falls to less than 70% of its nominal output
voltage, CSS begins discharging on the assumption that
the output is in an overcurrent and/or short-circuit condition. If the condition lasts for a long enough period as
determined by the size of CSS, the controller will be shut
down until the RUN/SS pin voltage is recycled. This builtin latchoff can be overridden by providing a current >5µA
at a compliance of 5V to the RUN/SS pin. This current
shortens the soft-start period but also prevents net discharge of CSS during an overcurrent and/or short-circuit
condition. Foldback current limiting is activated when the
output voltage falls below 70% of its nominal level whether
or not the short-circuit latchoff circuit is enabled.
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OPERATIO
(Refer to Functional Diagram)
INTVCC/EXTVCC POWER
Power for the top and bottom MOSFET drivers and most
of the internal circuitry of the LTC1735 is derived from the
INTVCC pin. When the EXTVCC pin is left open, an internal
5.2V low dropout regulator supplies the INTVCC power
from VIN. If EXTVCC is raised above 4.7V, the internal
regulator is turned off and an internal switch connects
EXTVCC to INTVCC. This allows a high efficiency source,
such as the primary or a secondary output of the converter
itself, to provide the INTVCC power. Voltages up to 7V can
be applied to EXTVCC for additional gate drive capability.
To provide clean start-up and to protect the MOSFETs,
undervoltage lockout is used to keep both MOSFETs off
until the input voltage is above 3.5V.
PGOOD (LTC1735F Only)
A window comparator monitors the output voltage and its
open-drain output is pulled low when the divided down
output voltage is not within ±7.5% of the reference voltage
of 0.8V.
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The basic LTC1735 application circuit is shown in Figure␣ 1
on the first page. External component selection is driven
by the load requirement and begins with the selection of
RSENSE. Once RSENSE is known, COSC and L can be chosen.
Next, the power MOSFETs and D1 are selected. The
operating frequency and the inductor are chosen based
largely on the desired amount of ripple current. Finally, CIN
is selected for its ability to handle the large RMS current
into the converter and COUT is chosen with low enough
ESR to meet the output voltage ripple and transient specifications. The circuit shown in Figure 1 can be configured
for operation up to an input voltage of 28V (limited by the
external MOSFETs).
RSENSE Selection for Output Current
RSENSE is chosen based on the required output current.
The LTC1735 current comparator has a maximum threshold of 75mV/RSENSE and an input common mode range of
SGND to 1.1(INTVCC). The current comparator threshold
sets the peak of the inductor current, yielding a maximum
average output current IMAX equal to the peak value less
half the peak-to-peak ripple current, ∆IL.
Allowing a margin for variations in the LTC1735 and
external component values yields:
RSENSE =
50mV
IMAX
COSC Selection for Operating Frequency and
Synchronization
The choice of operating frequency and inductor value is a
trade-off between efficiency and component size. Low
frequency operation improves efficiency by reducing
MOSFET switching losses, both gate charge loss and
transition loss. However, lower frequency operation requires more inductance for a given amount of ripple
current.
The LTC1735 uses a constant frequency architecture with
the frequency determined by an external oscillator capacitor COSC. Each time the topside MOSFET turns on, the
voltage on COSC is reset to ground. During the on-time,
COSC is charged by a fixed current. When the voltage on the
capacitor reaches 1.19V, COSC is reset to ground. The
process then repeats.
The value of COSC is calculated from the desired operating
frequency assuming no external clock input on the FCB
pin:
1.61(107 )
– 11
COSC (pF) =
Frequency
A graph for selecting COSC versus frequency is shown in
Figure 2. The maximum recommended switching frequency is 550kHz .
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The internal oscillator runs at its nominal frequency (fO)
when the FCB pin is pulled high to INTVCC or connected to
ground. Clocking the FCB pin above and below 0.8V will
cause the internal oscillator to injection lock to an external
clock signal applied to the FCB pin with a frequency
between 0.9fO and 1.3fO. The clock high level must exceed
1.3V for at least 0.3µs and the clock low level must be less
than 0.3V for at least 0.3µs. The top MOSFET turn-on will
synchronize with the rising edge of the clock.
Attempting to synchronize to too high an external frequency (above 1.3fO) can result in inadequate slope compensation and possible loop instability. If this condition
exists simply lower the value of COSC so fEXT = fO according
to Figure 2.
The inductor value has a direct effect on ripple current. The
inductor ripple current ∆IL decreases with higher inductance or frequency and increases with higher VIN or VOUT:
∆IL =
V
1
VOUT 1 – OUT
VIN
( f)(L)
Accepting larger values of ∆IL allows the use of low
inductances, but results in higher output voltage ripple
and greater core losses. A reasonable starting point for
setting ripple current is ∆IL = 0.3 to 0.4(IMAX). Remember,
the maximum ∆IL occurs at the maximum input voltage.
100.0
87.5
75.0
COSC VALUE (pF)
of smaller inductor and capacitor values. So why would
anyone ever choose to operate at lower frequencies with
larger components? The answer is efficiency. A higher
frequency generally results in lower efficiency because of
MOSFET gate charge losses. In addition to this basic trade
off, the effect of inductor value on ripple current and low
current operation must also be considered.
62.5
50.0
37.5
25.0
12.5
0
0
100
200
300
400
500
OPERATING FREQUENCY (kHZ)
600
1735 F02
Figure 2. Timing Capacitor Value
When synchronized to an external clock, Burst Mode
operation is disabled but the inductor current is not
allowed to reverse. The 25% minimum inductor current
clamp present in Burst Mode operation is removed,
providing constant frequency discontinuous operation
over the widest possible output current range. In this
mode the synchronous MOSFET is forced on once every
10 clock cycles to recharge the bootstrap capacitor. This
minimizes audible noise while maintaining reasonably
high efficiency.
Inductor Value Calculation
The operating frequency and inductor selection are interrelated in that higher operating frequencies allow the use
The inductor value also has an effect on low current
operation. The transition to low current operation begins
when the inductor current reaches zero while the bottom
MOSFET is on. Burst Mode operation begins when the
average inductor current required results in a peak current
below 25% of the current limit determined by RSENSE.
Lower inductor values (higher ∆IL) will cause this to occur
at higher load currents, which can cause a dip in efficiency
in the upper range of low current operation. In Burst Mode
operation, lower inductance values will cause the burst
frequency to decrease.
Inductor Core Selection
Once the value for L is known, the type of inductor must be
selected. High efficiency converters generally cannot afford the core loss found in low cost powdered iron cores,
forcing the use of more expensive ferrite, molypermalloy
or Kool Mµ® cores. Actual core loss is independent of
core size for a fixed inductor value, but it is very dependent
on inductance selected. As inductance increases, core
losses decrease. Unfortunately, increased inductance requires more turns of wire and therefore copper losses will
increase.
Kool Mµ is a registered trademark of Magnetics, Inc.
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Ferrite designs have very low core loss and are preferred
at high switching frequencies, so design goals can concentrate on copper loss and preventing saturation. Ferrite
core material saturates “hard,” which means that inductance collapses abruptly when the peak design current is
exceeded. This results in an abrupt increase in inductor
ripple current and consequent output voltage ripple. Do
not allow the core to saturate!
Molypermalloy (from Magnetics, Inc.) is a very good, low
loss core material for toroids, but it is more expensive than
ferrite. A reasonable compromise from the same manufacturer is Kool Mµ. Toroids are very space efficient,
especially when you can use several layers of wire. Because they generally lack a bobbin, mounting is more
difficult. However, designs for surface mount are available
that do not increase the height significantly.
Power MOSFET and D1 Selection
Two external power MOSFETs must be selected for use
with the LTC1735: An N-channel MOSFET for the top
(main) switch and an N-channel MOSFET for the bottom
(synchronous) switch.
The peak-to-peak gate drive levels are set by the INTVCC
voltage. This voltage is typically 5.2V during start-up (see
EXTVCC pin connection). Consequently, logic-level threshold MOSFETs must be used in most LTC1735 applications. The only exception is when low input voltage is
expected (VIN < 5V); then, sub-logic level threshold
MOSFETs (VGS(TH) < 3V) should be used. Pay close
attention to the BVDSS specification for the MOSFETs as
well; many of the logic level MOSFETs are limited to 30V
or less.
Selection criteria for the power MOSFETs include the “ON”
resistance RDS(ON), reverse transfer capacitance CRSS,
input voltage and maximum output current. When the
LTC1735 is operating in continuous mode the duty cycles
for the top and bottom MOSFETs are given by:
V
Main Switch Duty Cycle = OUT
VIN
V –V
Synchronous Switch Duty Cycle = IN OUT
VIN
The MOSFET power dissipations at maximum output
current are given by:
( )( )
2
V
PMAIN = OUT IMAX 1 + δ RDS(ON) +
VIN
( ) (IMAX )(CRSS )(f)
k VIN
2
( )( )
2
V –V
PSYNC = IN OUT IMAX 1 + δ RDS(ON)
VIN
where δ is the temperature dependency of RDS(ON) and k
is a constant inversely related to the gate drive current.
Both MOSFETs have I2R losses while the topside
N-channel equation includes an additional term for transition losses, which are highest at high input voltages. For
VIN < 20V the high current efficiency generally improves
with larger MOSFETs, while for VIN > 20V the transition
losses rapidly increase to the point that the use of a higher
RDS(ON) device with lower CRSS actually provides higher
efficiency. The synchronous MOSFET losses are greatest
at high input voltage or during a short-circuit when the
duty cycle in this switch is nearly 100%.
The term (1 + δ) is generally given for a MOSFET in the
form of a normalized RDS(ON) vs Temperature curve, but
δ␣ = 0.005/°C can be used as an approximation for low
voltage MOSFETs. CRSS is usually specified in the
MOSFET characteristics. The constant k = 1.7 can be
used to estimate the contributions of the two terms in the
main switch dissipation equation.
The Schottky diode D1 shown in Figure 1 conducts during the
dead-time between the conduction of the two power MOSFETs.
This prevents the body diode of the bottom MOSFET from
turning on and storing charge during the dead-time, which
could cost as much as 1% in efficiency. A 3A Schottky is
generally a good size for 10A to 12A regulators due to the
relatively small average current. Larger diodes can result in
additional transition losses due to their larger junction capacitance. The diode may be omitted if the efficiency loss can be
tolerated.
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CIN Selection
In continuous mode, the source current of the top
N-channel MOSFET is a square wave of duty cycle VOUT/
VIN. To prevent large voltage transients, a low ESR input
capacitor sized for the maximum RMS current must be
used. The maximum RMS capacitor current is given by:
V
V
IRMS ≅ IO(MAX ) OUT IN – 1
VIN VOUT
1/ 2
This formula has a maximum at VIN = 2VOUT, where
IRMS␣ =␣ IO(MAX)/2. This simple worst case condition is commonly used for design because even significant deviations do
not offer much relief. Note that capacitor manufacturers’
ripple current ratings are often based on only 2000 hours of
life. This makes it advisable to further derate the capacitor or
to choose a capacitor rated at a higher temperature than
required. Several capacitors may also be paralleled to meet
size or height requirements in the design. Always consult the
manufacturer if there is any question.
COUT Selection
The selection of COUT is primarily determined by the
effective series resistance (ESR) to minimize voltage
ripple. The output ripple (∆VOUT) in continuous mode is
determined by:
1
∆VOUT ≈ ∆IL ESR +
8 fCOUT
Where f = operating frequency, COUT = output capacitance and ∆IL = ripple current in the inductor. The output
ripple is highest at maximum input voltage since ∆IL
increases with input voltage. Typically, once the ESR
requirement for COUT has been met, the RMS current
rating generally far exceeds the IRIPPLE(P–P) requirement.
With ∆IL = 0.3IOUT(MAX) and allowing 2/3 of the ripple due
to ESR the output ripple will be less than 50mV at max VIN
assuming:
COUT required ESR < 2.2 RSENSE
COUT > 1/(8fRSENSE)
The first condition relates to the ripple current into the ESR
of the output capacitance while the second term guarantees that the output capacitance does not significantly
discharge during the operating frequency period due to
ripple current. The choice of using smaller output capacitance increases the ripple voltage due to the discharging
term but can be compensated for by using capacitors of
very low ESR to maintain the ripple voltage at or below
50mV. The ITH pin OPTI-LOOP compensation components can be optimized to provide stable, high performance transient response regardless of the output capacitors selected.
The selection of output capacitors for CPU or other applications with large load current transients is primarily
determined by the voltage tolerance specifications of the
load. The resistive component of the capacitor, ESR,
multiplied by the load current change plus any output
voltage ripple must be within the voltage tolerance of the
load (CPU).
The required ESR due to a load current step is:
RESR < ∆V/∆I
where ∆I is the change in current from full load to zero load
(or minimum load) and ∆V is the allowed voltage deviation
(not including any droop due to finite capacitance).
The amount of capacitance needed is determined by the
maximum energy stored in the inductor. The capacitance
must be sufficient to absorb the change in inductor current
when a high current to low current transition occurs. The
opposite load current transition is generally determined by
the control loop OPTI-LOOP components, so make sure
not to over compensate and slow down the response. The
minimum capacitance to assure the inductors’ energy is
adequately absorbed is:
COUT >
L( ∆I)2
2( ∆V)VOUT
where ∆I is the change in load current.
Manufacturers such as Nichicon, United Chemi-Con and
Sanyo can be considered for high performance throughhole capacitors. The OS-CON semiconductor electrolyte
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capacitor available from Sanyo has the lowest (ESR)(size)
product of any aluminum electrolytic at a somewhat
higher price. An additional ceramic capacitor in parallel
with OS-CON capacitors is recommended to reduce the
inductance effects.
In surface mount applications, ESR, RMS current handling and load step specifications may require multiple
capacitors in parallel. Aluminum electrolytic, dry tantalum
and special polymer capacitors are available in surface
mount packages. Special polymer surface mount capacitors offer very low ESR but have much lower capacitive
density per unit volume than other capacitor types. These
capacitors offer a very cost-effective output capacitor
solution and are an ideal choice when combined with a
controller having high loop bandwidth. Tantalum capacitors offer the highest capacitance density and are often
used as output capacitors for switching regulators having
controlled soft-start. Several excellent surge-tested choices
are the AVX TPS, AVX TPSV or the KEMET T510 series of
surface mount tantalums, available in case heights ranging from 1.5mm to 4.1mm. Aluminum electrolytic capacitors can be used in cost-driven applications, provided that
consideration is given to ripple current ratings, temperature and long-term reliability. A typical application will
require several to many aluminum electrolytic capacitors
in parallel. A combination of the above mentioned capacitors will often result in maximizing performance and
minimizing overall cost. Other capacitor types include
Nichicon PL series, NEC Neocap, Panasonic SP and
Sprague 595D series. Consult manufacturers for other
specific recommendations.
Like all components, capacitors are not ideal. Each capacitor has its own benefits and limitations. Combinations of different capacitor types have proven to be a very
cost effective solution. Remember also to include high
frequency decoupling capacitors. They should be placed
as close as possible to the power pins of the load. Any
inductance present in the circuit board traces negates
their usefulness.
INTVCC Regulator
An internal P-channel low dropout regulator produces the
5.2V supply that powers the drivers and internal circuitry
within the LTC1735. The INTVCC pin can supply a maximum RMS current of 50mA and must be bypassed to
ground with a minimum of 4.7µF tantalum, 10µF special
polymer or low ESR type electrolytic capacitor. A 1µF
ceramic capacitor placed directly adjacent to the INTVCC
and PGND IC pins is highly recommended. Good bypassing is required to supply the high transient currents
required by the MOSFET gate drivers.
Higher input voltage applications in which large MOSFETs
are being driven at high frequencies may cause the maximum junction temperature rating for the LTC1735 to be
exceeded. The system supply current is normally dominated by the gate charge current. Additional loading of
INTVCC also needs to be taken into account for the power
dissipation calculations. The total INTVCC current can be
supplied by either the 5.2V internal linear regulator or by
the EXTVCC input pin. When the voltage applied to the
EXTVCC pin is less than 4.7V, all of the INTVCC current is
supplied by the internal 5.2V linear regulator. Power
dissipation for the IC in this case is highest: (VIN)(IINTVCC)
and overall efficiency is lowered. The gate charge is
dependent on operating frequency as discussed in the
Efficiency Considerations section. The junction temperature can be estimated by using the equations given in
Note␣ 2 of the Electrical Characteristics. For example, the
LTC1735CS is limited to less than 17mA from a 30V
supply when not using the EXTVCC pin as follows:
TJ = 70°C + (17mA)(30V)(110°C/W) = 126°C
Use of the EXTVCC input pin reduces the junction temperature to:
TJ = 70°C + (17mA)(5V)(110°C/W) = 79°C
To prevent maximum junction temperature from being
exceeded, the input supply current must be checked
operating in continuous mode at maximum VIN.
EXTVCC Connection
The LTC1735 contains an internal P-channel MOSFET
switch connected between the EXTVCC and INTVCC pins.
Whenever the EXTVCC pin is above 4.7V, the internal 5.2V
regulator shuts off, the switch closes and INTVCC power is
supplied via EXTVCC until EXTVCC drops below 4.5V. This
allows the MOSFET gate drive and control power to be
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derived from the output or other external source during
normal operation. When the output is out of regulation
(start-up, short circuit) power is supplied from the internal
regulator. Do not apply greater than 7V to the EXTVCC pin
and ensure that EXTVCC ≤ VIN.
Significant efficiency gains can be realized by powering
INTVCC from the output, since the VIN current resulting
from the driver and control currents will be scaled by a
factor of (Duty Cycle)/(Efficiency). For 5V regulators this
simply means connecting the EXTVCC pin directly to VOUT.
However, for 3.3V and other lower voltage regulators,
additional circuitry is required to derive INTVCC power
from the output.
Output Voltage Programming
The output voltage is set by an external resistive divider
according to the following formula:
R2
VOUT = 0.8 V 1 +
R1
The resistive divider is connected to the output as shown
in Figure 4 allowing remote voltage sensing.
2. EXTVCC connected directly to VOUT. This is the normal
connection for a 5V output regulator and provides the
highest efficiency. For output voltages higher than 5V,
EXTVCC is required to connect to VOUT so the SENSE pins’
absolute maximum ratings are not exceeded.
3. EXTVCC connected to an output-derived boost network.
For 3.3V and other low voltage regulators, efficiency gains
can still be realized by connecting EXTVCC to an outputderived voltage that has been boosted to greater than
4.7V. This can be done with either the inductive boost
winding as shown in Figure 3a or the capacitive charge
pump shown in Figure 3b. The charge pump has the
advantage of simple magnetics.
4. EXTVCC connected to an external supply. If an external
supply is available in the 5V to 7V range (EXTVCC ≤ VIN),
such as notebook main 5V system power, it may be used
to power EXTVCC providing it is compatible with the
MOSFET gate drive requirements. This is the typical case
as the 5V power is almost always present and is derived by
another high efficiency regulator.
+
CIN
The following list summarizes the four possible connections for EXTVCC:
1. EXTVCC left open (or grounded). This will cause INTVCC
to be powered from the internal 5.2V regulator resulting in
an efficiency penalty of up to 10% at high input voltages.
VIN
OPTIONAL EXTVCC
CONNECTION
5V ≤ VSEC ≤ 7V
1N4148
VIN
LTC1735
VSEC
+
6.8V
1µF
N-CH
TG
RSENSE
VOUT
EXTVCC
SW
FCB
BG
L1
1:N
R4
+
COUT
N-CH
R3
SGND
PGND
1735 F03a
Figure 3a. Secondary Output Loop and EXTVCC Connection
+
VIN
1µF
+
CIN
BAT85
VIN
0.22µF
BAT85
LTC1735
TG
BAT85
VN2222LL
N-CH
RSENSE
VOUT
EXTVCC
L1
SW
+
BG
COUT
N-CH
PGND
1735 F03b
Figure 3b. Capacitive Charge Pump for EXTVCC
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VOUT
R2
VOSENSE
47pF
LTC1735
R1
SGND
1735 F04
Figure 4. Setting the LTC1735 Output Voltage
Topside MOSFET Driver Supply (CB, DB)
An external bootstrap capacitor CB connected to the
BOOST pin supplies the gate drive voltage for the topside
MOSFET. Capacitor CB in the Functional Diagram is charged
though external diode DB from INTVCC when the SW pin
is low. Note that the voltage across CB is about a diode
drop below INTVCC. When the topside MOSFET is to be
turned on, the driver places the CB voltage across the
gate-source of the MOSFET. This enhances the MOSFET
and turns on the topside switch. The switch node voltage
SW rises to VIN and the BOOST pin rises to VIN + INTVCC.
The value of the boost capacitor CB needs to be 100 times
greater than the total input capacitance of the topside
MOSFET. In most applications 0.1µF to 0.33µF is adequate. The reverse breakdown on DB must be greater
than VIN(MAX).
When adjusting the gate drive level, the final arbiter is the
total input current for the regulator. If you make a change
and the input current decreases, then you improved the
efficiency. If there is no change in input current, then there
is no change in efficiency.
SENSE +/SENSE – Pins
The common mode input range of the current comparator
is from 0V to 1.1(INTVCC). Continuous linear operation in
step-down applications is guaranteed throughout this
range allowing output voltages anywhere from 0.8V to 7V.
A differential NPN input stage is used and is biased with
internal resistors from an internal 2.4V source as shown
in the Functional Diagram. This causes current to either be
sourced or sunk by the sense pins depending on the
output voltage. If the output voltage is below 2.4V current
will flow out of both sense pins to the main output. This
forces a minimum load current that can be fulfilled by the
VOUT resistive divider. The maximum current flowing out
of the sense pins is:
ISENSE+ + ISENSE– = (2.4V – VOUT)/24k
Since VOSENSE is servoed to the 0.8V reference voltage, we
can choose R1 in Figure 4 to have a maximum value to
absorb this current:
0.8 V
R1(MAX ) = 24k
2.4V – VOUT
Regulating an output voltage of 1.8V, the maximum value
of R1 should be 32k. Note that at output voltages above
2.4V no maximum value of R1 is necessary to absorb the
sense pin currents; however, R1 is still bounded by the
VOSENSE feedback current.
Soft-Start/Run Function
The RUN/SS pin is a multipurpose pin that provides a softstart function and a means to shut down the LTC1735.
Soft-start reduces surge currents from VIN by gradually
increasing the controller’s current limit ITH(MAX). This pin
can also be used for power supply sequencing.
Pulling the RUN/SS pin below 1.5V puts the LTC1735 into
shutdown. This pin can be driven directly from logic as
shown in Figure 5. The VIN quiescent current is a function
of RUN/SS voltage (refer to Typical Performance Characteristics graphs on page 6). Releasing the RUN/SS pin
allows an internal 1.2µA current source to charge up the
external soft-start capacitor CSS. If RUN/SS has been
pulled all the way to ground there is a delay before starting
of approximately:
tDELAY =
1.5V
CSS = (1.25s /µF ) CSS
1.2µA
When the voltage on RUN/SS reaches 1.5V the LTC1735
begins operating with a current limit at approximately
25mV/RSENSE. As the voltage on the RUN/SS pin increases
from 1.5V to 3.0V, the internal current limit is increased
from 25mV/RSENSE to 75mV/RSENSE. The output current
limit ramps up slowly, taking an additional 1.25s/µF to
reach full current. The output current thus ramps up
slowly, reducing the starting surge current required from
the input power supply.
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Diode D1 in Figure 5 reduces the start delay while allowing
CSS to charge up slowly for the soft-start function. This
diode and CSS can be deleted if soft-start is not needed.
The RUN/SS pin has an internal 6V zener clamp (See
Functional Diagram).
3.3V OR 5V
RUN/SS
RUN/SS
D1
CSS
CSS
1735 F05
Figure 5. RUN/SS Pin Interfacing
Fault Conditions: Overcurrent Latchoff
The RUN/SS pin also provides the ability to shut off the
controller and latch off when an overcurrent condition is
detected. The RUN/SS capacitor, CSS, is used initially to
turn on and limit the inrush current of the controller. After
the controller has been started and given adequate time to
charge up the output capacitor and provide full load
current, CSS is used as a short-circuit timer. If the output
voltage falls to less than 70% of its nominal output voltage
after CSS reaches 4.1V, the assumption is made that the
output is in a severe overcurrent and/or short-circuit
condition, so CSS begins discharging. If the condition lasts
for a long enough period as determined by the size of CSS,
the controller will be shut down until the RUN/SS pin
voltage is recycled.
This built-in latchoff can be overridden by providing a
current >5µA at a compliance of 5V to the RUN/SS pin as
shown in Figure␣ 6. This current shortens the soft-start
period but also prevents net discharge of the RUN/SS
VIN
INTVCC
3.3V OR 5V
RUN/SS
RSS
D1
RSS
D1
RUN/SS
CSS
CSS
(a)
(b)
1735 F06
Figure 6. RUN/SS Pin Interfacing with Latchoff Defeated
capacitor during a severe overcurrent and/or short-circuit
condition. When deriving the 5µA current from VIN as in
Figure␣ 6a, current latchoff is always defeated. A diode
connecting this pull-up resistor to INTVCC , as in Figure␣ 6b,
eliminates any extra supply current during controller shutdown while eliminating the INTVCC loading from preventing controller start-up. If the voltage on CSS does not
exceed 4.1V, the overcurrent latch is not armed and the
function is disabled.
Why should you defeat overcurrent latchoff? During the
prototyping stage of a design, there may be a problem with
noise pickup or poor layout causing the protection circuit
to latch off. Defeating this feature will easily allow troubleshooting of the circuit and PC layout. The internal shortcircuit and foldback current limiting still remains active,
thereby protecting the power supply system from failure.
After the design is complete, a decision can be made
whether to enable the latchoff feature.
The value of the soft-start capacitor CSS will need to be
scaled with output current, output capacitance and load
current characteristics. The minimum soft-start capacitance is given by:
CSS > (COUT )(VOUT)(10 – 4)(RSENSE)
The minimum recommended soft-start capacitor of
CSS␣ =␣ 0.1µF will be sufficient for most applications.
Fault Conditions: Current Limit and Current Foldback
The LTC1735 current comparator has a maximum sense
voltage of 75mV resulting in a maximum MOSFET current
of 75mV/RSENSE.
The LTC1735 includes current foldback to help further
limit load current when the output is shorted to ground.
The foldback circuit is active even when the overload
shutdown latch described above is defeated. If the output
falls by more than half, then the maximum sense voltage
is progressively lowered from 75mV to 30mV. Under
short-circuit conditions with very low duty cycle, the
LTC1735 will begin cycle skipping in order to limit the
short-circuit current. In this situation the bottom MOSFET
will be conducting the peak current. The short-circuit
ripple current is determined by the minimum on-time
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∆IL(SC) = tON(MIN)VIN/L
The resulting short-circuit current is:
ISC =
30mV 1
+ ∆IL(SC)
RSENSE 2
The current foldback function is always active and is not
effected by the current latchoff function.
Fault Conditions: Output Overvoltage Protection
(Crowbar)
The output overvoltage crowbar is designed to blow a
system fuse in the input lead when the output of the
regulator rises much higher than nominal levels. This
condition causes huge currents to flow, much greater than
in normal operation. This feature is designed to protect
against a shorted top MOSFET; it does not protect against
a failure of the controller itself.
The comparator (OV in the Functional Diagram) detects
overvoltage faults greater than 7.5% above the nominal
output voltage. When this condition is sensed the top
MOSFET is turned off and the bottom MOSFET is forced
on. The bottom MOSFET remains on continuously for as
long as the 0V condition persists; if VOUT returns to a safe
level, normal operation automatically resumes.
Note that dynamically changing the output voltage may
cause overvoltage protection to be momentarily activated
during programmed output voltage decreases. This will
not cause permanent latchoff nor will it disrupt the desired
voltage change. With soft-latch overvoltage protection,
dynamically changing the output voltage is allowed and
the overvoltage protection tracks the newly programmed
output voltage, always protecting the load.
Minimum On-Time Considerations
Minimum on-time tON(MIN) is the smallest amount of time
that the LTC1735 is capable of turning the top MOSFET on
and off again. It is determined by internal timing delays and
the gate charge required to turn on the top MOSFET. Low
duty cycle applications may approach this minimum ontime limit and care should be taken to ensure that:
V
t ON(MIN) < OUT
VIN( f)
If the duty cycle falls below what can be accommodated by
the minimum on-time, the LTC1735 will begin to skip
cycles. The output voltage will continue to be regulated,
but the ripple current and voltage will increase.
The minimum on-time for the LTC1735 in a properly
configured application is generally less than 200ns. However, as the peak sense voltage decreases, the minimum
on-time gradually increases as shown in Figure 7. This is
of particular concern in forced continuous applications
with low ripple current at light loads. If the duty cycle drops
below the minimum on-time limit in this situation, a
significant amount of cycle skipping can occur with correspondingly larger current and voltage ripple.
If an application can operate close to the minimum ontime limit, an inductor must be chosen that is low enough
to provide sufficient ripple amplitude to meet the minimum on-time requirement. As a general rule, keep the
inductor ripple current equal or greater than 30% of
IOUT(MAX) at VIN(MAX).
250
200
MINIMUM ON-TIME (ns)
tON(MIN) of the LTC1735 (approximately 200ns), the input
voltage and inductor value:
150
100
50
0
0
10
20
30
40
∆IL /IOUT(MAX) (%)
1735 F07
Figure 7. Minimum On-Time vs ∆IL
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FCB Pin Operation
When the FCB pin drops below its 0.8V threshold, continuous mode operation is forced. In this case, the top and
bottom MOSFETs continue to be driven synchronously
regardless of the load on the main output. Burst Mode
operation is disabled and current reversal is allowed in the
inductor.
In addition to providing a logic input to force continuous
synchronous operation and external synchronization, the
FCB pin provides a means to regulate a flyback winding
output (refer to Figure 3a). During continuous mode,
current flows continuously in the transformer primary.
The secondary winding(s) draw current only when the
bottom, synchronous switch is on. When primary load
currents are low and/or the VIN/VOUT ratio is low, the
synchronous switch may not be on for a sufficient amount
of time to transfer power from the output capacitor to the
secondary load. Forced continuous operation will support
secondary windings provided there is sufficient synchronous switch duty factor. Thus, the FCB input pin removes
the requirement that power must be drawn from the
inductor primary in order to extract power from the
auxiliary windings. With the loop in continuous mode, the
auxiliary outputs may nominally be loaded without regard
to the primary output load.
The secondary output voltage VSEC is normally set as
shown in Figure␣ 3a by the turns ratio N of the transformer:
VSEC ≅ (N + 1)VOUT
However, if the controller goes into Burst Mode operation
and halts switching due to a light primary load current,
then VSEC will droop. An external resistive divider from
VSEC to the FCB pin sets a minimum voltage VSEC(MIN):
R4
VSEC(MIN) ≈ 0.8 V 1 +
R3
If VSEC drops below this level, the FCB voltage forces
continuous switching operation until VSEC is again above
its minimum.
In order to prevent erratic operation if no external connections are made to the FCB pin, the FCB pin has a 0.17µA
internal current source pulling the pin high. Remember to
include this current when choosing resistor values R3
and R4.
The internal LTC1735 oscillator can be synchronized to an
external oscillator by applying and clocking the FCB pin
with a signal above 1.5VP–P. When synchronized to an
external frequency, Burst Mode operation is disabled but
cycle skipping is allowed at low load currents since current
reversal is inhibited. The bottom gate will come on every
10 clock cycles to assure the bootstrap cap is kept refreshed. The rising edge of an external clock applied to the
FCB pin starts a new cycle. The FCB pin must not be driven
when the device is in shutdown (RUN/SS pin low).
The range of synchronization is from 0.9fO to 1.3fO, with
fO set by COSC. Attempting to synchronize to a higher
frequency than 1.3fO can result in inadequate slope compensation and cause loop instability with high duty cycles
(duty cycle > 50%). If loop instability is observed while
synchronized, additional slope compensation can be obtained by simply decreasing COSC.
The following table summarizes the possible states available on the FCB pin:
Table 1
FCB Pin
Condition
DC Voltage: 0V to 0.7V
Burst Disabled/Forced Continuous
Current Reversal Enabled
DC Voltage: ≥ 0.9V
Burst Mode Operation,
No Current Reversal
Feedback Resistors
Regulating a Secondary Winding
Ext Clock: (0V to VFCBSYNC)
(VFCBSYNC > 1.5V)
Burst Mode Operation Disabled
No Current Reversal
Efficiency Considerations
The percent efficiency of a switching regulator is equal to
the output power divided by the input power times 100%.
It is often useful to analyze individual losses to determine
what is limiting the efficiency and which change would
produce the most improvement. Percent efficiency can be
expressed as:
%Efficiency = 100% – (L1 + L2 + L3 + …)
where L1, L2, etc. are the individual losses as a percentage
of input power.
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Although all dissipative elements in the circuit produce
losses, 4 main sources usually account for most of the
losses in LTC1735 circuits: 1) VIN current, 2)␣ INTVCC
current, 3) I2R losses, 4) Topside MOSFET transition
losses.
4) Transition losses apply only to the topside MOSFET(s)
and only become significant when operating at high input
voltages (typically 12V or greater). Transition losses can
be estimated from:
1) The VIN current is the DC supply current given in the
electrical characteristics which excludes MOSFET driver
and control currents. VIN current results in a small (1µF) supply bypass capacitors. The
discharged bypass capacitors are effectively put in parallel
with COUT, causing a rapid drop in VOUT. No regulator can
alter its delivery of current quickly enough to prevent this
sudden step change in output voltage if the load switch
resistance is low and it is driven quickly. If the ratio of
CLOAD to COUT is greater than1:50, the switch rise time
should be controlled so that the load rise time is limited to
approximately (25)(CLOAD). Thus a 10µF capacitor would
require a 250µs rise time, limiting the charging current to
about 200mA.
Improve Transient Response and Reduce Output
Capacitance with Active Voltage Positioning
Fast load transient response, limited board space and low
cost are requirements of microprocessor power supplies.
Active voltage positioning improves transient response
and reduces the output capacitance required to power a
microprocessor where a typical load step can be from 0.2A
to 15A in 100ns or 15A to 0.2A in 100ns. The voltage at the
microprocessor must be held to about ±0.1V of nominal
in spite of these load current steps. Since the control loop
cannot respond this fast, the output capacitors must
supply the load current until the control loop can respond.
Capacitor ESR and ESL primarily determine the amount of
droop or overshoot in the output voltage. Normally, several capacitors in parallel are required to meet microprocessor transient requirements.
Active voltage positioning is a form of deregulation. It sets
the output voltage high for light loads and low for heavy
loads. When load current suddenly increases, the output
voltage starts from a level higher than nominal so the
output voltage can droop more and stay within the specified voltage range. When load current suddenly decreases
the output voltage starts at a level lower than nominal so
the output voltage can have more overshoot and stay
within the specified voltage range. Less output capacitance is required when voltage positioning is used because more voltage variation is allowed on the output
capacitors.
Active voltage positioning can be implemented using the
OPTI-LOOP architecture of the LTC1735 and two resistors
connected to the ITH pin. An input voltage offset is introduced when the error amplifier has to drive a resistive load.
This offset is limited to ±30mV at the input of the error
amplifier. The resulting change in output voltage is the
product of input offset and the feedback voltage divider
ratio.
Figure 8 shows a CPU-core-voltage regulator with active
voltage positioning. Resistors R1 and R4 force the input
voltage offset that adjusts the output voltage according to
the load current level. To select values for R1 and R4, first
determine the amount of output deregulation allowed. The
actual specification for a typical microprocessor allows
the output to vary ±0.112V. The LTC1735 reference accuracy is ±1%. Using 1% tolerance resistors, the total
feedback divider accuracy is about 1% because both
feedback resistors are close to the same value. The resulting setpoint accuracy is ±2% so the output transient
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R3
680k
R4
100k
R1
27k
C7
0.1µF
C1
39pF
1
C2
0.1µF
2
R2
100k
C3
100pF
3
C4
100pF
4
5
C5
47pF
6
C6
1000pF
COSC
TG
RUN/SS
BOOST
ITH
SW
U1
LTC1735
FCB
SGND
VOSENSE
VIN
INTVCC
BG
7
SENSE –
PGND
8
SENSE +
EXTVCC
16
M1
FDS6680A
15
C8
0.22µF
14
13
L1
1µH
D1
CMDSH-3
12
R5
0.003Ω
C9
1µF
10
+
C10
4.7µF
10V
C19
1µF
R6
10k
M2, M3
FDS6680A
×2
+
R7
11.5k
GND
VOUT
1.5V
15A
C11
330pF
D2
MBRS340
11
9
C9, C19: TAIYO YUDEN JMK107BJ105
C10: KEMET T494A475M010AS
C12 TO C14: TAIYO YUDEN GMK325F106
C15 TO C18: PANASONIC EEFUE0G181R
D1: CENTRAL SEMI CMDSH-3
D2: MOTOROLA MBRS340
L1: PANASONIC ETQP6F1R0SA
M1 TO M3: FAIRCHILD FDS6680A
R5: IRC LRF2512-01-R003-J
U1: LINEAR TECHNOLOGY LTC1735CS
C12 TO C14
10µF
35V
VIN
7.5V TO
24V
C15 TO
C18
180µF
4V
GND
5V (OPTIONAL)
1735 F08
Figure 8. CPU-Core-Voltage Regulator with Active Voltage Positioning
voltage cannot exceed ±0.082V. At VOUT = 1.5V, the
maximum output voltage change controlled by the ITH pin
would be:
Input Offset • VOUT
VREF
± 0.03V • 1.5
=
= ±56mV
0.8 V
∆VOSENSE =
With the optimum resistor values at the ITH pin, the output
voltage will swing from 1.55V at minimum load to 1.44V
at full load. At this output voltage, active voltage positioning provides an additional ±56mV to the allowable transient voltage on the output capacitors, a 68% improvement over the ±82mV allowed without active voltage
positioning.
The next step is to calculate the scale factor for VITH, the
ITH pin voltage. The VITH scale factor reflects the ITH pin
voltage required for a given load current. VITH controls the
peak sense resistor voltage, which represents the DC
output current plus one half of the peak-to-peak inductor
current. The no load to full load VITH range is from 0.3V to
2.4V, which controls the sense resistor voltage from 0V to
the ∆VSENSE(MAX) voltage of 75mV. The calculated VITH
scale factor with a 0.003Ω sense resistor is:
VITH Scale Factor =
=
VITH Range • Sense Re sistor Value
∆VSENSE(MAX)
(2.4V – 0.3V) • 0.003
= 0.084V/A
0.075V
VITH at any load current is:
∆I
VITH = IOUTDC + L • VITH Scale Factor
2
+ VITH Offset
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At full load current:
5A
VITH(MAX) = 15A + P−P • 0.084V/A + 0.3V
2
= 1.77 V
At minimum load current:
2A
VITH(MIN) = 0.2A + P−P • 0.084V/A + 0.3V
2
= 0.40 V
In this circuit, VITH changes from 0.40V at light load to
1.77V at full load, a 1.37V change. Notice that ∆IL, the
peak-to-peak inductor current, changes from light load to
full load. Increasing the DC inductor current decreases the
permeability of the inductor core material, which decreases the inductance and increases ∆IL. The amount of
inductance change is a function of the inductor design.
To create the ±30mV input offset, the gain of the error
amplifier must be limited. The desired gain is:
AV =
∆VITH
1.37 V
=
= 22.8
Input Offset Error 2(0.03V)
Connecting a resistor to the output of the transconductance
error amplifier will limit the voltage gain. The value of this
resistor is:
RITH =
AV
22.8
=
= 17.54k
Error Amplifier gm 1.3mS
To center the output voltage variation, VITH must be
centered so that no ITH pin current flows when the output
voltage is nominal. VITH(NOM) is the average voltage between VITH at maximum output current and minimum
output current:
The Thevenin equivalent of the gain limiting resistance
value of 17.54k is made up of a resistor R4 that sources
current into the ITH pin and resistor R1 that sinks current
to SGND.
To calculate the resistor values, first determine the ratio
between them:
k=
VINTVCC – VITH(NOM) 5.2V – 1.085V
=
= 3.79
1.085V
VITH(NOM)
VINTVCC is equal to VEXTVCC or 5.2V if EXTVCC is not used.
Resistor R4 is:
R4 = (k + 1) • RITH = (3.79 + 1) • 17.54 = 84.0k
Resistor R1 is:
R1 =
(k + 1) • RITH (3.79 + 1) • 17.54k
=
= 22.17k
k
3.79
Unfortunately, PCB noise can add to the voltage developed
across the sense resistor, R5, causing the ITH pin voltage
to be slightly higher than calculated for a given output
current. The amount of noise is proportional to the output
current level. This PCB noise does not present a serious
problem but it does change the effective value of R5 so the
calculated values of R1 and R4 may need to be adjusted to
achieve the required results. Since PCB noise is a function
of the layout, it will be the same on all boards with the same
layout.
Figures 9 and 10 show the transient response before and
after active voltage positioning is implemented. Notice
that active voltage positioning reduced the transient response from almost 200mVP-P to a little over 100mVP-P.
Refer to Design Solutions 10 for more information about
active voltage positioning.
VITH(MAX) – VITH(MIN)
+ VITH(MIN)
2
1.77 V – 0.40 V
=
+ 0.40 V = 1.085V
2
VITH(NOM) =
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VIN = 12V
VOUT = 1.5V
1.5V
100mV/DIV
OUTPUT
VOLTAGE
FIGURE 8 CIRCUIT
15A
LOAD
CURRENT
10A/DIV
0A
50µs/DIV
1735 F09
Figure 9. Normal Transient Response (Without R1, R4)
The network shown in Figure␣ 11 is the most straight
forward approach to protect a DC/DC converter from the
ravages of an automotive power line. The series diode
prevents current from flowing during reverse-battery,
while the transient suppressor clamps the input voltage
during load-dump. Note that the transient suppressor
should not conduct during double-battery operation, but
must still clamp the input voltage below breakdown of the
converter. Although the LTC1735 has a maximum input
voltage of 36V, most applications will be limited to 30V by
the MOSFET BVDSS.
50A IPK RATING
FIGURE 8 CIRCUIT
VIN = 12V
VOUT = 1.5V
1.582V
100mV/DIV 1.5V
12V
OUTPUT
VOLTAGE
VIN
TRANSIENT VOLTAGE
SUPPRESSOR
GENERAL INSTRUMENT
1.5KA24A
1.418V
15A
1735 F11
LOAD
CURRENT
10A/DIV
LTC1735
Figure 11. Plugging into the Cigarette Lighter
0A
50µs/DIV
1735 F10
Figure 10. Transient Response with Active Voltage Positioning
Automotive Considerations: Plugging into the
Cigarette Lighter
As battery-powered devices go mobile, there is a natural
interest in plugging into the cigarette lighter in order to
conserve or even recharge battery packs during operation.
But before you connect, be advised: you are plugging
into the supply from hell. The main power line in an
automobile is the source of a number of nasty potential
transients, including load-dump, reverse-battery and
double-battery.
Load-dump is the result of a loose battery cable. When the
cable breaks connection, the field collapse in the alternator
can cause a positive spike as high as 60V which takes
several hundred milliseconds to decay. Reverse-battery is
just what it says, while double-battery is a consequence of
tow-truck operators finding that a 24V jump start cranks
cold engines faster than 12V.
Design Example
As a design example, assume VIN = 12V(nominal),
VIN = 22V(max), VOUT = 1.8V, IMAX = 5A and f = 300kHz.
RSENSE and COSC can immediately be calculated:
RSENSE = 50mV/5A = 0.01Ω
COSC = 1.61(107)/(300kHz) – 11pF = 43pF
Assume a 3.3µH inductor and check the actual value of the
ripple current. The following equation is used:
V
V
∆IL = OUT 1 – OUT
( f)(L)
VIN
The highest value of the ripple current occurs at the
maximum input voltage:
∆IL =
1.8 V
1.8 V
1–
= 1.7 A
300kHz(3.3µH)
22V
The maximum ripple current is 33% of maximum output
current, which is about right.
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Next verify the minimum on-time of 200ns is not violated.
The minimum on-time occurs at maximum VIN:
tON(MIN) =
VOUT
VIN(MAX )f
=
1.8 V
= 273ns
22V(300kHz)
Since the output voltage is below 2.4V the output resistive
divider will need to be sized to not only set the output
voltage but also to absorb the sense pin current.
0.8 V
R1(MAX) = 24k
2.4V – VOUT
0.8 V
= 24k
= 32k
2.4V – 1.8 V
Choosing 1% resistors: R1 = 25.5k and R2 = 32.4k yields
an output voltage of 1.816V.
The power dissipation on the topside MOSFET can be
easily estimated. Choosing a Siliconix Si4412ADY results
in RDS(ON) = 0.035Ω, CRSS = 100pF. At maximum input
voltage with T(estimated) = 50°C:
PMAIN =
1.8 V 2
(5) [1+ (0.005)(50°C – 25°C)](0.035Ω)
22V
+1.7(22V ) (5A )(100pF )(300kHz)
2
= 204mW
Because the duty cycle of the bottom MOSFET is much
greater than the top, a larger MOSFET, Siliconix Si4410DY,
(RDS(ON) = 0.02Ω) is chosen. The power dissipation in the
bottom MOSFET, again assuming TA = 50°C, is:
22V – 1.8 V
2
5A ) (1.1)(0.02Ω)
(
22V
= 505mW
PSYNC =
Thanks to current foldback, the bottom MOSFET dissipation in short-circuit will be less than under full load
conditions.
CIN is chosen for an RMS current rating of at least 2.5A at
temperature. COUT is chosen with an ESR of 0.02Ω for low
output ripple. The output ripple in continuous mode will be
highest at the maximum input voltage. The worst-case
output voltage ripple due to ESR is approximately:
VORIPPLE = RESR( ∆IL ) = 0.02Ω(2.3A) = 46mVP −P
PC Board Layout Checklist
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of the
LTC1735. These items are also illustrated graphically in
the layout diagram of Figure␣ 12. Check the following in
your layout:
1) Are the signal and power grounds segregated? The
LTC1735 PGND pin should tie to the ground plane close to
the input capacitor(s). The SGND pin should then connect
to PGND, and all components that connect to SGND
should make a single point tie to the SGND pin. The
synchronous MOSFET source pins should connect to the
input capacitor(s) ground.
2) Does the VOSENSE pin connect directly to the feedback
resistors? The resistive divider R1, R2 must be connected
between the (+) plate of COUT and signal ground. The 47pF
to 100pF capacitor should be as close as possible to the
LTC1735. Be careful locating the feedback resistors too far
away from the LTC1735. The VOSENSE line should not be
routed close to any other nodes with high slew rates.
3) Are the SENSE – and SENSE + leads routed together with
minimum PC trace spacing? The filter capacitor between
SENSE + and SENSE – should be as close as possible to the
LTC1735. Ensure accurate current sensing with Kelvin
connections as shown in Figure 13. Series resistance can
be added to the SENSE lines to increase noise rejection.
4) Does the (+) terminal of CIN connect to the drain of the
topside MOSFET(s) as closely as possible? This capacitor
provides the AC current to the MOSFET(s).
5) Is the INTVCC decoupling capacitor connected closely
between INTVCC and the power ground pin? This capacitor carries the MOSFET driver peak currents. An additional 1µF ceramic capacitor placed immediately next to
1735fc
26
LTC1735
U
U
W
U
APPLICATIO S I FOR ATIO
the INTVCC and PGND pins can help improve noise
performance.
feedback pins. All of these nodes have very large and fast
moving signals and therefore should be kept on the
“output side” (Pin 9 to Pin 16) of the LTC1735 and occupy
minimum PC trace area.
6) Keep the switching node (SW), top gate node (TG) and
boost node (BOOST) away from sensitive small-signal
nodes, especially from the voltage and current sensing
+
COSC
M1
1
COSC
TG
16
CSS
BOOST
CIN
15
CC
3
ITH
SW
14
VIN
LTC1735
CC2
4
5
FCB
SGND
VIN
INTVCC
13
47pF
6
VOSENSE
BG
DB
12
11
CB
D1
+
4.7µF
M2
7
SENSE –
PGND
8
SENSE +
EXTVCC
10
–
1000pF
9
L1
–
R1
+
RC
RUN/SS
+
2
COUT
VOUT
RSENSE
R2
+
1735 F12
Figure 12. LTC1735 Layout Diagram
HIGH CURRENT PATH
1735 F13
SENSE + SENSE –
CURRENT SENSE
RESISTOR
(RSENSE)
Figure 13. Kelvin Sensing RSENSE
1735fc
27
LTC1735
U
TYPICAL APPLICATIO S
1.8V/5A Converter from Design Example with Burst Mode Operation Disabled
VIN
4.5V TO 22V
CIN
22µF
50V
CER
COSC
43pF
1
CSS
0.1µF
2
CC
470pF
RC
33k
3
CC2
220pF
COSC
TG
RUN/SS
BOOST
ITH
SW
16
M1
Si4412DY
CB
0.1µF
15
14
LTC1735
4
5
FCB
VIN
SGND
INTVCC
13
L1
3.3µH
DB
CMDSH-3
RSENSE
0.01Ω
R2
32.4k
1%
12
+
47pF
VOUT
1.8V
5A
COUT
150µF
6.3V
×2
PANASONIC SP
+
4.7µF
6
7
VOSENSE
BG
SENSE –
PGND
11
M2
Si4410DY
R1
25.5k
1%
MBRS140T3
10
SGND
1000pF
8
SENSE +
EXTVCC
9
OPTIONAL:
CONNECT TO 5V
COUT: PANASONIC EEFUEOG151R
CIN: MARCON THCR70LE1H226ZT
L1: PANASONIC ETQP6F3R3HFA
RSENSE: IRC LR 2010-01-R010F
1735 TA02
CPU Core Voltage Regulator for 2-Step Applications (VIN = 5V)
VIN
5V
100k*
COSC 39pF
1
CSS
0.1µF
2
RC
20k
CC
220pF
3
CC2
220pF
+
COSC
TG
RUN/SS
BOOST
ITH
SW
16
CB
0.22µF
15
14
LTC1735
4
5
FCB
SGND
VIN
INTVCC
13
DB
MBR0530
BG
11
1µF
7
SENSE –
PGND
RSENSE
0.004Ω
100pF
+
4.7µF
VOSENSE
L1
0.78µH
12
47pF
6
M1
FDS6680A
CIN
150µF
6.3V
×2
10
M2, M3
FDS6680A
×2
MBRD835L
R2
22.6k
1%
VOUT
1.5V
12A
+
R1
25.5k
1%
COUT
180µF
4V
×3
CO
47µF
10V
SGND
1000pF
8
SENSE +
EXTVCC
9
VIN
COUT: PANASONIC EEFUEOG181R
CIN: PANASONIC EEFUEOJ151R
CO: TAIYO YUDEN LMK550BJ476MM-B
L1: COILCRAFT 1705022P-781HC
RSENSE: IRC LRF 2512-01-R004-J
1735 TA03
*OPTIONAL TO DEFEAT OVERCURRENT LATCHOFF
1735fc
28
LTC1735
U
TYPICAL APPLICATIO S
Selectable Output Voltage Converter with Burst Mode Operation Disabled for CPU Power
VIN
4.5V TO 24V
0.1µF
1
NC
COSC
43pF
2
CSS
0.1µF
3
CC
RC 330pF
33k
4
CC2
47pF
NC
NC
COSC
TG
RUN/SS
BOOST
ITH
SW
20
4.7Ω
NC
19
M1
FDS6680A
CB
0.22µF
18
CIN: MARCON THCR70EIH226ZT
COUT: KEMET T510X447M006AS
L1: PANASONIC ETQP6F1R2HFA
RSENSE: IRC LRF2512-01-R004F
17
LTC1735F
5
6
FCB
VIN
SGND
INTVCC
CIN
22µF
×2
CER
L1
1.2µH
16
RSENSE
0.004Ω
DB
CMDSH-3
15
R2
10k
1%
47pF
+
47pF
VOUT
1.35V/1.60V
12A
COUT
470µF
6.3V
×3
KEMET
+
4.7µF
7
8
POWER GOOD
VOSENSE
BG
PGOOD
PGND
14
13
M2
FDS6680A
×2
MBRS340T3
1µF
CER
R3
33.2k
1%
47pF
R1
14.3k
1%
VN2222
9
SENSE –
EXTVCC
10
SENSE +
NC
12
OPTIONAL:
CONNECT TO 5V
ON: VOUT = 1.60V
OFF: VOUT = 1.35V
10k
1000pF
11
NC
SGND
10Ω
10Ω
1735 TA05
4V to 40V Input to 12V Flyback Converter
VIN
4V TO 40V
CMDSH-3
FMMT625
CIN
22µF
50V
×2
10k
6.2V
1
CSS
0.1µF
2
CC
2200pF RC
3.3k
3
CC2
100pF
COSC
TG
RUN/SS
BOOST
ITH
SW
16
6
7
R2
113k
1%
M1
IRL2910S
M2
Si4850EY
47Ω
MBRS1100
22Ω
15
14
RSENSE
0.004Ω
1nF
100V
R1
8.06k
1%
+
COUT
220µF
16V
×4
1nF
100V
LTC1735
4
5
FCB
SGND
VIN
INTVCC
13
12
47pF
6
VOSENSE
BG
7
SENSE –
PGND
8
SENSE +
EXTVCC
11
0.1µF
+
4.7µF
10
3300pF
100Ω
3
•
1M
COSC
150pF
VOUT
12V
3A
T1
10
•
VOUT
9
CIN: MARCON THCR70EIH226ZT
COUT: AVX TPSV227M016R0150
T1: COILTRONICS VP5-0155
RSENSE: IRC LRF2512-01-R004F
1735 TA07
1735fc
29
LTC1735
U
TYPICAL APPLICATIO S
5V/3.5A Converter with 12V/200mA Auxiliary Output
VIN
5.5V TO 28V
COSC
51pF
1
CSS
0.1µF
2
RC
33k
CIN
22µF
30V
OS-CON
22Ω
+
CC
470pF
3
CC2
220pF
COSC
TG
RUN/SS
BOOST
ITH
SW
16
M1
Si4802DY
CB
0.1µF
MBRS1100T3
14
+
LTC1735
4
5
FCB
VIN
SGND
INTVCC
1000pF
15
13
RSENSE
0.012Ω
DB
CMDSH-3
T1
1:1.8
10µH
12
+
100pF
CSEC
22µF
35V
AVX
R2
105k
1%
VOUT
5V
3.5A
COUT
100µF
10V
×3
AVX
+
4.7µF
6
7
VOSENSE
BG
SENSE –
PGND
11
M2
Si4802DY
R1
20k
1%
MBRS140T3
10
SGND
1000pF
8
SENSE
+
EXTVCC
9
CIN: SANYO OS-CON 30SC22M
COUT: AVX TPSD107M010R0068
T1: 1:8 DALE LPE6562-A262
100Ω
100Ω
10k
140k
1735 TA04
VOUT2
12V
120mA
UNREG
Dual Output 15W 3.3V/5V Power Supply
VIN
4.5V TO 28V
COSC
47pF
1
CSS
0.1µF
2
CC
470pF
TG
RUN/SS
BOOST
CB
0.1µF
CIN
22µF
50V
M1
Si4412DY
•
ITH
SW
0.01µF
CC2
100pF
FCB
VIN
SGND
INTVCC
4.7k
13
DB
CMDSH-3
•
T1A
MBRS140T3
RSENSE
0.01Ω
T1B
•
8 2
7
R2
62.6k
1%
12
+
100pF
VOSENSE
BG
7
SENSE –
PGND
8
SENSE +
EXTVCC
COUT2
100µF
10V
×2
VOUT1
3.3V
2.5A
+
4.7µF
6
+
VOUT2
5V
1.5A
14
1
5
6
M3
Si4412DY
15
LTC1735
4
T1C
3
CMDSH-3
3
RC
33k
COSC
16
11
M2
Si4412DY
R1
20k
1%
MBRS140T3
COUT1
100µF
10V
×2
10
1000pF
9
SGND
VOUT2
1735 TA08
CIN: MARCON THCR70EIH226ZT
COUT1, 2: AVX TPSD107M010R0065
T1: BI TECHNOLOGIES HM00-93839
RSENSE: IRC LRF2512-01-R010 F
1735fc
30
LTC1735
U
PACKAGE DESCRIPTION
F Package
20-Lead Plastic TSSOP (4.4mm)
(Reference LTC DWG # 05-08-1650)
1.10
(.0433)
MAX
4.30 – 4.50**
(.169 – .177)
6.40 – 6.60*
(.252 – .260)
20 19 18 17 16 15 14 13 12 11
1.05 ±0.10
0° – 8°
0.09 – 0.20
(.0036 – .0079)
0.65
(.0256)
BSC
0.45 – 0.75
(.018 – .030)
0.195 – 0.30
(.0077 – .0118)
6.40
6.60 ±0.10
BSC
0.05 – 0.15
(.002 – .006)
NOTE:
1. CONTROLLING DIMENSION: MILLIMETERS *DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED .152mm (.006") PER SIDE
MILLIMETERS
2. DIMENSIONS ARE IN
(INCHES)
**DIMENSIONS DO NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED .254mm (.010") PER SIDE
3. DRAWING NOT TO SCALE
4.50 ±0.10
F20 TSSOP 0502
0.45 ±0.05
1 2 3 4 5 6 7 8 9 10
0.65 TYP
RECOMMENDED SOLDER PAD LAYOUT
GN Package
16-Lead Plastic SSOP (Narrow .150 Inch)
(Reference LTC DWG # 05-08-1641)
.189 – .196*
(4.801 – 4.978)
.015 ± .004
× 45°
(0.38 ± 0.10)
.053 – .068
(1.351 – 1.727)
.007 – .0098
(0.178 – 0.249)
.009
(0.229)
REF
16 15 14 13 12 11 10 9
.004 – .0098
(0.102 – 0.249)
.045 ±.005
0° – 8° TYP
.229 – .244
.0250
(5.817 – 6.198)
(0.635)
BSC
NOTE:
*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
1. CONTROLLING DIMENSION: INCHES
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
INCHES
2. DIMENSIONS ARE IN
(MILLIMETERS) **DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
3. DRAWING NOT TO SCALE
.016 – .050
(0.406 – 1.270)
.150 – .157**
.254 MIN
(3.810 – 3.988)
.008 – .012
(0.203 – 0.305)
.150 – .165
GN16 (SSOP) 0502
1
2 3
5 6
4
7
.0165 ± .0015
8
.0250 TYP
RECOMMENDED SOLDER PAD LAYOUT
S Package
16-Lead Plastic Small Outline (Narrow .150 Inch)
(Reference LTC DWG # 05-08-1610)
.386 – .394
(9.804 – 10.008)
NOTE 3
.045 ±.005
.050 BSC
16
N
15
14
13
12
11
10
9
N
.245
MIN
.160 ±.005
.150 – .157
(3.810 – 3.988)
NOTE 3
.228 – .244
(5.791 – 6.197)
1
.030 ±.005
TYP
2
3
N/2
N/2
RECOMMENDED SOLDER PAD LAYOUT
1
.010 – .020
× 45°
(0.254 – 0.508)
.008 – .010
(0.203 – 0.254)
2
3
4
5
6
7
.053 – .069
(1.346 – 1.752)
.004 – .010
(0.101 – 0.254)
0° – 8° TYP
.016 – .050
(0.406 – 1.270)
.014 – .019
(0.355 – 0.483)
TYP
8
.050
(1.270)
BSC
S16 0502
NOTE:
1. DIMENSIONS IN
INCHES
3. THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS.
(MILLIMETERS)
MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED .006" (0.15mm)
2. DRAWING NOT TO SCALE
1735fc
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
31
LTC1735
U
TYPICAL APPLICATIO
3.3V to 2.5V/5A Converter with External Clock Synchronization Operating at 500kHz
VIN
3.3V
5V 0.1µF
COSC
20pF
+
1
CSS
0.1µF
2
CC
330pF
3
RC
33k
COSC
TG
RUN/SS
BOOST
ITH
SW
16
CB
0.1µF
M1
Si4410DY
15
CIN: SANYO OS-CON 10SL100M
COUT: AVX TPSD107M010R0065
L1: COILCRAFT DO3316P-152
RSENSE: IRC LR2010-01-R010-F
14
LTC1735
CC2
51pF
EXT
CLOCK
500kHz
4
5
FCB
VIN
SGND
INTVCC
CIN
100µF
10V
OS-CON
13
DB
CMDSH-3
L1
1.5µH
12
RSENSE
0.01Ω
47pF
+
47pF
R2
43.2k
1%
VOUT
2.5V
5A
COUT
100µF
10V
AVX
×3
+
4.7µF
6
VOSENSE
BG
7
SENSE –
PGND
8
SENSE +
11
10
M2
Si4410DY
R1
20k
1%
MBRS140T3
SGND
1000pF
EXTVCC
9
1735 TA06
RELATED PARTS
PART NUMBER
LTC1530
LTC1628/LTC3728
LTC1629/LTC3729
LTC1702
LTC1708-PG
LTC1736
LTC1778
LTC1929/
LTC1929-PG
LTC3711
LTC3729
LTC3730
LTC3732
LTC3778
DESCRIPTION
High Power Step-Down Synchronous DC/DC Controller
in SO-8
2-Phase, Dual Output Synchronous Step-Down
DC/DC Controllers
20A to 200A PolyPhase Synchronous Controllers
No RSENSETM 2-Phase Dual Synchronous Step-Down
Controller
2-Phase, Dual Synchronous Controller with Mobile VID
High Efficiency Synchronous Controller with 5-Bit Mobile
VID Control
No RSENSE Current Mode Synchronous Step-Down
Controller
2-Phase Synchronous Controllers
No RSENSE Current Mode Synchronous Step-Down
Controller with Digital 5-Bit Interface
20A to 200A, 550kHz PolyPhase Synchronous Controller
IMVP III 3-Phase Synchronous Controller
VRM 9.0/9.1 3-Phase DC/DC Synchronous Step-Down
Controller
Optional RSENSE Current Mode Synchronous Step-Down
Controller
Low VIN High Power Synchronous Controller
4A Multichemistry Multicell Battery Charger
COMMENTS
High Efficiency 5V to 3.3V Conversion at Up to 15A
Reduces CIN and COUT, Power Good Output Signal, Synchronizable,
3.5V ≤ VIN ≤ 36V, IOUT up to 20A, 0.8V ≤ VOUT ≤ 5V
Expandable from 2-Phase to 12-Phase, Uses All
Surface Mount Components, No Heat Sink, VIN up to 36V
550kHz, No Sense Resistor
3.5V ≤ VIN ≤ 36V, VID Sets VOUT1, PGOOD
Output Fault Protection, 24-Pin SSOP,
3.5V ≤ VIN ≤ 36V
Up to 97% Efficiency, 4V ≤ VIN ≤ 36V, 0.8V ≤ VOUT ≤ (0.9)(VIN),
IOUT up to 20A
Up to 42A, Uses All Surface Mount Components,
No Heat Sinks, 3.5V ≤ VIN ≤ 36V
Up to 97% Efficiency, Ideal for Pentium® III Processors,
0.925V ≤ VOUT ≤ 2V, 4V ≤ VIN ≤ 36V, IOUT up to 20A
Expandable from 2-Phase to 12-Phase, Uses all Surface Mount
Components, VIN up to 36V
IOUT Up to 60A, 0.6V ≤ VOUT ≤ 1.75V, Integrated MOSFET Drivers
1.1V ≤ VOUT ≤ 1.85V, 4.5V ≤ VIN ≤ 32V, SSOP-36
4V ≤ VIN ≤ 36V, Adjustable Frequency Up to 1.2MHz, TSSOP-20
VOUT ≥ 0.6V, IOUT ≤ 20A, 3V ≤ VIN ≤ 8V
NiCd, NiMH, Lead Acid, Li-Ion Batteries; 6V ≤ VIN ≤ 28V;
1.19V ≤ VOUT ≤ 28V
No RSENSE is a trademark of Linear Technology Corporation. Pentium is a registered trademark of Intel Corporation.
LTC3832
LTC4008
1735fc
32 Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com
LT/TP 0104 REV C 1K • PRINTED IN USA
LINEAR TECHNOLOGY CORPORATION 1998
工商网监
湘ICP备2023018690号
