AD
AD
S83
S8
17
317
ADS8317
www.ti.com .................................................................................................................................................... SBAS356D – JUNE 2007 – REVISED OCTOBER 2009
16-Bit, High-Speed, 2.7V to 5.5V microPower Sampling
ANALOG-TO-DIGITAL CONVERTER
Check for Samples: ADS8317
FEATURES
DESCRIPTION
•
The ADS8317 is a 16-bit, sampling, analog-to-digital
(A/D) converter specified for a supply voltage range
from 2.7V to 5.5V. It requires very little power, even
when operating at the full data rate. At lower data
rates, the high speed of the device enables it to
spend most of its time in the power-down mode. For
example, the average power dissipation is less than
0.2mW at a 10kHz data rate.
1
23
•
•
•
•
•
•
16 Bits No Missing Codes (Full-Supply Range,
High or Low Grade)
Very Low Noise: 5LSBPP
Excellent Linearity:
±0.8LSB typ, ±1.5LSB max INL
+0.7LSB typ, +1.25LSB max DNL
±1mV max Offset
±16LSB typ Gain Error
microPower:
10mW at 5V, 250kHz
4mW at 2.7V, 200kHz
2mW at 2.7V, 100kHz
0.2mW at 2.7V, 10kHz
Packages: MSOP-8, SON-8
Pin-Compatible with the ADS8321
Serial ( SPI™/SSI) Interface
The ADS8317 offers excellent linearity and very low
noise and distortion. It also features a synchronous
serial (SPI/SSI-compatible) interface and a differential
input. The reference voltage can be set to any level
within the range of 0.1V to VDD/2.
Low power and small size make the ADS8317 ideal
for portable and battery-operated systems. It is also
an excellent fit for remote data-acquisition modules,
simultaneous multichannel systems, and isolated
data acquisition. The ADS8317 is available in
MSOP-8 and SON-8 packages. The SON package
size is the same as a 3x3 QFN package.
APPLICATIONS
•
•
•
•
•
•
•
Battery-Operated Systems
Remote Data Acquisition
Isolated Data Acquisition
Simultaneous Sampling, Multichannel
Systems
Industrial Controls
Robotics
Vibration Analysis
SAR
REF
ADS8317
DOUT
+IN
Serial
Interface
CDAC
−IN
DCLOCK
S/H Amp
Comparator
CS/SHDN
1
2
3
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas
Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
SPI is a trademark of Motorola, Inc.
All other trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2007–2009, Texas Instruments Incorporated
ADS8317
SBAS356D – JUNE 2007 – REVISED OCTOBER 2009 .................................................................................................................................................... www.ti.com
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more
susceptible to damage because very small parametric changes could cause the device not to meet its published specifications.
ORDERING INFORMATION (1)
PRODUCT
MAXIMUM
INTEGRAL
LINEARITY
ERROR
(LSB) (2)
NO
MISSING
CODES
ERROR
(LSB)
PACKAGELEAD
PACKAGE
DESIGNATOR
SPECIFIED
TEMPERATURE
RANGE
PACKAGE
MARKING
ADS8317I
±2
16
MSOP-8
DGK
–40°C to +85°C
D17
ADS8317IB
±1.5
16
MSOP-8
DGK
–40°C to +85°C
D17
ADS8317I
±2
16
SON-8
DRB
–40°C to +85°C
D17
ADS8317IB
±1.5
16
SON-8
DRB
–40°C to +85°C
D17
(1)
(2)
ORDERING
NUMBER
TRANSPORT
MEDIA,
QUANTITY
ADS8317IDGKT
Tape and Reel, 250
ADS8317IDGKR
Tape and Reel, 2500
ADS8317IBDGKT
Tape and Reel, 250
ADS8317IBDGKR
Tape and Reel, 2500
ADS8317IDRBT
Tape and Reel, 250
ADS8317IDRBR
Tape and Reel, 2500
ADS8317IBDRBT
Tape and Reel, 250
ADS8317IBDRBR
Tape and Reel, 2500
For the most current package and ordering information, see the Package Option Addendum located at the end of this data sheet, or see
the TI website at www.ti.com.
Maximum Integral Linearity Error specifies a 5V power supply and 2.5V reference voltage.
ABSOLUTE MAXIMUM RATINGS (1)
Over operating free-air temperature range (unless otherwise noted).
ADS8317
UNIT
–0.3 to +7
V
–0.3 to VDD + 0.3
V
–0.3 to VDD + 0.3
V
Supply voltage, VDD to GND
Analog input voltage (2)
Reference input voltage
(2)
Digital input voltage (2)
–0.3 to VDD + 0.3
V
–20 to +20
mA
Input current to any pin except supply
Power dissipation
See Dissipation Ratings Table
Operating virtual junction temperature range, TJ
–40 to +150
°C
Operating free-air temperature range, TA
–40 to +85
°C
Storage temperature range, TSTG
–65 to +150
°C
(1)
(2)
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, and functional operation of the device at these or any other conditions beyond those indicated under Recommended Operating
Conditions is not implied. Exposure to absolute-maximum rated conditions for extended periods may affect device reliability.
All voltage values are with respect to ground terminal.
DISSIPATION RATINGS
2
PACKAGE
R θ JC
R θ JA
DERATING
FACTOR ABOVE
TA = +25°C
DGK
+39.1°C/W
+206.3°C/W
4.847mW/°C
606mW
388mW
315mW
DRB
+5°C/W
+45.8°C/W
3.7mW/°C
370mW
204mW
148mW
TA ≤ +25°C
POWER RATING
TA = +70°C
POWER RATING
TA = +85°C
POWER RATING
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Copyright © 2007–2009, Texas Instruments Incorporated
Product Folder Link(s): ADS8317
ADS8317
www.ti.com .................................................................................................................................................... SBAS356D – JUNE 2007 – REVISED OCTOBER 2009
RECOMMENDED OPERATING CONDITIONS
MIN
Supply voltage, GND to VDD
MAX
UNIT
3.6
V
5.5
V
1
VDD/2
V
–IN to GND
–0.2
VDD + 0.2
V
+IN to GND
–0.2
VDD + 0.2
V
+IN – (–IN)
–VREF
+VREF
V
–40
+125
°C
Low-voltage levels
2.7
5V logic levels
4.5
Reference input voltage
Analog input voltage
Operating junction temperature, TJ
TYP
5.0
ELECTRICAL CHARACTERISTICS: VDD = +5V
At –40°C to +85°C, VREF = +2.5V, –IN = +2.5V, fSAMPLE = 250kHz, and fCLK = 24 × fSAMPLE, unless otherwise noted.
ADS8317I
PARAMETER
TEST CONDITIONS
MIN
TYP
ADS8317IB
MAX
MIN
VREF
–VREF
VDD + 0.1
–0.1
TYP
MAX
UNIT
VREF
V
ANALOG INPUT
Full-scale range
FSR +IN – (–IN)
Absolute input range
Input resistance
–VREF
+IN
RON
Input capacitance
–0.1
Hold
5
V
100
50
GΩ
Ω
Sampling
50
During sampling
24
24
pF
nA
Input leakage current
100
±50
±50
+IN to –IN, during sampling
20
20
pF
FSBW fS sinewave, SINAD = 60dB
500
500
kHz
Differential input capacitance
Full-power bandwidth
VDD + 0.1
5
DC ACCURACY
Resolution
No missing codes
Integral linearity error
16
NMC
16
16
16
±1.5
+2
Bits
16
16
Bits
–1.5
±0.8
+1.5
LSB
LSB
INL
–2
Differential linearity error
DNL
–1
±1
+2
–1
+0.7,–0.5
+1.25
Offset error
VOS
–2
±0.75
+2
–1
±0.5
+1
Offset error drift
Gain error
Gain error drift
TCVOS
GERR
±3
Positive
–32
±16
+32
–32
±16
+32
Negative
–32
±16
+32
–32
±16
+32
TCGERR
±0.1
Bipolar zero error
–2
±0.75
mV
μV/°C
±3
±0.1
+2
–1
±0.5
LSB
LSB
ppm/°C
+1
mV
Bipolar zero error drift
±3
±3
μV/°C
Noise
50
50
μVRMS
1
1
LSB
Power-supply rejection
PSRR 4.75V ≤ VDD ≤ 5.25V
SAMPLING DYNAMICS
Conversion time
(16 DCLOCKs)
Acquisition time
(4.5 DCLOCKs)
tCONV 24kHz ≤ fCLK ≤ 6.0MHz
tAQ fCLK = 6.0MHz
2.667
0.75
Throughput rate
(22 DCLOCKs)
Clock frequency
666.7
2.667
6.0
0.024
250
kSPS
6.0
MHz
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Product Folder Link(s): ADS8317
μs
μs
0.75
250
0.024
666.7
3
ADS8317
SBAS356D – JUNE 2007 – REVISED OCTOBER 2009 .................................................................................................................................................... www.ti.com
ELECTRICAL CHARACTERISTICS: VDD = +5V (continued)
At –40°C to +85°C, VREF = +2.5V, –IN = +2.5V, fSAMPLE = 250kHz, and fCLK = 24 × fSAMPLE, unless otherwise noted.
ADS8317I
PARAMETER
TEST CONDITIONS
MIN
TYP
ADS8317IB
MAX
MIN
TYP
MAX
UNIT
AC ACCURACY
Total harmonic distortion
Spurious-free dynamic range
Signal-to-noise ratio
Signal-to-noise + distortion
Effective number of bits
THD
SFDR
SNR
SINAD
ENOB
5VPP sinewave at 2kHz
–102
–106
dB
5VPP sinewave at 10kHz
–100
–104
dB
5VPP sinewave at 2kHz
106
110
dB
5VPP sinewave at 10kHz
104
109
dB
5VPP sinewave at 2kHz
89.6
90
dB
5VPP sinewave at 10kHz
89.6
90
dB
5VPP sinewave at 2kHz
89.5
89.9
dB
5VPP sinewave at 10kHz
89.4
89.8
dB
5VPP sinewave at 2kHz
14.57
14.65
Bits
5VPP sinewave at 10kHz
14.56
14.63
Bits
VOLTAGE REFERENCE INPUT
Reference voltage
0.5
CS = GND, fSAMPLE = 0Hz
Reference input resistance
CS = VDD
Reference input capacitance
Reference input current
VDD/2
0.5
5
VDD/2
V
5
GΩ
5
5
GΩ
24
24
pF
fS = 250kHz
35
52
35
52
μA
fS = 200kHz
25
38
25
38
μA
fS = 100kHz
10
15
10
15
μA
fS = 10kHz
1
2
1
2
μA
CS = VDD
0.1
μA
0.1
DIGITAL INPUTS (1)
Logic family
CMOS
CMOS
High-level input voltage
VIH
0.7 × VDD
VDD + 0.3
0.7 × VDD
VDD + 0.3
Low-level input voltage
VIL
–0.3
0.3 × VDD
–0.3
0.3 × VDD
V
Input current
IIN VI = VDD or GND
–50
+50
–50
+50
nA
Input capacitance
CI
5
5
V
pF
DIGITAL OUTPUTS (1)
Logic family
CMOS
High-level output voltage
VOH VDD = 4.5V, IOH = –100μA
Low-level output voltage
VOL VDD = 4.5V, IOL = 100μA
High-impedance state output
current
IOZ CS = VDD, VI = VDD or GND
Output capacitance
CO
Load capacitance
CL
Data format
(1)
4
CMOS
4.44
4.44
V
0.5
–50
+50
5
–50
0.5
V
+50
nA
5
30
Binary twos complement
pF
30
pF
Binary twos complement
Applies for 5.0V nominal supply: VDD (min) = 4.5V and VDD (max) = 5.5V.
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Product Folder Link(s): ADS8317
ADS8317
www.ti.com .................................................................................................................................................... SBAS356D – JUNE 2007 – REVISED OCTOBER 2009
ELECTRICAL CHARACTERISTICS: VDD = +2.7V
At –40°C to +85°C, VREF = +1.25V, –IN = 1.25V, fSAMPLE = 200kHz, and fCLK = 24 × fSAMPLE, unless otherwise noted.
ADS8317I
PARAMETER
TEST CONDITIONS
MIN
TYP
ADS8317IB
MAX
MIN
–VREF
VREF
–0.1
VDD + 0.1
TYP
MAX
UNIT
–VREF
VREF
V
–0.1
VDD + 0.1
ANALOG INPUT
Full-scale range
FSR +IN – (–IN)
Absolute input range
Input resistance
+IN
RON
Input capacitance
Hold
5
Sampling
100
During sampling
Input leakage current
150
100
GΩ
Ω
150
24
24
pF
nA
±50
±50
+IN to –IN, during sampling
20
20
pF
FSBW fS sinewave, SINAD = 60dB
1000
1000
kHz
Differential input capacitance
Full-power bandwidth
V
5
DC ACCURACY
Resolution
No missing codes
Integral linearity error
Differential linearity error
Offset error
Offset error drift
Gain error
Gain error drift
16
NMC
INL
16
±2
+3
–2
–1 +1.5,–1
+2.5
+2
–3
DNL
VOS
–2
TCVOS
GERR
±1
Positive
–32
Negative
–32
TCGERR
16
Bits
±1.5
+2
LSB
–1
±1
+2
LSB
–1
±0.5
+1
–2
±16
+32
–32
±16
+32
–32
Noise
PSRR 2.7V ≤ VDD ≤ 3.6V
±0.8
mV
μV/°C
±0.4
±0.15
Bipolar zero error drift
Bits
16
±0.4
Bipolar zero error
Power-supply rejection
16
16
±16
+32
±16
+32
±0.15
+2
–1
±0.4
LSB
LSB
ppm/°C
+1
mV
μV/°C
±0.2
±0.2
50
50
μVRMS
1
1
LSB
SAMPLING DYNAMICS
Conversion time (16 DCLOCKs)
Acquisition time (4.5 DCLOCKs)
tCONV 24kHz ≤ fCLK ≤ 4.8MHz
tAQ fCLK = 4.8MHz
3.333
666.7
0.9375
3.333
666.7
Throughput rate (22 DCLOCKs)
200
Clock frequency
0.024
4.8
μs
μs
0.9375
0.024
200
kSPS
4.8
MHz
AC ACCURACY
Total harmonic distortion
Spurious-free dynamic range
Signal-to-noise ratio
THD
SFDR
SNR
Signal-to-noise + distortion
SINAD
Effective number of bits
ENOB
2.5VPP sinewave at 2kHz
–104
–107
dB
2.5VPP sinewave at 10kHz
–101
–106
dB
2.5VPP sinewave at 2kHz
106
108
dB
2.5VPP sinewave at 10kHz
104
107
dB
2.5VPP sinewave at 2kHz
84.8
85
dB
2.5VPP sinewave at 10kHz
84.8
85
dB
2.5VPP sinewave at 2kHz
84.7
84.9
dB
2.5VPP sinewave at 10kHz
84.7
84.8
dB
2.5VPP sinewave at 2kHz
13.77
13.8
Bits
2.5VPP sinewave at 10kHz
13.77
13.79
Bits
VOLTAGE REFERENCE INPUT
Reference voltage
Reference input resistance
1
CS = GND, fSAMPLE = 0Hz
CS = VDD
Reference input capacitance
fS = 200kHz
Reference input current
VDD/2
5
1
VDD/2
V
5
kΩ
5
5
GΩ
20
20
9
14
pF
9
14
μA
3
5
3
5
μA
fS = 10kHz
0.5
1
0.5
1
μA
CS = VDD
0.1
fS = 100kHz
0.1
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μA
5
ADS8317
SBAS356D – JUNE 2007 – REVISED OCTOBER 2009 .................................................................................................................................................... www.ti.com
ELECTRICAL CHARACTERISTICS: VDD = +2.7V (continued)
At –40°C to +85°C, VREF = +1.25V, –IN = 1.25V, fSAMPLE = 200kHz, and fCLK = 24 × fSAMPLE, unless otherwise noted.
ADS8317I
PARAMETER
TEST CONDITIONS
MIN
ADS8317IB
TYP
MAX
MIN
TYP
MAX
UNIT
DIGITAL INPUTS (1)
Logic family
LVCMOS
LVCMOS
High-level input voltage
VIH VDD = 3.6V
2
VDD + 0.3
2
VDD + 0.3
Low-level input voltage
VIL VDD = 2.7V
–0.3
0.8
–0.3
0.3 × VDD
V
Input current
IIN VI = VDD or GND
–50
+50
–50
+50
nA
Input capacitance
CI
5
5
V
pF
DIGITAL OUTPUTS (1)
Logic family
LVCMOS
High-level output voltage
VOH VDD = 2.7V, IOH = –100μA
Low-level output voltage
VOL VDD = 2.7V, IOL = 100μA
High-impedance state output
current
IOZ CS = VDD, VI = VDD or GND
Output capacitance
CO
Load capacitance
CL
VDD – 0.2
V
0.2
–50
+50
–50
5
0.2
V
+50
nA
5
pF
30
Data format
(1)
LVCMOS
VDD – 0.2
Binary twos complement
30
pF
Binary twos complement
Applies for 5.0V nominal supply: VDD (min) = 2.7V and VDD (max) = 3.6V.
ELECTRICAL CHARACTERISTICS: GENERAL
At –40°C to +85°C, –IN = GND, and fDCLOCK = 24 × fSAMPLE, unless otherwise noted.
ADS8317I
PARAMETER
TEST CONDITIONS
MIN
TYP
ADS8317IB
MAX
MIN
TYP
MAX
UNIT
ANALOG INPUT
Power supply
VDD
Low-voltage levels
2.7
3.6
2.7
3.6
V
5V logic levels
4.5
5.5
4.5
5.5
V
VDD = 2.7V, fS = 10kHz,
fDCLOCK = 4.8MHz
Operating supply current
Power-down supply current
Power dissipation
Power dissipation in power-down
6
IDD
IDD
0.065
0.085
0.065
0.085
mA
VDD = 2.7V, fS = 100kHz,
fDCLOCK = 4.8MHz
0.7
1.0
0.7
1.0
mA
VDD = 2.7V, fS = 200kHz,
fDCLOCK = 4.8MHz
1.4
2.0
1.4
2.0
mA
VDD = 5V, fS = 200kHz,
fDCLOCK = 6MHz
1.5
2.5
1.5
2.5
mA
VDD = 5V, fS = 250kHz,
fDCLOCK = 6MHz
2.0
3.0
2.0
3.0
mA
VDD = 2.7V
0.1
0.1
μA
VDD = 5V
0.2
0.2
μA
VDD = 2.7V, fS = 10kHz,
fDCLOCK = 4.8MHz
0.18
0.23
0.18
0.23
mW
VDD = 2.7V, fS = 100kHz,
fDCLOCK = 4.8MHz
1.9
2.7
1.9
2.7
mW
VDD = 2.7V, fS = 200kHz,
fDCLOCK = 4.8MHz
3.8
5.4
3.8
5.4
mW
VDD = 5V, fS = 200kHz,
fDCLOCK = 6MHz
7.5
12.5
7.5
12.5
mW
VDD = 5V, fS = 250kHz,
fDCLOCK = 6MHz
10
15
10
15
mW
VDD = 2.7V, CS = VDD
0.3
0.3
μW
VDD = 5V, CS = VDD
0.6
0.6
μW
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Product Folder Link(s): ADS8317
ADS8317
www.ti.com .................................................................................................................................................... SBAS356D – JUNE 2007 – REVISED OCTOBER 2009
PIN CONFIGURATION
DGK PACKAGE
MSOP-8
(TOP VIEW)
REF
1
+IN
2
8
VDD
7
DCLOCK
ADS8317
-IN
3
6
DOUT
GND
4
5
CS/SHDN
DRB PACKAGE
SON-8
(TOP VIEW)
REF
1
+IN
2
8
VDD
7
DCLOCK
6
DOUT
5
CS/SHDN
ADS8317
(1)
-IN
3
GND
4
(Thermal Pad)
The DRB package thermal pad must be soldered to the printed circuit board for proper thermal and mechanical
performance.
TERMINAL FUNCTIONS
TERMINAL
NAME
NO.
I/O
DESCRIPTION
REF
1
Analog input
Reference input
+IN
2
Analog input
Noninverting analog input
–IN
3
Analog input
Inverting analog input
GND
4
Power-supply connection
CS/SHDN
5
Digital input
DOUT
6
Digital output
DCLOCK
7
Digital input
VDD
8
Power-supply connection
Ground
Chip select when low; Shutdown mode when high.
Serial output data word
Data clock synchronizes the serial data transfer and determines conversion speed.
Power supply
Equivalent Input Circuits (VDD = 5.0V)
VDD
VDD
RON
50W
C(SAMPLE)
24pF
ANALOG IN
GND
Diode Turn-On Voltage: 0.35V
Equivalent Analog Input Circuit
VDD
RON
50W
REF
GND
Equivalent Reference Input Circuit
24pF
I/O
GND
Equivalent Digital Input/Output Circuit
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7
ADS8317
SBAS356D – JUNE 2007 – REVISED OCTOBER 2009 .................................................................................................................................................... www.ti.com
TIMING INFORMATION
tCYC
CS/SHDN
Sample
Power Down
Conversion
tSUCS
DCLOCK
tCSD
Use positive clock edge for data transfer
Hi-Z
DOUT
0
tSMPL
B7
B15 B14 B13 B12 B11 B10 B9 B8
(MSB)
tCONV
B6
B5 B4
B3
B2
Hi-Z
(1)
B1 B0
(LSB)
NOTE: (1) A minimum of 22 clock cycles are required for 16-bit conversion; 24 clock cycles are shown.
If CS remains low at the end of conversion, a new data stream is shifted out with LSB-first data followed by zeroes indefinitely.
tCYC
CS/SHDN
tSUCS
Power Down
DCLOCK
tCSD
Hi-Z
DOUT
0
tSMPL
B15 B14 B6
(MSB)
B5
B4
B3
tCONV
B2
B1
B0
B1
B2
B3
B4
B5
(LSB)
B0
(2)
Hi-Z
B11 B12 B13 B14 B15
(MSB)
NOTE: (2) After completing the data transfer, if further clocks are applied with CS low, the A/D converter will output zeroes indefinitely.
1.4V
3kW
DOUT
90%
DOUT
10%
Test Point
tr
100pF
CLOAD
tf
Voltage Waveforms for DOUT Rise and Fall Times, tr, tf
Load Circuit for tdDO, tr, and tf
Test Point
DCLOCK
VDD
DOUT
tdDO
tdis Waveform 2, ten
3kW
tdis Waveform 1
100pF
CLOAD
DOUT
thDO
Load Circuit for tdis and ten
Voltage Waveforms for DOUT Delay Times, tdDO
90%
CS/SHDN
DOUT
Waveform 1(3)
CS/SHDN
90%
DCLOCK
1
4
5
tdis
DOUT
Waveform 2(4)
10%
DOUT
B15
ten
Voltage Waveforms for tdis
Voltage Waveforms for ten
NOTES: (3) Waveform 1 is for an output with internal conditions such that
the output is high unless disabled by the output control.
(4) Waveform 2 is for an output with internal conditions such that
the output is low unless disabled by the output control.
Figure 1. Timing Diagrams
8
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TIMING INFORMATION (continued)
Timing Characteristics
SYMBOL
DESCRIPTION
MIN
TYP
4.5
MAX
5.0
UNIT
tSMPL
Analog input sample time
tCONV
Conversion time
tCYC
Complete cycle time
tCSD
CS falling to DCLOCK low
tSUCS
CS falling to DCLOCK rising
tHDO
DCLOCK falling to current DOUT not valid
tDIS
CS rising to DOUT 3-state
70
100
ns
tEN
DCLOCK falling to DOUT enabled
20
50
ns
tF
DOUT fall time
5
25
ns
tR
DOUT rise time
7
25
ns
16
DCLOCKs
22
DCLOCKs
0
20
5
DCLOCKs
ns
ns
15
ns
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TYPICAL CHARACTERISTICS: VDD = +5V
At TA = 25°C, VREF = 2.5V, fSAMPLE = 250kHz, fCLK = 24 × fSAMPLE, unless otherwise noted.
DIFFERENTIAL LINEARITY ERROR vs CODE
3
2
2
1
1
DLE (LSB)
ILE (LSB)
INTEGRAL LINEARITY ERROR vs CODE
3
0
0
-1
-1
-2
-2
-3
8000h
C000h
0000h
4000h
-3
8000h
7FFFh
C000h
0000h
Output Code
Figure 2.
7FFFh
Figure 3.
SUPPLY CURRENT vs TEMPERATURE
CHANGE IN OFFSET vs TEMPERATURE
3.0
1.745
2.5
Delta from +25°C (LSB)
1.750
1.740
Supply Current (mA)
4000h
Output Code
1.735
1.730
1.725
1.720
1.715
2.0
1.5
1.0
0.5
0
-0.5
-1.0
-1.5
1.710
-2.0
-50
-25
0
25
50
75
100
-50
-25
0
Temperature (°C)
25
50
75
100
75
100
Temperature (°C)
Figure 4.
Figure 5.
CHANGE IN BIPOLAR ZERO ERROR vs TEMPERATURE
CHANGE IN GAIN vs TEMPERATURE
3.0
0.50
2.0
Delta from +25°C (LSB)
Delta from +25°C (LSB)
2.5
1.5
1.0
0.5
0
-0.5
-1.0
0.25
0
-0.25
-0.50
-1.5
-2.0
-0.75
-50
-25
0
25
50
75
100
-50
Temperature (°C)
0
25
50
Temperature (°C)
Figure 6.
10
-25
Figure 7.
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TYPICAL CHARACTERISTICS: VDD = +5V (continued)
At TA = 25°C, VREF = 2.5V, fSAMPLE = 250kHz, fCLK = 24 × fSAMPLE, unless otherwise noted.
SUPPLY CURRENT vs SAMPLING RATE
10
1
Supply Current (mA)
Power-Down Current (nA)
POWER-DOWN CURRENT vs TEMPERATURE
38
37
36
35
34
33
32
31
30
29
28
27
26
25
24
0.1
0.01
0.001
-50
-25
0
25
50
75
100
1
10
100
250
Sampling Rate (kHz)
Temperature (°C)
Figure 8.
Figure 9.
REFERENCE CURRENT vs SAMPLING RATE
FREQUENCY SPECTRUM
(8192 point FFT, fIN = 1.9836kHz, –0.2dB)
0
100
10
-40
Amplitude (dB)
Supply Current (mA)
-20
1
0.1
-60
-80
-100
-120
-140
0.01
-160
1
10
100
250
0
Sampling Rate (kHz)
25
50
75
125
Figure 10.
Figure 11.
FREQUENCY SPECTRUM
(8192 Point FFT, fIN = 9.9792kHz, –0.2dB)
SIGNAL-TO-NOISE RATIO AND
SIGNAL-TO-NOISE + DISTORTION vs INPUT FREQUENCY
0
95
-20
SNR and SINAD (dB)
SNR
-40
Amplitude (dB)
100
Frequency (kHz)
-60
-80
-100
-120
90
SINAD
85
80
-140
75
-160
0
25
50
75
100
125
1
10
100
200
Frequency (kHz)
Frequency (kHz)
Figure 12.
Figure 13.
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TYPICAL CHARACTERISTICS: VDD = +5V (continued)
At TA = 25°C, VREF = 2.5V, fSAMPLE = 250kHz, fCLK = 24 × fSAMPLE, unless otherwise noted.
SPURIOUS-FREE DYNAMIC RANGE AND
TOTAL HARMONIC DISTORTION vs INPUT FREQUENCY
120
-120
115
-115
SFDR
15.0
14.5
-105
100
-100
95
-95
THD(1)
90
-90
85
-85
80
-80
75
-75
70
10
100
13.5
13.0
12.5
12.0
-65
1
14.0
-70
NOTE: (1) First nine harmonics of the input frequency.
65
ENOB (Bits)
-110
105
THD (dB)
SFDR (dB)
110
EFFECTIVE NUMBER OF BITS vs INPUT FREQUENCY
200
1
10
Frequency (kHz)
100
200
Frequency (kHz)
Figure 14.
Figure 15.
CHANGE IN SIGNAL-TO-NOISE + DISTORTION
vs TEMPERATURE
CHANGE IN SIGNAL-TO-NOISE + DISTORTION
vs INPUT LEVEL
100
0.7
fIN = 1.98364kHz, -0.2dB
0.6
90
fIN = 1.98364kHz
80
0.4
0.3
SINAD (dB)
Delta from +25°C (LSB)
0.5
0.2
0.1
0
70
60
50
-0.1
40
-0.2
30
-0.3
20
-0.4
10
-0.5
-50
-25
0
25
75
50
100
-80
-70
-60
Temperature (°C)
-50
-40
-30
-20
-10
0
Input Level (dB)
Figure 16.
Figure 17.
PEAK-TO-PEAK NOISE FOR A DC INPUT
vs REFERENCE VOLTAGE
OUTPUT CODE HISTOGRAM FOR A DC INPUT
(8192 Conversions)
100
Peak-to-Peak Noise (LSB)
VDD = 5V
4835
10
1673
1608
34
42
1
1
0.1
2
2.5
FFFE
Figure 18.
12
FFFF
0000
0001
0002
Code (Hex)
Reference Voltage (V)
Figure 19.
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TYPICAL CHARACTERISTICS: VDD = +2.7V
At TA = 25°C, VREF = 1.25V, fSAMPLE = 200kHz, fCLK = 24 × fSAMPLE, unless otherwise noted.
DIFFERENTIAL LINEARITY ERROR vs CODE
3
2
2
1
1
DLE (LSB)
ILE (LSB)
INTEGRAL LINEARITY ERROR vs CODE
3
0
0
-1
-1
-2
-2
-3
8000h
C000h
0000h
4000h
-3
8000h
7FFFh
C000h
0000h
Output Code
Figure 20.
7FFFh
Figure 21.
SUPPLY CURRENT vs TEMPERATURE
CHANGE IN OFFSET vs TEMPERATURE
1.310
1.00
1.305
0.75
Delta from +25°C (LSB)
1.300
Supply Current (mA)
4000h
Output Code
1.295
1.290
1.285
1.280
1.275
1.270
0.50
0.25
0
-0.25
-0.50
-0.75
1.265
1.260
-1.00
-50
-25
0
25
75
50
100
-50
-25
0
Temperature (°C)
25
50
75
100
75
100
Temperature (°C)
Figure 22.
Figure 23.
CHANGE IN BIPOLAR ZERO ERROR
vs TEMPERATURE
CHANGE IN GAIN vs TEMPERATURE
0.50
0.50
Delta from +25°C (LSB)
Delta from +25°C (LSB)
0.25
0.25
0
-0.25
0
-0.25
-0.50
-0.75
-0.50
-1.00
-50
-25
0
25
50
75
100
-50
Temperature (°C)
-25
0
25
50
Temperature (°C)
Figure 24.
Figure 25.
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TYPICAL CHARACTERISTICS: VDD = +2.7V (continued)
At TA = 25°C, VREF = 1.25V, fSAMPLE = 200kHz, fCLK = 24 × fSAMPLE, unless otherwise noted.
POWER-DOWN CURRENT vs TEMPERATURE
SUPPLY CURRENT vs SAMPLING RATE
10
25
1
23
Supply Current (mA)
Power-Down Current (nA)
24
22
21
20
19
0.1
0.01
18
0.001
17
0.0001
16
-50
-25
0
25
75
50
100
10
1
100
200
Sampling Rate (kHz)
Temperature (°C)
Figure 26.
Figure 27.
REFERENCE CURRENT vs SAMPLING RATE
FREQUENCY SPECTRUM
(8192 Point FFT, fIN = 1.9775kHz, –0.2dB)
0
100
10
-40
Amplitude (dB)
Reference Current (mA)
-20
1
0.1
-60
-80
-100
-120
-140
0.01
-160
10
1
100
200
0
20
30
40
50
60
70
80
90
100
Figure 28.
Figure 29.
FREQUENCY SPECTRUM
(8192 Point FFT, fIN = 9.9854kHz, –0.2dB)
SIGNAL-TO-NOISE RATIO AND
SIGNAL-TO-NOISE + DISTORTION vs INPUT FREQUENCY
0
86
-20
85
-40
84
-60
-80
-100
-120
SNR
83
82
81
SINAD
80
79
-140
78
-160
0
10
20
30
40
50
60
70
80
90
100
1
10
100
200
Frequency (kHz)
Frequency (kHz)
Figure 30.
14
10
Frequency (kHz)
SNR and SINAD (dB)
Amplitude (dB)
Sampling Rate (kHz)
Figure 31.
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TYPICAL CHARACTERISTICS: VDD = +2.7V (continued)
At TA = 25°C, VREF = 1.25V, fSAMPLE = 200kHz, fCLK = 24 × fSAMPLE, unless otherwise noted.
SPURIOUS-FREE DYNAMIC RANGE AND
TOTAL HARMONIC DISTORTION vs INPUT FREQUENCY
110
EFFECTIVE NUMBER OF BITS vs INPUT FREQUENCY
-110
15.0
-105
14.5
-100
14.0
SFDR (dB)
100
THD(1)
ENOB (Bits)
105
THD (dB)
SFDR
95
-95
90
-90
13.0
-85
12.5
-80
12.0
85
13.5
NOTE: (1) First nine harmonics of the input frequency.
80
1
10
100
200
1
10
Frequency (kHz)
Figure 32.
200
Figure 33.
CHANGE IN SIGNAL-TO-NOISE + DISTORTION vs
TEMPERATURE
SIGNAL-TO-NOISE + DISTORTION vs INPUT LEVEL
0.6
100
fIN = 1.97754kHz, -0.2dB
0.5
fIN = 1.97754kHz
90
0.4
80
0.3
70
0.2
SINAD (dB)
Delta from +25°C (LSB)
100
Frequency (kHz)
0.1
0
-0.1
-0.2
60
50
40
30
-0.3
20
-0.4
10
-0.5
0
-0.6
-50
-25
0
25
75
50
100
-80
-70
-60
Temperature (°C)
-50
-40
-30
-20
-10
0
Input Level (dB)
Figure 34.
Figure 35.
OUTPUT CODE HISTOGRAM FOR A DC INPUT
(8192 Conversions)
3920
1596
1504
581
2
497
50
41
1
FFFC FFFD FFFE FFFF 0000 0001 0002 0003 0004
Code (Hex)
Figure 36.
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THEORY OF OPERATION
The ADS8317 is a classic Successive Approximation
Register (SAR) analog-to-digital (A/D) converter. The
architecture is based on capacitive redistribution that
inherently includes a sample-and-hold function. The
converter is fabricated on a 0.6μ CMOS process. The
architecture and fabrication process allow the
ADS8317 to acquire and convert an analog signal at
up to 250,000 conversions per second while
consuming less than 10mW from VDD.
Differential
linearity
for
the
ADS8317
is
factory-adjusted via a package-level trim procedure.
The state of the trim elements is stored in non-volatile
memory and is continuously updated after each
acquisition cycle, just prior to the start of the
successive approximation operation. This process
ensures that one complete conversion cycle always
returns the part to its factory-adjusted state in the
event of a power interruption.
The ADS8317 requires an external reference, an
external clock, and a single power source (VDD). The
external reference can be any voltage between 0.1V
and VDD/2. The value of the reference voltage directly
sets the range of the analog input. The reference
input current depends on the conversion rate of the
ADS8317.
The external clock can vary between 24kHz (1kHz
throughput) and 6.0MHz (250kHz throughput). The
duty cycle of the clock is not significant, as long as
the minimum high and low times are at least 200ns
(VDD = 4.75V or greater). The minimum clock
frequency is set by the leakage on the internal
capacitors to the ADS8317.
The analog input is provided to two input pins: +IN
and –IN. When a conversion is initiated, the
differential input on these pins is sampled on the
internal capacitor array. While a conversion is in
progress, both inputs are disconnected from any
internal function.
The digital data that are provided on the DOUT pin are
for the conversion currently in progress—there is no
pipeline delay. It is possible to continue to clock the
ADS8317 after the conversion is complete and to
obtain the serial data least significant bit first. See the
Digital Timing section for more information.
ANALOG INPUT
The analog input is bipolar and fully differential. There
are two general methods of driving the analog input
of the ADS8317: single-ended or differential, as
shown in Figure 37. When the input is single-ended,
the –IN input is held at a fixed voltage. The +IN input
swings around the same voltage and the
peak-to-peak amplitude is 2 × VREF. The value of
VREF determines the range over which the common
voltage may vary, as shown in Figure 38 and
Figure 39.
Single-Ended Input
2 ´ VREF
Peak-to-Peak
ADS8317
Common
Voltage
Differential Input
VREF
Peak-to-Peak
ADS8317
Common
Voltage
VREF
Peak-to-Peak
Figure 37. Methods of Driving the
ADS8317—Single-Ended or Differential
The digital result of the conversion is clocked out by
the DCLOCK input and is provided serially (most
significant bit first) on the DOUT pin.
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5
5
4
3.8
2.8
3
Single-Ended Input
2
2.2
1
0.2
0
VDD = 5V
Common Voltage Range (V)
Common Voltage Range (V)
VDD = 5V
4
3.8
2.75
3
2
Differential Input
1
0.95
0
-0.3
-0.3
-1
-1
0
0.5
1.0
1.5
2.0
2.5
0
0.5
VREF (V)
1.0
1.5
Figure 38. Single-Ended 5V Input,
Common Voltage Range vs VREF
3.0
2.5
2.0
1.5
1.5
1.0
Single-Ended Input
1.05
0.2
0
-0.5
-0.2
2.5
2.0
1.5
1.5
1.5
1.0
Differential Input
0.5
0.5
0
-0.3
-0.5
0.5
-1.0
0.10
VDD = 2.7V
Common Voltage Range (V)
Common Voltage Range (V)
VDD = 2.7V
0.5
2.5
Figure 40. Differential 5V Input,
Common Voltage Range vs VREF
3.0
1.5
2.0
VREF (V)
-1.0
0.35
0.60
0.85
1.10
1.35
0
VREF (V)
0.25
0.50
0.75
1.00
1.25
VREF (V)
Figure 39. Single-Ended 2.7V Input,
Common Voltage Range vs VREF
Figure 41. Differential 2.7V Input,
Common Voltage Range vs VREF
When the input is differential, the amplitude of the
input is the difference between the +IN and –IN input,
or +IN – (–IN). A voltage or signal is common to both
of these inputs. The peak-to-peak amplitude of each
input is VREF about this common voltage. However,
since the inputs are 180° out-of-phase, the
peak-to-peak amplitude of the difference voltage is 2
× VREF. The value of VREF also determines the range
of the voltage that may be common to both inputs, as
shown in Figure 41 and Figure 40.
In each case, care should be taken to ensure that the
output impedance of the sources driving the +IN and
–IN inputs are matched. If this matching is not
observed, the two inputs could have different settling
times. This difference may result in offset error, gain
error, and linearity error that change with both
temperature and input voltage. If the impedance
cannot be matched, the errors can be lessened by
giving the ADS8317 additional acquisition time.
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The input current on the analog inputs depends on a
number of factors: sample rate, input voltage, and
source impedance. Essentially, the current into the
ADS8317 charges the internal capacitor array during
the sample period. After this capacitance has been
fully charged, there is no further input current. The
source of the analog input voltage must be able to
charge the input capacitance (24pF) to 16-bit settling
level within 4.5 clock cycles. When the converter
goes into the hold mode, or while it is in the
power-down mode, the input impedance is greater
than 1GΩ.
Care must be taken regarding the absolute analog
input voltage. The +IN input should always remain
within the range of GND – 300mV to VDD + 300mW.
The –IN input should always remain within the range
of GND – 300mV to 4V. Outside of these ranges, the
converter linearity may not meet specifications. To
obtain maximum performance from the ADS8317, an
input circuit such as that shown in Figure 42 is
recommended.
Single-Ended
10W
+IN
OPA365
50W
24pF
1000pF
ADS8317
-IN
VCM
50W
24pF
+
10W
+IN
50W
24pF
1000pF
ADS8317
1nF
10W
-IN
OPA365
The external reference sets the analog input range.
The ADS8317 operates with a reference in the range
of 0.1V to VDD/2. There are several important
implications to this specification.
As the reference voltage is reduced, the analog
voltage weight of each digital output code is reduced.
This reduction is often referred to as the least
significant bit (LSB) size and is equal to the reference
voltage divided by 65,536. This relationship means
that any offset or gain error inherent in the A/D
converter appears to increase (in terms of LSB size)
as the reference voltage is reduced. For a reference
voltage of 2.5V, the value of the LSB is 76.3μV, and
for a reference voltage of 1.25V, the LSB is 38.15μV.
The noise inherent in the converter also appears to
increase with a lower LSB size. With a 2.5V
reference, the internal noise of the converter typically
contributes only 5LSB peak-to-peak of potential error
to the output code. When the external reference is
1.25V, the potential error contribution from the
internal noise is almost two times larger (9LSB). The
errors arising from the internal noise are Gaussian in
nature and can be reduced by averaging consecutive
conversion results.
For more information regarding noise, consult
Figure 18, Peak-to-Peak Noise for a DC Input vs
Reference Voltage. Note that the Effective Number Of
Bits (ENOB) figure is calculated based on the
converter signal-to-(noise + distortion) ratio with a
2kHz, 0dB input signal. SINAD is related to ENOB as
follows:
SINAD = 6.02 × ENOB + 1.76
Differential
OPA365
REFERENCE INPUT
50W
24pF
With lower reference voltages, extra care should be
taken to provide a clean layout including adequate
bypassing, a clean power supply, a low-noise
reference, and a low-noise input signal. Due to the
lower LSB size, the converter is also more sensitive
to external sources of error, such as nearby digital
signals and electromagnetic interference.
1000pF
Figure 42. Single-Ended and Differential Methods
of Interfacing the ADS8317
18
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The equivalent input circuit for the reference voltage
is presented in Figure 43. At the same time, an
equivalent capacitor of 24pF is switched. To obtain
optimum performance from the ADS8317, special
care must be taken in designing the interface circuit
to the reference input pin. To ensure a stable
reference voltage, a 47μF tantalum capacitor with low
ESR should be connected as close as possible to the
input pin. If a high output impedance reference
source is used, an additional operational amplifier
with a current-limiting resistor must be placed in front
of the capacitors.
ADS8317
VREF
50W
24pF
OPA350
47mF
Figure 43. Input Reference Circuit and Interface
When the ADS8317 is in power-down mode, the input
resistance of the reference pin has a value of 5GΩ.
Because the input capacitors must be recharged
before the next conversion starts, an operational
amplifier with good dynamic characteristics, such as
the OPA350, should be used to buffer the reference
input.
Noise
The transition noise of the ADS8317 itself is
extremely low, as shown in Figure 19 and Figure 36;
it is much lower than competing A/D converters.
These histograms were generated by applying a
low-noise DC input and initiating 8192 conversions.
The digital output of the A/D converter varies in
output code because of the internal noise of the
ADS8317. This variance is true for all 16-bit,
SAR-type A/D converters. Using a histogram to plot
the output codes, the distribution should appear
bell-shaped with the peak of the bell curve
representing the nominal code for the input value.
The ±1σ, ±2σ, and ±3σ distributions represent 68.3%,
95.5%, and 99.7%, respectively, of all codes. The
transition noise can be calculated by dividing the
number of codes measured by 6, which yields the
±3σ distribution, or 99.7%, of all codes. Statistically,
up to three codes could fall outside the distribution
when executing 1000 conversions. The ADS8317,
with five output codes for the ±3σ distribution, yields
less than ±0.8LSB of transition noise. Remember that
to achieve this low-noise performance, the
peak-to-peak noise of the input signal and reference
must be less than 50μV.
Averaging
The noise of the A/D converter can be compensated
by averaging the digital codes. By averaging
conversion results, transition noise is reduced by a
factor of 1/√n , where n is the number of averages.
For example, averaging four conversion results
reduces the transition noise from ±0.8LSB to
±0.4LSB. Averaging should only be used for input
signals with frequencies near DC.
For AC signals, a digital filter can be used to
low-pass filter and decimate the output codes. This
configuration works in a similar manner to averaging;
for every decimation by 2, the signal-to-noise ratio
improves by 3dB.
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ADS8317
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DIGITAL INTERFACE
Signal Levels
The ADS8317 has a wide range of power-supply
voltage. The A/D converter, as well as the digital
interface circuit, is designed to accept and operate
from 2.7V up to 5.5V. This voltage range
accommodates different logic levels. When the
ADS8317 power-supply voltage is in the range of
4.5V to 5.5V (5V logic level), the ADS8317 can be
connected directly to another 5V, CMOS-integrated
circuit. When the ADS8317 power-supply voltage is in
the range of 2.7V to 3.6V (3V logic level), the
ADS8317 can be connected directly to another 3.3V
LVCMOS integrated circuit.
Serial Interface
The ADS8317 communicates with microprocessors
and other digital systems via a synchronous 3-wire
serial interface, as illustrated in the Timing
Information section and Timing Characteristics . The
DCLOCK signal synchronizes the data transfer, with
each bit being transmitted on the falling edge of
DCLOCK. Most receiving systems capture the
bitstream on the rising edge of DCLOCK. However, if
the minimum hold time for DOUT is acceptable, the
system can use the falling edge of DCLOCK to
capture each bit.
A falling CS signal initiates the conversion and data
transfer. The first 4.5 to 5.0 clock periods of the
conversion cycle are used to sample the input signal.
20
After the fifth falling DCLOCK edge, DOUT is enabled
and outputs a low value for one clock period. For the
next 16 DCLOCK periods, DOUT outputs the
conversion result, most significant bit first. After the
least significant bit (B0) has been output, subsequent
clocks repeat the output data, but in a least significant
bit first format.
After the most significant bit (B15) has been
repeated, DOUT will 3-state. Subsequent clocks have
no effect on the converter. A new conversion is
initiated only when CS has been taken high and
returned low.
Data Format
The output data from the ADS8317 are in binary twos
complement format, as shown in Table 1 and
Figure 44. The table and figure represent the ideal
output code for the given input voltage and do not
include the effects of offset, gain error, or noise.
Table 1. Ideal Input Voltages and Output Codes
DESCRIPTION
ANALOG VALUE
Full-scale range
2 × VREF
Least significant
bit (LSB)
2 × VREF/65536
Binary
Code
Hex
Code
+Full scale
+VREF – 1 LSB
0111 1111 1111 1111
7FFF
0V
0000 0000 0000 0000
0000
0V – 1 LSB
1111 1111 1111 1111
FFFF
–VREF
1000 0000 0000 0000
8000
Midscale
Midscale – 1 LSB
–Full scale
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DIGITAL OUTPUT
BINARY TWOS COMPLEMENT
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0111 1111 1111 1111
65535
0111 1111 1111 1110
65534
0111 1111 1111 1101
65533
0000 0000 0000 0001
32769
0000 0000 0000 0000
32768
1111 1111 1111 1111
32767
1000 0000 0000 0010
2
1000 0000 0000 0001
1
1000 0000 0000 0000
Step
Digital Output Code
Binary Twos Complement
0
+38.15mV
-38.15mV
V-FS = -2.5V
V+FS = VREF = 2.5V
V+FS - 1LSB = 2.499992V
0V
-2.49996V
2.499985V
-2.49992V
Bipolar Analog Input Voltage = V(+IN) - V(-IN)
-2.49985V
1LSB = 76.29mV
VCM = 2.5V
16-BIT
-Full-Scale Code
Midscale Code
+Full-Scale Code
Twos Complement Output
V-FS = 8000h
VMS = 0000h
V+FS = 7FFFh
Bipolar Analog Input
VCODE = -VREF
VCODE = 0V
VCODE = VREF - 1LSB
VREF = 2.5V
Figure 44. Ideal Conversion Characteristics (Conditions: VCM = 2.5V, VREF = 2.5V)
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ADS8317
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POWER DISSIPATION
Short Cycling
The architecture of the converter, the semiconductor
fabrication process, and a careful design allow the
ADS8317 to convert at up to a 250kHz rate while
requiring very little power. However, for the absolute
lowest power dissipation, there are several things to
keep in mind.
Another way to save power is to use the CS signal to
short-cycle the conversion. The ADS8317 places the
latest data bit on the DOUT line as it is generated;
therefore, the converter can easily be short-cycled.
This term means that the conversion can be
terminated at any time. For example, if only 14 bits of
the conversion result are needed, then the conversion
can be terminated (by pulling CS high) after the 14th
bit has been clocked out.
The power dissipation of the ADS8317 scales directly
with conversion rate. Therefore, the first step to
achieving the lowest power dissipation is to find the
lowest conversion rate that satisfies the system
requirements.
In addition, the ADS8317 goes into Power-Down
mode under two conditions: when the conversion is
complete and whenever CS is high (see the Timing
Characteristics section). Ideally, each conversion
should occur as quickly as possible, preferably at a
6.0MHz clock rate. This way, the converter spends
the longest possible time in power-down mode. This
is very important because the converter not only uses
power on each DCLOCK transition (as is typical for
digital CMOS components), but also uses some
current for the analog circuitry, such as the
comparator. The analog section dissipates power
continuously until power-down mode is entered.
Figure 9 and Figure 27 illustrate the current
consumption of the ADS8317 versus sample rate. For
these graphs, the converter is clocked at maximum
speed regardless of the sample rate. CS is held high
during the remaining sample period.
This technique can also be used to lower the power
dissipation (or to increase the conversion rate) in
those applications where an analog signal is being
monitored until some condition becomes true. For
example, if the signal is outside a predetermined
range, the full 16-bit conversion result may not be
needed. If so, the conversion can be terminated after
the first n bits, where n might be as low as 3 or 4.
This technique results in lower power dissipation in
both the converter and the rest of the system
because they spend more time in power-down mode.
POWER-ON RESET
The ADS8317 bias circuit is self-starting. There may
be a static current (approximately 1.5mA with VDD =
5V) after power-on, unless the circuit is powered
down. It is recommended to run a single test
conversion (configured the same as any regular
conversion) after the power supply reaches at least
2.4V to ensure the device is put into power-down
mode.
There is an important distinction between the
power-down mode that is entered after a conversion
is complete and the full power-down mode that is
enabled when CS is high. CS low only shuts down
the analog section. The digital section completely
shuts down only when CS is high. Thus, if CS is left
low at the end of a conversion, and the converter is
continually clocked, the power consumption is not as
low as when CS is high.
22
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LAYOUT
For optimum performance, care should be taken with
the physical layout of the ADS8317 circuitry. This
caution is particularly true if the reference voltage is
low and/or the conversion rate is high. At a 250kHz
conversion rate, the ADS8317 makes a bit decision
every 167ns. That is, for each subsequent bit
decision, the digital output must be updated with the
results of the last bit decision, the capacitor array
appropriately switched and charged, and the input to
the comparator settled to a 16-bit level, all within one
clock cycle.
The basic SAR architecture is sensitive to spikes on
the power supply, reference, and ground connections
that occur just prior to latching the comparator output.
Thus, during any single conversion for an n-bit SAR
converter, there are n windows in which large
external transient voltages can easily affect the
conversion result. Such spikes might originate from
switching power supplies, digital logic, and
high-power devices, to name a few potential sources.
This particular source of error can be very difficult to
track down if the glitch is almost synchronous to the
converter DCLOCK signal because the phase
difference between the glitch and DCLOCK changes
with time and temperature, causing sporadic
misoperation.
With these considerations in mind, power to the
ADS8317 should be clean and well-bypassed. A
0.1μF ceramic bypass capacitor should be placed as
close as possible to the ADS8317 package. In
addition, a 1μF to 10μF capacitor and a 5Ω or 10Ω
series resistor may be used to low-pass filter a noisy
supply.
The reference should be similarly bypassed with a
47μF capacitor. Again, a series resistor and large
capacitor can be used to low-pass filter the reference
voltage. If the reference voltage originates from an op
amp, make sure that the op amp can drive the
bypass capacitor without oscillation (the series
resistor can help in this case). Keep in mind that
while the ADS8317 draws very little current from the
reference on average, there are still instantaneous
current demands placed on the external input and
reference circuitry.
Texas Instruments' OPA365 op amp provides
optimum performance for buffering the signal inputs;
the OPA350 can be used to effectively buffer the
reference input.
Also, keep in mind that the ADS8317 offers no
inherent rejection of noise or voltage variation in
regards to the reference input. This characteristic is
of particular concern when the reference input is tied
to the power supply. Any noise and ripple from the
supply appears directly in the digital results. While
high-frequency noise can be filtered out, as described
in the previous paragraph, voltage variation resulting
from the line frequency (50Hz or 60Hz) can be
difficult to remove.
The GND pin on the ADS8317 should be placed on a
clean ground point. In many cases, this point is the
analog ground. Avoid connecting the GND pin too
close to the grounding point for a microprocessor,
microcontroller, or digital signal processor. If needed,
run a ground trace directly from the converter to the
power-supply connection point. The ideal layout
includes an analog ground plane for the converter
and associated analog circuitry.
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ADS8317
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APPLICATION CIRCUITS
Figure 45 shows an example of a basic data
acquisition system. The ADS8317 input range is
connected to 2.5V or 4.096V. The 5Ω resistor and
1μF to 10μF capacitor filters the microcontroller noise
on the supply, as well as any high-frequency noise
from the supply itself. The exact values should be
picked such that the filter provides adequate rejection
of noise. Operational amplifiers and voltage reference
are connected to the analog power supply, AVDD.
DVDD
2.7V to 3.6V
0.1mF
AVDD
2.7V to 5V
10mF
5W
REF3225
REF
OPA350
10W
IN
OUT
VDD
47mF
2.2mF
0.47mF
+
0.1mF
GND
+
10mF
ADS8317
DSP
10W
VCM + (0V to 2.5V)
1000pF
CS
1nF
DOUT
DCLOCK
10W
-IN
OPA365
VCM
TMS320C6xx
or
TMS320C5xx
or
TMS320C2xx
+IN
OPA365
GND
GND
1000pF
Figure 45. Example of a Basic Data Acquisition System
24
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REVISION HISTORY
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision C (June 2008) to Revision D .................................................................................................... Page
•
Changed SON-8 (DRB) package availability ........................................................................................................................ 1
•
Deleted lead temperature specification ................................................................................................................................ 2
•
Added missing mu symbol to High-level output voltage test condition ................................................................................. 4
•
Added missing mu symbol to Low-level output voltage test condition ................................................................................. 4
•
Changed X-axis unit from mA to μA in Figure 10 ............................................................................................................... 11
•
Changed Figure 38 ............................................................................................................................................................. 17
•
Changed Figure 39 ............................................................................................................................................................. 17
•
Changed Figure 40 title (typo) ............................................................................................................................................ 17
•
Added missing mu symbol to the value of the LSB, 76.3V ................................................................................................ 18
Changes from Revision B (May 2008) to Revision C ...................................................................................................... Page
•
Changed 2nd timing diagram from the top in Figure 1 ......................................................................................................... 8
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PACKAGE OPTION ADDENDUM
www.ti.com
7-Oct-2021
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
ADS8317IBDGKR
ACTIVE
VSSOP
DGK
8
2500
RoHS & Green
Call TI
Level-2-260C-1 YEAR
-40 to 85
D17
ADS8317IBDGKT
ACTIVE
VSSOP
DGK
8
250
RoHS & Green
Call TI
Level-2-260C-1 YEAR
-40 to 85
D17
ADS8317IBDRBR
ACTIVE
SON
DRB
8
3000
RoHS & Green
Call TI
Level-2-260C-1 YEAR
-40 to 85
D17
ADS8317IBDRBT
ACTIVE
SON
DRB
8
250
RoHS & Green
Call TI
Level-2-260C-1 YEAR
-40 to 85
D17
ADS8317IDGKR
ACTIVE
VSSOP
DGK
8
2500
RoHS & Green
Call TI
Level-2-260C-1 YEAR
-40 to 85
D17
ADS8317IDGKT
ACTIVE
VSSOP
DGK
8
250
RoHS & Green
Call TI
Level-2-260C-1 YEAR
-40 to 85
D17
ADS8317IDRBR
ACTIVE
SON
DRB
8
3000
RoHS & Green
Call TI
Level-2-260C-1 YEAR
-40 to 85
D17
ADS8317IDRBT
ACTIVE
SON
DRB
8
250
RoHS & Green
Call TI
Level-2-260C-1 YEAR
-40 to 85
D17
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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