LM20333
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SNVS558D – MAY 2008 – REVISED APRIL 2013
LM20333 36V, 3A Synchronous Buck Regulator with Frequency Synchronization
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FEATURES
DESCRIPTION
•
•
•
•
The LM20333 is a full featured synchronous buck
regulator capable of delivering up to 3A of load
current. The current mode control loop is externally
compensated with only two components, offering both
high performance and ease of use. The device is
optimized to work over the input voltage range of
4.5V to 36V making it well suited for high voltage
systems.
1
2
•
•
•
•
•
•
•
•
•
•
•
•
4.5V to 36V Input Voltage Range
3A Output Current, 5.2A Peak Current
130 mΩ/110 mΩ Integrated Power MOSFETs
94% Peak Efficiency with Synchronous
Rectification
1.5% Feedback Voltage Accuracy
Current Mode Control, Selectable
Compensation
Oscillator Synchronization from 250kHz to
1.5MHz
Adjustable Output Voltage Down to 0.8V
Compatible with Pre-biased Loads
Programmable Soft-start with External
Capacitor
Precision Enable Pin with Hysteresis
OVP, UVLO Inputs and PGOOD Output
Internally Protected with Peak Current Limit,
Thermal Shutdown and Restart
Accurate Current Limit Minimizes Inductor
Size
Non-linear Current Mode Slope Compensation
20-Pin HTSSOP Exposed Pad Package
The device features internal Over Voltage Protection
(OVP) and Over Current Protection (OCP) circuits for
increased system reliability. A precision Enable pin
and integrated UVLO allows the turn on of the device
to be tightly controlled and sequenced. Startup inrush
currents are limited by both an internally fixed and
externally adjustable soft-start circuit. Fault detection
and supply sequencing are possible with the
integrated power good (PGOOD) circuit.
The LM20333 is designed to work well in multi-rail
power supply architectures. The output voltage of the
device can be configured to track a higher voltage rail
using the SS/TRK pin. If the output of the LM20333 is
pre-biased at startup it will not sink current to pull the
output low until the internal soft-start ramp exceeds
the voltage at the feedback pin.
The switching frequency of the LM20333 can be
synchronized to an external clock by use of the
SYNC pin. The SYNC pin is capable of synchronizing
to input signals ranging from 250 kHz to 1.5 MHz.
APPLICATIONS
•
•
•
Simple to Design, High Efficiency Point of
Load Regulation from a 4.5V to 36V Bus
High Performance DSPs, FPGAs, ASICs and
Microprocessors
Communications Infrastructure, Automotive
The LM20333 is offered in an exposed pad 20-pin
HTSSOP package that can be soldered to the PCB,
eliminating the need for bulky heatsinks.
Simplified Application Circuit
LM20333
BOOT
VIN
VIN
CBOOT
L
VOUT
SW
D1
(Optional)
CIN
EN
SYNC
COMP
RC1
CC1
SS/TRK
AGND
RFB1
COUT
FB
RFB2
PGOOD
VCC
GND
CVCC
1
2
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
All trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2008–2013, Texas Instruments Incorporated
LM20333
SNVS558D – MAY 2008 – REVISED APRIL 2013
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Connection Diagram
SS/TRK
1
20 SYNC
FB
2
19 EN
PGOOD
3
18
COMP
4
17 BOOT
VCC
VIN
5
VIN
6
SW
7
14
SW
SW
8
13
SW
GND
9
12
AGND
GND
10
11 GND
EP
16
VIN
15
VIN
Figure 1. 20-Pin HTSSOP, Top View
See PWP0020A Package
PIN DESCRIPTIONS
Pin(s)
Name
Description
1
SS/TRK
Application Information
Soft-Start or Tracking control input
An internal 4.5 µA current source charges an external capacitor to set
the soft-start rate. The PWM can track to an external voltage ramp with
a low impedance source. If left open, an internal 1 ms SS ramp is
activated.
2
FB
Feedback input to the error amplifier
from the regulated output
This pin is connected to the inverting input of the internal
transconductance error amplifier. An 800 mV reference is internally
connected to the non-inverting input of the error amplifier.
3
PGOOD
Power good output signal
Open drain output indicating the output voltage is regulating within
tolerance. A pull-up resistor of 10 kΩ to 100 kΩ is recommended if this
function is used.
4
COMP
Output of the internal error amplifier and
input to the Pulse Width Modulator
The loop compensation network should be connected between the
COMP pin and the AGND pin.
5,6,15,16
7,8,13,14
VIN
Input supply voltage
Nominal operating range: 4.5V to 36V.
SW
Switch pin
The drain terminal of the internal Synchronous Rectifier power
NMOSFET and the source terminal of the internal Control power
NMOSFET.
9,10,11
GND
Ground
Internal reference for the power MOSFETs.
12
AGND
Analog ground
Internal reference for the regulator control functions.
17
BOOT
Boost input for bootstrap capacitor
An internal diode from VCC to BOOT charges an external capacitor
required from SW to BOOT to power the Control MOSFET gate driver.
18
VCC
Output of the high voltage linear
regulator. The VCC voltage is regulated
to approximately 5.5V.
VCC tracks VIN up to about 7.2V. Above VIN = 7.2V, VCC is regulated
to approximately 5.5 Volts. A 0.1 µF to 1 µF ceramic decoupling
capacitor is required. The VCC pin is an output only.
19
EN
Enable or UVLO input
An external voltage divider can be used to set the line undervoltage
lockout threshold. If the EN pin is left unconnected, a 2 µA pull-up
current source pulls the EN pin high to enable the regulator.
20
SYNC
Frequency synchronization input
An external clock connected to this pin will set the switching frequency.
If left open the device will operate at approximately 200 kHz.
EP
Exposed
Pad
Exposed pad
Exposed metal pad on the underside of the package with a weak
electrical connection to GND. Connect this pad to the PC board ground
plane in order to improve heat dissipation.
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
2
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Absolute Maximum Ratings (1) (2)
VIN to GND
-0.3V to +38V
BOOT to GND
-0.3V to +43V
BOOT to SW
-0.3V to +7V
SW to GND
-0.5V to +38V
SW to GND (Transient)
-1.5V (< 20 ns)
FB, EN, SS/TRK, COMP, SYNC, PGOOD to GND
-0.3V to +6V
VCC to GND
-0.3V to +8V
Storage Temperature
-65°C to 150°C
ESD Rating
Human Body Model (3)
(1)
(2)
(3)
2kV
Absolute Maximum Ratings indicate limits beyond witch damage to the device may occur. Operating Ratings indicate conditions for
which the device is intended to be functional, but do not ensure specific performance limits. For ensured specifications and test
conditions, see the Electrical Characteristics.
If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/Distributors for availability and
specifications.
The human body model is a 100 pF capacitor discharged through a 1.5 kΩ resistor to each pin.
Operating Ratings
VIN to GND
+4.5V to +36V
−40°C to + 125°C
Junction Temperature
Electrical Characteristics
Unless otherwise stated, the following conditions apply: VVIN = 12V. Limits in standard type are for TJ = 25°C only, limits in
bold face type apply over the junction temperature (TJ) range of -40°C to +125°C. Minimum and maximum limits are
specified through test, design, or statistical correlation. Typical values represent the most likely parametric norm at
TJ = 25°C, and are provided for reference purposes only.
Symbol
Parameter
Conditions
Feedback Pin Voltage
VVIN = 4.5V to 36V
RHSW-DS(ON)
High-Side MOSFET On-Resistance
RLSW-DS(ON)
VFB
Min
Typ
Max
0.788
0.8
0.812
Units
V
ISW = 3A
130
225
mΩ
Low-Side MOSFET On-Resistance
ISW = 3A
110
190
mΩ
IQ
Operating Quiescent Current
VVIN = 4.5V to 36V
2.3
3
mA
ISD
Shutdown Quiescent Current
VEN = 0V
150
180
µA
VIN Under Voltage Lockout
Rising VVIN
VUVLO
VUVLO(HYS)
VVCC
4
VIN Under Voltage Lockout Hysteresis
4.25
4.5
V
350
450
mV
VCC Voltage
IVCC = -5 mA, VEN = 5V
ISS
Soft-Start Pin Source Current
VSS = 0V
VTRKACC
Soft-Start/Track Pin Accuracy
VSS = 0.4V
BOOT Diode Leakage
VBOOT = 4V
10
BOOT Diode Forward Voltage
IBOOT = -100 mA
0.9
1.1
V
Over Voltage Protection Rising Threshold
VFB(OVP) / VFB
110
112
%
Over Voltage Protection Hysteresis
ΔVFB(OVP) / VFB
2
3
%
PGOOD Threshold, VOUT Rising
VFB(PG) / VFB
95
97
%
PGOOD Hysteresis
ΔVFB(PG) / VFB
2
3
IBOOT
VF-BOOT
5.5
V
2
4.5
7
µA
-10
5
15
mV
nA
Powergood
VFB(OVP)
VFB(OVP-HYS)
VFB(PG)
VFB(PG-HYS)
TPGOOD
107
93
PGOOD Delay
20
IPGOOD(SNK)
PGOOD Low Sink Current
VPGOOD = 0.5V
IPGOOD(SRC)
PGOOD High Leakage Current
VPGOOD = 5V
FOSC
Oscillator Frequency
VSYNC = 0V
FOSCH
Maximum SYNC Frequency
0.6
%
µs
1
mA
5
200
nA
200
240
kHz
Oscillator
160
1500
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Electrical Characteristics (continued)
Unless otherwise stated, the following conditions apply: VVIN = 12V. Limits in standard type are for TJ = 25°C only, limits in
bold face type apply over the junction temperature (TJ) range of -40°C to +125°C. Minimum and maximum limits are
specified through test, design, or statistical correlation. Typical values represent the most likely parametric norm at
TJ = 25°C, and are provided for reference purposes only.
Symbol
FOSCL
Parameter
Conditions
Min
Minimum SYNC Frequency
VIH_SYNC
SYNC pin Logic High
VIL_SYNC
SYNC pin Logic Low
Typ
Max
250
Units
kHz
2
V
0.8
V
ISYNC
SYNC pin input leakage
VSYNC = 5V
10
nA
TMIN
Minimum Off Time
ILOAD = 3A
170
ns
Error Amplifier
Feedback Pin Bias Current
VFB = 1V
50
nA
ICOMP(SRC)
IFB
COMP Output Source Current
VFB = 0V
VCOMP = 0V
200
400
µA
ICOMP(SNK)
COMP Output Sink Current
VFB = 1.6V
VCOMP = 1.6V
200
350
µA
450
Error Amplifier DC Transconductance
ICOMP = -50 µA to +50 µA
AVOL
gm
Error Amplifier Voltage Gain
COMP pin open
2000
515
600
µmho
V/V
GBW
Error Amplifier Gain-Bandwidth Product
COMP pin open
7
MHz
Current Limit
ILIM
Cycle By Cycle Positive Current Limit
ILIMNEG
Cycle By Cycle Negative Current Limit
2.8
A
Cycle By Cycle Current Limit Delay
150
ns
TILIM
4.3
5.2
6.0
A
Enable
VEN(RISING)
EN Pin Rising Threshold
VEN(HYS)
EN Pin Hysteresis
IEN
EN Source Current
1.2
1.25
1.3
V
50
mV
2
µA
Thermal Shutdown
170
°C
Thermal Shutdown Hysteresis
20
°C
5.6
°C/W
27
°C/W
VEN = 0V, VVIN = 12V
Thermal Shutdown
TSD
TSD(HYS)
Thermal Resistance
θJC
θJA
(1)
4
Junction to Case
Junction to Ambient
(1)
0 LFM airflow
Measured on a 4 layer 2" x 2" PCB with 1 oz. copper weight inner layers and 2 oz. outer layers.
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Typical Performance Characteristics
Unless otherwise specified: VVIN = 12V, VOUT = 3.3V, L= 4.7µH, fSW=750kHz, CSS = 100nF, TA = 25°C for efficiency curves,
loop gain plots and waveforms, and TJ = 25°C for all others.
Efficiency
vs.
Load Current
fSW = 350 kHz
Efficiency
vs.
Load Current
fSW = 500 kHz
Figure 2.
Figure 3.
Efficiency
vs.
Load Current
fSW = 750 kHz
Error Amplifier Gain
Figure 4.
Figure 5.
Error Amplifier Phase
Line Regulation
Figure 6.
Figure 7.
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Typical Performance Characteristics (continued)
Unless otherwise specified: VVIN = 12V, VOUT = 3.3V, L= 4.7µH, fSW=750kHz, CSS = 100nF, TA = 25°C for efficiency curves,
loop gain plots and waveforms, and TJ = 25°C for all others.
6
Load Regulation
VCC
vs.
VIN
Figure 8.
Figure 9.
Non-Switching IQ
vs.
VIN
Shutdown IQ
vs.
VIN
Figure 10.
Figure 11.
PGOOD Output Low Level Voltage
vs.
IPGOOD
Enable Threshold and Hysteresis
vs.
Temperature
Figure 12.
Figure 13.
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Typical Performance Characteristics (continued)
Unless otherwise specified: VVIN = 12V, VOUT = 3.3V, L= 4.7µH, fSW=750kHz, CSS = 100nF, TA = 25°C for efficiency curves,
loop gain plots and waveforms, and TJ = 25°C for all others.
UVLO Threshold and Hysteresis
vs.
Temperature
Enable Current
vs.
Temperature
Figure 14.
Figure 15.
Clock Synchronization
High-Side FET Resistance
vs.
Temperature
Figure 16.
Figure 17.
Low-Side FET Resistance
vs.
Temperature
Load Transient Response
Figure 18.
Figure 19.
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Typical Performance Characteristics (continued)
Unless otherwise specified: VVIN = 12V, VOUT = 3.3V, L= 4.7µH, fSW=750kHz, CSS = 100nF, TA = 25°C for efficiency curves,
loop gain plots and waveforms, and TJ = 25°C for all others.
Peak Current Limit
vs.
Temperature
Startup with prebiased output
Figure 20.
Figure 21.
Startup with CSS = 0
Startup with CSS = 100 nF
Figure 22.
Figure 23.
Startup with applied Track Signal
Figure 24.
8
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Block Diagram
VCC_INT
BOOT
INTERNAL
+5.5V
REGULATOR
BOOT
VCC
ENABLE_INT
VIN
VCC
2.7V
UVLO
+
-
4.25V
+5.5V
REGULATOR
+2.7V
REGULATOR
SLOPE COMP
COMP
2.7V
CURRENT SENSE
+
4.5 PA
DISCHARGE
ERROR AMP
gm = 515 Pmho
SS/TRK
+
FB
DISCHARGE
CURRENT LIMIT
5.2A
VREF
+ 800 mV
-
BOOT
+
-
+
+
PWM COMPARATOR
880 mV
740 mV
+
-
PG-L
+
-
VCC_INT
NEGATIVE
CURRENT LIMIT
OVERVOLTAGE
UNDERVOLTAGE
+
-
CONTROL
LOGIC
-2.8A
SW
VCC
THERMAL
PROTECTION
2 PA
EN
1.25V
+
-
ENABLE_INT
PHASE LOCK
LOOP
GND
SYNC
PG-L
PGOOD
AGND
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OPERATION DESCRIPTION
GENERAL
The LM20333 switching regulator features all of the functions necessary to implement an efficient buck regulator
using a minimum number of external components. This easy to use regulator features two integrated switches
and is capable of supplying up to 3A of continuous output current. The regulator utilizes peak current mode
control with nonlinear slope compensation to optimize stability and transient response over the entire output
voltage range. Peak current mode control also provides inherent line feed-forward, cycle-by-cycle current limiting
and easy loop compensation. The switching frequency can be synchronized to an external oscillator over the
range of 250 kHz to 1.5 MHz. The SYNC function allows the device to operate at high switching frequencies
minimizing the size of the inductor while still achieving efficiencies as high as 94%. The precision internal voltage
reference allows the output to be set as low as 0.8V. Fault protection features include: current limiting, thermal
shutdown, over voltage protection, and shutdown capability. The device is available in the HTSSOP package
featuring an exposed pad to aid thermal dissipation. The typical application circuit for the LM20333 is shown in
Figure 25 in the design guide.
PRECISION ENABLE
The enable (EN) pin allows the output of the device to be enabled or disabled with an external control signal.
This pin is a precision analog input that enables the device when the voltage exceeds 1.25V (typical). The EN pin
has 50 mV of hysteresis and will disable the output when the enable voltage falls below 1.2V (typical). If the EN
pin is not used, it should be disconnected so the internal 2 µA pull-up will default this function to the enabled
condition. Since the enable pin has a precise turn-on threshold it can be used along with an external resistor
divider network from VIN to configure the device to turn-on at a precise input voltage. The precision enable
circuitry will remain active even when the device is disabled.
FREQUENCY SYNCHRONIZATION
The frequency sychronization pin(SYNC) allows the switching frequency of the device to be controlled with an
external clock signal. This feature allows the user to sychronize multiple converters, avoiding undesirable
frequency bands of operation.
The switching frequency of the device will synchronize to the rising edge of the clock source that is driving the
SYNC pin. The logic low level for the input clock must be below 0.8V and the logic high level must exceed 2.0V
to ensure proper operation. The device will synchronize to frequencies from 250 kHz to 1.5 MHz. If the
synchronization clock is removed or not present during startup, the oscillator of the device will run at
approximately 200 kHz. If the SYNC pin is not used it should be connected to ground.
PEAK CURRENT MODE CONTROL
In most cases, the peak current mode control architecture used in the LM20333 only requires two external
components to achieve a stable design. The compensation can be selected to accommodate any capacitor type
or value. The external compensation also allows the user to set the crossover frequency and optimize the
transient performance of the device.
For duty cycles above 50% all peak current mode control buck converters require the addition of an artificial
ramp to avoid sub-harmonic oscillation. This artificial linear ramp is commonly referred to as slope compensation.
What makes the LM20333 unique is the amount of slope compensation will change depending on the output
voltage. When operating at high output voltages the device will have more slope compensation than when
operating at lower output voltages. This is accomplished in the LM20333 by using a non-linear parabolic ramp for
the slope compensation. The parabolic slope compensation of the LM20333 is an improvement over the
traditional linear slope compensation because it optimizes the stability of the device over the entire output voltage
range.
10
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CURRENT LIMIT
The precise current limit enables the device to operate with smaller inductors that have lower saturation currents.
When the peak inductor current reaches the current limit threshold, an over current event is triggered and the
internal high-side FET turns off and the low-side FET turns on, allowing the inductor current to ramp down until
the next switching cycle. For each sequential over-current event, the reference voltage is decremented and PWM
pulses are skipped resulting in a current limit that does not aggressively fold back for brief over-current events,
while at the same time providing frequency and voltage foldback protection during hard short circuit conditions.
SOFT-START AND VOLTAGE TRACKING
The SS/TRK pin is a dual function pin that can be used to set the startup time or track an external voltage
source. The startup or soft-start time can be adjusted by connecting a capacitor from the SS/TRK pin to ground.
The soft-start feature allows the regulator output to gradually reach the steady state operating point, thus
reducing stresses on the input supply and controlling startup current. If no soft-start capacitor is used the device
defaults to the internal soft-start circuitry resulting in a startup time of approximately 1 ms. For applications that
require a monotonic startup or utilize the PGOOD pin, an external soft-start capacitor is recommended. The
SS/TRK pin can also be set to track an external voltage source. The tracking behavior can be adjusted by two
external resistors connected to the SS/TRK pin as shown in Figure 30 in the design guide.
PRE-BIAS STARTUP CAPABILITY
The LM20333 is in a pre-biased state when it starts up with an output voltage greater than zero. This often
occurs in many multi-rail applications such as when powering an FPGA, ASIC, or DSP. In these applications the
output can be pre-biased through parasitic conduction paths from one supply rail to another. Even though the
LM20333 is a synchronous converter, it will not pull the output low when a pre-bias condition exists. During start
up the LM20333 will not sink current until the soft-start voltage exceeds the voltage on the FB pin. Since the
device cannot sink current, it protects the load from damage that might otherwise occur if current is conducted
through the parasitic paths of the load.
POWER GOOD AND OVER VOLTAGE FAULT HANDLING
The LM20333 has built in under and over voltage comparators that control the power switches. Whenever there
is an excursion in output voltage above the set OVP threshold, the part will terminate the present on-pulse, turnon the low-side FET, and pull the PGOOD pin low. The low-side FET will remain on until either the FB voltage
falls back into regulation or the negative current limit is triggered which in turn tri-states the FETs. If the output
reaches the UVP threshold the part will continue switching and the PGOOD pin will be deasserted and go low.
Typical values for the PGOOD resistor are on the order of 100 kΩ or less. To avoid false tripping during transient
glitches the PGOOD pin has 20 µs of built in deglitch time to both rising and falling edges.
UVLO
The LM20333 has an internal under-voltage lockout protection circuit that keeps the device from switching until
the input voltage reaches 4.25V (typical). The UVLO threshold has 350 mV of hysteresis that keeps the device
from responding to power-on glitches during start up. If desired the turn-on point of the supply can be changed
by using the precision enable pin and a resistor divider network connected to VIN as shown in Figure 29 in the
design guide.
THERMAL PROTECTION
Internal thermal shutdown circuitry is provided to protect the integrated circuit in the event that the maximum
junction temperature is exceeded. When activated, typically at 170°C, the LM20333 tri-states the power FETs
and resets soft-start. After the junction cools to approximately 150°C, the part starts up using the normal start up
routine. This feature is provided to prevent catastrophic failures from accidental device overheating.
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Design Guide
This section walks the designer through the steps necessary to select the external components to build a fully
functional power supply. As with any DC-DC converter numerous trade-offs are possible to optimize the design
for efficiency, size, or performance. These will be taken into account and highlighted throughout this discussion.
To facilitate component selection discussions the circuit shown in Figure 25 below may be used as a reference.
Unless otherwise indicated all formulas assume units of amps (A) for current, farads (F) for capacitance, henries
(H) for inductance and volts (V) for voltages.
LM20333
BOOT
VIN
VIN
CIN1
CBOOT
L
VOUT
SW
D1
(Optional)
CIN2
EN
FB
VPULLUP
SYNC
COMP
RC1
CC1
CSS
PGOOD
SS/TRK
VCC
AGND GND
RFB1
COUT
RFB2
RPG
CVCC
Figure 25. Typical Application Circuit
The first equation to calculate for any buck converter is duty-cycle. Ignoring conduction losses associated with
the FETs and parasitic resistances it can be approximated by:
D=
VOUT
VIN
(1)
INDUCTOR SELECTION (L)
The inductor value is determined based on the operating frequency, load current, ripple current and duty cycle.
The inductor selected should have a saturation current rating greater than the peak current limit of the device.
Keep in mind the specified current limit does not account for delay of the current limit comparator, therefore the
current limit in the application may be higher than the specified value. To optimize the performance and prevent
the device from entering current limit at maximum load, the inductance is typically selected such that the ripple
current, ΔiL, is not greater than 30% of the rated output current. Figure 26 illustrates the switch and inductor
ripple current waveforms. Once the input voltage, output voltage, operating frequency and desired ripple current
are known, the minimum value for the inductor can be calculated by the formula shown below:
LMIN =
12
(VIN - VOUT) x D
'iL x fSW
(2)
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VSW
VIN
Time
IL
IL AVG = IOUT
'iL
Time
Figure 26. Switch and Inductor Current Waveforms
If needed, slightly smaller value inductors can be used, however, the peak inductor current, IOUT + ΔiL/2, should
be kept below the peak current limit of the device. In general, the inductor ripple current, ΔiL, should be more
than 10% of the rated output current to provide adequate current sense information for the current mode control
loop. If the ripple current in the inductor is too low, the control loop will not have sufficient current sense
information and can be prone to instability.
OUTPUT CAPACITOR SELECTION (COUT)
The output capacitor, COUT, filters the inductor ripple current and provides a source of charge for transient load
conditions. A wide range of output capacitors may be used with the LM20333 that provide excellent performance.
The best performance is typically obtained using ceramic, SP or OSCON type chemistries. Typical trade-offs are
that the ceramic capacitor provides extremely low ESR to reduce the output ripple voltage and noise spikes,
while the SP and OSCON capacitors provide a large bulk capacitance in a small volume for transient loading
conditions.
When selecting the value for the output capacitor, the two performance characteristics to consider are the output
voltage ripple and transient response. The output voltage ripple can be approximated by using the following
formula:
'VOUT = 'iL x RESR +
1
8 x fSW x COUT
where
•
•
•
•
ΔVOUT (V) is the amount of peak to peak voltage ripple at the power supply output
RESR (Ω) is the series resistance of the output capacitor
fSW(Hz) is the switching frequency
COUT (F) is the output capacitance used in the design
(3)
The amount of output ripple that can be tolerated is application specific; however a general recommendation is to
keep the output ripple less than 1% of the rated output voltage. Keep in mind ceramic capacitors are sometimes
preferred because they have very low ESR; however, depending on package and voltage rating of the capacitor
the value of the capacitance can drop significantly with applied voltage. The output capacitor selection will also
affect the output voltage droop during a load transient. The peak droop on the output voltage during a load
transient is dependent on many factors; however, an approximation of the transient droop ignoring loop
bandwidth can be obtained using the following equation:
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VDROOP = 'IOUTSTEP x RESR +
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L x 'IOUTSTEP2
COUT x (VIN - VOUT)
where
•
•
•
•
•
•
•
COUT (F) is the minimum required output capacitance
L (H) is the value of the inductor
VDROOP (V) is the output voltage drop ignoring loop bandwidth considerations
ΔIOUTSTEP (A) is the load step change
RESR (Ω) is the output capacitor ESR
VIN (V) is the input voltage
VOUT (V) is the set regulator output voltage
(4)
Both the tolerance and voltage coefficient of the capacitor should be examined when designing for a specific
output ripple or transient droop target.
INPUT CAPACITOR SELECTION
Good quality input capacitors are necessary to limit the ripple voltage at the VIN pin while supplying most of the
switch current during the on-time. In general it is recommended to use a ceramic capacitor for the input as they
provide both a low impedance and small footprint. One important note is to use a good dielectric for the ceramic
capacitor such as X5R or X7R. These provide better over temperature performance and also minimize the DC
voltage derating that occurs on Y5V capacitors. The input capacitors CIN1 and CIN2 should be placed as close as
possible to the VIN and GND pins on both sides of the device.
Non-ceramic input capacitors should be selected for RMS current rating and minimum ripple voltage. A good
approximation for the required ripple current rating is given by the relationship:
IIN-RMS = IOUT D(1 - D)
(5)
As indicated by the RMS ripple current equation, highest requirement for RMS current rating occurs at 50% duty
cycle. For this case, the RMS ripple current rating of the input capacitor should be greater than half the output
current. For best performance, low ESR ceramic capacitors should be placed in parallel with higher capacitance
capacitors to provide the best input filtering for the device.
SETTING THE OUTPUT VOLTAGE (RFB1, RFB2)
The resistors RFB1 and RFB2 are selected to set the output voltage for the device. Table 1 provides suggestions
for RFB1 and RFB2 for common output voltages.
Table 1. Suggested Values for RFB1 and RFB2
RFB1(kΩ)
RFB2(kΩ)
VOUT
short
open
0.8
4.99
10
1.2
8.87
10.2
1.5
12.7
10.2
1.8
21.5
10.2
2.5
31.6
10.2
3.3
52.3
10
5.0
If different output voltages are required, RFB2 should be selected to be between 4.99 kΩ to 49.9 kΩ and RFB1 can
be calculated using the equation below.
RFB1 =
14
VOUT
0.8
- 1 x RFB2
(6)
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LOOP COMPENSATION (RC1, CC1)
The purpose of loop compensation is to meet static and dynamic performance requirements while maintaining
adequate stability. Optimal loop compensation depends on the output capacitor, inductor, load and the device
itself. Table 2 below gives values for the compensation network that will result in a stable system when using a
150 µF, 6.3V POSCAP output capacitor (6TPB150MAZB).
Table 2. Recommended Compensation for
COUT = 150 µF, IOUT = 3A, fSW= 500kHz
VIN
VOUT
L (µH)
RC (kΩ)
CC1 (nF)
12
5
6.8
30.9
4.7
12
3.3
5.6
33.2
3.3
12
2.5
4.7
40.2
2.2
12
1.5
3.3
22.1
2.2
12
1.2
2.2
18.2
2.2
12
0.8
1.5
8.45
3.3
5
3.3
2.2
38.3
2.2
5
2.5
3.3
38.3
2.2
5
1.5
2.2
30.1
2.2
5
1.2
2
18.2
2.2
5
0.8
1.5
13
2.2
Output Filter Pole, fP(FIL)
AM
0 dB
Output Filter Zero, fZ(FIL)
Complex Double Pole, fP(MOD)
Modulator and Output Filter
Transfer Function
If the desired solution differs from the table above the loop transfer function should be analyzed to optimize the
loop compensation. The overall loop transfer function is the product of the power stage and the feedback network
transfer functions. For stability purposes, the objective is to have a loop gain slope that is -20dB/decade from a
very low frequency to beyond the crossover frequency. Figure 27 shows the transfer functions for power stage,
feedback/compensation network, and the resulting compensated loop for the LM20333.
Pole, fP2(EA)
0 dB
Error Amp Zero, fZ(EA)
AEA + AM
Error Amp Pole, fP(EA)
0 dB
Complex Double Pole, fP(MOD)
fC
Error Amplifier
Transfer Function
Optional Error Amp
Compensated Open
Loop Transfer Function
GAIN (dB)
Error Amp Pole, fP1(EA)
AEA
fSW/2
FREQUENCY (Hz)
Figure 27. LM20333 Loop Compensation
The power stage transfer function is dictated by the modulator, output LC filter, and load; while the feedback
transfer function is set by the feedback resistor ratio, error amp gain and external compensation network.
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To achieve a -20dB/decade slope, the error amplifier zero, located at fZ(EA), should be positioned to cancel the
output filter pole (fP(FIL)).
Compensation of the LM20333 is achieved by adding an RC network as shown in Figure 28 below.
LM20333
COMP
RC1
CC2
(optional)
CC1
Figure 28. Compensation Network for LM20333
A good starting value for CC1 for most applications is 2.2 nF. Once the value of CC1 is chosen the value of RC
should be approximated using the equation below to cancel the output filter pole (fP(FIL)) as shown in Figure 27.
RC1 =
CC1
IOUT
2xD
x
+
COUT
VOUT fSW x L
-1
(7)
A higher crossover frequency can be obtained, usually at the expense of phase margin, by lowering the value of
CC1 and recalculating the value of RC1. Likewise, increasing CC1 and recalculating RC1 will provide additional
phase margin at a lower crossover frequency. As with any attempt to compensate the LM20333 the stability of
the system should be verified for desired transient droop and settling time.
For low duty cycle operation, when the on time of the switch node is less than 200ns, an additional capacitor
(CC2) should be added from the COMP pin to AGND. The recommended value of this capacitor is 20pF. If low
duty cycle jitter on the switch node is observed, the value of this capacitor can be increased to improve noise
immunity; however, values much larger than 100pF will cause the pole fP2(EA) to move to a lower frequency
degrading loop stability.
BOOT CAPACITOR (CBOOT)
The LM20333 integrates an N-channel buck switch and associated floating high voltage level shift / gate driver.
This gate driver circuit works in conjunction with an internal diode and an external bootstrap capacitor. A 0.1 µF
ceramic capacitor, connected with short traces between the BOOT pin and SW pin, is recommended. During the
off-time of the buck switch, the SW pin voltage is approximately 0V and the bootstrap capacitor is charged from
VCC through the internal bootstrap diode.
SUB-REGULATOR BYPASS CAPACITOR (CVCC)
The capacitor at the VCC pin provides noise filtering for the internal sub-regulator. The recommended value of
CVCC should be no smaller than 0.1 µF and no greater than 1 µF. The capacitor should be a good quality ceramic
X5R or X7R capacitor. In general, a 1 µF ceramic capacitor is recommended for most applications. The VCC
regulator should not be used for other functions since it isn't protected against short circuit.
SETTING THE START UP TIME (CSS)
The addition of a capacitor connected from the SS pin to ground sets the time at which the output voltage will
reach the final regulated value. Larger values for CSS will result in longer start up times. Table 3, shown below
provides a list of soft start capacitors and the corresponding typical start up times.
Table 3. Start Up Times for Different Soft-Start Capacitors
16
Start Up Time (ms)
CSS (nF)
1
none
5
33
10
68
15
100
20
120
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If different start up times are needed the equation shown below can be used to calculate the start up time.
tSS =
0.8V x CSS
ISS
22222
(8)
As shown above, the start up time is influenced by the value of the soft-start capacitor CSS and the 4.5 µA softstart pin current ISS.
While the soft-start capacitor can be sized to meet many start up requirements, there are limitations to its size.
The soft-start time can never be faster than 1 ms due to the internal default 1 ms start up time. When the device
is enabled there is an approximate time interval of 50 µs when the soft-start capacitor will be discharged just
prior to the soft-start ramp. If the enable pin is rapidly pulsed or the soft-start capacitor is large there may not be
enough time for CSS to completely discharge resulting in start up times less than predicted. To aid in discharging
of soft-start capacitor during long disable periods an external 1MΩ resistor from SS/TRK to ground can be used
without greatly affecting the start up time.
USING PRECISION ENABLE AND POWER GOOD
The precision enable (EN) and power good (PGOOD) pins of the LM20333 can be used to address many
sequencing requirements. The turn-on of the LM20333 can be controlled with the precision enable pin by using
two external resistors as shown in Figure 29 .
External
Power Supply
VOUT1
LM20333
RA
VOUT2
EN
RB
Figure 29. Sequencing LM20333 with Precision Enable
The value for resistor RB can be selected by the user to control the current through the divider. Typically this
resistor will be selected to be between 10 kΩ and 49.9 kΩ. Once the value for RB is chosen the resistor RA can
be solved using the equation below to set the desired turn-on voltage.
RA =
VTO
VIH_EN
- 1 x RB
(9)
When designing for a specific turn-on threshold (VTO) the tolerance on the input supply, enable threshold
(VIH_EN), and external resistors need to be considered to ensure proper turn-on of the device.
The LM20333 features an open drain power good (PGOOD) pin to sequence external supplies or loads and to
provide fault detection. This pin requires an external resistor (RPG) to pull PGOOD high when the output is within
the PGOOD tolerance window. Typical values for this resistor range from 10 kΩ to 100 kΩ.
TRACKING AN EXTERNAL SUPPLY
By using a properly chosen resistor divider network connected to the SS/TRK pin, as shown in Figure 30, the
output of the LM20333 can be configured to track an external voltage source to obtain a simultaneous or
ratiometric start up.
External
Power Supply
EN
VOUT1
R1
LM20333
VOUT2
SS/TRK
R2
Figure 30. Tracking an External Supply
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Since the soft-start charging current ISS is always present on the SS/TRK pin, the size of R2 should be less than
10 kΩ to minimize the errors in the tracking output. Once a value for R2 is selected the value for R1 can be
calculated using appropriate equation in Figure 31, to give the desired start up. Figure 30 shows two common
start up sequences; the top waveform shows a simultaneous start up while the waveform at the bottom illustrates
a ratiometric start up.
SIMULTANEOUS START UP
VOLTAGE
VOUT1
VOUT2
§VOUT2 ·
-1¸¸ x R2
R1 = ¨¨
© 0.8V ¹
VEN
VOUT2 < 0.8 x VOUT1
TIME
RATIOMETRIC START UP
VOUT1
VOLTAGE
VOUT2
R1 = ( VOUT1 -1) x R2
VEN
TIME
Figure 31. Common Start Up Sequences
A simultaneous start up is preferred when powering most FPGAs, DSPs, or other microprocessors. In these
systems the higher voltage, VOUT1, usually powers the I/O, and the lower voltage, VOUT2, powers the core. A
simultaneous start up provides a more robust power up for these applications since it avoids turning on any
parasitic conduction paths that may exist between the core and the I/O pins of the processor.
The second most common power on behavior is known as a ratiometric start up. This start up is preferred in
applications where both supplies need to be at the final value at the same time.
Similar to the soft-start function, the fastest start up possible is 1ms regardless of the rise time of the tracking
voltage. When using the track feature the final voltage seen by the SS/TRACK pin should exceed 1V to provide
sufficient overdrive and transient immunity.
BENEFIT OF AN EXTERNAL SCHOTTKY
The LM20333 employs a 40ns dead time between conduction of the control and synchronous FETs in order to
avoid the situation where both FETs simultaneously conduct, causing shoot-through current. During the dead
time, the body diode of the synchronous FET acts as a free-wheeling diode and conducts the inductor current.
The structure of the high voltage DMOS is optimized for high breakdown voltage, but this typically leads to
inefficient body diode conduction due to the reverse recovery charge. The loss associated with the reverse
recovery of the body diode of the synchronous FET manifests itself as a loss proportional to load current and
switching frequency. The additional efficiency loss becomes apparent at higher input voltages and switching
frequencies. One simple solution is to use a small 1A external Schottky diode between SW and GND as shown
in Figure 38. The external Schottky diode effectively conducts all inductor current during the dead time,
minimizing the current passing through the synchronous MOSFET body diode and eliminating reverse recovery
losses.
18
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The external Schottky conducts currents for a very small portion of the switching cycle, therefore the average
current is low. An external Schottky rated for 1A will improve efficiency by several percent in some applications.
A Schottky rated at a higher current will not significantly improve efficiency and may be worse due to the
increased reverse capacitance. The forward voltage of the synchronous MOSFET body diode is approximately
700 mV, therefore an external Schottky with a forward voltage less than or equal to 700 mV should be selected
to ensure the majority of the dead time current is carried by the Schottky.
THERMAL CONSIDERATIONS
The thermal characteristics of the LM20333 are specified using the parameter θJA, which relates the junction
temperature to the ambient temperature. Although the value of θJA is dependant on many variables, it still can be
used to approximate the operating junction temperature of the device.
To obtain an estimate of the device junction temperature, one may use the following relationship:
TJ = PD x θJA + TA
(10)
and
PD = PIN x (1 - Efficiency) - 1.1 x (IOUT)2 x DCR
where
•
•
•
•
•
•
TJ is the junction temperature in °C
PIN is the input power in Watts (PIN = VIN x IIN)
θJA is the junction to ambient thermal resistance for the LM20333
TA is the ambient temperature in °C
IOUT is the output load current
DCR is the inductor series resistance
(11)
It is important to always keep the operating junction temperature (TJ) below 125°C for reliable operation. If the
junction temperature exceeds 170°C the device will cycle in and out of thermal shutdown. If thermal shutdown
occurs it is a sign of inadequate heatsinking or excessive power dissipation in the device.
Figure 32, Figure 33, Figure 34 and Figure 35 can be used as a guide to avoid exceeding the maximum junction
temperature of 125°C provided an external 1A Schottky diode, such as Central Semiconductor's CMMSH1-40NST, is used to improve reverse recovery losses.
Figure 32. Safe Thermal Operating Areas (IOUT = 3A, fSW = 350kHz)
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Figure 33. Safe Thermal Operating Areas (IOUT = 3A, fSW = 500kHz)
Figure 34. Safe Thermal Operating Areas (IOUT = 3A, fSW = 750kHz)
Figure 35. Safe Thermal Operating Areas (IOUT = 2.5A, fSW = 500kHz)
The dashed lines in the figures above show an approximation of the minimum and maximum duty cycle
limitations; while, the solid lines define areas of operation for a given ambient temperature. This data for the
figure was derived assuming the device is operating at 3A continuous output current on a 4 layer PCB with a
copper area greater than 4 square inches exhibiting a thermal characteristic less than 27 °C/W. Since the
internal losses are dominated by the FETs a slight reduction in current by 500mA allows for much larger regions
of operation, as shown in Figure 35.
20
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Figure 36, shown below, provides a better approximation of the θJA for a given PCB copper area. The PCB used
in this test consisted of 4 layers: 1oz. copper was used for the internal layers while the external layers were
plated to 2oz. copper weight. To provide an optimal thermal connection, a 5 x 4 array of 12 mil thermal vias
located under the thermal pad was used to connect the 4 layers.
Figure 36. Thermal Resistance vs PCB Area (4 Layer Board)
PCB LAYOUT CONSIDERATIONS
PC board layout is an important part of DC-DC converter design. Poor board layout can disrupt the performance
of a DC-DC converter and surrounding circuitry by contributing to EMI, ground bounce, and resistive voltage loss
in the traces. These can send erroneous signals to the DC-DC converter resulting in poor regulation or instability.
Good layout can be implemented by following a few simple design rules.
1. Minimize area of switched current loops. In a buck regulator there are two loops where currents are switched
at high slew rates. The first loop starts from the input capacitor, to the regulator VIN pin, to the regulator SW pin,
to the inductor then out to the output capacitor and load. The second loop starts from the output capacitor
ground, to the regulator GND pins, to the inductor and then out to the load (see Figure 37). To minimize both
loop areas the input capacitor should be placed as close as possible to the VIN pin. Grounding for both the input
and output capacitor should consist of a small localized top side plane that connects to GND and the exposed
pad (EP). The inductor should be placed as close as possible to the SW pin and output capacitor.
2. Minimize the copper area of the switch node. Since the LM20333 has the SW pins on opposite sides of the
package it is recommended that the SW pins should be connected with a trace that runs around the package.
The inductor should be placed at an equal distance from the SW pins using 100 mil wide traces to minimize
capacitive and conductive losses.
3. Have a single point ground for all device grounds located under the EP. The ground connections for the
compensation, feedback, and soft-start components should be connected together then routed to the EP pin of
the device. The AGND pin should connect to GND under the EP. If not properly handled poor grounding can
result in degraded load regulation or erratic switching behavior.
4. Minimize trace length to the FB pin. Since the feedback node can be high impedance the trace from the output
resistor divider to FB pin should be as short as possible. This is most important when high value resistors are
used to set the output voltage. The feedback trace should be routed away from the SW pin and inductor to avoid
contaminating the feedback signal with switch noise.
5. Make input and output bus connections as wide as possible. This reduces any voltage drops on the input or
output of the converter and can improve efficiency. Voltage accuracy at the load is important so make sure
feedback voltage sense is made at the load. Doing so will correct for voltage drops at the load and provide the
best output accuracy.
6. Provide adequate device heatsinking. For most 3A designs a four layer board is recommended. Use as many
vias as is possible to connect the EP to the power plane heatsink. For best results use a 5x4 via array with a
minimum via diameter of 12 mils. "Via tenting" with the solder mask may be necessary to prevent wicking of the
solder paste applied to the EP. See the THERMAL CONSIDERATIONS section to ensure enough copper
heatsinking area is used to keep the junction temperature below 125°C.
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LM20333
L
SW
PVIN
VOUT
CIN
COUT
PGND
LOOP1
LOOP2
Figure 37. Schematic of LM20333 Highlighting Layout Sensitive Nodes
VIN
C1
C3
C2
GND
C2, C3 should be placed at VIN
pins 5,6 and 15,16 respectively.
U1
VOUT
18
C4
VCC
R1
3
ENABLE
19
20
1
4
2
12
C8
SYNC
C6
PGOOD BOOT
EN
SYNC
SS
COMP
FB
LM20333
PGOOD
5
VIN
6
VIN
15
VIN
16
VIN
AGND
17
7
SW
8
SW
13
SW
14
SW
9
GND
10
GND
11
GND
C5
L1
D1
(OPTIONAL
for improved
Efficiency)
VOUT
C9
GND
EP
R3
C7
R2
R4
Figure 38. Typical Application Schematic
22
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Table 4. Bill of Materials (VIN = 12V, VOUT = 3.3V, IOUT = 3A, fSW = 500kHz)
ID
Qty
Part Number
Size
Description
Vendor
U1
1
LM20333MH
HTSSOP
IC, Switching Regulator
TI
C1
1
C3225X5R1E226M
1210
22µF, X5R, 25V, 20%
TDK
C2, C3
2
GRM21BR61E475KA12L
0805
4.7µF, X5R, 25V, 10%
MuRata
C5, C6
1
C1608X7R1H104K
0603
100nF, X7R, 50V, 10%
TDK
C4
1
C1608X5R1A105K
0603
1µF, X7R, 10V, 10%
TDK
C7
1
C1608C0G1H100J
0603
10pF, C0G, 50V, 5%
TDK
C8
1
C1608C0G1H152J
0603
1.5nF, C0G, 50V, 5%
TDK
C9
1
6TPB150MAZB
B
150µF,POSCAP, 6.3V, 20%
Sanyo
D1
1
CMMSH1-40-NST
SOD123
Vr = 40V, Io = 1A, Vf = 0.55V
Central
Semiconductor
L1
1
IHLP4040DZER5R6M01
IHLP4040
5.6µH, 0.018 Ohms, 16A
Vishay
R1, R4
2
CRCW06031002F
0603
10kΩ, 1%
Vishay
R2
1
CRCW06031502F
0603
15kΩ, 1%
Vishay
R3
1
CRCW06033092F
0603
30.9kΩ, 1%
Vishay
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REVISION HISTORY
Changes from Revision C (April 2013) to Revision D
•
24
Page
Changed layout of National Data Sheet to TI format .......................................................................................................... 23
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PACKAGE OPTION ADDENDUM
www.ti.com
10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
LM20333MH/NOPB
ACTIVE
HTSSOP
PWP
20
73
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 125
20333MH
LM20333MHE/NOPB
ACTIVE
HTSSOP
PWP
20
250
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 125
20333MH
LM20333MHX/NOPB
ACTIVE
HTSSOP
PWP
20
2500
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 125
20333MH
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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