LM2696MXA

LM2696 www.ti.com SNVS375B – OCTOBER 2005 – REVISED APRIL 2013 LM2696 3A, Constant On Time Buck Regulator Check for Samples: LM2696 FEATURES DESCRIPTION • • • • • • • • • The LM2696 is a pulse width modulation (PWM) buck regulator capable of delivering up to 3A into a load. The control loop utilizes a constant on-time control scheme with input voltage feed forward. This provides a topology that has excellent transient response without the need for compensation. The input voltage feed forward ensures that a constant switching frequency is maintained across the entire VIN range. 1 2 Input Voltage Range of 4.5V–24V Constant On-Time No Compensation Needed Maximum Load Current of 3A Switching Frequency of 100 kHz–500 kHz Constant Frequency Across Input Range TTL Compatible Shutdown Thresholds Low Standby Current of 12 µA 130 mΩ Internal MOSFET Switch The LM2696 is capable of switching frequencies in the range of 100 kHz to 500 kHz. Combined with an integrated 130 mΩ high side NMOS switch the LM2696 can utilize small sized external components and provide high efficiency. An internal soft-start and power-good flag are also provided to allow for simple sequencing between multiple regulators. APPLICATIONS • • • High Efficiency Step-Down Switching Regulators LCD Monitors Set-Top Boxes The LM2696 is available with an adjustable output in an exposed pad HTSSOP-16 package. Typical Application Circuit LM2696 VPGOOD PGOOD EXTVCC VSD CEXT SD CSD RON CBOOT RON VIN CBOOT AVIN L SW VOUT PVIN CIN CAVIN CSS RFB1 GND SS DCATCH FB COUT RFB2 1 2 Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. All trademarks are the property of their respective owners. PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of the Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. Copyright © 2005–2013, Texas Instruments Incorporated LM2696 SNVS375B – OCTOBER 2005 – REVISED APRIL 2013 www.ti.com Connection Diagram Top View SW 1 16 PVIN SW 2 15 PVIN SW 3 14 PVIN CBOOT 4 13 SD AVIN 5 12 RON EXTVCC 6 11 PGOOD FB 7 10 SS N/C 8 9 GND Figure 1. HTSSOP-16 Package See Package Number PWP0016A PIN DESCRIPTIONS Pin # Name Function 1, 2, 3 SW 4 CBOOT 5 AVIN 6 EXTVCC 7 FB Feedback signal from output 8 N/C No connect Ground 9 GND 10 SS 11 PGOOD 12 RON 13 SD 14, 15, 16 PVIN - Exposed Pad Switching node Bootstrap capacitor input Analog voltage input Output of internal regulator for decoupling Soft-start pin Power-good flag, open drain output Sets the switch on-time dependent on current Shutdown pin Power voltage input Must be connected to ground These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam during storage or handling to prevent electrostatic damage to the MOS gates. 2 Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: LM2696 LM2696 www.ti.com SNVS375B – OCTOBER 2005 – REVISED APRIL 2013 ABSOLUTE MAXIMUM RATINGS (1) (2) (3) Voltages from the indicated pins to GND AVIN −0.3V to +26V PVIN −0.3V to (AVIN+0.3V) −0.3V to +33V CBOOT −0.3V to +7V CBOOT to SW −0.3V to +7V FB, SD, SS, PGOOD −65°C to +150°C Storage Temperature Range Junction Temperature +150°C Lead Temperature (Soldering, 10 sec.) 260°C Minimum ESD Rating 1.5 kV (1) Absolute Maximum Ratings indicate limits beyond which damage may occur to the device. Operating Ratings indicate conditions for which the device is intended to be functional, but do not ensure specific performance limits. For ensured specifications, see Electrical Characteristics. If Military/Aerospace specified devices are required, please contact the TI Sales Office/ Distributors for availability and specifications. Without PCB copper enhancements. The maximum power dissipation must be derated at elevated temperatures and is limited by TJMAX (maximum junction temperature), θJ-A (junction to ambient thermal resistance) and TA (ambient temperature). The maximum power dissipation at any temperature is: PDissMAX = (TJMAX - TA) /θJ-A up to the value listed in the Absolute Maximum Ratings. θJ-A for HTSSOP16 package is 38.1°C/W, TJMAX = 125°C. (2) (3) OPERATING RANGE −40°C to +125°C Junction Temperature AVIN to GND 4.5V to 24V PVIN 4.5V to 24V ELECTRICAL CHARACTERISTICS Specifications with standard typeface are for TJ = 25°C, and those in boldface type apply over the full Operating Temperature Range (TJ = −40°C to +125°C). Minimum and Maximum limits are specified through test, design or statistical correlation. Typical values represent the most likely parametric norm at TJ = 25°C and are provided for reference purposes only. Unless otherwise specified VIN = 12V. Symbol Parameter Condition VFB Feedback Pin Voltage VIN = 4.5V to 24V ISW = 0A to 3A ICL Switch Current Limit VCBOOT = VSW + 5V RDS_ON Switch On Resistance ISW = 3A IQ Operating Quiescent Current VFB = 1.5V VUVLO AVIN Under Voltage Lockout Rising VIN VUVLO HYS Typ Max Units 1.225 1.254 1.282 V 3.6 Shutdown Quiescent Current VSD = 0V kON Switch On-Time Constant ION = 50 µA to 100 µA VD ON RON Voltage TOFF_MIN Minimum Off Time TON Minimum On-time 4.9 6.4 A 0.13 0.22 Ω 1.3 2 mA 4.125 4.3 V 60 120 mV 12 25 µA 50 66 82 µA µs 0.35 0.65 0.95 V 165 12 250 30 ns µs 3.9 AVIN Under Voltage Lockout Hysteresis ISD MIN Min FB = 1.24V FB = 0V 400 VEXTV EXTVCC Voltage ΔVEXTV EXTVCC Load Regulation IEXTV = 0 µA to 50 µA 3.30 VPWRGD PGOOD Threshold (PGOOD Transition from Low to High) With respect to VFB 91.5 VPG_HYS PGOOD Hysteresis IOL PGOOD Low Sink Current VPGOOD = 0.4V 0.5 IOH PGOOD High Leakage Current IFB Feedback Pin Bias Current VFB = 1.2V ISS_SOURCE Soft-Start Pin Source Current VSS = 0V ns 3.65 4.00 V 0.03 0.5 % 93.5 95.5 % 1 2.1 mA 50 nA 0 0.7 1 nA 1.4 Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: LM2696 % 2 µA 3 LM2696 SNVS375B – OCTOBER 2005 – REVISED APRIL 2013 www.ti.com ELECTRICAL CHARACTERISTICS (continued) Specifications with standard typeface are for TJ = 25°C, and those in boldface type apply over the full Operating Temperature Range (TJ = −40°C to +125°C). Minimum and Maximum limits are specified through test, design or statistical correlation. Typical values represent the most likely parametric norm at TJ = 25°C and are provided for reference purposes only. Unless otherwise specified VIN = 12V. Symbol Parameter Condition Min Typ ISS SINK Soft-Start Pin Sink Current VSS = 1.2V VSD = 0V 15 ISD Shutdown Pull-Up Current VSD = 0V 1 VIH SD Pin Minimum High Input Level VIL SD Pin Maximum Low Input Level θJ-A Thermal Resistance 4 Max mA 3 1.8 µA V 0.6 35.1 Submit Documentation Feedback Units V °C/W Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: LM2696 LM2696 www.ti.com SNVS375B – OCTOBER 2005 – REVISED APRIL 2013 TYPICAL PERFORMANCE CHARACTERISTICS IQ vs VIN 1.5 1.30 1.45 1.28 1.4 1.26 1.35 IQ (PA) IQ (mA) IQ vs Temp 1.32 1.24 1.22 1.3 1.25 1.2 1.2 1.18 1.15 1.16 1.1 1.14 1.05 1.12 -40 -20 0 20 40 60 1 4.5 80 100 120 7.5 10.5 13.5 16.5 19.5 22.5 TEMPERATURE (oC) Figure 2. Figure 3. IQ in Shutdown vs Temp IQ vs VIN in Shutdown 20 14 18 12 16 14 10 12 IQ (PA) IQ (SHUTDOWN) (PA) 16 24 VIN (V) 8 6 10 8 6 4 4 2 2 0 -40 -20 0 20 40 60 80 0 4.5 100 120 o 7.5 10.5 13.5 16.5 19.5 22.5 TEMPERATURE ( C) Figure 4. Figure 5. Shutdown Thresholds vs Temp EXTVCC vs Temp 1.5 3.67 1.4 3.665 1.3 VIH (V) 3.66 1.2 1.1 EXT VCC (V) VIL AND VIH (V) 24 VIN (V) VIL (V) 1.0 0.9 3.655 3.65 3.645 0.8 3.64 0.7 0.6 -60 -40 -20 0 20 40 60 80 100 120 140 3.635 -40 -20 0 20 40 60 80 100 120 TEMPERATURE (oC) TEMPERATURE (oC) Figure 6. Figure 7. Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: LM2696 5 LM2696 SNVS375B – OCTOBER 2005 – REVISED APRIL 2013 www.ti.com TYPICAL PERFORMANCE CHARACTERISTICS (continued) EXTVCC vs VIN EXTVCC vs Load Current 3.660 3.66 3.658 3.65 EXT VCC (V) EXTVCC (V) 3.656 3.654 3.652 3.650 3.64 3.63 3.648 3.646 3.644 4.5 3.62 10.5 13.5 16.5 19.5 22.5 7.5 0 24 10 20 66.5 1.256 66.3 1.255 66.1 TON . ION (Ps . PA) VFB (V) kON vs Temp 1.257 1.254 1.253 1.252 1.251 1.25 65.9 65.7 65.5 65.3 65.1 1.249 64.9 1.248 -40 -20 64.7 -40 -20 0 20 40 60 80 100 120 0 o Figure 10. Switch ON Time vs RON Pin Current 178 2 176 TOFF (ns) 2.5 1.5 172 0.5 170 0 200 80 100 120 174 1 150 60 Min Off-Time vs Temp 180 100 40 Figure 11. 3 50 20 TEMPERATURE (°C) TEMPERATURE ( C) TON (Ps) 50 Figure 9. Feedback Threshold Voltage vs Temp 6 40 EXT VCC (PA) VIN (V) Figure 8. 0 30 250 300 168 -40 -20 0 20 40 60 80 100 120 o ION (PA) TEMPERATURE ( C) Figure 12. Figure 13. Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: LM2696 LM2696 www.ti.com SNVS375B – OCTOBER 2005 – REVISED APRIL 2013 TYPICAL PERFORMANCE CHARACTERISTICS (continued) Max and Min Duty-Cycle vs Freq (Min TON = 400 ns, Min TOFF = 200 ns) FET Resistance vs Temp 0.25 1.0 0.2 0.6 0.15 RDS_ON (:) DUTY CYCLE Max Duty Cycle 0.8 0.4 0.2 0.05 Min Duty Cycle 0.0 100 200 0.1 300 400 0 -40 -20 500 0 20 40 60 80 100 120 o TEMPERATURE ( C) FREQUENCY (kHz) Figure 14. Figure 15. RON Pin Voltage vs Temp Current Limit vs Temp 1 5.4 5.2 CURRENT LIMIT (A) VON (V) 0.8 0.6 0.4 0.2 0 -40 -20 5 4.8 4.6 4.4 4.2 0 20 40 60 80 100 120 4 -40 -20 0 20 40 60 80 100 120 o TEMPERATURE ( C) o TEMPERATURE ( C) Figure 16. Figure 17. Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: LM2696 7 LM2696 SNVS375B – OCTOBER 2005 – REVISED APRIL 2013 www.ti.com BLOCK DIAGRAM LM2696 AVIN EXTVCC 3.65V INTERNAL LDO SD RON UVLO ON TIMER THERMAL SHUTDOWN 6V INTERNAL SUB REGULATOR Ron Q SS 1 PA CBOOT 1.25V SD PGOOD PVIN OFF TIMER Q DRIVER 94% x Vbg 1.25V LEVEL SHIFT UNDER-VOLTAGE COMPARATOR FB REGULATION COMPARATOR FB S Q R Q SW COMPLETE CURRENT LIMIT START OFF TIMER SD 4.8A 1 PA BUCK SWITCH CURRENT SENSE Shutdown GND APPLICATION INFORMATION CONSTANT ON-TIME CONTROL OVERVIEW The LM2696 buck DC-DC regulator is based on the constant on-time control scheme. This topology relies on a fixed switch on-time to regulate the output. The on-time can be set manually by adjusting the size of an external resistor (RON). The LM2696 automatically adjusts the on-time inversely with the input voltage (AVIN) to maintain a constant frequency. In continuous conduction mode (CCM) the frequency depends only on duty cycle and ontime. This is in contrast to hysteretic regulators where the switching frequency is determined by the output inductor and capacitor. In discontinuous conduction mode (DCM), experienced at light loads, the frequency will vary according to the load. This leads to high efficiency and excellent transient response. The on-time will remain constant for a given VIN unless an over-current or over-voltage event is encountered. If these conditions are encountered the FET will turn off for a minimum pre-determined time period. This minimum TOFF (250 ns) is internally set and cannot be adjusted. After the TOFF period has expired the FET will remain off until the comparator trip-point has been reached. Upon passing this trip-point the FET will turn back on, and the process will repeat. Switchers that regulate using an internal comparator to sense the output voltage for switching decisions, such as hysteretic or constant on-time, require a minimum ESR. A minimum ESR is required so that the control signal will be dominated by ripple that is in phase with the switchpin. Using a control signal dominated by voltage ripple that is in phase with the switchpin eliminates the need for compensation, thus reducing parts count and simplifying design. Alternatively, an RC feed forward scheme may be used to eliminate the need for a minimum ESR. 8 Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: LM2696 LM2696 www.ti.com SNVS375B – OCTOBER 2005 – REVISED APRIL 2013 INTERNAL OPERATION UNDER-VOLTAGE COMPARATOR An internal comparator is used to monitor the feedback pin for sensing under-voltage output events. If the output voltage drops below the UVP threshold the power-good flag will fall. ON-TIME GENERATOR SHUTDOWN The on-time for the LM2696 is inversely proportional to the input voltage. This scheme of on-time control maintains a constant frequency over the input voltage range. The on-time can be adjusted by using an external resistor connected between the PVIN and RON pins. CURRENT LIMIT The LM2696 contains an intelligent current limit off-timer. If the peak current in the internal FET exceeds 4.9A the present on-time is terminated; this is a cycle-by-cycle current limit. Following the termination of the on-time, a non-resetable extended off timer is initiated. The length of the off-time is proportional to the feedback voltage. When FB = 0V the off-time is preset to 20 µs. This condition is often a result of in short circuit operation when a maximum amount of off-time is required. This amount of time ensures safe short circuit operation up to the maximum input voltage of 24V. In cases of overload (not complete short circuit, FB > 0V) the current limit off-time is reduced. Reduction of the off-time during smaller overloads reduces the amount of fold back. This also reduces the initial startup time. N-CHANNEL HIGH SIDE SWITCH AND DRIVER The LM2696 utilizes an integrated N-Channel high side switch and associated floating high voltage gate driver. This gate driver circuit works in conjunction with an external bootstrap capacitor and an internal diode. The minimum off-time (165 ns) is set to ensure that the bootstrap capacitor has sufficient time to charge. THERMAL SHUTDOWN An internal thermal sensor is incorporated to monitor the die temperature. If the die temp exceeds 165ºC then the sensor will trip causing the part to stop switching. Soft-start will restart after the temperature falls below 155ºC. COMPONENT SELECTION As with any DC-DC converter, numerous trade-offs are present that allow the designer to optimize a design for efficiency, size and performance. These trade-offs are taken into consideration throughout this section. The first calculation for any buck converter is duty cycle. Ignoring voltage drops associated with parasitic resistances and non-ideal components, the duty cycle may be expressed as: VOUT D= VIN (1) A duty cycle relationship that considers the voltage drop across the internal FET and voltage drop across the external catch diode may be expressed as: VOUT + VD D= VIN + VD - VSW Where • • VD is the forward voltage of the external catch diode (DCATCH) VSW is the voltage drop across the internal FET. (2) FREQUENCY SELECTION Switching frequency affects the selection of the output inductor, capacitor, and overall efficiency. The trade-offs in frequency selection may be summarized as; higher switching frequencies permit use of smaller inductors possibly saving board space at the trade-off of lower efficiency. It is recommended that a nominal frequency of 300 kHz should be used in the initial stages of design and iterated if necessary. Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: LM2696 9 LM2696 SNVS375B – OCTOBER 2005 – REVISED APRIL 2013 www.ti.com The switching frequency of the LM2696 is set by the resistor connected to the RON pin. This resistor controls the current flowing into the RON pin and is directly related to the on-time pulse. Connecting a resistor from this pin to PVIN allows the switching frequency to remain constant as the input voltage changes. In normal operation this pin is approximately 0.65V above GND. In shutdown, this pin becomes a high impedance node to prevent current flow. The ON time may be expressed as: kON RON TON = VIN - VD 10-3 Ps where • • • • VIN is the voltage at the high side of the RON resistor (typically PVIN) VD is the diode voltage present at the RON pin (0.65V typical) RON is in kΩ kON is a constant value set internally (66 µA•µs nominal). (3) This equation can be re-arranged such that RON is a function of switching frequency: (VIN - VD) ‡ ' 106 k RON = kON ‡ ISW where • fSW is in kHz. (4) In CCM the frequency may be determined using the relationship: D 103 kHz fSW = TON (5) (TON is in µs) Which is typically used to set the switching frequency. Under no condition should a bypass capacitor be connected to the RON pin. Doing so couples any AC perturbations into the pin and prevents proper operation. INDUCTOR SELECTION Selecting an inductor is a process that may require several iterations. The reason for this is that the size of the inductor influences the amount of ripple present at the output that is critical to the stability of an adaptive on-time circuit. Typically, an inductor is selected such that the maximum peak-to-peak ripple current is equal to 30% of the maximum load current. The inductor current ripple (ΔIL) may be expressed as: (VIN - VOUT) D 'IL = L fSW (6) Therefore, L can be initially set to the following by applying the 30% guideline: L= (VIN - VOUT) D 0.3 fSW IOUT (7) The other features of the inductor that should be taken into account are saturation current and core material. A shielded inductor or low profile unshielded inductor is recommended to reduce EMI. OUTPUT CAPACITOR The output capacitor size and ESR have a direct affect on the stability of the loop. This is because the adaptive on-time control scheme works by sensing the output voltage ripple and switching appropriately. The output voltage ripple on a buck converter can be approximated by assuming that the AC inductor ripple current flows entirely into the output capacitor and the ESR of the capacitor creates the voltage ripple. This is expressed as: ΔVOUT≈ ΔIL • RESR 10 (8) Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: LM2696 LM2696 www.ti.com SNVS375B – OCTOBER 2005 – REVISED APRIL 2013 To ensure stability, two constraints need to be met. These constraints are the voltage ripple at the feedback pin must be greater than some minimum value and the voltage ripple must be in phase with the switch pin. The ripple voltage necessary at the feedback pin may be estimated using the following relationship: ΔVFB > −0.057 • fSW + 35 where • • fSW is in kHz ΔVFB is in mV. (9) This minimum ripple voltage is necessary in order for the comparator to initiate switching. The voltage ripple at the feedback pin must be in-phase with the switch. Because the ripple due to the capacitor charging and capacitor ESR are out of phase, the ripple due to capacitor ESR must dominate. The ripple at the output may be calculated by multiplying the feedback ripple voltage by the gain seen through the feedback resistors. This gain H may be expressed as: VOUT H= VFB VOUT = 1.25V (10) To simplify design and eliminate the need for high ESR output capacitors, an RC network may be used to feed forward a signal from the switchpin to the feedback (FB) pin. See the ‘RIPPLE FEED FORWARD’ section for more details. Typically, the best performance is obtained using POSCAPs, SP CAPs, tantalum, Niobium Oxide, or similar chemistry type capacitors. Low ESR ceramic capacitors may be used in conjunction with the RC feed forward scheme; however, the feed forward voltage at the feedback pin must be greater than 30 mV. RIPPLE FEED FORWARD An RC network may be used to eliminate the need for high ESR capacitors. Such a network is connected as shown in Figure 18. L SW VOUT Rff RFB1 FB COUT Cff RFB2 Figure 18. RC Feed Forward Network The value of Rff should be large in order to prevent any potential offset in VOUT. Typically the value of Rff is on the order of 1 MΩ and the value of RFB1 should be less than 10 kΩ. The large difference in resistor values minimizes output voltage offset errors in DCM. The value of the capacitor may be selected using the following relationship: (VIN_MIN - VFB) x TON_MIN Cff_MAX = 0.03V x Rff pF where • • on-time (TON_MIN) is in µs resistance (Rff) is in MΩ. (11) Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: LM2696 11 LM2696 SNVS375B – OCTOBER 2005 – REVISED APRIL 2013 www.ti.com FEEDBACK RESISTORS The feedback resistors are used to scale the output voltage to the internal reference value such that the loop can be regulated. The feedback resistors should not be made arbitrarily large as this creates a high impedance node at the feedback pin that is more susceptible to noise. Typically, RFB2 is on the order of 1 kΩ. To calculate the value of RFB1, one may use the relationship: RFB1 = RFB2 § VOUT ¨ VFB © · ¹ - 1¸ Where • VFB is the internal reference voltage that can be found in the ELECTRICAL CHARACTERISTICS table (1.254V typical). (12) The output voltage value can be set in a precise manner by taking into account the fact that the reference voltage is regulating the bottom of the output ripple as opposed to the average value. This relationship is shown in Figure 19. VOUT VOUT_AVG 'VOUT VREF Time Figure 19. Average and Ripple Output Voltages It can be seen that the average output voltage is higher than the gained up reference by exactly half the output voltage ripple. The output voltage may then be appended according to the voltage ripple. The appended VOUT term may be expressed using the relationship: VOUT = VOUT_AVG - 1 'I R 1 'VOUT = VOUT_AVG L ESR 2 2 (13) One should note that for high output voltages (>5V), a load of approximately 15 mA may be required for the output voltage to reach the desired value. INPUT CAPACITOR Because PVIN is the power rail from which the output voltage is derived, the input capacitor is typically selected according to the load current. In general, package size and ESR determine the current capacity of a capacitor. If these criteria are met, there should be enough capacitance to prevent impedance interactions with the source. In general, it is recommended to use a low ESR, high capacitance electrolytic and ceramic capacitor in parallel. Using two capacitors in parallel ensures adequate capacitance and low ESR over the operating range. The Sanyo MV-WX series electrolytic capacitors and a ceramic capacitor with X5R or X7R dielectric are an excellent combination. To calculate the input capacitor RMS, one may use the following relationship: 2 ICIN_RMS = IOUT 'IL § · D ¨1 - D + 2¸ 12 IOUT © ¹ (14) that can be approximated by, ICIN_RMS = IOUT D(1 - D) (15) Typical values are 470 µF for the electrolytic capacitor and 0.1 µF for the ceramic capacitor. 12 Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated Product Folder Links: LM2696 LM2696 www.ti.com SNVS375B – OCTOBER 2005 – REVISED APRIL 2013 AVIN CAPACITOR AVIN is the analog bias rail of the device. It should be bypassed externally with a small (1 µF) ceramic capacitor to prevent unwanted noise from entering the device. In a shutdown state the current needed by AVIN will drop to approximately 12 µA, providing a low power sleep state. In most cases of operation, AVIN is connected to PVIN; however, it is possible to have split rail operation where AVIN is at a higher voltage than PVIN. AVIN should never be lower than PVIN. Splitting the rails allows the power conversion to occur from a lower rail than the AVIN operating range. SOFT-START CAPACITOR The SS capacitor is used to slowly ramp the reference from 0V to its final value of 1.25V (during shutdown this pin will be discharged to 0V). This controlled startup ability eliminates large in-rush currents in an attempt to charge up the output capacitor. By changing the value of this capacitor, the duration of the startup may be changed accordingly. The startup time may be calculated using the following relationship: tSS = 1.25V CSS ISS Where • ISS is the soft-start pin source current (1 µA typical) that may be found in the ELECTRICAL CHARACTERISTICS table. (16) While the CSS capacitor can be sized to meet the startup requirements, there are limitations to its size. If the capacitor is too small, the soft-start will have little effect as the reference voltage is rising faster than the output capacitor can be charged causing the part to go into current limit. Therefore a minimum soft-start time should be taken into account. This can be determined by: COUTVOUT tSS_MIN = 3A (17) While COUT and VOUT control the slew rate of the output voltage, the total amount of time the LM2696 takes to startup is dependent on two other terms. See the “ Startup” section for more information. EXTVCC CAPACITOR External VCC is a 3.65V rail generated by an internal sub-regulator that powers the parts internal circuitry. This rail should be bypassed with a 1 µF ceramic capacitor (X5R or equivalent dielectric). Although EXTVCC is for internal use, it can be used as an external rail for extremely light loads (