Product
Folder
Order
Now
Support &
Community
Tools &
Software
Technical
Documents
LM2743
SNVS276I – APRIL 2004 – REVISED FEBRUARY 2019
LM2743 2.2-V to 16-V Input, voltage mode,
synchronous buck controller with tracking
1 Features
3 Description
•
•
•
•
•
•
•
•
•
The LM2743 is a high-speed synchronous buck
regulator controller with an accurate feedback voltage
accuracy of ±2%. It can provide simple down
conversion to output voltages as low as 0.6 V.
Though the control sections of the IC are rated for 3
V to 6 V, the driver sections are designed to accept
input supply rails as high as 16 V. The use of
adaptive non-overlapping MOSFET gate drivers helps
avoid potential shoot-through problems while
maintaining high efficiency. The IC is designed for the
more cost-effective option of driving only N-channel
MOSFETs in both the high-side and low-side
positions. It senses the low-side switch voltage drop
for providing a simple, adjustable current limit.
1
•
•
•
Power stage input voltage from 1 V to 16 V
Control stage input voltage from 3 V to 6 V
Output voltage adjustable down to 0.6 V
Power-good flag and shutdown
Output overvoltage and undervoltage detection
±2% Feedback voltage accuracy over temperature
Low-side adjustable current sensing
Adjustable soft start
Tracking and sequencing with shutdown and softstart pins
Switching frequency from 50 kHz to 1 MHz
TSSOP-14 package
Create a custom design using the LM2743 with
the WEBENCH® Power Designer
The fixed-frequency voltage-mode PWM control
architecture is adjustable from 50 kHz to 1 MHz with
one external resistor. This wide range of switching
frequency gives the power supply designer the
flexibility to make better tradeoffs between
component size, cost and efficiency.
2 Applications
•
•
•
•
•
3.3-V Buck regulation
Cable modem, DSL and ADSL
Laser Jet and ink jet printers
Low-voltage power modules
DSP, ASIC, core, and I/O
Features include soft start, input undervoltage lockout
(UVLO) and Power Good (based on both
undervoltage and overvoltage detection). In addition,
the shutdown pin of the IC can be used for providing
start-up delay, and the soft-start pin can be used for
implementing precise tracking, for the purpose of
sequencing with respect to an external rail.
Device Information(1)
PART NUMBER
LM2743
PACKAGE
TSSOP (14)
BODY SIZE (NOM)
5.00 mm × 4.40 mm
(1) For all available packages, see the orderable addendum at
the end of the data sheet.
Typical Application Diagram
VCC = 3.3V
RCC
VIN = 3.3V
D1
CBOOT
RPULL-UP
VCC
BOOT
PWGD
CSS
RCS
L1
VOUT = 1.2V@4A
ISEN
LM2743
FREQ
RFADJ
CIN1,2
HG
SD
CCC
+
Q1
LG
SS/TRACK
PGND
SGND
PGND
EAO
FB
RFB2
RC2
CC3
+
CO1,2
RFB1
CC1
CC2
RC1
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
LM2743
SNVS276I – APRIL 2004 – REVISED FEBRUARY 2019
www.ti.com
Table of Contents
1
2
3
4
5
6
7
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Pin Configuration and Functions .........................
Specifications.........................................................
1
1
1
2
3
4
6.1
6.2
6.3
6.4
6.5
6.6
4
4
4
4
5
7
Absolute Maximum Ratings ......................................
ESD Ratings ............................................................
Recommended Operating Conditions.......................
Thermal Information ..................................................
Electrical Characteristics...........................................
Typical Characteristics ..............................................
Detailed Description ............................................ 10
7.1 Overview ................................................................. 10
7.2 Functional Block Diagram ....................................... 10
7.3 Feature Description................................................. 10
7.4 Device Functional Modes........................................ 18
8
Application and Implementation ........................ 19
8.1 Application Information............................................ 19
8.2 Typical Applications ................................................ 19
9 Power Supply Recommendations...................... 33
10 Layout................................................................... 33
10.1 Layout Guidelines ................................................. 33
10.2 Layout Example .................................................... 33
11 Device and Documentation Support ................. 34
11.1
11.2
11.3
11.4
11.5
11.6
Device Support ....................................................
Receiving Notification of Documentation Updates
Community Resources..........................................
Trademarks ...........................................................
Electrostatic Discharge Caution ............................
Glossary ................................................................
34
34
34
34
34
34
12 Mechanical, Packaging, and Orderable
Information ........................................................... 35
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision H (October 2015) to Revision I
•
Editorial updates; add links for WEBENCH; no technical changes ....................................................................................... 1
Changes from Revision G (March 2013) to Revision H
•
2
Page
Added ESD Ratings table, Feature Description section, Device Functional Modes, Application and Implementation
section, Power Supply Recommendations section, Layout section, Device and Documentation Support section, and
Mechanical, Packaging, and Orderable Information section. ................................................................................................ 1
Changes from Revision F (March 2013) to Revision G
•
Page
Page
Changed layout of National Data Sheet to TI format ........................................................................................................... 32
Submit Documentation Feedback
Copyright © 2004–2019, Texas Instruments Incorporated
Product Folder Links: LM2743
LM2743
www.ti.com
SNVS276I – APRIL 2004 – REVISED FEBRUARY 2019
5 Pin Configuration and Functions
PW Package
14-Pin TSSOP
Top View
1
2
3
5
6
7
HG
PGND
SGND
VCC
PWGD
ISEN
PGND
LM2743
4
BOOT
LG
SD
FREQ
FB
SS/TRACK
EAO
14
13
12
11
10
9
8
Pin Functions
PIN
NAME
DESCRIPTION
NO.
BOOT
1
Bootstrap pin. This is the supply rail for the gate drivers. When the high-side MOSFET turns on, the
voltage on this pin should be at least one gate threshold above the regulator input voltage VIN to
properly turn on the MOSFET. See MOSFET Gate Drivers for more details on how to select
MOSFETs.
LG
2
Low-gate drive pin. This is the gate drive for the low-side N-channel MOSFET. This signal is
interlocked with the high-side gate drive HG (Pin 14), so as to avoid shoot-through.
3
Power ground. This is also the ground for the low-side MOSFET driver. Both the pins must be
connected together on the PCB and form a ground plane, which is usually also the system ground.
PGND
13
SGND
4
Signal ground. It should be connected appropriately to the ground plane with due regard to good layout
practices in switching power regulator circuits.
VCC
5
Supply rail for the control sections of the IC.
PWGD
6
Power Good pin. This is an open drain output, which is typically meant to be connected to VCC or any
other low voltage source through a pull-up resistor. Choose the pull-up resistor so that the current
going into this pin is kept below 1 mA. For most applications a recommended value for the pull-up
resistor is 100 kΩ. The voltage on this pin is thus pulled low under output under-voltage or overvoltage fault conditions and also under input UVLO.
ISEN
7
Current limit threshold setting pin. This sources a fixed 40 µA current. A resistor of appropriate value
should be connected between this pin and the drain of the low-side MOSFET (switch node).
EAO
8
Output of the error amplifier. The voltage level on this pin is compared with an internally generated
ramp signal to determine the duty cycle. This pin is necessary for compensating the control loop.
SS/TRACK
9
Soft-start and tracking pin. This pin is internally connected to the non-inverting input of the error
amplifier during soft-start, and in fact any time the SS/TRACK pin voltage happens to be below the
internal reference voltage. For the basic soft-start function, a capacitor of minimum value 1 nF is
connected from this pin to ground. To track the rising ramp of another power supply’s output, connect
a resistor divider from the output of that supply to this pin as described in Application and
Implementation.
FB
10
Feedback pin. This is the inverting input of the error amplifier, which is used for sensing the output
voltage and compensating the control loop.
FREQ
11
Frequency adjust pin. The switching frequency is set by connecting a resistor of suitable value
between this pin and ground. The equation for calculating the exact value is provided in Application
and Implementation, but some typical values (rounded up to the nearest standard values) are 324 kΩ
for 100 kHz, 97.6 kΩ for 300 kHz, 56.2 kΩ for 500 kHz, 24.9 kΩ for 1 MHz.
SD
12
IC shutdown pin. Pull this pin to VCC to ensure the IC is enabled. Connect to ground to disable the IC.
Under shutdown, both high-side and low-side drives are off. This pin also features a precision
threshold for power supply sequencing purposes, as well as a low threshold to ensure minimal
quiescent current.
HG
14
High-gate drive pin. This is the gate drive for the high-side N-channel MOSFET. This signal is
interlocked with LG (Pin 2) to avoid shoot-through.
Submit Documentation Feedback
Copyright © 2004–2019, Texas Instruments Incorporated
Product Folder Links: LM2743
3
LM2743
SNVS276I – APRIL 2004 – REVISED FEBRUARY 2019
www.ti.com
6 Specifications
6.1 Absolute Maximum Ratings
VCC
BOOT voltage
ISEN
All other pins
TJ
Tstg
(2)
MAX
UNIT
7
V
–0.3
21
V
–0.3
9.5
V
–0.3
VCC + 0.3
V
Junction temperature
Soldering information
(1)
MIN
–0.3
150
°C
Lead temperature (soldering, 10 s)
260
°C
Infrared or convection (20 s)
235
°C
150
°C
Storage temperature
–65
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended
Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/ Distributors for availability and
specifications.
6.2 ESD Ratings
Electrostatic discharge (1)
V(ESD)
(1)
(2)
VALUE
UNIT
2000
V
Human-body model (HBM), per ANSI/ESDA/JEDEC JS-001 (2)
The human body model is a 100 pF capacitor discharged through a 1.5-kΩ resistor into each pin.
JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
6.3 Recommended Operating Conditions
MIN
NOM
MAX
UNIT
VCC
Supply voltage range
3
6
V
TJ
Junction temperature
–40
125
°C
6.4 Thermal Information
LM2743
THERMAL METRIC (1)
TSSOP (PW)
UNIT
14 PINS
RθJA
Junction-to-ambient thermal resistance
107.9
°C/W
RθJC(top)
Junction-to-case (top) thermal resistance
33.7
°C/W
RθJB
Junction-to-board thermal resistance
50.7
°C/W
ψJT
Junction-to-top characterization parameter
1.8
°C/W
ψJB
Junction-to-board characterization parameter
50
°C/W
(1)
4
For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report, SPRA953.
Submit Documentation Feedback
Copyright © 2004–2019, Texas Instruments Incorporated
Product Folder Links: LM2743
LM2743
www.ti.com
SNVS276I – APRIL 2004 – REVISED FEBRUARY 2019
6.5 Electrical Characteristics
Typical limits are for TJ = 25°C only, represent the most likely parametric norm at TJ = 25°C, and are provided for reference
purposes only; minimum and maximum limits apply over the junction temperature range of –40°C to 125°C. Unless otherwise
specified, VCC = 3.3 V. Data sheet minimum and maximum specification limits are specified by design, test, or statistical
analysis. (See (1))
PARAMETER
VFB
VON
IQ_VCC
TEST CONDITIONS
FB Pin Voltage
VCC = 3 V to 6 V
UVLO Thresholds
Rising
Falling
Operating VCC Current
MIN
0.588
TYP
MAX
UNIT
0.6
0.612
V
2.76
2.42
V
VCC = 3.3 V, VSD = 3.3 V
Fsw = 600 kHz
1
1.5
2.1
VCC = 5V, VSD = 3.3 V
Fsw = 600 kHz
1
1.7
2.1
110
185
mA
Shutdown VCC Current
VCC = 3.3 V, VSD = 0 V
tPWGD1
PWGD Pin Response Time
VFB Rising
6
µs
tPWGD2
PWGD Pin Response Time
VFB Falling
6
µs
ISS-ON
SS Pin Source Current
VSS = 0 V
ISS-OC
SS Pin Sink Current During Over
Current
VSS = 2.5 V
ISEN-TH
ISEN Pin Source Current Trip Point
7
10
14
90
25
40
µA
µA
µA
55
µA
ERROR AMPLIFIER
GBW
Error Amplifier Unity Gain
Bandwidth
G
Error Amplifier DC Gain
106
dB
SR
Error Amplifier Slew Rate
3.2
V/µs
IEAO
EAO Pin Current Sourcing and
Sinking Capability
VEAO = 1.5, FB = 0.55 V
VEAO = 1.5, FB = 0.65 V
2.6
9.2
mA
Error Amplifier Output Voltage
Minimum
1
V
Maximum
2
V
VEA
9
MHz
GATE DRIVE
IQ-BOOT
BOOT Pin Quiescent Current
VBOOT = 12 V, VSD = 0
RHG_UP
High-Side MOSFET Driver Pull-Up
ON resistance
18
90
µA
VBOOT = 5 V at 350 mA Sourcing
3
Ω
RHG_DN
High-Side MOSFET Driver PullDown ON resistance
HG = 5 V at 350 mA Sourcing
2
Ω
RLG_UP
Low-Side MOSFET Driver Pull-Up
ON resistance
VBOOT = 5 V at 350 mA Sourcing
3
Ω
RLG_DN
Low-Side MOSFET Driver PullDown ON resistance
LG = 5 V at 350 mA Sourcing
2
Ω
OSCILLATOR
RFADJ = 702.1 kΩ
fSW
PWM Frequency
Max High-Side Duty Cycle
D
50
RFADJ = 98.74 kΩ
RFADJ = 45.74 kΩ
300
475
600
RFADJ = 24.91 kΩ
1000
fSW = 300 kHz
fSW = 600 kHz
fSW = 1 MHz
80%
76%
73%
725
kHz
LOGIC INPUTS AND OUTPUTS
V STBY-IH
V STBY-IL
V SD-IH
(1)
Standby High Trip Point
VFB = 0.575 V, VBOOT = 3.3 V, VSD
Rising
Standby Low Trip Point
VFB = 0.575 V, VBOOT = 3.3 V, VSD
Falling
SD Pin Logic High Trip Point
VSD Rising
1.1
0.232
V
V
1.3
V
The power MOSFETs can run on a separate 1-V to 16-V rail (Input voltage, VIN). Practical lower limit of VIN depends on selection of the
external MOSFET.
Submit Documentation Feedback
Copyright © 2004–2019, Texas Instruments Incorporated
Product Folder Links: LM2743
5
LM2743
SNVS276I – APRIL 2004 – REVISED FEBRUARY 2019
www.ti.com
Electrical Characteristics (continued)
Typical limits are for TJ = 25°C only, represent the most likely parametric norm at TJ = 25°C, and are provided for reference
purposes only; minimum and maximum limits apply over the junction temperature range of –40°C to 125°C. Unless otherwise
specified, VCC = 3.3 V. Data sheet minimum and maximum specification limits are specified by design, test, or statistical
analysis. (See (1))
PARAMETER
TEST CONDITIONS
V SD-IL
SD Pin Logic Low Trip Point
VSD Falling
VPWGD-TH-LO
PWGD Pin Trip Points
FB Falling
VPWGD-TH-HI
PWGD Pin Trip Points
FB Rising
VPWGD-HYS
PWGD Hysteresis
6
MIN
TYP
MAX
0.408
0.434
0.457
V
0.677
0.710
0.742
V
0.8
V
FB Falling
60
FB Rising
90
Submit Documentation Feedback
UNIT
mV
Copyright © 2004–2019, Texas Instruments Incorporated
Product Folder Links: LM2743
LM2743
www.ti.com
SNVS276I – APRIL 2004 – REVISED FEBRUARY 2019
6.6 Typical Characteristics
100
100
VIN = 3.3V
90
90
EFFICIENCY (%)
EFFICIENCY (%)
80
VIN = 12V
70
VIN = 5V
60
50
80
70
VIN = 5V
50
40
40
30
30
20
VIN = 3.3V
60
VIN = 12V
20
0.0 0.4 0.8 1.2 1.6 2.0 2.4 2.8 3.2 3.6 4.0
0.0 0.4 0.8 1.2 1.6 2.0 2.4 2.8 3.2 3.6 4.0
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
VCC = 3.3 V
fSW = 300 kHz
VCC = 3.3 V
Figure 1. Efficiency VOUT = 1.2 V
fSW = 300 kHz
Figure 2. Efficiency VOUT = 2.5 V
100
25
VCC OPERATING CURRENT
EFFICIENCY (%)
80
70
VIN = 5V
60
50
VIN = 12V
40
PLUS BOOT CURRENT (mA)
90
19
13
6
30
0
0.05
20
0.0 0.4 0.8 1.2 1.6 2.0 2.4 2.8 3.2 3.6 4.0
FDS6898A FET
fSW = 300 kHz
6.7
10
6.6
9.9
BOOT PIN CURRENT (mA)
BOOT PIN CURRENT (mA)
0.65
0.85
1.05
TA = 25°C
Figure 4. VCC Operating Current Plus BOOT Current vs
Frequency
Figure 3. Efficiency VOUT = 3.3 V
6.5
6.4
6.3
6.2
6.1
9.8
9.7
9.6
9.5
9.4
6
9.3
-40 -25 -10 5 20 35 50 65 80 95 110 125
5.9
-40 -25 -10 5 20 35 50 65 80 95 110 125
TEMPERATURE (oC)
TEMPERATURE (oC)
fSW = 300 kHz
0.45
FREQUENCY (MHz)
OUTPUT CURRENT (A)
VCC = 5 V
0.25
FDS6898A FET
No-Load
Figure 5. BOOT Pin Current vs Temperature for BOOT
Voltage = 3.3 V
fSW = 300 kHz
FDS6898A FET
No-Load
Figure 6. BOOT Pin Current vs Temperature for BOOT
Voltage = 5 V
Submit Documentation Feedback
Copyright © 2004–2019, Texas Instruments Incorporated
Product Folder Links: LM2743
7
LM2743
SNVS276I – APRIL 2004 – REVISED FEBRUARY 2019
www.ti.com
Typical Characteristics (continued)
24.9
0.2
0.1
24.7
INTERNAL REFERENCE
VOLTAGE ('%)
BOOT PIN CURRENT (mA)
24.8
24.6
24.5
24.4
24.3
24.2
24.1
0
-0.1
-0.2
-0.3
-0.4
-0.5
24
23.9
-40 -25 -10 5 20 35 50 65 80 95 110125
-0.6
-40 -25 -10 5 20 35 50 65 80 95 110 125
TEMPERATURE (oC)
fSW = 300 kHz
TEMPERATURE (oC)
FDS6898A FET
No-Load
Figure 7. BOOT Pin Current vs Temperature for BOOT
Voltage = 12 V
Figure 8. Internal Reference Voltage vs Temperature
1.210
640
1.209
1.208
OUTPUT VOLTAGE (V)
FREQUENCY (kHz)
620
600
580
560
1.207
1.206
1.205
1.204
1.203
1.202
540
1.201
1.200
0
520
-55 -35 -15
5
25
45
65 85 105 125
o
TEMPERATURE ( C)
0.4 0.8 1.2 1.6
2
2.4 2.8 3.2 3.6
OUTPUT CURRENT (A)
Figure 9. Frequency vs Temperature
Figure 10. Output Voltage vs Output Current
1V/div
HG
LG
2V/div
1V/div
2V/div
HG
LG
SW
SW
5V/div
5V/div
100 ns/DIV
VCC = 3.3 V
IOUT = 4 A
VIN = 5 V
CSS = 12 nF
100 ns/DIV
VOUT = 1.2 V
fSW = 300 kHz
VCC = 3.3 V
IOUT = 4 A
Figure 11. Switch Waveforms (HG Rising)
8
4
Submit Documentation Feedback
VIN = 5 V
CSS = 12 nF
VOUT = 1.2 V
fSW = 300 kHz
Figure 12. Switch Waveforms (HG Falling)
Copyright © 2004–2019, Texas Instruments Incorporated
Product Folder Links: LM2743
LM2743
www.ti.com
SNVS276I – APRIL 2004 – REVISED FEBRUARY 2019
Typical Characteristics (continued)
VOUT
50 mV/div
2A/div
IOUT
40 Ps/DIV
VCC = 3.3 V
CSS = 12 nF
VIN = 5V
fSW = 300 kHz
VOUT = 1.2V
IOUT = 0 A to 4 A
VOUT = 1.2 V
VCC = 3.3 V
CSS = 12nF
VIN = 5 V
fSW = 300 kHz
Figure 14. Load Transient Response
Figure 13. Start-Up (No-Load)
VOUT
VOUT
20 mV/div
20mV/div
IOUT
2A/div
2A/div
IOUT
40 Ps/DIV
IOUT = 4 A to 0 A
VOUT = 1.2 V
100 Ps/DIV
VCC = 3.3 V
CSS = 12 nF
VIN = 5 V
fSW = 300 kHz
VCC = 3.3 V
CSS = 12 nF
Figure 15. Load Transient Response
VIN = 5 V
fSW = 300 kHz
VOUT = 1.2 V
Figure 16. Load Transient Response
100mV/div
100 mV/div
VOUT
VOUT
5V/div
VIN
5V/div
VIN
100 Ps/DIV
VIN = 3 V to 9 V
Line () , IOUT = 2 A
VCC = 3.3 V
fSW = 300 kHz
100 Ps/DIV
VOUT = 1.2 V
VIN = 9 V to 3 V
IOUT = 2 A
Figure 17. Line Transient Response
VCC = 3.3 V
fSW = 300 kHz
VOUT = 1.2 V
Figure 18. Transient Response
Submit Documentation Feedback
Copyright © 2004–2019, Texas Instruments Incorporated
Product Folder Links: LM2743
9
LM2743
SNVS276I – APRIL 2004 – REVISED FEBRUARY 2019
www.ti.com
7 Detailed Description
7.1 Overview
The LM2743 is a voltage-mode, high-speed synchronous buck regulator with a PWM control scheme. It has
output shutdown (SD), input under-voltage lock-out (UVLO) mode and power good (PWGD) flag (based on overvoltage and under-voltage detection). The over-voltage and under-voltage signals are logically OR'ed to drive the
power good signal and provide a logic signal to the system if the output voltage goes out of regulation. Current
limit is achieved by sensing the voltage VDS across the low-side MOSFET.
7.2 Functional Block Diagram
VCC
SD
FREQ
CLOCK &
RAMP
PGND
PGND
SGND
UVLO
SHUT DOWN
LOGIC
BOOT
10 Ps
DELAY
HG
SSDONE
PWGD
SYNCHRONOUS
DRIVER LOGIC
OV
UV
0.71V
LG
0.434V
10 PA
SS/TRACK
1VPP
PWM LOGIC
PWM
Soft-Start
Comparator
+ Logic
REF
40 PA
-
+
90 PA
ISEN
ILIM
EA
-
+
VREF = 0.6V
FB
EAO
7.3 Feature Description
7.3.1 Start Up and Soft-Start
When VCC exceeds 2.76V and the shutdown pin (SD) sees a logic high, the soft-start period begins. Then an
internal, fixed 10 µA source begins charging the soft-start capacitor. During soft-start the voltage on the soft-start
capacitor CSS is connected internally to the non-inverting input of the error amplifier. The soft-start period lasts
until the voltage on the soft-start capacitor exceeds the LM2743 reference voltage of 0.6V. At this point the
reference voltage takes over at the non-inverting error amplifier input. The capacitance of CSS determines the
length of the soft-start period, and can be approximated by:
CSS =
tSS
60
where
10
Submit Documentation Feedback
Copyright © 2004–2019, Texas Instruments Incorporated
Product Folder Links: LM2743
LM2743
www.ti.com
SNVS276I – APRIL 2004 – REVISED FEBRUARY 2019
Feature Description (continued)
•
CSS is in µF and tSS is in ms
(1)
During soft start the Power Good flag is forced low and it is released when the FB pin voltage reaches 70% of
0.6V. At this point the chip enters normal operation mode, and the output overvoltage and undervoltage
monitoring starts.
7.3.2 Normal Operation
While in normal operation mode, the LM2743 regulates the output voltage by controlling the duty cycle of the
high side and low side MOSFETs (see Typical Application Diagram). The equation governing output voltage is:
RFB1 + RFB2
VOUT =
VFB
RFB1
(VFB = 0.6V)
(2)
The PWM frequency is adjustable between 50 kHz and 1 MHz and is set by an external resistor, RFADJ, between
the FREQ pin and ground. The resistance needed for a desired frequency is approximately:
RFADJ = -5.93 + 3.06
12
107
10
+ 0.24
2
fSW
(fSW)
(3)
Where fSW is in Hz and RFADJ is in kΩ.
7.3.3 Tracking a Voltage Level
The LM2743 can track the output of a master power supply during soft-start by connecting a resistor divider to
the SS/TRACK pin. In this way, the output voltage slew rate of the LM2743 will be controlled by the master
supply for loads that require precise sequencing. When the tracking function is used no soft-start capacitor
should be connected to the SS/TRACK pin. Otherwise, a CSS value of at least 1 nF between the soft-start pin
and ground should be used.
Master Power
Supply
VOUT1 = 5V
RT2
1 k:
VOUT2 = 1.8V
SS/TRACK
VSS = 0.65V
RT1
150:
LM2743
FB
RFB2
10 k:
VFB
RFB1
5 k:
Figure 19. Tracking Circuit
One way to use the tracking feature is to design the tracking resistor divider so that the master supply’s output
voltage (VOUT1) and the LM2743’s output voltage (represented symbolically in Figure 19 as VOUT2, that is, without
explicitly showing the power components) both rise together and reach their target values at the same time. For
this case, the equation governing the values of the tracking divider resistors RT1 and RT2 is:
0.65 = VOUT1
RT1
RT1 + RT2
(4)
Submit Documentation Feedback
Copyright © 2004–2019, Texas Instruments Incorporated
Product Folder Links: LM2743
11
LM2743
SNVS276I – APRIL 2004 – REVISED FEBRUARY 2019
www.ti.com
Feature Description (continued)
The current through RT1 should be about 3 mA to 4 mA for precise tracking. The final voltage of the SS/TRACK
pin should be set higher than the feedback voltage of 0.6 V (say about 0.65 V as in the above equation). If the
master supply voltage was 5 V and the LM2743 output voltage was 1.8 V, for example, then the value of RT1
needed to give the two supplies identical soft-start times would be 150 Ω. A timing diagram for the equal softstart time case is shown in Figure 20.
5V
VOUT1
1.8V
VOUT2
Figure 20. Tracking with Equal Soft-Start Time
7.3.4 Tracking Voltage Slew Rate
The tracking feature can alternatively be used not to make both rails reach regulation at the same time but rather
to have similar rise rates (in terms of output dV/dt). This method ensures that the output voltage of the LM2743
always reaches regulation before the output voltage of the master supply. Because the output of the master
supply is divided down, in order to track properly the output voltage of the LM2743 must be lower than the
voltage of the master supply. In this case, the tracking resistors can be determined based on the following
equation:
RT1
VOUT2 = VOUT1
RT1 + RT2
(5)
For the example case of VOUT1 = 5 V and VOUT2 = 1.8 V, with RT1 set to 150 Ω as before, RT2 is calculated from
the above equation to be 265 Ω. A timing diagram for the case of equal slew rates is shown in Figure 21.
5V
1.8V
VOUT1
1.8V
VOUT2
Figure 21. Tracking with Equal Slew Rates
7.3.5 Sequencing
The start up/soft-start of the LM2743 can be delayed for the purpose of sequencing by connecting a resistor
divider from the output of a master power supply to the SD pin, as shown in Figure 22.
12
Submit Documentation Feedback
Copyright © 2004–2019, Texas Instruments Incorporated
Product Folder Links: LM2743
LM2743
www.ti.com
SNVS276I – APRIL 2004 – REVISED FEBRUARY 2019
Feature Description (continued)
Master Power
Supply
VOUT1
RS2
VOUT2
SD
LM2743
RS1
RFB2
FB
VFB
RFB1
Figure 22. Sequencing Circuit
A desired delay time tDELAY between the startup of the master supply output voltage and the LM2743 output
voltage can be set based on the SD pin low-to-high threshold VSD-IH and the slew rate of the voltage at the SD
pin, SRSD:
tDELAY = VSD-IH / SRSD
(6)
Note again, that in Figure 22, the output voltage of the LM2743 has been represented symbolically as VOUT2,.
without explicitly showing the power components.
VSD-IH is typically 1.08V and SRSD is the slew rate of the SD pin voltage. The values of the sequencing divider
resistors RS1 and RS2 set the SRSD based on the master supply output voltage slew rate, SROUT1, using the
following equation:
SRSD = SROUT1
RS1
RS1 + RS2
(7)
For example, if the master supply output voltage slew rate was 1V/ms and the desired delay time between the
startup of the master supply and LM2743 output voltage was 5ms, then the desired SD pin slew rate would be
(1.08V/5 ms) = 0.216 V/ms. Due to the internal impedance of the SD pin, the maximum recommended value for
RS2 is 1 kΩ. To achieve the desired slew rate, RS1 would then be 274 Ω. A timing diagram for this example is
shown in Figure 23.
5V
VSD-IH
1.08V
VOUT1
1.8V
VOUT2
t = 5 ms
Figure 23. Delay for Sequencing
Submit Documentation Feedback
Copyright © 2004–2019, Texas Instruments Incorporated
Product Folder Links: LM2743
13
LM2743
SNVS276I – APRIL 2004 – REVISED FEBRUARY 2019
www.ti.com
Feature Description (continued)
7.3.6 SD Pin Impedance
When connecting a resistor divider to the SD pin of the LM2743 some care has to be taken. Once the SD voltage
goes above VSD-IH, a 17-µA pull-up current is activated as shown in Figure 24. This current is used to create the
internal hysteresis (≊170 mV); however, high external impedances will affect the SD pin logic thresholds as well.
The external impedance used for the sequencing divider network should preferably be a small fraction of the
impedance of the SD pin for good performance (around 1 kΩ).
17 PA
8 PA
Bias Enable
SD
10k
Soft-Start Enable
1.25V
+
-
Figure 24. SD Pin Logic
7.3.7 MOSFET Gate Drivers
The LM2743 has two gate drivers designed for driving N-channel MOSFETs in a synchronous mode. Note that
unlike most other synchronous controllers, the bootstrap capacitor of the LM2743 provides power not only to the
driver of the upper MOSFET, but the lower MOSFET driver too (both drivers are ground referenced, i.e. no
floating driver). To fully turn the top MOSFET on, the BOOT voltage must be at least one gate threshold greater
than VIN when the high-side drive goes high. This bootstrap voltage is usually supplied from a local charge pump
structure. But looking at the Typical Application schematic, this also means that the difference voltage VCC - VD1,
which is the voltage the bootstrap capacitor charges up to, must be always greater than the maximum tolerance
limit of the threshold voltage of the upper MOSFET. Here VD1 is the forward voltage drop across the bootstrap
diode D1. This therefore may place restrictions on the minimum input voltage and/or type of MOSFET used.
The most basic charge bootstrap pump circuit can be built using one Schottky diode and a small capacitor, as
shown in Figure 25. The capacitor CBOOT serves to maintain enough voltage between the top MOSFET gate and
source to control the device even when the top MOSFET is on and its source has risen up to the input voltage
level. The charge pump circuitry is fed from VCC, which can operate over a range from 3.0V to 6.0V. Using this
basic method the voltage applied to the gates of both high-side and low-side MOSFETs is VCC - VD. This method
works well when VCC is 5 V±10%, because the gate drives will get at least 4.0V of drive voltage during the worst
case of VCC-MIN = 4.5 V and VD-MAX = 0.5 V. Logic level MOSFETs generally specify their on-resistance at VGS =
4.5 V. When VCC = 3.3 V ±10%, the gate drive at worst case could go as low as 2.5 V. Logic level MOSFETs are
not specified to turn on, or may have much higher on-resistance at 2.5 V. Sub-logic level MOSFETs, usually
specified at VGS = 2.5 V, will work, but are more expensive, and tend to have higher on-resistance. The circuit in
Figure 25 works well for input voltages ranging from 1 V up to 16 V and VCC = 5 V ±10%, because the drive
voltage depends only on VCC.
14
Submit Documentation Feedback
Copyright © 2004–2019, Texas Instruments Incorporated
Product Folder Links: LM2743
LM2743
www.ti.com
SNVS276I – APRIL 2004 – REVISED FEBRUARY 2019
Feature Description (continued)
LM2743
D1
BOOT
VCC
CBOOT
VIN
HG
+
VO
+
LG
Figure 25. Basic Charge Pump (Bootstrap)
Note that the LM2743 can be paired with a low cost linear regulator like the LM78L05 to run from a single input
rail between 6.0 and 14 V. The 5-V output of the linear regulator powers both the VCC and the bootstrap circuit,
providing efficient drive for logic level MOSFETs. An example of this circuit is shown in Figure 26.
LM2743
VCC
5V
LM78L05
D1
BOOT
CBOOT
HG
VIN
+
VO
LG
+
Figure 26. LM78L05 Feeding Basic Charge Pump
Figure 27 shows a second possibility for bootstrapping the MOSFET drives using a doubler. This circuit provides
an equal voltage drive of VCC - 3VD + VIN to both the high-side and low-side MOSFET drives. This method should
only be used in circuits that use 3.3 V for both VCC and VIN. Even with VIN = VCC = 3.0 V (10% lower tolerance on
3.3 V) and VD = 0.5 V both high-side and low-side gates will have at least 4.5 V of drive. The power dissipation of
the gate drive circuitry is directly proportional to gate drive voltage, hence the thermal limits of the LM2743 IC will
quickly be reached if this circuit is used with VCC or VIN voltages over 5V.
Submit Documentation Feedback
Copyright © 2004–2019, Texas Instruments Incorporated
Product Folder Links: LM2743
15
LM2743
SNVS276I – APRIL 2004 – REVISED FEBRUARY 2019
www.ti.com
Feature Description (continued)
LM2743
BOOT
D3
D2
D1
VCC
VIN
HG
+
VO
+
LG
Figure 27. Charge Pump with Added Gate Drive
All the gate drive circuits shown in Figure 25 through Figure 27 typically use 100-nF ceramic capacitors in the
bootstrap locations.
7.3.8 Power Good Signal
The open drain output on the Power Good pin needs a pull-up resistor to a low voltage source. The pull-up
resistor should be chosen so that the current going into the Power Good pin is less than 1 mA. A 100-kΩ resistor
is recommended for most applications.
The Power Good signal is an OR-gated flag which takes into account both output over-voltage and under-voltage
conditions. If the feedback pin (FB) voltage is 18% above its nominal value (118% x VFB = 0.708V) or falls 28%
below that value (72 %x VFB = 0.42V) the Power Good flag goes low. The Power Good flag can be used to signal
other circuits that the output voltage has fallen out of regulation, however the switching of the LM2743 continues
regardless of the state of the Power Good signal. The Power Good flag will return to logic high whenever the
feedback pin voltage is between 72% and 118% of 0.6V.
7.3.9 UVLO
The 2.76V turn-on threshold on VCC has a built in hysteresis of about 300 mV. If VCC drops below 2.42V, the chip
enters UVLO mode. UVLO consists of turning off the top and bottom MOSFETS and remaining in that condition
until VCC rises above 2.76V. As with shutdown, the soft-start capacitor is discharged through an internal
MOSFET, ensuring that the next start-up will be controlled by the soft-start circuitry.
7.3.10 Current Limit
Current limit is realized by sensing the voltage across the low-side MOSFET while it is on. The RDS(ON) of the
MOSFET is a known value; hence the current through the MOSFET can be determined as:
VDS = IOUT x RDS(ON)
(8)
The current through the low-side MOSFET while it is on is also the falling portion of the inductor current. The
current limit threshold is determined by an external resistor, RCS, connected between the switching node and the
ISEN pin. A constant current of 40 µA is forced through RCS, causing a fixed voltage drop. This fixed voltage is
compared against VDS and if the latter is higher, the current limit of the chip has been reached. To obtain a more
accurate value for RCS you must consider the operating values of RDS(ON) and ISEN-TH at their operating
temperatures in your application and the effect of slight parameter differences from part to part. RCS can be found
by using the following equation using the RDS(ON) value of the low side MOSFET at it's expected hot temperature
and the absolute minimum value expected over the full temperature range for the for the ISEN-TH which is 25 µA:
RCS = RDSON-HOT x ILIM / 40 µA
16
(9)
Submit Documentation Feedback
Copyright © 2004–2019, Texas Instruments Incorporated
Product Folder Links: LM2743
LM2743
www.ti.com
SNVS276I – APRIL 2004 – REVISED FEBRUARY 2019
Feature Description (continued)
For example, a conservative 15-A current limit in a 10-A design with a minimum RDS(ON) of 10 mΩ would require
a 6-kΩ resistor. To prevent the ISEN pin from sinking too much current when the switch node goes above 9.5 V,
the value of the current limit setting resistor RCS should not be too low. The criterion is as follows,
VIN ± 9.5V
RCS t
10 mA
(10)
where the 10 mA is the maximum current ISEN pin is allowed to sink. For example if VIN = 13.2 V, the minimum
value of RCS is 370 Ω. Because current sensing is done across the low-side MOSFET, no minimum high-side ontime is necessary. The LM2743 enters current limit mode if the inductor current exceeds the current limit
threshold at the point where the high-side MOSFET turns off and the low-side MOSFET turns on. (The point of
peak inductor current, see Figure 28). Note that in normal operation mode the high-side MOSFET always turns
on at the beginning of a clock cycle. In current limit mode, by contrast, the high-side MOSFET on-pulse is
skipped. This causes inductor current to fall. Unlike a normal operation switching cycle, however, in a current
limit mode switching cycle the high-side MOSFET will turn on as soon as inductor current has fallen to the
current limit threshold. The LM2743 will continue to skip high-side MOSFET pulses until the inductor current peak
is below the current limit threshold, at which point the system resumes normal operation.
Normal Operation
Current Limit
ILIM
IL
D
Figure 28. Current Limit Threshold
Unlike a high-side MOSFET current sensing scheme, which limits the peaks of inductor current, low-side current
sensing is only allowed to limit the current during the converter off-time, when inductor current is falling.
Therefore in a typical current limit plot the valleys are normally well defined, but the peaks are variable, according
to the duty cycle. The PWM error amplifier and comparator control the off-pulse of the high-side MOSFET, even
during current limit mode, meaning that peak inductor current can exceed the current limit threshold. Assuming
that the output inductor does not saturate, the maximum peak inductor current during current limit mode can be
calculated with the following equation:
IPK-CL = ILIM + (TSW - 200 ns)
VIN - VO
L
(11)
Where TSW is the inverse of switching frequency fSW. The 200 ns term represents the minimum off-time of the
duty cycle, which ensures enough time for correct operation of the current sensing circuitry.
Submit Documentation Feedback
Copyright © 2004–2019, Texas Instruments Incorporated
Product Folder Links: LM2743
17
LM2743
SNVS276I – APRIL 2004 – REVISED FEBRUARY 2019
www.ti.com
Feature Description (continued)
In order to minimize the time period in which peak inductor current exceeds the current limit threshold, the IC
also discharges the soft-start capacitor through a fixed 90-µA sink. The output of the LM2743 internal error
amplifier is limited by the voltage on the soft-start capacitor. Hence, discharging the soft-start capacitor reduces
the maximum duty cycle D of the controller. During severe current limit this reduction in duty cycle will reduce the
output voltage if the current limit conditions last for an extended time. Output inductor current will be reduced in
turn to a flat level equal to the current limit threshold. The third benefit of the soft-start capacitor discharge is a
smooth, controlled ramp of output voltage when the current limit condition is cleared.
7.3.11
Foldback Current Limit
In the case where extra protection is used to help an output short condition, a current foldback resistor (RCLF)
should be considered, see Figure 29. First select the percentage of current limit foldback (PLIM):
PLIM = ILIM x P
(12)
where P is a ratio between 0 and 1.
VOUT
40 PA
RCLF
RCS
ILIM
+
SW
ISEN
Figure 29. Foldback Current Limit Circuit
Obtain the RCS with the following equation:
PLIM x RDS(ON)
= RCS
ISEN
(13)
where ISEN = 40 μA. If the switch node goes above 9.5 V the following criterion must be satisfied:
VIN ± 9.5V
RCS t
10 mA
(14)
The equation for calculating the foldback resistance value is:
RCS x VOUT
RCLF =
(ILIM x RDS(ON)) ± (ISEN x RCS)
(15)
7.4 Device Functional Modes
7.4.1 Shutdown
If the shutdown pin is pulled low, (below 0.8 V) the LM2743 enters shutdown mode, and discharges the soft-start
capacitor through a MOSFET switch. The high and low-side MOSFETs are turned off. The LM2743 remains in
this state as long as VSD sees a logic low (see the Electrical Characteristics table). To assure proper IC start-up
the shutdown pin should not be left floating. For normal operation this pin should be connected directly to VCC or
to another voltage between 1.3 V to VCC (see the Electrical Characteristics table).
18
Submit Documentation Feedback
Copyright © 2004–2019, Texas Instruments Incorporated
Product Folder Links: LM2743
LM2743
www.ti.com
SNVS276I – APRIL 2004 – REVISED FEBRUARY 2019
8 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
8.1 Application Information
The LM2743 is a voltage-mode, high-speed synchronous buck regulator with a PWM control scheme. It is
designed for use in set-top boxes, thin clients, DSL/Cable modems, and other applications that require highefficiency buck converters. Use the following design procedure to select component values for the LM2743
device. Use the WEBENCH to generate a complete design (see Custom Design With WEBENCH® Tools.
8.2 Typical Applications
8.2.1 Synchronous Buck Converter Typical Application using LM2743
VCC = 3.3V
RCC
VIN = 3.3V
D1
CBOOT
RPULL-UP
+
Q1
VCC
SD
CCC
BOOT
PWGD
CSS
RCS
L1
VOUT = 1.2V@4A
ISEN
LM2743
FREQ
RFADJ
CIN1,2
HG
LG
SS/TRACK
PGND
SGND
PGND
EAO
FB
RFB2
RC2
CC3
+
CO1,2
RFB1
CC1
CC2
RC1
Figure 30. 3.3 V to 1.2 V at 4 A, fSW = 300 kHz
8.2.1.1 Design Requirements
The following section provides a step-by-step design guide of a voltage-mode synchronous buck converter using
the LM2743. This design converts 3.3 V (VIN) to 1.2 V (VOUT) at a maximum load of 4 A, with an efficiency of
89% and a switching frequency of 300 kHz. The same procedures can be followed to create many other designs
with varying input voltages, output voltages, and load currents.
8.2.1.2 Detailed Design Procedure
8.2.1.2.1 Custom Design With WEBENCH® Tools
Click here to create a custom design using the LM2743 device with the WEBENCH® Power Designer.
1. Start by entering the input voltage (VIN), output voltage (VOUT), and output current (IOUT) requirements.
2. Optimize the design for key parameters such as efficiency, footprint, and cost using the optimizer dial.
3. Compare the generated design with other possible solutions from Texas Instruments.
The WEBENCH Power Designer provides a customized schematic along with a list of materials with real-time
pricing and component availability.
In most cases, these actions are available:
• Run electrical simulations to see important waveforms and circuit performance
• Run thermal simulations to understand board thermal performance
Submit Documentation Feedback
Copyright © 2004–2019, Texas Instruments Incorporated
Product Folder Links: LM2743
19
LM2743
SNVS276I – APRIL 2004 – REVISED FEBRUARY 2019
www.ti.com
Typical Applications (continued)
•
•
Export customized schematic and layout into popular CAD formats
Print PDF reports for the design, and share the design with colleagues
Get more information about WEBENCH tools at www.ti.com/WEBENCH.
8.2.1.2.2 Duty Cycle Calculation
The complete duty cycle for a buck converter is defined with Equation 16:
VOUT + VSWL
D=
VIN - VSWH + VSWL
(16)
where VSWL and VSWH are the respective forward voltage drops that develop across the low side and high side
MOSFETs. Assuming the inductor ripple current is 20% to 30% of the output current, therefore:
VSWL = IOUT x RDS(ON)LOW (Low-Side MOSFET)
VSWH = IOUT x RDS(ON)HIGH (High-Side MOSFET)
(17)
(18)
To calculate the maximum duty cycle use the estimated 'hot' RDS(on) value of the MOSFETs, the minimum input
voltage, and maximum load. As shown in Figure 31, the worst case maximum duty cycles of the LM2743 occurs
at 125°C junction temperature vs VCC (IC control section voltage). Ensure that the operating duty cycle is below
the curve in Figure 31, if this condition is not satisfied, the system will be unable to develop the required duty
cycle to derive the necessary system power and so the output voltage will fall out of regulation.
Figure 31. Maximum Duty Cycle vs VCC
TJ = 125°C
8.2.1.2.3 Input Capacitor
The input capacitors in a Buck converter are subjected to high stress due to the input current trapezoidal
waveform. Input capacitors are selected for their ripple current capability and their ability to withstand the heat
generated since that ripple current passes through their ESR. Input rms ripple current is approximately:
IRMS_RIP = IOUT x
D(1 - D)
(19)
The power dissipated by each input capacitor is:
(IRMS_RIP)2 x ESR
PCAP =
20
n2
(20)
Submit Documentation Feedback
Copyright © 2004–2019, Texas Instruments Incorporated
Product Folder Links: LM2743
LM2743
www.ti.com
SNVS276I – APRIL 2004 – REVISED FEBRUARY 2019
Typical Applications (continued)
where n is the number of capacitors, and ESR is the equivalent series resistance of each capacitor. The equation
above indicates that power loss in each capacitor decreases rapidly as the number of input capacitors increases.
The worst-case ripple for a Buck converter occurs during full load and when the duty cycle (D) is 0.5. For this
3.3V to 1.2V design the duty cycle is 0.364. For a 4A maximum load the ripple current is 1.92A.
8.2.1.2.4 Output Inductor
The output inductor forms the first half of the power stage in a Buck converter. It is responsible for smoothing the
square wave created by the switching action and for controlling the output current ripple (ΔIOUT). The inductance
is chosen by selecting between tradeoffs in efficiency and response time. The smaller the output inductor, the
more quickly the converter can respond to transients in the load current. However, as shown in the efficiency
calculations, a smaller inductor requires a higher switching frequency to maintain the same level of output current
ripple. An increase in frequency can mean increasing loss in the MOSFETs due to the charging and discharging
of the gates. Generally the switching frequency is chosen so that conduction loss outweighs switching loss. The
equation for output inductor selection is:
VIN - VOUT
L=
L=
xD
'IOUT x fSW
(21)
3.3V - 1.2V
1.2V
x
3.3V
0.4 x 4A x 300 kHz
(22)
(23)
L = 1.6 µH
Here we have plugged in the values for output current ripple, input voltage, output voltage, switching frequency,
and assumed a 40% peak-to-peak output current ripple. This yields an inductance of 1.6 µH. The output inductor
must be rated to handle the peak current (also equal to the peak switch current), which is (IOUT + (0.5 x ΔIOUT)) =
4.8 A, for a 4 A design. The Coilcraft DO3316P-222P is 2.2 µH, is rated to 7.4-A peak, and has a direct current
resistance (DCR) of 12 mΩ.
After selecting an output inductor, inductor current ripple should be re-calculated with the new inductance value,
as this information is needed to select the output capacitor. Re-arranging the equation used to select inductance
yields the following:
VIN(MAX) - VO
'IOUT =
FSW x LACTUAL
xD
(24)
VIN(MAX) is assumed to be 10% above the steady state input voltage, or 3.6V. The actual current ripple will then
be 1.2A. Peak inductor/switch current will be 4.6A.
8.2.1.2.5 Output Capacitor
The output capacitor forms the second half of the power stage of a Buck switching converter. It is used to control
the output voltage ripple (ΔVOUT) and to supply load current during fast load transients.
In this example the output current is 4 A and the expected type of capacitor is an aluminum electrolytic, as with
the input capacitors. Other possibilities include ceramic, tantalum, and solid electrolyte capacitors, however the
ceramic type often do not have the large capacitance needed to supply current for load transients, and tantalums
tend to be more expensive than aluminum electrolytic. Aluminum capacitors tend to have very high capacitance
and fairly low ESR, meaning that the ESR zero, which affects system stability, will be much lower than the
switching frequency. The large capacitance means that at the switching frequency, the ESR is dominant, hence
the type and number of output capacitors is selected on the basis of ESR. One simple formula to find the
maximum ESR based on the desired output voltage ripple, ΔVOUT and the designed output current ripple, ΔIOUT,
is:
ESRMAX =
'VOUT
'IOUT
(25)
Submit Documentation Feedback
Copyright © 2004–2019, Texas Instruments Incorporated
Product Folder Links: LM2743
21
LM2743
SNVS276I – APRIL 2004 – REVISED FEBRUARY 2019
www.ti.com
Typical Applications (continued)
In this example, in order to maintain a 2% peak-to-peak output voltage ripple and a 40% peak-to-peak inductor
current ripple, the required maximum ESR is 20 mΩ. The Sanyo 4SP560M electrolytic capacitor will give an
equivalent ESR of 14 mΩ. The capacitance of 560 µF is enough to supply energy even to meet severe load
transient demands.
8.2.1.2.6 MOSFETs
Selection of the power MOSFETs is governed by a tradeoff between cost, size, and efficiency. One method is to
determine the maximum cost that can be endured, and then select the most efficient device that fits that price.
Breaking down the losses in the high-side and low-side MOSFETs and then creating spreadsheets is one way to
determine relative efficiencies between different MOSFETs. Good correlation between the prediction and the
bench result is not specified, however. Single-channel buck regulators that use a controller IC and discrete
MOSFETs tend to be most efficient for output currents of 2A to 10A.
Losses in the high-side MOSFET can be broken down into conduction loss, gate charging loss, and switching
loss. Conduction loss, or I2R loss, is approximately:
PC = D ((IO)2 x RDSON-HI x 1.3) (High-Side MOSFET)
PC = (1 - D) x ((IO)2 x RDSON-LO x 1.3) (Low-Side MOSFET)
(26)
(27)
In the above equations, the factor 1.3 accounts for the increase in MOSFET RDSON due to heating. Alternatively,
the 1.3 can be ignored and the RDSON of the MOSFET estimated using the RDSON Vs. Temperature curves in the
MOSFET datasheets.
Gate charging loss results from the current driving the gate capacitance of the power MOSFETs, and is
approximated as:
PGC = n x (VDD) x QG x fSW
(28)
where ‘n’ is the number of MOSFETs (if multiple devices have been placed in parallel), VDD is the driving voltage
(see MOSFET Gate Drivers section) and QGS is the gate charge of the MOSFET. If different types of MOSFETs
are used, the n term can be ignored and their gate charges simply summed to form a cumulative QG. Gate
charge loss differs from conduction and switching losses in that the actual dissipation occurs in the LM2743, and
not in the MOSFET itself.
Switching loss occurs during the brief transition period as the high-side MOSFET turns on and off, during which
both current and voltage are present in the channel of the MOSFET. It can be approximated as:
PSW = 0.5 x VIN x IO x (tr + tf) x fSW
(29)
where tR and tF are the rise and fall times of the MOSFET. Switching loss occurs in the high-side MOSFET only.
For this example, the maximum drain-to-source voltage applied to either MOSFET is 3.6V. The maximum drive
voltage at the gate of the high-side MOSFET is 3.1V, and the maximum drive voltage for the low-side MOSFET
is 3.3V. Due to the low drive voltages in this example, a MOSFET that turns on fully with 3.1V of gate drive is
needed. For designs of 5A and under, dual MOSFETs in SOIC-8 package provide a good trade-off between size,
cost, and efficiency.
8.2.1.2.7 Support Components
CIN2 - A small value (0.1-µF to 1-µF) ceramic capacitor should be placed as close as possible to the drain of the
high-side MOSFET and source of the low-side MOSFET (dual MOSFETs make this easy). This capacitor should
be X5R type dielectric or better.
RCC, CCC- These are standard filter components designed to ensure smooth DC voltage for the chip supply. RCC
should be 1 Ω to 10 Ω. CCC should 1 µF, X5R type or better.
CBOOT- Bootstrap capacitor, typically 100 nF.
RPULL-UP – This is a standard pull-up resistor for the open-drain power good signal (PWGD). The recommended
value is 10 kΩ connected to VCC. If this feature is not necessary, the resistor can be omitted.
22
Submit Documentation Feedback
Copyright © 2004–2019, Texas Instruments Incorporated
Product Folder Links: LM2743
LM2743
www.ti.com
SNVS276I – APRIL 2004 – REVISED FEBRUARY 2019
Typical Applications (continued)
D1 - A small Schottky diode should be used for the bootstrap. It allows for a minimum drop for both high and lowside drivers. The MBR0520 or BAT54 work well in most designs.
RCS - Resistor used to set the current limit. Since the design calls for a peak current magnitude (IOUT+ (0.5 x
ΔIOUT)) of 4.8 A, a safe setting would be 6A. (This is below the saturation current of the output inductor, which is
7 A.) Following the equation from the Current Limit section, a 1.3-kΩ resistor should be used.
RFADJ - This resistor is used to set the switching frequency of the chip. The resistor value is calculated from
equation in Normal Operation section. For 300-kHz operation, a 97.6-kΩ resistor should be used.
CSS - The soft-start capacitor depends on the user requirements and is calculated based on the equation given in
the section titled Start Up and Soft-Start. Therefore, for a 700-μs delay, a 12-nF capacitor is suitable.
8.2.1.2.8 Control Loop Compensation
The LM2743 uses voltage-mode (‘VM’) PWM control to correct changes in output voltage due to line and load
transients. One of the attractive advantages of voltage mode control is its relative immunity to noise and layout.
However VM requires careful small signal compensation of the control loop for achieving high bandwidth and
good phase margin.
The control loop is comprised of two parts. The first is the power stage, which consists of the duty cycle
modulator, output inductor, output capacitor, and load. The second part is the error amplifier, which for the
LM2743 is a 9-MHz op-amp used in the classic inverting configuration. Figure 32 shows the regulator and control
loop components.
L
RL
+ C
O
VIN
RO
+
-
RC
+
VRAMP
RC2
CC2
RC1
10 k:
CC3
CC1
+
10 k:
+
-
VREF
Figure 32. Power Stage and Error Amplifier
One popular method for selecting the compensation components is to create Bode plots of gain and phase for
the power stage and error amplifier. Combined, they make the overall bandwidth and phase margin of the
regulator easy to see. Software tools such as Excel, MathCAD, and Matlab are useful for showing how changes
in compensation or the power stage affect system gain and phase.
The power stage modulator provides a DC gain ADC that is equal to the input voltage divided by the peak-to-peak
value of the PWM ramp. This ramp is 1.0VP-P for the LM2743. The inductor and output capacitor create a
double pole at frequency fDP, and the capacitor ESR and capacitance create a single zero at frequency fESR. For
this example, with VIN = 3.3 V, these quantities are:
VIN
ADC =
fDP =
VRAMP
1
2S
=
3.3
= 10.4 dB
1.0
RO + RL
LCO(RO + ESR)
(30)
= 4.5 kHz
(31)
Submit Documentation Feedback
Copyright © 2004–2019, Texas Instruments Incorporated
Product Folder Links: LM2743
23
LM2743
SNVS276I – APRIL 2004 – REVISED FEBRUARY 2019
www.ti.com
Typical Applications (continued)
fESR =
1
2SCOESR
= 20.3 kHz
(32)
In the equation for fDP, the variable RL is the power stage resistance, and represents the inductor DCR plus the
on resistance of the top power MOSFET. RO is the output voltage divided by output current. The power stage
transfer function GPS is given by the following equation, and Figure 34 shows Bode plots of the phase and gain in
this example.
VIN x RO
GPS =
VRAMP
x
sCORC + 1
a x s2 + b x s + c
where
a = LCO(RO + RC)
b = L + CO(RORL + RORC + RCRL)
c = RO + RL
(33)
20
0
4
-30
PHASE (o)
GAIN (dB)
•
•
•
-12
-28
-44
-60
-90
-120
-60
-150
100
1k
10k
100k
100
1M
1k
10k
100k
1M
FREQUENCY (Hz)
FREQUENCY (Hz)
Figure 33. Gain vs Frequency
Figure 34. Power Stage Gain and Phase
The double pole at 4.5 kHz causes the phase to drop to approximately -130° at around 10 kHz. The ESR zero, at
20.3 kHz, provides a +90° boost that prevents the phase from dropping to -180º. If this loop were left
uncompensated, the bandwidth would be approximately 10 kHz and the phase margin 53°. In theory, the loop
would be stable, but would suffer from poor DC regulation (due to the low DC gain) and would be slow to
respond to load transients (due to the low bandwidth.) In practice, the loop could easily become unstable due to
tolerances in the output inductor, capacitor, or changes in output current, or input voltage. Therefore, the loop is
compensated using the error amplifier and a few passive components.
For this example, a Type III, or three-pole-two-zero approach gives optimal bandwidth and phase.
In most voltage mode compensation schemes, including Type III, a single pole is placed at the origin to boost DC
gain as high as possible. Two zeroes fZ1 and fZ2 are placed at the double pole frequency to cancel the double
pole phase lag. Then, a pole, fP1 is placed at the frequency of the ESR zero. A final pole fP2 is placed at one-half
of the switching frequency. The gain of the error amplifier transfer function is selected to give the best bandwidth
possible without violating the Nyquist stability criteria. In practice, a good crossover point is one-fifth of the
switching frequency, or 60 kHz for this example. The generic equation for the error amplifier transfer function is:
s
+1
2SfZ1
GEA = AEA x
s
24
s
+1
2SfP1
s
+1
2SfZ2
s
+1
2SfP2
(34)
Submit Documentation Feedback
Copyright © 2004–2019, Texas Instruments Incorporated
Product Folder Links: LM2743
LM2743
www.ti.com
SNVS276I – APRIL 2004 – REVISED FEBRUARY 2019
Typical Applications (continued)
In this equation, the variable AEA is a ratio of the values of the capacitance and resistance of the compensation
components, arranged as shown in Figure 32. AEA is selected to provide the desired bandwidth. A starting value
of 80,000 for AEA should give a conservative bandwidth. Increasing the value will increase the bandwidth, but will
also decrease phase margin. Designs with 45° to 60° are usually best because they represent a good trade-off
between bandwidth and phase margin. In general, phase margin is lowest and gain highest (worst-case) for
maximum input voltage and minimum output current. One method to select AEA is to use an iterative process
beginning with these worst-case conditions.
1. Increase AEA
2. Check overall bandwidth and phase margin
3. Change VIN to minimum and recheck overall bandwidth and phase margin
4. Change IO to maximum and recheck overall bandwidth and phase margin
The process ends when the both bandwidth and the phase margin are sufficiently high. For this example input
voltage can vary from 3.0 to 3.6 V and output current can vary from 0 to 4 A, and after a few iterations a
moderate gain factor of 101 dB is used.
The error amplifier of the LM2743 has a unity-gain bandwidth of 9 MHz. In order to model the effect of this
limitation, the open-loop gain can be calculated as:
OPG =
2S x 9 MHz
s
(35)
The new error amplifier transfer function that takes into account unity-gain bandwidth is:
GEA x OPG
HEA =
1 + GEA + OPG
(36)
60
50
48
20
PHASE (o)
GAIN (dB)
The gain and phase of the error amplifier are shown in Figure 36.
36
24
-10
-40
-70
12
-100
0
100
1k
10k
100k
1M
100
1k
10k
100k
1M
FREQUENCY (Hz)
FREQUENCY (Hz)
Figure 35. Gain vs Frequency
Figure 36. Error Amplifier Gain and Phase
In VM regulators, the top feedback resistor RFB2 forms a part of the compensation. Setting RFB2 to 10 kΩ, ±1%
usually gives values for the other compensation resistors and capacitors that fall within a reasonable range.
(Capacitances > 1 pF, resistances < 1 MΩ) CC1, CC2, CC3, RC1, and RC2 are selected to provide the poles and
zeroes at the desired frequencies, using the following equations:
fZ1
CC1 =
CC2 =
AEA x 10,000 x fP2
1
AEA x 10,000
= 27 pF
(37)
- CC1 = 882 pF
(38)
Submit Documentation Feedback
Copyright © 2004–2019, Texas Instruments Incorporated
Product Folder Links: LM2743
25
LM2743
SNVS276I – APRIL 2004 – REVISED FEBRUARY 2019
www.ti.com
Typical Applications (continued)
CC3 =
RC1 =
RC2 =
1
2S x 10,000
x
1
2S x CC2 x fZ1
1
2S x CC3 x fP1
1
1
= 2.73 nF
fZ2 fP1
(39)
= 39.8 k:
(40)
= 2.55 k:
(41)
In practice, a good trade off between phase margin and bandwidth can be obtained by selecting the closest
±10% capacitor values above what are suggested for CC1 and CC2, the closest ±10% capacitor value below the
suggestion for CC3, and the closest ±1% resistor values below the suggestions for RC1, RC2. Note that if the
suggested value for RC2 is less than 100Ω, it should be replaced by a short circuit. Following this guideline, the
compensation components will be:
CC1 = 27 pF ±10%
CC2 = 820 pF ±10%
CC3 = 2.7 nF ±10%
RC1 = 39.2 kΩ ±1%
RC2 = 2.55 kΩ ±1%
The transfer function of the compensation block can be derived by considering the compensation components as
impedance blocks ZF and ZI around an inverting op-amp:
ZF
GEA-ACTUAL =
ZI
(42)
1
1
x 10,000 +
sCC2
sCC1
ZF =
10,000 +
1
1
+
sCC1 sCC2
(43)
RC1 RC2 +
1
sCC3
Z1 =
RC1 + RC2 +
1
sCC3
(44)
As with the generic equation, GEA-ACTUAL must be modified to take into account the limited bandwidth of the error
amplifier. The result is:
GEA-ACTUAL x OPG
HEA =
1 + GEA-ACTUAL+ OPG
(45)
The total control loop transfer function H is equal to the power stage transfer function multiplied by the error
amplifier transfer function.
H = GPS x HEA
26
(46)
Submit Documentation Feedback
Copyright © 2004–2019, Texas Instruments Incorporated
Product Folder Links: LM2743
LM2743
www.ti.com
SNVS276I – APRIL 2004 – REVISED FEBRUARY 2019
Typical Applications (continued)
60
-60
40
-84
PHASE (o)
GAIN (dB)
The bandwidth and phase margin can be read graphically from Bode plots of HEA are shown in Figure 38.
20
0
-20
-108
-132
-156
-40
-180
100
1k
10k
100k
1M
100
FREQUENCY (Hz)
1k
10k
100k
1M
FREQUENCY (Hz)
Figure 37. Gain vs Frequency
Figure 38. Overall Loop Gain and Phase
The bandwidth of this example circuit is 59 kHz, with a phase margin of 60°.
8.2.1.2.9 Efficiency Calculations
The following is a sample calculation.
A reasonable estimation of the efficiency of a switching buck controller can be obtained by adding together the
Output Power (POUT) loss and the Total Power (PTOTAL) loss:
POUT
K=
x 100%
POUT + PTOTAL
(47)
The Output Power (POUT) for the Figure 30 design is (1.2 V x 4 A) = 4.8 W. The Total Power (PTOTAL), with an
efficiency calculation to complement the design, is shown below.
The majority of the power losses are due to low and high side of MOSFET’s losses. The losses in any MOSFET
are group of switching (PSW) and conduction losses(PCND).
PFET = PSW + PCND = 61.38 mW + 270.42 mW
PFET = 331.8 mW
(48)
(49)
The following equations show FET Switching Loss (PSW).
PSW =
PSW =
PSW =
PSW =
PSW(ON) + PSW(OFF)
0.5 x VIN x IOUT x (tr + tf) x fSW
0.5 x 3.3 V x 4 A x 300 kHz x 31 ns
61.38 mW
(50)
(51)
(52)
(53)
The FDS6898A has a typical turn-on rise time tr and turn-off fall time tf of 15 ns and 16 ns, respectively. The
switching losses for this type of dual N-Channel MOSFETs are 0.061 W.
The following equations show FET Conduction Loss (PCND).
PCND = PCND1 + PCND2
PCND1 = (IOUT)2 x RDS(ON) x k x D
PCND2 = (IOUT)2 x RDS(ON) x k x (1-D)
(54)
(55)
(56)
RDS(ON) = 13 mΩ and the factor is a constant value (k = 1.3) to account for the increasing RDS(ON) of a FET due to
heating.
PCND1 = (4A)2 x 13 mΩ x 1.3 x 0.364
(57)
Submit Documentation Feedback
Copyright © 2004–2019, Texas Instruments Incorporated
Product Folder Links: LM2743
27
LM2743
SNVS276I – APRIL 2004 – REVISED FEBRUARY 2019
www.ti.com
Typical Applications (continued)
PCND2 = (4A)2 x 13 mΩ x 1.3 x (1 - 0.364)
PCND = 98.42 mW + 172 mW = 270.42 mW
(58)
(59)
There are few additional losses that are taken into account:
The following equations show IC Operating Loss (PIC).
PIC = IQ_VCC x VCC,
(60)
where IQ-VCC is the typical operating VCC current
PIC= 1.5 mA x 3.3V = 4.95 mW
(61)
The following equations show FET Gate Charging Loss (PGATE).
PGATE = n x VCC x QGS x fSW
PGATE = 2 x 3.3 V x 3 nC x 300 kHz
PGATE = 5.94 mW
(62)
(63)
(64)
The value n is the total number of FETs used and QGS is the typical gate-source charge value, which is 3 nC. For
the FDS6898A the gate charging loss is 5.94 mW.
The following equations show Input Capacitor Loss (PCAP).
(IRMS_RIP)2 x ESR
PCAP =
n2
where
(65)
IRMS_RIP = IOUT x
D(1 - D)
(66)
Here n is the number of paralleled capacitors, ESR is the equivalent series resistance of each, and PCAP is the
dissipation in each. So for example if we use only one input capacitor of 24 mΩ.
PCAP =
(1.924A)2 x 24 m:
12
(67)
(68)
PCAP = 88.8 mW
The following equation shows Output Inductor Loss (PIND).
PIND = I2OUT x DCR
(69)
where DCR is the DC resistance. Therefore, for example
PIND = (4A)2 x 11 mΩ
PIND = 176 mW
(70)
(71)
The following equations show Total System Efficiency.
PTOTAL = PFET + PIC + PGATE + PCAP + PIND
(72)
POUT
K=
K=
28
x 100%
POUT + PTOTAL
(73)
4.8W
= 89%
4.8W + 0.6W
(74)
Submit Documentation Feedback
Copyright © 2004–2019, Texas Instruments Incorporated
Product Folder Links: LM2743
LM2743
www.ti.com
SNVS276I – APRIL 2004 – REVISED FEBRUARY 2019
Typical Applications (continued)
8.2.1.3 Application Curves
VCC = 3.3V
IOUT = 4A
VOUT = 1.2V
fSW = 300 kHz
CSS = 12nF
VCC = 3.3 V
IOUT = 4 A
Figure 39. Start-Up (Full-Load)
CSS = 12 nF
VOUT = 1.2 V
fSW = 300 kHz
Figure 40. Shutdown (Full-Load)
8.2.2 Example Circuit 1
VCC = VIN= 3.3V
D1
RCC
CBOOT
RPULL-UP
+ CIN1,2
Q1
VCC
HG
SD
CCC
BOOT
LM2743
FREQ
RFADJ
CSS
RCS
VOUT = 1.8V@2A
L1
ISEN
PWGD
LG
SS/TRACK
PGND
SGND
PGND
EAO
FB
RFB2
RC2
CC3
+
CO1,2
RFB1
CC1
CC2
RC1
Figure 41. 3.3 V to 1.8 V at 2 A, fSW = 300 kHz
8.2.2.1 Design Requirements
This design converts 3.3 V (VIN) to 1.8 V (VOUT) at a maximum load of 2 A, with a switching frequency of 300
kHz.
8.2.2.2 Detailed Design Procedure
Follow the detailed design procedure in Detailed Design Procedure.
Submit Documentation Feedback
Copyright © 2004–2019, Texas Instruments Incorporated
Product Folder Links: LM2743
29
LM2743
SNVS276I – APRIL 2004 – REVISED FEBRUARY 2019
www.ti.com
Typical Applications (continued)
8.2.2.3 Bill of Materials
Table 1. Bill of Materials
PART
PART NUMBER
TYPE
PACKAGE
U1
LM2743
Synchronous
Controller
TSSOP-14
Q1
FDS6898A
Dual N-MOSFET
SOIC-8
D1
MBR0520LTI
Schottky Diode
SOD-123
L1
DO3316P-472
Inductor
CIN1
16SP100M
Aluminum Electrolytic
CO1
6SP220M
CCC, CBOOT,
CIN2, CO2
DESCRIPTION
VENDOR
TI
20 V, 10 mΩ at 4.5 V,
16 nC
Fairchild
4.7 µH, 4.8 Arms, 18
mΩ
Coilcraft
10mm x 6mm
100 µF, 16 V, 2.89
Arms
Sanyo
Aluminum Electrolytic
10mm x 6mm
220 µF, 6.3 V, 3.1
Arms
Sanyo
VJ1206Y104KXXA
Capacitor
1206
0.1 µF, 10%
Vishay
CC3
VJ0805Y332KXXA
Capacitor
805
3300 pF, 10%
Vishay
CSS
VJ0805A123KXAA
Capacitor
805
12 nF, 10%
Vishay
CC2
VJ0805A821KXAA
Capacitor
805
820 pF 10%
Vishay
CC1
VJ0805A220KXAA
Capacitor
805
22 pF, 10%
Vishay
RFB2
CRCW08051002F
Resistor
805
10.0 kΩ 1%
Vishay
RFB1
CRCW08054991F
Resistor
805
4.99 kΩ1%
Vishay
RFADJ
CRCW08051103F
Resistor
805
110 kΩ 1%
Vishay
RC2
CRCW08052101F
Resistor
805
2.1 kΩ 1%
Vishay
RCS
CRCW08052101F
Resistor
805
2.1 kΩ 1%
Vishay
RCC
CRCW080510R0F
Resistor
805
10.0 Ω 1%
Vishay
RC1
CRCW08055492F
Resistor
805
54.9 kΩ 1%
Vishay
RPULL-UP
CRCW08051003J
Resistor
805
100 kΩ 5%
Vishay
8.2.3 Example Circuit 2
VCC = 5V
D1
RCC
CBOOT
RPULL-UP
HG
SD
BOOT
PWGD
RFADJ
CSS
+ CIN1,2
Q1
VCC
CCC
VIN = 5V
LM2743
RCS
L1
VOUT = 2.5V@2A
ISEN
FREQ
LG
SS/TRACK
PGND
SGND
PGND
EAO
FB
RFB2
RC2
CC3
+
CO1,2
RFB1
CC1
CC2
RC1
Figure 42. 5 V to 2.5 V at 2A, fSW = 300kHz
8.2.3.1 Design Requirements
This design converts 5 V (VIN) to 2.5 V (VOUT) at a maximum load of 2 A, with a switching frequency of 300 kHz.
30
Submit Documentation Feedback
Copyright © 2004–2019, Texas Instruments Incorporated
Product Folder Links: LM2743
LM2743
www.ti.com
SNVS276I – APRIL 2004 – REVISED FEBRUARY 2019
8.2.3.2 Detailed Design Procedure
Follow the detailed design procedure in Detailed Design Procedure.
8.2.3.3 Bill of Materials
Table 2. Bill of Materials
PART
PART NUMBER
TYPE
PACKAGE
U1
LM2743
Synchronous
Controller
TSSOP-14
Q1
FDS6898A
Dual N-MOSFET
SOIC-8
D1
MBR0520LTI
Schottky Diode
SOD-123
L1
DO3316P-682
Inductor
CIN1
16SP100M
Aluminum
Electrolytic
10mm x 6mm
CO1
10SP56M
Aluminum
Electrolytic
CCC, CBOOT,
CIN2, CO2
VJ1206Y104KXXA
CC3
CSS
DESCRIPTION
VENDOR
TI
20 V, 10 mΩ at 4.5 V, 16
nC
Fairchild
6.8 µH, 4.4 Arms, 27 mΩ
Coilcraft
100 µF, 16 V, 2.89 Arms
Sanyo
6.3mm x 6mm
56 µF, 10 V, 1.7 Arms
Sanyo
Capacitor
1206
0.1 µF, 10%
Vishay
VJ0805Y182KXXA
Capacitor
805
1800 pF, 10%
Vishay
VJ0805A123KXAA
Capacitor
805
12 nF, 10%
Vishay
CC2
VJ0805A821KXAA
Capacitor
805
820 pF 10%
Vishay
CC1
VJ0805A330KXAA
Capacitor
805
33 pF, 10%
Vishay
RFB2
CRCW08051002F
Resistor
805
10.0 kΩ 1%
Vishay
RFB1
CRCW08053161F
Resistor
805
3.16 kΩ 1%
Vishay
RFADJ
CRCW08051103F
Resistor
805
110 kΩ 1%
Vishay
RC2
CRCW08051301F
Resistor
805
1.3 kΩ 1%
Vishay
RCS
CRCW08052101F
Resistor
805
2.1 kΩ 1%
Vishay
RCC
CRCW080510R0F
Resistor
805
10.0 Ω 1%
Vishay
RC1
CRCW08053322F
Resistor
805
33.2 kΩ 1%
Vishay
RPULL-UP
CRCW08051003J
Resistor
805
100 kΩ 5%
Vishay
8.2.4 Example Circuit 3
VCC = 5V
D1
RCC
CBOOT
RPULL-UP
HG
SD
BOOT
PWGD
RFADJ
CSS
+ CIN1,2
Q1
VCC
CCC
VIN = 12V
LM2743
RCS
L1
VOUT = 3.3V@4A
ISEN
FREQ
LG
SS/TRACK
PGND
SGND
PGND
EAO
FB
RFB2
RC2
CC3
+
CO1,2
RFB1
CC1
CC2
RC1
Figure 43. 12 V to 3.3 V at 4 A, fSW = 300 kHz
8.2.4.1 Design Requirements
This design converts 12 V (VIN) to 3.3 V (VOUT) at a maximum load of 4 A, with a switching frequency of 300 kHz.
Submit Documentation Feedback
Copyright © 2004–2019, Texas Instruments Incorporated
Product Folder Links: LM2743
31
LM2743
SNVS276I – APRIL 2004 – REVISED FEBRUARY 2019
www.ti.com
8.2.4.2 Detailed Design Procedure
Follow the detailed design procedure in Detailed Design Procedure.
8.2.4.3 Bill of Materials
Table 3. Bill of Materials
PART
PART NUMBER
TYPE
PACKAGE
U1
LM2743
Synchronous
Controller
TSSOP-14
Q1
FDS6898A
Dual N-MOSFET
SOIC-8
D1
MBR0520LTI
Schottky Diode
SOD-123
L1
DO3316P-332
Inductor
CIN1
16SP100M
Aluminum Electrolytic
10 mm x 6 mm
CO1
6SP220M
Aluminum Electrolytic
CCC, CBOOT,
CIN2, CO2
VJ1206Y104KXXA
CC3
DESCRIPTION
VENDOR
TI
20 V, 10 mΩ at 4.5 V, 16
nC
Fairchild
3.3 µH, 5.4 Arms, 15 mΩ
Coilcraft
100 µF, 16 V, 2.89 Arms
Sanyo
10 mm x 6 mm
220 µF, 6.3 V 3.1 Arms
Sanyo
Capacitor
1206
0.1 µF, 10%
Vishay
VJ0805Y222KXXA
Capacitor
805
2200 pF, 10%
Vishay
CSS
VJ0805A123KXAA
Capacitor
805
12 nF, 10%
Vishay
CC2
VJ0805Y332KXXA
Capacitor
805
3300 pF 10%
Vishay
CC1
VJ0805A820KXAA
Capacitor
805
82 pF, 10%
Vishay
RFB2
CRCW08051002F
Resistor
805
10.0 kΩ 1%
Vishay
RFB1
CRCW08052211F
Resistor
805
2.21 kΩ 1%
Vishay
RFADJ
CRCW08051103F
Resistor
805
110 kΩ 1%
Vishay
RC2
CRCW08052611F
Resistor
805
2.61 kΩ 1%
Vishay
RCS
CRCW08054121F
Resistor
805
4.12 kΩ 1%
Vishay
RCC
CRCW080510R0F
Resistor
805
10.0 Ω 1%
Vishay
RC1
CRCW08051272F
Resistor
805
12.7 kΩ 1%
Vishay
RPULL-UP
CRCW08051003J
Resistor
805
100 kΩ 5%
Vishay
32
Submit Documentation Feedback
Copyright © 2004–2019, Texas Instruments Incorporated
Product Folder Links: LM2743
LM2743
www.ti.com
SNVS276I – APRIL 2004 – REVISED FEBRUARY 2019
9 Power Supply Recommendations
The LM2743 is a power management device. The power supply for the device is any DC voltage source within
the specified input range (see Design Requirements).
10 Layout
10.1 Layout Guidelines
In a buck regulator, the primary switching loop consists of the input capacitor and MOSFET switches. Minimixing
the area of this loop reduces the stray inductance, and minimizes noise and possible erratic operation. High
quality input capacitors should be placed as close as possible to the MOSFET switches, with the positive side of
the input capacitor connected directly to the high-side MOSFET drain, and the ground side of the capacitor
connected as close as possible to the low-side MOSFET switch ground connection. Connect all of the low power
ground connections directly to the SGND pin. Connect the VCC capacitor directly to the PGND pin.
10.2 Layout Example
Controller
Place controller as
close to the switches
Inductor
QL
QH
CIN
COUT
CIN
VIN
COUT
GND GND
VOUT
Figure 44. Layout Recommendation
Submit Documentation Feedback
Copyright © 2004–2019, Texas Instruments Incorporated
Product Folder Links: LM2743
33
LM2743
SNVS276I – APRIL 2004 – REVISED FEBRUARY 2019
www.ti.com
11 Device and Documentation Support
11.1 Device Support
11.1.1 Development Support
11.1.1.1 Custom Design With WEBENCH® Tools
Click here to create a custom design using the LM2743 device with the WEBENCH® Power Designer.
1. Start by entering the input voltage (VIN), output voltage (VOUT), and output current (IOUT) requirements.
2. Optimize the design for key parameters such as efficiency, footprint, and cost using the optimizer dial.
3. Compare the generated design with other possible solutions from Texas Instruments.
The WEBENCH Power Designer provides a customized schematic along with a list of materials with real-time
pricing and component availability.
In most cases, these actions are available:
• Run electrical simulations to see important waveforms and circuit performance
• Run thermal simulations to understand board thermal performance
• Export customized schematic and layout into popular CAD formats
• Print PDF reports for the design, and share the design with colleagues
Get more information about WEBENCH tools at www.ti.com/WEBENCH.
11.2 Receiving Notification of Documentation Updates
To receive notification of documentation updates, navigate to the device product folder on ti.com. In the upper
right corner, click on Alert me to register and receive a weekly digest of any product information that has
changed. For change details, review the revision history included in any revised document.
11.3 Community Resources
The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective
contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of
Use.
TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration
among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help
solve problems with fellow engineers.
Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and
contact information for technical support.
11.4 Trademarks
E2E is a trademark of Texas Instruments.
WEBENCH is a registered trademark of Texas Instruments.
All other trademarks are the property of their respective owners.
11.5 Electrostatic Discharge Caution
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
11.6 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
34
Submit Documentation Feedback
Copyright © 2004–2019, Texas Instruments Incorporated
Product Folder Links: LM2743
LM2743
www.ti.com
SNVS276I – APRIL 2004 – REVISED FEBRUARY 2019
12 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
Submit Documentation Feedback
Copyright © 2004–2019, Texas Instruments Incorporated
Product Folder Links: LM2743
35
PACKAGE OPTION ADDENDUM
www.ti.com
30-Sep-2021
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
LM2743MTC
NRND
TSSOP
PW
14
94
Non-RoHS
& Green
Call TI
Level-1-260C-UNLIM
-40 to 125
2743
MTC
LM2743MTC/NOPB
ACTIVE
TSSOP
PW
14
94
RoHS & Green
NIPDAU | SN
Level-1-260C-UNLIM
-40 to 125
2743
MTC
LM2743MTCX/NOPB
ACTIVE
TSSOP
PW
14
2500
RoHS & Green
NIPDAU | SN
Level-1-260C-UNLIM
-40 to 125
2743
MTC
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
工商网监
湘ICP备2023018690号
