LM2833
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LM2833 1.5MHz/3MHz 3.0A Step-Down DC-DC Switching Regulator
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FEATURES
DESCRIPTION
•
•
•
The LM2833 regulator is a monolithic, high frequency,
PWM step-down DC/DC converter available in a 10pin WSON or MSOP-PowerPAD package. It contains
all the active functions to provide local DC/DC
conversion with fast transient response and accurate
regulation in the smallest possible PCB area. With a
minimum of external components, the LM2833 is
easy to use. The ability to drive 3.0A loads with an
internal 56 mΩ PMOS switch using state-of-the-art
0.5µm BiCMOS technology results in the best power
density available. The world-class control circuitry
allows on-times as low as 30ns, thus supporting
exceptionally high frequency conversion over the
entire 3V to 5.5V input operating range down to the
minimum output voltage of 0.6V. Switching frequency
is internally set to 1.5MHz or 3.0MHz, allowing the
use of extremely small surface mount inductors and
capacitors. Even though the operating frequency is
high, efficiencies up to 93% are easy to achieve.
External shutdown is included, featuring an ultra-low
stand-by current of 300nA. The LM2833 utilizes peak
current-mode control and internal compensation to
provide high-performance regulation over a wide
range of operating conditions. Additional features
include internal soft-start circuitry to reduce inrush
current, cycle-by-cycle current limit, frequency
foldback, thermal shutdown, and output over-voltage
protection.
1
2
•
•
•
•
•
•
•
•
•
•
•
Input Voltage Range of 3.0V to 5.5V
Output Voltage Range of 0.6V to 4.5V
Tiny MSOP-PowerPAD 10 or WSON-10
Package
3.0A Steady-State Output Current
High Switching Frequencies
– 1.5MHz (LM2833X)
– 3.0MHz (LM2833Z)
Enable Pin
56mΩ PMOS Switch
0.6V, 2% Internal Voltage Reference Over Line
and Temperature
Internal Soft-Start
Internally Compensated Peak Current-Mode
Control
Cycle-by-Cycle Current Limit and Thermal
Shutdown
Frequency Foldback Protection
Input Voltage UVLO (Under-Voltage Lockout)
Output Over-Voltage Protection
APPLICATIONS
•
•
•
•
•
•
Multimedia Set Top Box
Broadband Communications
Core Power in HDDs
Data Acquisition/Telemetry
USB Powered Devices
DSL Modems
1
2
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
All trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
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LM2833
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Typical Application Circuit
9,10
VIN
SW
VIND
L1
7,8
VOUT
D1
R1
R3
2
1
C1
EN
LM2833
FB
5
C2
VINC
NC
C3
SGND
3
R2
4
PGND
6
Connection Diagrams
VINC
1
10
VIND
VINC
1
8 10
VIND
EN
2
9
VIND
EN
2
7
9
VIND
SGND
3
8
SW
SGND
3
6
8
SW
NC
4
7
SW
NC
4
5 7
SW
FB
5
6
PGND
FB
5
6
DAP
Figure 1. 10-Pin WSON
See Package DSC
DAP
PGND
Figure 2. 10-pin MSOP-PowerPAD
See Package DGQ
PIN DESCRIPTIONS
2
Pin(s)
Name
Description
1
VINC
Input supply for internal bias and control circuitry. Need to locally bypass this pin to GND.
2
EN
3
SGND
4
NC
No user function, connect this pin to GND.
5
FB
Feedback pin. Connect this pin to the external resistor divider to set output voltage.
6
PGND
7, 8
SW
9, 10
VIND
DAP
Die Attach Pad
Enable control input. Logic high enables operation. Do not allow this pin to float or subject to
voltages greater than VIN + 0.3V.
Signal (analog) ground. Place the bottom resistor of the feedback network as close as possible to
this pin for good load regulation.
Power ground pin. Provides ground return path for the internal driver.
Switch pins. Connect these pins to the inductor and catch diode.
Input supply voltage. Connect a bypass capacitor locally from these pins to PGND.
Connect to system ground for low thermal impedance, but it cannot be used as a primary GND
connection.
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These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
Absolute Maximum Ratings (1) (2)
VINC, VIND
-0.5V to 7V
FB Voltage
-0.5V to 3V
EN Voltage
-0.5V to 7V
SW Voltage
-0.5V to 7V
ESD Susceptibility (3)
2kV
Junction Temperature (4)
150°C
−65°C to +150°C
Storage Temperature
Soldering Information
(1)
(2)
(3)
(4)
Infrared/Convection Reflow (15sec)
220°C
Absolute Maximum Ratings indicate limits beyond which damage to the device may occur, including inoperability and degradation of
device reliability and/or performance. Functional operation of the device and/or non-degradation at the Absolute Maximum Ratings or
other conditions beyond those indicated in the recommended Operating Ratings is not implied. The recommended Operating Ratings
indicate conditions at which the device is functional and should not be operated beyond such conditions.
If Military/Aerospace specified devices are required, please contact Texas Instruments Sales Office/Distributors for availability and
specifications.
Human body model, 1.5kΩ in series with 100pF.
Thermal shutdown will occur if the junction temperature exceeds the maximum junction temperature of the device.
Operating Ratings
VINC, VIND
3V to 5.5V
−40°C to +125°C
Junction Temperature
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Electrical Characteristics
Unless otherwise specified under the Conditions column, VIN = 5V. Limits in standard type are for TJ = 25°C only; limits in
boldface type apply over the junction temperature (TJ) range of -40°C to +125°C. Minimum and Maximum limits are specified
through test, design, or statistical correlation. Typical values represent the most likely parametric norm, and are provided for
reference purposes only.
Symbol
VFB
ΔVFB/(ΔVINxVFB)
IB
UVLO
Parameter
Conditions
Min
Typ
Max
WSON-10 Package
0.588
0.600
0.612
Feedback Voltage
MSOP-PowerPAD-10
Package
0.584
0.600
0.616
Feedback Voltage Line Regulation
VIN = 3V to 5.5V
0.08
Feedback Input Bias Current
VIN Rising
Undervoltage Lockout
fSW
Switching Frequency
DMAX
Maximum Duty Cycle
DMIN
Minimum Duty Cycle
RDS(ON)
ICL
VEN_TH
(1)
4
2.90
LM2833X
1.1
1.5
1.95
LM2833Z
2.25
3.0
3.75
LM2833X
86
95
LM2833Z
80
90
2.35
V
%
7
WSON-10 Package
58
MSOP-PowerPAD 10
Package
56
mΩ
4.4
A
Shutdown Threshold Voltage
%
90
0.4
100
Quiescent Current (switching)
MHz
LM2833Z
1.8
Enable Pin Current
V
5
3.4
Switch Leakage
nA
LM2833X
Enable Threshold Voltage
IEN
VFB_F
2.70
1.85
Switch Current Limit
ISW
IQ
100
0.35
Switch On Resistance
V
%/V
0.1
VIN Falling
UVLO Hysteresis
Units
V
nA
Sink/Source
100
LM2833X, VFB = 0.55
3.2
5
nA
LM2833Z, VFB = 0.55
4.3
6.5
mA
Quiescent Current (shutdown)
All Options VEN = 0V
300
nA
FB Frequency Foldback Threshold
All Options
0.32
V
LM2833X, VFB = 0V
400
LM2833Z, VFB = 0V
800
WSON-10 Package
53
MSOP-PowerPAD-10
Package
50
WSON-10 Package
12
MSOP-PowerPAD-10
Package
11
fFB
Foldback Frequency
kHz
θJA
Junction to Ambient
0 LFPM Air Flow (1)
θJC
Junction to Case (1)
TSD
Thermal Shutdown Threshold
Junction Temperature
Rising
165
°C
TSD_HYS
Thermal Shutdown Hysteresis
Junction Temperature
Falling
15
°C
°C/W
°C/W
Applies for packages soldered directly onto a 4” x 3” 4-layer standard JEDEC board in still air.
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Typical Performance Characteristics
Unless otherwise specified, VIN = 5V and TA = 25°C.
Efficiency vs Load Current - "LM2833X" and "LM2833Z"
Efficiency vs Load Current - "LM2833X"
Figure 3.
Figure 4.
Efficiency vs Load Current - "LM2833Z"
Oscillator Frequency vs Temperature - "LM2833X"
Figure 5.
Figure 6.
Oscillator Frequency vs Temperature - "LM2833Z"
Current Limit vs Temperature
Figure 7.
Figure 8.
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Typical Performance Characteristics (continued)
Unless otherwise specified, VIN = 5V and TA = 25°C.
6
RDS(ON) vs Temperature (WSON-10 Package)
RDS(ON) vs Temperature (MSOP-PowerPAD-10 Package)
Figure 9.
Figure 10.
LM2833X IQ (Quiescent Current)
LM2833Z IQ (Quiescent Current)
Figure 11.
Figure 12.
VFB vs Temperature
Frequency Foldback
Figure 13.
Figure 14.
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Typical Performance Characteristics (continued)
Unless otherwise specified, VIN = 5V and TA = 25°C.
Loop Gain and Phase - "LM2833X"
Loop Gain and Phase - "LM2833Z"
Figure 15.
Figure 16.
Load Step Response - "LM2833X"
Line Transient Response - "LM2833X"
Figure 17.
Figure 18.
Startup by EN - "LM2833X"
Shutdown by EN - "LM2833X"
Figure 19.
Figure 20.
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Typical Performance Characteristics (continued)
Unless otherwise specified, VIN = 5V and TA = 25°C.
8
Startup with EN tied to VIN - "LM2833X"
Short-circuit Triggering - "LM2833X"
Figure 21.
Figure 22.
Short-circuit Release - "LM2833X"
Recovery from Thermal Shutdown - "LM2833X"
Figure 23.
Figure 24.
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Block Diagram
Figure 25. Simplified Block Diagram
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APPLICATION INFORMATION
THEORY OF OPERATION
The LM2833 is a constant frequency PWM buck regulator IC that delivers a 3.0A load current. The regulator is
available in preset switching frequencies of 1.5MHz or 3.0MHz. This high frequency allows the LM2833 to
operate with small surface mount capacitors and inductors, resulting in a DC/DC converter that requires a
minimum amount of board space. The LM2833 is internally compensated, therefore it is simple to use and
requires few external components. The LM2833 uses peak current-mode control to regulate the output voltage.
The following description of operation of the LM2833 will refer to the Typical Application Circuit, to the waveforms
in Figure 26 and simplified block diagram in Figure 25. The LM2833 supplies a regulated output voltage by
switching the internal PMOS power switch at a constant frequency and variable duty cycle. A switching cycle
begins at the falling edge of the reset pulse generated by the internal oscillator. When this pulse goes low, the
output control logic turns on the internal PMOS power switch. During this on-time, the SW pin voltage (VSW)
swings up to approximately VIN, and the inductor current (IL) increases with a linear slope. IL is measured by the
current sense amplifier, which generates an output proportional to the switch current. The sense signal is
summed with the regulator’s corrective ramp and compared to the error amplifier’s output, which is proportional
to the difference between the feedback voltage and VREF. When the PWM comparator output goes high, the
internal power switch turns off until the next switching cycle begins. During the switch off-time, the inductor
current discharges through the catch diode D1, which forces the SW pin to swing below ground by the forward
voltage (VD) of the catch diode. The regulator loop adjusts the duty cycle (D) to maintain a constant output
voltage.
VSW
D = TON/TSW
VIN
SW
Voltage
TOFF
TON
0
-VD
IL
t
TSW
ILPK
IOUT
'iL
Inductor
Current
t
0
Figure 26. SW Pin Voltage and Inductor Current Waveforms
SOFT-START/SHUTDOWN
The LM2833 has both enable and shutdown modes that are controlled by the EN pin. Connecting a voltage
source greater than 1.8V to the EN pin enables the operation of the LM2833, while reducing this voltage below
0.4V places the part in a low quiescent current (300nA typical) shutdown mode. There is no internal pull-up on
EN pin, therefore an external signal is required to initiate switching. Do not allow this pin to float or rise to 0.3V
above VIN. It should be noted that when the EN pin voltage rises above 1.8V while the input voltage is greater
than UVLO, there is 15µs delay before switching starts. During this delay the LM2833 will go through a power on
reset state after which the internal soft-start process commences. During soft-start, the error amplifier’s reference
voltage ramps from 0V to its nominal value of 0.6V in approximately 600µs. This forces the regulator output to
ramp up in a controlled fashion, which helps reduce inrush current seen at the input and minimizes output
voltage overshoot.
The simplest way to enable the operation of the LM2833 is to connect the EN pin to VIN which allows self start-up
of the LM2833 whenever the input voltage is applied. However, when an input voltage of slow rise time is used to
power the application and if both the input voltage and the output voltage are not fully established before the softstart time elapses, the control circuit will command maximum duty cycle operation of the internal power switch to
bring up the output voltage rapidly. When the feedback pin voltage exceeds 0.6V, the duty cycle will have to
10
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reduce from the maximum value accordingly, to maintain regulation. It takes a finite amount of time for this
reduction of duty cycle and this can result in a transient in output voltage for a short duration, as shown in
Figure 27. In applications where this output voltage overshoot is undesirable, one simple solution is to add a
feed-forward capacitor (CFF) across the top feedback resistor R1 to speed Gm Amplifier recovery. In practice, a
27nF to 100nF ceramic capacitor is usually a good choice to remove the overshoot completely or limit the
overshoot to an insignificant level during startup, as shown in Figure 28. Another more effective solution is to
control EN pin voltage by a separate logic signal, and pull the signal high only after VIN is fully established. In this
way, the chip can execute a normal, complete soft start process, minimizing any output voltage overshoot. Under
some circumstances at cold temperature, this approach may also be required to minimize any unwanted output
voltage transients that may occur when the input voltage rises slowly. For a fast rising input voltage (100µs for
example), there is no need to control EN separately or add a feed-forward capacitor since the soft-start can bring
up output voltage smoothly as shown in Figure 29.
During startup, the LM2833 gradually increases the switching frequency from 400kHz (LM2833X) or 800kHz
(LM2833Z) to the nominal fixed value, as the feedback voltage increases (see FREQUENCY FOLDBACK section
for more information). Since the internal corrective ramp signal adjusts its slope dynamically, and is proportional
to the switching frequency during startup, a larger output capacitance may be required to insure a smooth output
voltage rise, at low programmed output voltage and high output load current.
Figure 27. Startup Response to VIN
Figure 28. Startup Response to VIN with CFF
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Figure 29. Startup Response to VIN with 100µs rise time
FREQUENCY FOLDBACK
The LM2833 uses frequency foldback to help limit switch current and power dissipation during start-up, shortcircuit and over load conditions by sensing if the feedback voltage is below 0.32V (typical). The LM2833 will
reduce the switching frequency from the nominal fixed value (1.5MHz or 3.0MHz) down to 400kHz (LM2833X) or
800kHz (LM2833Z) when the feedback voltage drops to 0V. See Figure 14 in the Typical Performance
Characteristics section.
LOAD STEP RESPONSE
The LM2833 has a fixed internal loop compensation, which results in a small-signal loop bandwidth highly related
to the output voltage level. In general, the loop bandwidth at low voltage is larger than at high voltage due to the
increased overall loop gain. The limited bandwidth at high output voltage may pose a challenge when loop step
response is concerned. In this case, one effective approach to improving loop step response is to add a feedforward capacitor (CFF) in the range of 27nF to 100nF in parallel with the upper feedback resistor (assuming the
lower feedback resistor is 2kΩ), as shown in Figure 30. The feed-forward capacitor introduces a zero-pole pair
which helps compensate the loop. The position of the zero-pole pair is a function of the feedback resistors and
capacitor:
(1)
(2)
Note the factor in parenthesis is the ratio of the output voltage to the feedback voltage. As the output voltage
gets close to 0.6V, the pole moves towards the zero, tending to cancel it out. Consequently, adding CFF will have
less effect on the step response at lower output voltages.
As an example, Figure 32 shows that at the output voltage of 3.3V, a 47nF of CFF can boost the loop bandwidth
to 117kHz, from the original 23kHz as shown in Figure 31. Correspondingly, the responses to a load step
between 0.3A and 3A without and with CFF are shown in Figure 33 and Figure 34 respectively. The higher loop
bandwidth as a result of CFF reduces the total output excursion by more than half.
Aside from the above approach, increasing the output capacitance is generally also effective to reduce the
excursion in output voltage caused by a load step. This approach remains valid for applications where the
desired output voltages are close to the feedback voltage.
12
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Figure 30. Adding a CFF Capacitor
Figure 31. Loop Gain and Phase without CFF
Figure 32. Loop Gain and Phase with CFF
Figure 33. Load Step Response without CFF
Figure 34. Load Step Response with CFF
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OUTPUT OVER-VOLTAGE PROTECTION
The LM2833 has a built in output over-voltage comparator that compares the FB pin voltage to a threshold
voltage that is 15% higher than the internal reference VREF. Once the FB pin voltage exceeds this threshold level
(typically 0.69V), the internal PMOS power switch is turned off, which allows the output voltage to decrease
towards regulation.
UNDER-VOLTAGE LOCKOUT
Under-voltage lockout (UVLO) prevents the LM2833 from operating until the input voltage exceeds 2.70V
(typical). The UVLO threshold has approximately 350mV of hysteresis, so the part will operate until VIN drops
below 2.35V (typical). Hysteresis prevents the part from turning off during power up if VIN is non-monotonic.
CURRENT LIMIT
The LM2833 uses cycle-by-cycle current limiting to protect the internal power switch. During each switching
cycle, a current limit comparator detects if the power switch current exceeds 4.4A (typical), and turns off the
switch until the next switching cycle begins.
THERMAL SHUTDOWN
Thermal shutdown limits total power dissipation by turning off the internal power switch when the IC junction
temperature typically exceeds 165°C. After thermal shutdown occurs, the power switch does not turn on again
until the junction temperature drops below approximately 150°C.
Design Guide
INDUCTOR SELECTION
The Duty Cycle (D) can be approximated quickly using the ratio of output voltage (VOUT) to input voltage (VIN):
D=
VOUT
VIN
(3)
The catch diode (D1) forward voltage drop and the voltage drop across the internal PMOS must be included to
calculate a more accurate duty cycle. Calculate D by using the following formula:
VOUT + VD
D=
VIN + VD - VSW
(4)
VSW can be approximated by:
VSW = IOUT x RDS(ON)
where
•
IOUT is output load current.
(5)
The diode forward drop (VD) can range from 0.3V to 0.7V depending on the quality of the diode. The lower the
VD, the higher the operating efficiency of the converter.
The inductor value determines the output ripple current (ΔiL, as defined in Figure 26). Lower inductor values
decrease the size of the inductor, but increase the output ripple current. An increase in the inductor value will
decrease the output ripple current. In general, the ratio of ripple current to the output current is optimized when it
is set between 0.2 and 0.4 for output currents above 2A. This ratio r is defined as:
r=
'iL
lOUT
(6)
One must ensure that the minimum current limit (3.4A) is not exceeded, so the peak current in the inductor must
be calculated. The peak current (ILPK) in the inductor is calculated by:
ILPK = IOUT + ΔiL/2
14
(7)
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When the designed maximum output current is reduced, the ratio r can be increased. At a current of 0.1A, r can
be made as high as 0.9. The ripple ratio can be increased at lighter loads because the net ripple is actually quite
low, and if r remains constant the inductor value can be made quite large. An equation empirically developed for
the maximum ripple ratio at any current below 2A is:
r = 0.387 x IOUT-0.3667
(8)
Note that this is just a guideline, and it needs to be combined with two important factors for proper selection of
inductance values at any operating condition. The first consideration is at output voltage above 2.5V, one needs
to ensure that the inductance given by the above guideline should not be less than 1µH for the LM2833X or
0.5µH for the LM2833Z. Since the LM2833 has a fixed internal corrective ramp signal, a very low inductance
value at high output voltage will generate a very steep down slope of inductor current, which will result in an
insufficient slope compensation, and cause instability known as sub-harmonic oscillation. Another consideration
is at low load current, one needs to ensure that the inductance value given by the guideline should not exceed
10µH for the LM2833X and 4.7µH for the LM2833Z, since too much inductance effectively flattens the down
slope of the inductor current, and may significantly limit the system bandwidth and phase margin resulting in
instability.
The LM2833 operates at frequencies allowing the use of ceramic output capacitors without compromising
transient response. Ceramic capacitors allow higher inductor ripple without significantly increasing output ripple.
See the OUTPUT CAPACITOR section for more details on calculating output voltage ripple.
Now that the ripple current is determined, the inductance is calculated by:
L=
VOUT + VD
IOUT x r x fSW
x (1-D)
where
•
fSW is the switching frequency.
(9)
When selecting an inductor, make sure that it is capable of supporting the peak output current without saturating.
Inductor saturation will result in a sudden reduction in inductance and prevent the regulator from operating
properly. Because of the operating frequency of the LM2833, ferrite based inductors are preferred to minimize
core losses. This presents little restriction since the variety and availability of ferrite-based inductors is large.
Lastly, inductors with lower series resistance (DCR) will provide better operating efficiency. For recommended
inductor selection, refer to Design Examples.
INPUT CAPACITOR
An input capacitor is necessary to ensure that VIN does not drop excessively during switching transients. The
primary specifications of the input capacitor are capacitance, voltage rating, RMS current rating, and ESL
(Equivalent Series Inductance). The input voltage rating is specifically stated by the capacitor manufacturer.
Make sure to check any recommended deratings and also verify if there is any significant change in capacitance
at the operating input voltage and the operating temperature. The input capacitor maximum RMS input current
rating (IRMS-IN) must be greater than:
2
IRMS-IN = IOUT x D x 1 - D + r
12
(10)
Neglecting inductor ripple simplifies the above equation to:
IRMS-IN = IOUT x D x 1 - D
(11)
It can be shown from the above equation that maximum RMS capacitor current occurs when D = 0.5. Always
calculate the RMS at the point where the duty cycle D is closest to 0.5. The ESL of an input capacitor is usually
determined by the effective cross sectional area of the current path. As a rule of thumb, a large leaded capacitor
will have high ESL and a 1206 ceramic chip capacitor will have very low ESL. At the operating frequencies of the
LM2833, leaded capacitors may have an ESL so large that the resulting impedance (2πfL) will be higher than
that required to provide stable operation. It is strongly recommended to use ceramic capacitors due to their low
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ESR and low ESL. A 22µF multilayer ceramic capacitor (MLCC) is a good choice for most applications. In cases
where large capacitance is required, use surface mount capacitors such as Tantalum capacitors and place at
least a 4.7µF ceramic capacitor close to the VIN pin. For MLCCs it is recommended to use X7R or X5R
dielectrics. Consult capacitor manufacturer datasheet to see how rated capacitance varies over operating
conditions.
OUTPUT CAPACITOR
The output capacitor is selected based upon the desired output ripple and transient response. The initial current
of a load transient is provided mainly by the output capacitor. The output ripple of the converter is:
'VOUT = 'IL RESR +
1
8 x fSW x COUT
(12)
When using MLCCs, the ESR is typically so low that the capacitive ripple may dominate. When this occurs, the
output ripple will be approximately sinusoidal and 90° phase shifted from the switching action. Given the
availability and quality of MLCCs and the expected output voltage of designs using the LM2833, there is really no
need to review any other capacitor technologies. Another benefit of ceramic capacitors is their ability to bypass
high frequency noise. A certain amount of switching edge noise will couple through parasitic capacitances in the
inductor to the output. A ceramic capacitor will bypass this noise while a tantalum will not. Since the output
capacitor is one of the two external components that control the stability of the regulator control loop, most
applications will require a minimum of 22µF output capacitance. In the case of low output voltage, a larger output
capacitance is required to ensure sufficient phase margin. Capacitance can often, but not always, be increased
significantly with little detriment to the regulator stability. Like the input capacitor, recommended multilayer
ceramic capacitors are X7R or X5R types. Again, verify actual capacitance at the desired operating voltage and
temperature. Check the RMS current rating of the capacitor. The maximum RMS current rating of the capacitor
is:
(13)
One may select a 1206 size MLCC for output capacitor, since its current rating is typically above 1A, more than
enough for the requirement.
CATCH DIODE
The catch diode conducts during the switch off-time. A Schottky diode is recommended for its fast switching time
and low forward voltage drop. The catch diode should be chosen such that its current rating is greater than:
ID = IOUT x (1-D)
(14)
The reverse breakdown rating of the diode must be at least the maximum input voltage plus appropriate margin.
To improve efficiency, choose a Schottky diode with a low forward voltage drop.
OUTPUT VOLTAGE
The output voltage is set using the following equation where R2 is connected between the FB pin and GND, and
R1 is connected between VOUT and the FB pin. A good value for R2 is 2kΩ.
VOUT
- 1 x R2
R1 =
VREF
(15)
VREF = 0.60V
16
(16)
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EFFICIENCY ESTIMATION
The complete LM2833 DC/DC converter efficiency can be calculated in the following manner:
K=
POUT
PIN
(17)
Or
K=
POUT
POUT + PLOSS
(18)
Calculations for determining the most significant power losses are shown below. Other losses totaling less than
2% are not discussed.
The main power loss (PLOSS) in the converter includes two basic types of losses: switching loss and conduction
loss. In addition, there is loss associated with the power required for the internal circuitry of IC. Conduction
losses usually dominate at higher output loads, whereas switching losses dominate at lower output loads. The
first step in determining the losses is to calculate the duty cycle (D):
D=
VOUT + VD
VIN + VD - VSW
(19)
VSW is the voltage drop across the internal power switch when it is on, and is equal to:
VSW = IOUT x RDS(ON)
(20)
VD is the forward voltage drop across the catch diode. It can be obtained from the diode manufactures Electrical
Characteristics section. If the DC voltage drop across the inductor (VDCR) is accounted for, the equation
becomes:
D=
VOUT + VD + VDCR
VIN + VD - VSW
(21)
The conduction losses in the catch diode are calculated as follows:
PDIODE = VD x IOUT x (1-D)
(22)
Often this is the single most significant power loss in the circuit. Care should be taken to choose a Schottky
diode with a low forward voltage drop.
Another significant external power loss is the conduction loss in the output inductor. The equation can be
simplified to:
PIND = IOUT2 x RDCR
(23)
The LM2833 conduction loss is mainly associated with the internal power switch:
PCOND =
2
(IOUT
'iL
1
x D) x 1 + x
3
IOUT
2
x RDS (ON)
(24)
If the inductor ripple current is fairly small, the conduction losses can be simplified to:
PCOND = IOUT2 x RDS(ON) x D
(25)
Switching losses are also associated with the internal power switch. They occur during the switch on and off
transition periods, where voltages and currents overlap resulting in power loss. The simplest means to determine
this loss is to empirically measuring the rise and fall times (10% to 90%) of the switch at the switch node.
Switching Power Loss is calculated as follows:
PSWR = 0.5 x (VIN x IOUT x fSW x TRISE)
PSWF = 0.5 x (VIN x IOUT x fSW x TFALL)
PSW = PSWR + PSWF
(26)
(27)
(28)
The power loss required for operation of the internal circuitry is given by:
PQ = IQ x VIN
(29)
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IQ is the quiescent operating current, and is typically around 3.2mA for the LM2833X, and 4.3mA for the
LM2833Z.
An example of efficiency calculation for a typical application is shown in Table 1:
Table 1. Power Loss Tabulation
Conditions
Power loss
VIN
5V
VOUT
3.3V
IOUT
3.0A
POUT
9.9W
VD
0.33V
PDIODE
277mW
RDS(ON)
56mΩ
PCOND
363mW
fSW
1.5MHz
PSW
225mW
TRISE
10ns
TFALL
10ns
INDDCR
28mΩ
PIND
252mW
IQ
3.2mA
PQ
16mW
η
89.7%
D is calculated to be 0.72
PLOSS = Σ ( PCOND + PSW + PQ + PIND + PDIODE )
PLOSS = 1.133W
(30)
(31)
PCB LAYOUT CONSIDERATIONS
When planning layout there are a few things to consider to achieve a clean, regulated output. The most important
consideration is the close coupling of the GND connections of the input capacitor C1 and the catch diode D1.
These ground ends should be close to one another and be connected to the GND plane with at least two
through-holes. Place these components as close to the IC as possible. The next consideration is the location of
the GND connection of the output capacitor C2, which should be near the GND connections of C1 and D1. There
should be a continuous ground plane on the bottom layer of a two-layer board except under the switching node
island. The signal ground SGND (pin 3) and power ground PGND (pin 6) should be tied together and connected
to ground plane through vias.
The FB pin is a high impedance node and care should be taken to make the FB trace short to avoid noise pickup
that causes inaccurate regulation. The feedback resistors should be placed as close as possible to the IC, with
the GND of R2 placed as close as possible to the SGND of the IC. The VOUT trace to R1 should be routed away
from the inductor and any other traces that are switching.
High AC currents flow through the VIN, SW and VOUT traces, so they should be as short and wide as possible.
Radiated noise can be decreased by choosing a shielded inductor.
The remaining components should also be placed as close as possible to the IC. Please see Application Note
AN-1229 SNVA054 for further considerations and the LM2833 demo board as an example of a four-layer layout.
18
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LM2833X Design Example 1
VIN
9,10
SW
VIND
L1
7,8
VOUT
D1
R3
2
1
C1
R1
EN
LM2833X
FB
VINC
C3
NC
SGND
3
5
C2
4
R2
PGND
6
Figure 35. LM2833X (1.5MHz): VIN = 3.3V, Output = 1.2V/3.0A
Table 2. Bill of Materials
Part ID
Part Value
Manufacturer
Part Number
U1
3.0A Buck Regulator
TI
LM2833X
C3216X5R0J226M
C1, Input Cap
22µF, 6.3V, X5R
TDK
C2, Output Cap
47µF, 6.3V, X5R
TDK
C3216X5R0J476M
C3, Bypass Cap
0.22µF, 10V, X7R
Murata
GRM216R71A224KC01D
D1, Catch Diode
Schottky, 0.33V at 3A, VR=30V
Toshiba
CMS01
L1
1.8µH, 3.6A
TDK
LTF5022T-1R8N3R6
R1
2.0kΩ, 1%
Vishay
CRCW08052K00FKEA
R2
2.0kΩ, 1%
Vishay
CRCW08052K00FKEA
R3
10Ω, 1%
Vishay
CRCW080510R0FKEA
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LM2833X Design Example 2
Figure 36. LM2833X (1.5MHz): VIN = 5V, Output = 3.3V/3.0A
Table 3. Bill of Materials
20
Part ID
Part Value
Manufacturer
Part Number
U1
3.0A Buck Regulator
TI
LM2833X
C1, Input Cap
22µF, 6.3V, X5R
TDK
C3216X5R0J226M
C2, Output Cap
47µF, 6.3V, X5R
TDK
C3216X5R0J476M
C3, Bypass Cap
0.22µF, 10V, X7R
Murata
GRM216R71A224KC01D
0805ZC473JAZ2A
CFF, Feed-forward Cap
47nF, 10V, X7R
AVX
D1, Catch Diode
Schottky, 0.43V at 3A, VR=30V
Vishay
SSA33L-E3/61T
L1
1.2µH, 4.2A
TDK
LTF5022T-1R2N4R2
R1
10.2kΩ, 1%
Vishay
CRCW080510K2FKEA
R2
2.26kΩ, 1%
Vishay
CRCW08052K26FKEA
R3
10Ω, 1%
Vishay
CRCW080510R0FKEA
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LM2833Z Design Example 3
VIN
9,10
SW
VIND
L1
7,8
VOUT
D1
R3
2
1
C1
R1
EN
LM2833Z
FB
VINC
C3
NC
SGND
5
C2
4
R2
PGND
3
6
Figure 37. LM2833Z (3MHz): VIN = 3.3V, Output = 1.2V/3.0A
Table 4. Bill of Materials
Part ID
Part Value
Manufacturer
Part Number
U1
3.0A Buck Regulator
TI
LM2833Z
C3216X5R0J226M
C1, Input Cap
22µF, 6.3V, X5R
TDK
C2, Output Cap
47µF, 6.3V, X5R
TDK
C3216X5R0J476M
C3, Bypass Cap
0.22µF, 10V, X7R
Murata
GRM216R71A224KC01D
D1, Catch Diode
Schottky, 0.33V at 3A, VR=30V
Toshiba
CMS01
L1
1.0µH, 4.0A
Taiyo Yuden
NP04SZB1R0N
R1
2.0kΩ, 1%
Vishay
CRCW08052K00FKEA
R2
2.0kΩ, 1%
Vishay
CRCW08052K00FKEA
R3
10Ω, 1%
Vishay
CRCW080510R0FKEA
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LM2833Z Design Example 4
Figure 38. LM2833Z (3MHz): VIN = 5V, Output = 3.3V/3.0A
Table 5. Bill of Materials
22
Part ID
Part Value
Manufacturer
Part Number
U1
3.0A Buck Regulator
TI
LM2833Z
C1, Input Cap
22µF, 6.3V, X5R
TDK
C3216X5R0J226M
C2, Output Cap
47µF, 6.3V, X5R
TDK
C3216X5R0J476M
C3, Bypass Cap
0.22µF, 10V, X7R
Murata
GRM216R71A224KC01D
CFF, Feed-forward Cap
47nF, 10V, X7R
AVX
0805ZC473JAZ2A
D1, Catch Diode
Schottky, 0.43V at 3A, VR=30V
Vishay
SSA33L-E3/61T
L1
1.0µH, 4.0A
Taiyo Yuden
NP04SZB1R0N
R1
10.2kΩ, 1%
Vishay
CRCW080510K2FKEA
R2
2.26kΩ, 1%
Vishay
CRCW08052K26FKEA
R3
10Ω, 1%
Vishay
CRCW080510R0FKEA
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REVISION HISTORY
Changes from Revision D (April 2013) to Revision E
•
Page
Changed layout of National Data Sheet to TI format .......................................................................................................... 22
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PACKAGE OPTION ADDENDUM
www.ti.com
10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
LM2833XMY/NOPB
ACTIVE
HVSSOP
DGQ
10
1000
RoHS & Green
SN
Level-3-260C-168 HR
-40 to 125
SPYB
LM2833XSD/NOPB
ACTIVE
WSON
DSC
10
1000
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 125
2833X
LM2833ZMY/NOPB
ACTIVE
HVSSOP
DGQ
10
1000
RoHS & Green
SN
Level-3-260C-168 HR
-40 to 125
SPZB
LM2833ZMYX/NOPB
ACTIVE
HVSSOP
DGQ
10
3500
RoHS & Green
SN
Level-3-260C-168 HR
-40 to 125
SPZB
LM2833ZSD/NOPB
ACTIVE
WSON
DSC
10
1000
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 125
2833Z
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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