LM3401
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LM3401 Hysteretic PFET Controller for High Power LED Drive
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FEATURES
DESCRIPTION
•
•
•
•
•
•
The LM3401 is a switching controller designed to
provide constant current to high power LEDs. The
LM3401 drives an external P-MOSFET switch for
step-down (Buck) regulators. The LM3401 delivers
constant current within ±6% accuracy to a wide
variety and number of series connected LEDs. Output
current is adjusted with an external current sensing
resistor to drive high power LEDs in excess of 1A.
1
2
•
•
•
•
Hysteretic Control for Speed and Simplicity
Input Operating Range of 4.5V-35V
1.5MHz Maximum Switching Frequency
Low 200mV Reference Voltage
Programmable Current Limit
High speed CMOS Compatible
Enable/Dimming
Adjustable Hysteresis
Input UVLO
No output Capacitor Required
8-pin VSSOP Package
For improved accuracy and efficiency, the LM3401
features dual-side hysteresis, very low reference
voltage, and short propagation delay. A cycle by
cycle current limit provides protection against over
current and short circuit failures. Additional features
include adjustable hysteresis and a CMOS
compatible input pin for PWM dimming.
APPLICATIONS
•
•
LED Driver
Battery Charger
TYPICAL APPLICATION CIRCUIT
VIN = 4.5V to 35V
R3
VIN
ILIM
C1
HG
Q1
HYS
CS
D1
GND
SNS
DIM
LM3401
L1
LED
current
R2
R1
1
2
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
All trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2007–2013, Texas Instruments Incorporated
LM3401
SNVS516C – AUGUST 2007 – REVISED MAY 2013
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These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
CONNECTION DIAGRAM
CS
1
8
ILIM
DIM
2
7
VIN
SNS
3
6
HG
HYS
4
5
GND
Figure 1. Top View
8-Lead VSSOP
See DGK Package
PIN DESCRIPTIONS
2
Pin #
Pin
Name
Description
1
CS
Current sense pin. Connect to the PFET drain
2
DIM
Dimming input pin. When DIM is low, the HG drive is off. Can be connected to a logic level PWM signal
3
SNS
Current feedback pin – to LED cathode. Connect a resistor from this pin to ground to set the DC LED current
4
HYS
Hysteresis adjust pin. Connect a resistor from this pin to GND to set the hysteresis window
5
GND
Ground pin
6
HG
Gate drive output. Connect to the PFET gate
7
VIN
Power supply input
8
ILIM
Current limit adjust pin. Connect a resistor from this pin to the PFET source to set the current limit threshold
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ABSOLUTE MAXIMUM RATINGS
(1)
If Military/Aerospace specified devices are required, contact the Texas Instruments Semiconductor Sales Office/
Distributors for availability and specifications.
VALUE / UNIT
VIN
-0.3V to 36V
CS
-2.0V to 36V
SNS
-0.3V to 8V
ILIM
-0.3V to 36V
DIM
-0.3V to 36V
HYS
-0.3V to 4V
Storage Temperature
-65°C to +150°C
Lead Temperature
Vapor Phase (60sec)
215°C
Infrared (15sec)
220°C
ESD Rating Human Body Model
(1)
(2)
(2)
2.5kV
Absolute Maximum Ratings indicate limits beyond which damage may occur to the device. Operating Ratings indicate conditions for
which the device is intended to be functional, but do not include specific performance limits. For specified specifications, see Electrical
Characteristics.
The human body model is a 100 pF capacitor discharged through a 1.5 kΩ resistor into each pin.
RECOMMENDED OPERATING CONDITIONS (1)
VALUE / UNIT
VIN
4.5V to 35V
Junction Temperature Range
-40°C to +125°C
Thermal Resistance θJA (2)
Power Dissispation
(1)
(2)
151°C/W
(2)
0.66W
Absolute Maximum Ratings indicate limits beyond which damage may occur to the device. Operating Ratings indicate conditions for
which the device is intended to be functional, but do not include specific performance limits. For specified specifications, see Electrical
Characteristics.
The maximum allowable power dissipation is a function of the maximum junction temperature, TJ_MAX, the junction-to-ambient thermal
resistance, θJA, and the ambient temperature, TA. The maximum allowable power dissipation at any ambient temperature is calculated
using: PD_MAX = (TJ_MAX - TA)/θJA. The maximum power dissipation of 0.66W is determined using TA = 25°C, θJA = 151°C/W, and
TJ_MAX = 125°C. θJA will vary with board size and copper area. The θJA spec is based on a JEDEC standard 4-layer board.
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ELECTRICAL CHARACTERISTICS
Specifications in standard type are for TJ = 25°C only, and limits in boldface type apply over the junction temperature (TJ)
range of -40°C to +125°C. Unless otherwise stated, VIN = 24V. Minimum and Maximum limits are specified through test,
design, or statistical correlation. Typical values represent the most likely parametric norm at TJ = 25°C, and are provided for
reference purposes only (1).
Symbol
Parameter
Conditions
Min
Typ
Max
200
212
Units
SYSTEM
VREF
ΔVREF / ΔVIN
IQ
IHYS
Reference Voltage
188
Line regulation
Operating VIN Current
5V < VIN < 35V
(2)
HYS pin = 50 mV to 500 mV
SNS Comparator Hysteresis Multiplier
HYS pin = 250 mV
TDLY
SNS Comparator to HG Delay
SNS rising
TDIM
DIM to HG Delay
DIM rising
IILIM
ILIM Pin Sink Current
VCL_OFF
Current Limit Comparator Offset
VZC
Zero Cross Comparator Threshold
VDIM
DIM Threshold Voltage
Measured at CS pin
ISNS
mA
15
20
25
µA
0.168
0.20
0.224
-
46
80
ns
69
120
ns
4
5.5
8
µA
-10
0
+10
mV
-70
-130
-200
mV
1.85
2.0
2.25
Hysteresis
UVLO
mV
mV/V
1.05
Hysteresis Pin Source Current
SNSHYS_MU
0.002
286
V
mV
SNS pin Bias Current
VSNS = 200 mV
300
780
nA
UVLO threshold
Vin rising
4.3
4.48
V
Hysteresis
0.5
V
DRIVER
Ton_min
RHG
IHG
VHG
(1)
(2)
4
Minimum on-time
Gate Drive Resistance
Driver Output Current
HG on voltage
150
ns
Source Current = 100 mA
5.3
Ω
Sink Current = 100 mA
10.5
Ω
Source, HG = VIN -2.5V
0.41
A
Sink, HG = VIN -2.5V
0.33
Referenced to VIN, steady state voltage
-4.2
-4.7
A
-5.5
V
All room temperature limits are 100% production tested. All limits at temperature extremes are specified through correlation using
standard Statistical Quality Control (SQC) methods. All limits are used to calculate Average Outgoing Quality Level (AOQL).
IQ specifies the current into the VIN pin and applies to non-switching operation.
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TYPICAL PERFORMANCE CHARACTERISTICS
Unless otherwise specified, VIN = 24V, TA = 25°C.
VREF vs Temperature
HYS Current vs VIN
0.205
20.85
20.6
HYS CURRENT (PA)
VREF (V)
0.203
0.201
0.199
20.35
20.1
19.85
0.197
0.195
-50
19.6
0
-25
25
50
75
100
125
0
5
10
TEMPERATURE (qC)
15
20
25
30
35
INPUT VOLTAGE (V)
Figure 2.
Figure 3.
HYS Multiplier vs Temperature
Startup Waveforms
0.23
ILED
100 mA/Div
HYS MULTIPLIER
0.22
0.21
VIN
5V/Div
500 mV
0.20
0.19
50 mV
0.18
0.17
-50
VCS
10V/Div
-25
0
25
50
75
100
4 Ps/DIV
125
TEMPERATURE (qC)
Figure 4.
Figure 5.
HYS Current vs Temperature
21
5.8
20.6
HYS CURRENT (PA)
ILIM SINK CURRENT (PA)
ILIM Sink Current vs Temperature
6.0
5.6
5.4
5.2
5.0
-50
20.2
19.8
19.4
-25
0
25
50
75
100
19
-50
125
-25
0
25
50
75
100
125
TEMPERATURE (°C)
TEMPERATURE (qC)
Figure 6.
Figure 7.
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TYPICAL PERFORMANCE CHARACTERISTICS (continued)
Unless otherwise specified, VIN = 24V, TA = 25°C.
UVLO Threshold vs Temperature
SNS to HG Propagation Delay vs VIN
70
4.50
TURN-ON
60
DELAY TIME (ns)
INPUT VOLTAGE (V)
4.30
4.10
TURN-OFF
3.90
3.70
50
40
30
3.50
3.30
-50
-25
0
25
50
75
100
20
125
0
5
10
15
20
25
30
TEMPERATURE (°C)
INPUT VOLTAGE (V)
Figure 8.
Figure 9.
Line Transient Response
35
40
Initial HG Voltage vs Input Voltage
8.0
180 pF
6.5
ILED
100 mA/Div
VIN_HG (V)
470 pF
VIN
10V/Div
5.0
1000 pF
3.5
2.0
20 Ps/DIV
0
5
10
15
20
25
30
35
INPUT VOLTAGE (V)
Figure 10.
Figure 11.
350 mA Efficiency
700 mA and 1A Efficiency
100
100
95
2 series (700 mA)
95
3 series
90
EFFICIENCY (%)
EFFICIENCY (%)
90
85
2 series
80
1 series
75
70
80
1 series (1A)
75
70
65
65
60
60
0
5
10
15
20
25
30
35
INPUT VOLTAGE (V)
0
5
10
15
20
25
30
35
INPUT VOLTAGE (V)
Figure 12.
6
1 series (700 mA)
85
Figure 13.
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BLOCK DIAGRAM
VIN
on
UVLO
VDD
regulator
-0.13V
Q
Q
S
DIM
R
HG
GND
-
+
+
CS
SNS
+
ILIM
ILIM
X 0.2
20 PA
+
-
VREF
200 mV
5.5 PA
HYS
Figure 14. Block Diagram
Operational Description
HYSTERETIC CONTROL
The LM3401 is a step-down DC-DC controller designed to provide a constant current source for driving a high
power LED string.
The LM3401 uses comparator-based voltage mode hysteretic control for a simple and stable design. Hysteretic
control does not utilize an internal oscillator, but relies on output conditions to directly control switching. The
LM3401 controls LED current within the adjustable hysteresis window by monitoring peak and valley voltage at
the SNS pin. A dual sided hysteresis window is used to optimize accuracy.
Regulated LED current flows to ground through a sense resistor at the SNS pin. The voltage generated at the
SNS pin is compared to the 200 mV internal reference. When the SNS voltage falls below the reference voltage
minus hysteresis, the output of the hysteretic comparator goes low. This results in the driver output, HG, pulling
the gate of the PFET low and turning on the PFET.
With the PFET on, LED current ramps up through the PFET and the inductor. As the LED current increases, the
SNS voltage reaches its upper threshold (reference voltage plus hysteresis). This forces the output of the
comparator and HG to go high, which turns the PFET off. When the PFET turns off, current flows through the
catch diode, and LED and inductor current ramp down. When the SNS voltage falls to its lower threshold, the
cycle repeats. The resulting LED current, SNS, and switch node waveforms are shown in Figure 15.
ILED
100 mA/Div
VSNS
100 mV/Div
VCS
10V/Div
400 ns/DIV
Figure 15. Hysteretic Switching Waveforms
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OUTPUT CURRENT SETTING
The LED average current is programmed using a resistor between SNS and GND, shown as R1 in the typical
application schematic. The SNS resistor (RSNS) can be calculated as follows:
RSNS =
VSNS
ILED
(1)
Where VSNS is 200 mV typically, and ILED is the DC average LED current.
The sense resistor power rating must be higher than its power dissipation. The required power rating can be
calculated (in Watts) as follows:
WRSNS = VSNS x ILED
(2)
When selecting a sense resistor, thermal de-rating must also be taken into consideration.
While RSNS sets the DC LED current, the AC peak LED current will be higher than the DC setting. This peak
current must not be greater than the maximum peak current rating of the LED, ILED_max. Peak LED current can be
calculated from the equation below:
ILED_PK = ILED +
ILED_RIP
2
(3)
The LED ripple current, ILED_RIP, is described below in the Hysteresis Adjust section.
HYSTERESIS ADJUST
Adjustable hysteresis (via the HYS pin) provides direct control over the LED ripple current. The HYS pin also
gives some control over the switching frequency. Although the hysteresis value can be set after the inductor is
selected, a preliminary value must be set in order to begin the frequency calculation. The hysteresis window
must be set small enough to keep the peak LED current below its maximum rating, ILED_max.
The maximum SNS pin hysteresis can be calculated as shown:
SNSHYS_MAX = (ILED_max - ILED) x RSNS
(4)
Any SNSHYS value between 10mV and this maximum is acceptable.
The SNSHYS value is set with a single resistor from the HYS pin to GND, shown as R2 in the typical application
schematic. The HYS pin voltage, VHYS, is internally multiplied by SNSHYS_MU (0.2 typical) to generate the
hysteresis at the SNS pin, SNSHYS. The hysteresis setting resistor can be determined from the following
equation:
R2 =
SNSHYS x 5
20 PA
(5)
Where 20 µA is the typical HYS source current, and SNSHYS is the resulting SNS pin hysteresis voltage. The
hysteresis voltage can be set within a range of 10 mV to 100 mV (50 mV to 500 mV at the HYS pin). The
SNSHYS value defines both the upper and lower threshold of the SNS pin. For example, with a VHYS setting of
100 mV, SNSHYS will be 20 mV. Therefore, the total hysteretic window will be 40 mV, or 20 mV above and 20mV
below the 200 mV reference voltage. This directly correlates to peak-to-peak inductor and LED ripple current,
approximated by the following equation:
ILED_RIP =
SNSHYS x 2
RSNS
(6)
If LED ripple current is a design priority, the preliminary R2 value can be determined using a target LED ripple
current as shown:
R2 =
ILED_RIP x RSNS x 5
40 PA
(7)
A more precise equation for ripple current is given in the Inductor Selection section. Once an inductor is selected
the more precise equation should be used to determine the actual ripple current and LED peak current. Larger
hysteresis values will result in lower switching frequency and higher ripple current for a given inductor. Typical
examples are shown in Figure 16 below.
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1800
1800
22 PH freq
1500
22 PH rip
1200
1200
68 PH freq
900
900
600
600
RIPPLE CURRENT (mA)
FREQUENCY (kHz)
1500
300
300
68 PH rip
0
0
110
50
175
255
340
425
505
VHYS (mV)
Figure 16. Switching Frequency and Ripple Current vs Hysteresis
SWITCHING FREQUENCY
Although hysteretic control is a simple control method, the switching frequency depends on application conditions
and components. If the inductance, input voltage, or LED forward voltage is changed, there will be a change in
the switching frequency. Therefore, care must be taken to select components which will provide the desired
frequency range. Usually, the best approach is to determine a nominal switching frequency for the application
and then select the inductor accordingly. Once the inductor is selected, VHYS can be adjusted to set the
frequency range more precisely. This design process usually involves a few iterations to select appropriate
standard values that will result in the desired frequency and ripple current.
Switching frequency can be approximately calculated using the formula:
fSW =
D
2 x SNSHYS x L
RSNS x (VIN - VANODE)
+ (2 x delay)
(8)
Where D is the duty cycle, defined as (VOUT + VDIODE) / VIN, VANODE is 200 mV plus the sum of LED forward
voltages, and delay is the sum of the LM3401 propagation delay time and the PFET delay time. The LM3401
propagation delay is 46 ns typically (See the Propagation Delay curve). Alternately, the inductor value can be
calculated from a known frequency by re-arranging the same equation:
D
- (2 x delay) x RSNS x (VIN - VANODE)
fSW
L=
2 x SNSHYS
(9)
Switching frequency for a typical application is shown in Figure 17 along with the calculated frequency.
1500
= calculated value
1250
FREQUENCY (kHz)
22 PH
1000
750
500
68 PH
250
0
0
5
10
15
20
25
30
35
INPUT VOLTAGE (V)
Figure 17. Frequency vs Input Voltage
Maximum switching frequency will typically occur around the input voltage which corresponds to 25% duty cycle.
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If the input voltage falls below the forward voltage of the LED string, the LM3401 will operate at 100% duty cycle.
In this state, the anode voltage will be equal to the input voltage and LED current is determined by the v-i curve
of the LED. At 100% duty cycle, LED current will be continuous with a maximum value equal to the ILED_PK level
set at the HYS pin.
In some applications, maximum operating frequency will be limited by the minimum on-time as shown in the
following equation:
ton =
D
fSW
(10)
When the on-time reaches minimum (150 ns typical) due to increasing input voltage, the frequency will be
reduced in order to maintain the proper duty cycle.
INDUCTOR SELECTION
The most important parameters for the inductor are the inductance and the current rating. The LM3401 operates
over a wide frequency range and can use a wide range of inductance values.
Once an inductance value is determined from the frequency equation, the maximum operating current must be
verified.
Although peak-to-peak ripple current is controlled by the hysteresis value, there is some variation due to
propagation delay. This means that the inductance has a direct effect on LED current line regulation.
In general, a larger inductor will result in lower frequency and better line regulation. The actual peak-to-peak
inductor current can be calculated as follows:
ILED_RIP = IL_RIP =
2 x SNSHYS
RSNS
+
(VIN - VANODE) x 2 x delay
L
(11)
Use the maximum value of Vin to determine the worst case ILED_RIP value. This value should be used to
determine the peak current, ILED_PK, shown in the Output Current Setting section.
Since the LM3401 can operate at 100% duty cycle, the inductor must be rated to handle ILED_PK continuously.
RIPPLE REDUCTION CAPACITOR
The typical application schematic shown on the front page is the simplest application of the LM3401. In this
schematic, inductor current is equal to LED current. Therefore, the equations for ripple current apply to both LED
ripple and inductor ripple.
However, LED ripple current can be reduced without altering the inductor current by using a bypass capacitor
placed in parallel with the LED string (shown as C2 in the Figure 24 schematic).
This allows larger hysteresis values to be used while significantly reducing ripple current in the LED string.
Figure 18 below shows this effect: inductor ripple current is unaffected while LED ripple current is dramatically
reduced.
ILED
50 mA /Div
IL
50 mA/Div
VCS
20V/Div
400 ns/DIV
Figure 18. LED Ripple Current Reduction with a 1 µF Ripple Reduction Capacitor
If a ripple reduction capacitor is used, the equation for SNSHYS_MAX only applies for 100% duty operation.
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LED average DC current and peak inductor current are not affected by the ripple reduction capacitor. However,
LED peak current is reduced and switching frequency may shift slightly.
Any type of capacitor can be used, provided the working voltage rating is sufficient. In general, higher
capacitance and lower ESR will provide more ripple reduction; a typical value greater than 100 nF is
recommended. Smaller capacitance values will be less effective, and large ESR values may actually increase
LED ripple current.
Despite its effectiveness to smooth LED ripple current, there are two notable disadvantages to using a ripple
reduction capacitor.
First, when used, care must be taken to avoid shorting the LED anode to ground. If this occurs, the capacitor will
force a large negative voltage spike at the SNS pin which could damage the IC.
Second, this capacitor will limit the maximum PWM dimming frequency because it takes some additional time to
charge and discharge. Additionally, ceramic capacitors can generate audible noise due to fast voltage changes
during dimming. To reduce audible noise, use the smallest possible case size, use dimming frequencies below
500 Hz, or use a non-ceramic ripple reduction capacitor.
A small bypass capacitor, in the range of 100 pF to 200 pF can also be used to reduce high frequency switching
noise. This is recommended in higher current applications, where switching noise can adversely affect the SNS
or DIM pins. A small capacitor for noise reduction will have little to no effect on the LED ripple current or dimming
but may help solve potential EMI problems.
HG AND PFET SELECTION
When switching, the HG pin swings from VIN (off state) to 4.7V below VIN (typical). As long as the DIM pin is high
and the SNS pin is below the upper threshold, HG will stay low, driving the PFET on.
The PFET should be selected based on the maximum Drain-Source voltage (VDS), Drain current rating (Id),
maximum Gate-Source voltage (VGS), on-resistance (RDS(on)), and Gate capacitance.
The voltage across the PFET in the off state is equal to the sum of the input voltage and the diode forward
voltage. The VDS must therefore be selected to provide some margin beyond this voltage.
Since the peak current through the PFET is equal to the peak current through the inductor, Id must be rated
higher than the maximum ILED_PK. The LM3401 is capable of 100% duty cycle, therefore, the PFET drain current
should be rated to handle ILED_PK continuously. In this case there is no ripple, so IPK = IAVE.
Although the typical HG voltage is VIN - 4.7V, this voltage can go much lower during the initial PFET turn-on time.
How far HG swings at turn-on depends on several factors including the gate capacitance, on-time, and input
voltage. As shown in the Typical Performance Characteristics, the initial HG voltage swing increases with
decreasing PFET gate capacitance. Therefore, A PFET must be selected with a maximum VGS rating larger than
the initial HG voltage. Conversely, when driving PFETs with larger gate capacitance, the initial HG voltage will be
lower. In some cases, a low VGS threshold PFET may be required to ensure complete turn-on. Use the Typical
Performance curve as a guideline to selecting a proper PFET.
Note that HG will eventually settle around the typical voltage of VIN - 4.7V regardless of the PFET gate
capacitance.
HG has an absolute minimum voltage of 1.2V typically. When the input voltage is below approximately 6V, this
minimum limit causes a reduction in drive voltage. At 5V input, for example, HG will swing to 1.2V (or a gate
drive voltage of -3.8V). This may not be sufficient to drive some PFETs, and at this reduced HG voltage, RDS(on)
is likely to increase and trigger current limit. Therefore, a low VGS threshold PFET is also recommended for lower
input voltage applications.
The power loss in the PFET consists of switching losses and conducting losses. Although switching losses are
difficult to precisely calculate, the equations below can be used to estimate total power dissipation, which is the
sum of PDCOND and PDSW.
PDFET_COND = RDS(on) x ILED2 x D
(12)
fSW x ILED x VIN x (Pon + Poff)
PDFET_SW =
2
(13)
Where Pon = PFET turn-on time, Poff = PFET turn-off time, and D is the duty cycle. A value of 10 ns to 50 ns is
typical for ton and toff. Longer PFET on and off times will degrade both efficiency and accuracy.
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Increasing RDS(on) will increase power losses and degrade efficiency. FET RDS(on) has a positive temperature
coefficient. At 125°C, the RDS(on) may be as much as 150% higher than the value at 25°C. The Gate capacitance
of the PFET has a direct impact on both PFET transition time and the power dissipation in the LM3401. Most of
the power dissipated in the LM3401 is used to drive the PFET switch. This power can be calculated as follows:
The average amount of gate driver current required during switching (IG) is:
IG = Qg x fSW
(14)
Where Qg is the PFET gate charge.
And the total power dissipated in the IC is:
PD = (Iq x VIN) + (IG x 4.7V)
(15)
Where Iq is typically 1.05 mA and 4.7V is the typical HG voltage.
Maximum power dissipation within the LM3401 is limited by ambient temperature. Use the following equation to
determine maximum allowable power dissipation, or maximum allowable ambient temperature:
PDmax =
(125°C ± Ta_max)
TJA
(16)
Where θJA is the typical thermal resistance of 151°C/W. In general, keeping the gate capacitance below 2000 pF
is recommended to keep propagation delay, switching losses, and power losses low. PFETs with very fast rise
times may cause excessive ringing at the HG node when combined with the inductance of a long HG trace. To
reduce this ringing, a small resistor can be added between HG and the PFET gate. A typical value of 10Ω is
usually sufficient.
CURRENT LIMIT OPERATION
The LM3401 current limit monitors inductor current at each switching cycle. Current is sensed across the RDS(on)
of the PFET at the CS pin. When the PFET current exceeds the current limit threshold, HG is turned off and the
current limit latch is set. In current limit mode, the PFET is held off until the inductor current falls to near zero.
IL
500 PA /Div
VCS
10V/Div
10 Ps/DIV
Figure 19. Typical Current Limit Operation
The current limit threshold is adjusted with a setting resistor, shown as R3 in the typical application schematic,
connected from ILIM to the VIN node of the PFET.
An internal 5.5 µA (typical) current sink at the ILIM pin creates a voltage across the setting resistor. This voltage
is compared to the sensed voltage at the CS pin. Current limit is activated and latched when the voltage at the
CS pin drops below the voltage at the ILIM pin.
The current limit setting resistor, R3, can be calculated from the equation below. The minimum current limit
occurs at maximum RDS(on) and minimum IILIM value.
R3 =
ILIM_PK x RDS(on)
4 PA
(17)
Where 4 µA is the minimum IILIM value and ILIM_PK is the peak inductor current limit threshold. ILIM_PK should be
set somewhat higher than the maximum LED current, ILED_PK, to avoid false current limit triggering. The
temperature variation of the PFET RDS(on) will result in an equivalent variation in current limit. To ensure that
current limit is not falsely triggered, use the highest RDS(on) value over the temperature range to set the R3 value.
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When current limit is activated, the HG driver remains off until the CS voltage rises to -130 mV (typical). This
ensures that inductor current is close to 0A when the current limit latch is released. The actual minimum inductor
current will depend on the catch diode forward voltage characteristic, which determines the CS pin negative
voltage.
Although the LM3401 monitors voltage at the CS pin to reset the current limit, there is also a minimum off time of
typically 3 µs. When current limit is triggered, HG will be turned off for at least this amount of time, regardless of
the inductor current.
The current limit comparator imposes typically 150 ns of blanking time at the beginning of each switching cycle.
This ensures that the PFET is fully on and any switch node ringing has dissipated when the current is sensed.
However a slower PFET may not fully turn on within the blanking time. In this case, the current limit threshold
must be increased or a faster PFET must be used.
Because the current limit comparator has a limited differential voltage capability, a maximum of 1MΩ is
recommended for R3.
PWM DIMMING
The DIM pin is a CMOS compatible input for a PWM (Pulse Width Modulation) dimming signal. PWM dimming
adjusts LED brightness by varying the duty cycle, which varies the average LED current. This type of dimming is
recommended, because LED peak current remains constant regardless of brightness, which results in more
predictable LED color and performance as compared to analog dimming. Figure 20 shows a typical PWM
dimming waveform.
When DIM is high (above 2V typically) the LM3401 operates normally and the LED string will be driven at the set
current. When pulled low, DIM will disable HG and switching will stop. The PFET will remain off as long as DIM is
low. When the LM3401 is powered up or enabled with the DIM pin, the LED current will very rapidly increase to
its set point.
There is minimal delay time between a DIM logic change and HG switching. Also, because the LM3401 requires
no output capacitor, minimal time is required to ramp-up the LED current. This allows for low duty cycle, high
frequency PWM dimming signals to be used.
A dimming frequency greater than 100 Hz is recommended to avoid visible flicker. The LM3401 is capable of
PWM dimming frequencies up to 10 kHz with a duty cycle between 1 and 100%. Any DIM signal pulse width
longer than 100 ns can be used. In most cases, the maximum dimming frequency is limited by the inductor size
and input voltage to anode voltage ratio. Less inductance and higher VIN/VANODE ratios will allow the inductor and
LED current to increase faster, thus allowing for a faster PWM frequency, or lower dimming duty cycle.
ILED
200 mA/Div
VCS
20V/Div
DIM
2V/Div
2 Ps/DIV
Figure 20. Typical PWM DIM Signal and LED Current L = 22 µH
DIM is a high impedance pin, which is somewhat sensitive to noise. If there is excessive switching noise at the
DIM pin, a small bypass filter capacitor can be used. See the Ripple Reduction Capacitor section. VIN can also
be used for PWM dimming when a logic signal is not available. In this mode of operation DIM should be
connected to VIN through a 10 kΩ resistor. There is typically 10 us of startup delay time when using VIN for
dimming. Depending on the application, this delay limits the maximum dimming frequency to typically several
hundred Hz.
Higher dimming frequency and lower dimming duty cycle can be achieved by using a FET switch in parallel with
the LED string. This is shown in Figure 21 below.
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L1
DIM signal
to
SNS
R1
Figure 21. Parallel FET Dimming
When the FET switches on, inductor current flows through the FET and the regulated average inductor current is
unchanged. Using this method, inductor current rise time does not limit the dimming frequency. A ripple reduction
capacitor should not be used with the parallel FET dimming method since it significantly slows the LED current
rise time. However, a small noise filter capacitor can be used.
INPUT CAPACITOR SELECTION
An input bypass capacitor is required between VIN and ground. The input capacitor prevents large voltage
transients at the input and provides the instantaneous current when the PFET turns on. The important
parameters for the input capacitor are the voltage rating and the RMS current rating. Follow the manufacturer’s
recommended voltage de-rating. RMS current can be calculated with the equation below. The highest RMS
current will occur around 50% duty cycle.
Irms = ILED x
VANODE
VANODE
x 1VIN
VIN
(18)
A ceramic input capacitor must be placed close to the drain of the PFET. This minimizes the trace inductance
between VIN and the PFET, which is a source of switching noise. If the input capacitor is not properly located,
switching noise can cause current limit and stability problems.
CATCH DIODE SELECTION
The catch diode provides the current path to the LED string during the PFET off-time and must be rated higher
than the average current through the diode, which can be calculated as shown:
IDIODE = ILED x (1-D)
(19)
The peak reverse voltage across the catch diode is approximately equal to the input voltage. Therefore, the
diode’s peak reverse voltage rating should be larger than the maximum input voltage, plus some safety margin.
A Schottky diode is recommended because its low forward voltage maximizes efficiency. For high temperature
applications, diode leakage current may become significant and require a higher reverse voltage rating or a low
leakage diode to achieve acceptable performance.
LED CURRENT ACCURACY
The total accuracy of average LED current is affected by several factors, both internal and external to the
LM3401. Total static accuracy is the part-to-part variation and can be calculated from the equation below:
ILED_Acc% = RSNS%2 + VSNS%2
(20)
Where the worst case VSNS% is ±6%, and RSNS% is the sense resistor accuracy. Because these factors are not
correlated, the RSS (root-sum-square) method of calculation is used.
The LED current will also show some variation with input voltage. This is primarily due to propagation delay and
the dynamic resistance of the LED. In longer on-time operation, the error due to dynamic resistance tends to
dominate, while at shorter on-time, the propagation delay will dominate. These two effects counteract each other,
resulting in typical regulation curves similar to those shown in Figure 22. A larger inductor will reduce the error
due to propagation delay and will result in better overall line regulation.
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LED AVERAGE CURRENT (mA)
395
100%
Duty
385
375
50 %
Duty
365
33 PH
68 PH
355
100 PH
345
335
5
10
15
20
25
30
35
INPUT VOLTAGE (V)
Figure 22. LED DC current line regulation LED Vf = 7.0V
For most applications, the average LED current will be the highest at the maximum input voltage and lowest at a
duty cycle somewhat greater than 50%. The maximum LED current variation can be estimated as:
ILED_reg (A) =
(VIN_max ± VIN_60%) x delay
2xL
(21)
Where VIN_60% is the input voltage corresponding to a duty cycle of 60%. Since the actual input voltage where
minimum LED current occurs varies with the application, this is an approximation. As the duty cycle approaches
100%, the average LED current will approach ILED_PK. The average LED current will be the highest at the point
that 100% duty cycle is reached. In the case that 100% duty cycle can occur, maximum LED current variation is
calculated as:
ILED_var_100% (A) =
SNSHYS
RSNS
(22)
PCB LAYOUT
PCB layout is very important in all switching regulator designs. Poor layout can cause EMI problems, excess
switching noise, and improper device operation. The following key points should be followed to ensure a quality
layout.
Traces carrying large AC currents should be as wide and short as possible to minimize trace inductance. These
areas, shown as darker regions in Figure 23, are:
• VIN between the input capacitor and PFET
• GND between the input capacitor and catch diode
• The switch node
As shown in Figure 23, place the input capacitor ground as close as possible to the anode of the catch diode.
The VIN side of the input capacitor should be placed close to the top of the PFET.
The CS node (the node connecting the catch diode cathode, inductor, and PFET source) should be kept as small
as possible. This node is one of the main sources for radiated EMI. The SNS and HYS pins are sensitive to
noise. Be sure to route the SNS trace away from the inductor and the switch node, which are sources of noise.
The SNS and HYS resistors should be placed close to their respective pins and grounded close to the GND pin.
An isolated ground area shown as SGND in Figure 23 is recommended for the SNS, HYS, and GND pin
connections. The two ground areas, GND and SGND, should be connected on an inner or bottom layer. This
connection is shown as two vias in Figure 23.
A large, continuous ground plane can also be used, as long as the input capacitor and catch diode ground area
is somewhat isolated.
The HG trace should be kept as short as possible to minimize inductance and gate ringing (See HG and PFET
selection section).
Finally, for accurate current limit sensing, the CS pin and ILIM resistor connections should be made at the PFET
pads, via separate traces.
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LED+
GND
CS
IC
SGND
VIN
Figure 23. Example PCB Layout
DESIGN EXAMPLE
VIN = 18V to 35V
VIN
C1
2.2 PF
50V
ILIM
R3
46k
DIM
HG
LM3401
HYS
CS
GND
SNS
Q1
40V
1.8A
L1
D1
40V 33 PH
1A
R2
5.6k
1%
* Optional Component
C2*
1 PF
50V
+
690 mA
± 5%
R1
290 m
1%
0.25W
Figure 24. Example Circuit
The following design example is intended to illustrate the step-by-step design process described in the previous
sections. The example refers to the circuit in Figure 24, and the results are summaized in Table 1. The resulting
circuit will drive a string of 2 Luxeon V Star LEDs at 700 mA from an input voltage between 18V and 35V.
The example LEDs have a maximum DC current rating of 700 mA, a forward voltage of 5.4V to 8.3V, and a
maximum peak current rating of 1.0A.
First, set the LED DC current with R1:
R1 =
200 mV
= 286 m:
700 mA
(23)
And the required wattage is:
WRSNS = 700 mA2 x 0.286 = 140 mW
(24)
Select a standard value of 290 mΩ, 1/4W resistor, which will result in a 690 mA LED DC current.
To keep the peak LED current below ILED_MAX, the maximum hysteresis is determined by:
SNSHYS_MAX = (1.0A - .690A) x 0.29Ω = 90 mV
(25)
Which gives a maximum R2 value of:
R2 =
90 mV x 5
= 22.48 k:
20 PA
(26)
Next, a preferred switching frequency of 1 MHz is selected for this example.
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Since this is a relatively high switching frequency, a low starting point of 25 mV is selected for the comparator
hysteresis to maintain good line regulation. This will allow a larger inductor at the same operating frequency and
is well below the calculated maximum. Set a preliminary hysteresis value with R2:
25 mV x 5
= 6.25 k:
20 PA
R2 =
(27)
For 1 MHz switching frequency and 25 mV hysteresis, inductance can be calculated. Because frequency varies
with input voltage and LED forward voltage, for this calculation, assume typical values of 24V and 13.6V
respectively, and a PFET delay time of 15 ns.
0.60
-(2 x 60 ns) x [0.29 x (24V - 13.8V)]
1 MHz
L=
= 29.6 PH
2 x 25 mV
(28)
Select a value of 33 µH and the hysteresis can be adjusted downward by re-arranging the same frequency
equation:
0.60
-(2 x 60 ns) x [0.29 x (24V - 13.8V)]
1 MHz
SNSHYS =
= 22.4 mV
2 x 33 PH
(29)
This gives a new R2 value of 5.6k. This will result in a typical operating frequency of 1 MHz at 24VIN. Next, it
must be verified that the peak LED current is within the maximum allowed.
For now, the design is created without using a ripple reduction capacitor. Therefore, LED ripple current is equal
to inductor ripple current. Maximum LED ripple current is calculated as:
ILED_RIP =
2 x 22.4 mV
0.29:
+
(35V ± 11V) x 2 x 60 ns
33 PH
= 227 mA
(30)
Note that the maximum input voltage and minimum anode voltage were used for this worst case calculation.
Now peak LED current can be determined:
ILED_PK = 690 mA +
227 mA
= 804 mA
2
(31)
This confirms that the component selections will keep LED peak current below the maximum LED rating. Notice if
a ripple reduction capacitor is chosen, the peak inductor current is still 804mA, but the LED peak current is
reduced. Therefore, the inductor must be rated for a DC current greater than 804 mA. Now that the inductor
value has been selected and verified, the operating frequency range can be determined. Lowest operating
frequency occurs at minimum input voltage and maximum anode voltage. For this example the values are 18V
minimum input and 16.8V maximum anode voltage (200 mV SNS voltage plus the maximum LED forward
voltages) and calculate:
fSW =
0.96
2 x 22.4 mV x 33 PH
0.29: x (18V ± 16.8V)
= 219 kHz
+ (2 x 60 ns)
(32)
At duty cycles close to 100% (96% in this case) the frequency equation becomes less accurate. Actual switching
frequency will typically be lower than the calculated value.
To estimate maximum operating frequency, calculate using a VIN which corresponds to a duty cycle of 25%. In
this particular example, 25% duty cycle would occur above 35VIN (VIN = (VOUT + VDIODE)/0.25 = 69.9V), therefore
maximum frequency will occur at the maximum input voltage corresponding to a duty cycle of 50% (D = (VOUT +
VDIODE)/35V = 0.50):
fSW =
0.50
2 x 22.4 mV x 33 PH
+ (2 x 60 ns)
0.29: x (35V ± 16.8V)
= 1.25 MHz
(33)
Using the equation in the Switching Frequency section, it can be verified that this maximum frequency is within
the minimum on-time limited frequency (and below the maximum operating frequency).
The maximum frequency calculation is only an estimate, the actual maximum should be verified on the bench.
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The next step is to select a PFET. The critical PFET parameters must meet the minimum circuit requirements of
35V input, 804 mA DC current, and adequate gate drive voltage rating.
Therefore, select a PFET with the following ratings:
• 40V maximum VDS
• –20V maximum VGS
• 1.8A continuous Id
• 130mohm maximum RDS(on)
Typically the PFET may be only sourcing 690 mA for about 50% duty cycle. However, at minimum input voltage
the duty cycle will increase close to 100%. Therefore, the PFET Id rating should be based on its continuous, not
pulsed, current capability.
Now the power dissipation should be verified. Assume the selected PFET has a gate capacitance of 200 pF,
which is within recommendation, and a gate charge of 15 nC. Maximum frequency and input voltage are used for
a worst case calculation:
IG = 15 nC x 1.25 MHz = 18.75 mA PD = (1.05 mA x 35V) + (18.75 mA x 4.7V) = 0.125W Ta_max = 125°C - (151°C/W x
0.125W) = 106°C
(34)
With the selected components, the maximum ambient temperature is above 100°C, sufficient for most
applications. Note that this limit applies to the IC only and depends on the pcb type and size. Lower ambient
temperature limits may apply to the PFET and other components.
Now the current limit threshold is set with R3 at 0.95A, which is 120% of the maximum peak current. The worst
case RDS(on) value at 125°C is used, which is 150% of nominal, and the worst case ILIM pin sink current.
R3 =
0.95A x 195 m:
= 46.3 k:
4 PA
(35)
The typical current limit threshold will be higher than 1A and can be determined by using typical values for RDS(on)
and ILIM sink current. The PFET, inductor, and catch diode must be able to handle this current for short periods of
time.
The next component is the input capacitor, C1. A low ESR ceramic capacitor must be used and properly located
on the PCB. For this design, the capacitor working voltage must be rated to at least 40V, and 50V is
recommended. A 2.2 µF input capacitor should be sufficient, assuming a good PCB layout. The worst case input
RMS current is calculated below 50% at duty cycle:
Irms = 690 mA x
13.8V
x 1 - 13.8V = 345 mA
27.6V
27.6V
(36)
It must be verified that the selected input capacitor can tolerate this current. An additional bulk capacitor placed
at the input voltage connection is also recommended.
Next, select D1, the catch diode. A Schottky diode should be used. The reverse voltage rating must be greater
than 35V and the average forward current rating must be greater than:
IDIODE = 690 mA x (1 - 0.31) = 480 mA
(37)
This calculation assumes the minimum duty cycle, which is maximum input voltage and minimum anode voltage.
The diode must also be able to handle peak currents as high as the current limit threshold for short periods. We
select a 1A diode to ensure adequate capability over temperature.
If desired, a ripple reduction capacitor can be added at C2 to reduce the LED ripple current. A minimum starting
value of 100 nF is recommended for C2, and a value of 1 µF will work well in most applications. In case of an
open LED failure, the ripple reduction capacitor must be rated to the maximum input voltage of 35V. If C2 is
used, LED ripple current is reduced and the calculated maximum R2 value no longer applies as a limit.
Finally, check the accuracy. The static accuracy is calculated below using a 1% sense resistor.
ILED_Acc% =
2
2
0.01 + 0.06 = 6.1%
(38)
To estimate line regulation, maximum input voltage and 60% duty cycle input voltage is used. For this example
60% duty cycle occurs at (13.6V + 0.2V)/0.60 or 23V input.
ILED_reg =
18
(35V ± 23V) x 60 ns
2 x 33 PH
= 11 mA
(39)
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This is the estimated amount of LED current variation over the input voltage range. If the minimum input voltage
was below 17V, the LED current variation would be calculated using the 100% duty cycle equation.
Table 1. Design Example Summary
Parameter
Value
Result
Comment
R1
290 mΩ
690 mA DC
1%
R2
5.6 kΩ
±22.4 mV hysteresis
VHYS = 112 mV(adjustable)
L1
33 µH, >804 mA
fsw
-
1MHz typical
219 kHz min
Iripple
-
227 mA p-p
worst case
ILED_PK
-
804 mA
worst case
PFET
40V, 1.8A, 130 mΩ
adjustable
R3
46 kΩ
0.95A minimum peak current limit
C1
2.2 µF, 50V ceramic
>345 mA rms
D1
40V, 1A
Schottky
Accuracy
±6.1%
±42 mA max variation
Regulation
1.6%
11 mA variation
Ta_max
108°C
C2
1.0 µF, 50V
part-to-part
vs VIN
worst case
LED ripple reduction
optional
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REVISION HISTORY
Changes from Revision B (May 2013) to Revision C
•
20
Page
Changed layout of National Data Sheet to TI format .......................................................................................................... 19
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PACKAGE OPTION ADDENDUM
www.ti.com
6-Dec-2022
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
Samples
(4/5)
(6)
LM3401MM/NOPB
ACTIVE
VSSOP
DGK
8
1000
RoHS & Green Call TI | SN | NIPDAUAG Level-1-260C-UNLIM
-40 to 125
SNHB
Samples
LM3401MMX/NOPB
ACTIVE
VSSOP
DGK
8
3500
RoHS & Green Call TI | SN | NIPDAUAG Level-1-260C-UNLIM
-40 to 125
SNHB
Samples
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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