LM359M/NOPB

LM359 www.ti.com SNOSBT4C – MAY 1999 – REVISED MARCH 2013 LM359 Dual, High Speed, Programmable, Current Mode (Norton) Amplifiers Check for Samples: LM359 FEATURES APPLICATIONS • • • • • 1 2 • • • • • • User Programmable Gain Bandwidth Product, Slew Rate, Input Bias Current, Output Stage Biasing Current and Total Device Power Dissipation High Gain Bandwidth Product (ISET = 0.5 mA) – 400 MHz for AV = 10 to 100 – 30 MHz for AV = 1 High Slew Rate (ISET = 0.5 mA) – 60 V/μs for AV = 10 to 100 – 30 V/μs for AV = 1 Current Differencing Inputs Allow High Common-Mode Input Voltages Operates from a Single 5V to 22V Supply Large Inverting Amplifier Output Swing, 2 mV to VCC − 2V Low Spot Noise, 6 nV /√Hz, for f > 1 kHz Typical Application • AV = 20 dB • −3 dB bandwidth = 2.5 Hz to 25 MHz • Differential phase error < 1° at 3.58 MHz • Differential gain error < 0.5% at 3.58 MHz • General Purpose Video Amplifiers High Frequency, High Q Active Filters Photo-Diode Amplifiers Wide Frequency Range Waveform Generation Circuits All LM3900 AC Applications Work to Much Higher Frequencies DESCRIPTION The LM359 consists of two current differencing (Norton) input amplifiers. Design emphasis has been placed on obtaining high frequency performance and providing user programmable amplifier operating characteristics. Each amplifier is broadbanded to provide a high gain bandwidth product, fast slew rate and stable operation for an inverting closed loop gain of 10 or greater. Pins for additional external frequency compensation are provided. The amplifiers are designed to operate from a single supply and can accommodate input common-mode voltages greater than the supply. Connection Diagram Figure 1. PDIP/SOIC Package Top View See Package Number D0014A or NFF0014A 1 2 Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. All trademarks are the property of their respective owners. PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of the Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. Copyright © 1999–2013, Texas Instruments Incorporated LM359 SNOSBT4C – MAY 1999 – REVISED MARCH 2013 www.ti.com These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam during storage or handling to prevent electrostatic damage to the MOS gates. Absolute Maximum Ratings (1) (2) Supply Voltage 22 VDC or ±11 VDC D Package Power Dissipation (3) 1W NFF Package 750 mW D Package Maximum TJ +150°C NFF Package +125°C 147°C/W still air D Package θjA 110°C/W with 400 linear feet/min air flow Thermal Resistance 100°C/W still air NFF Package θjA 75°C/W with 400 linear feet/min air flow Input Currents, IIN(+) or IIN(−) 10 mADC Set Currents, ISET(IN) or ISET(OUT) 2 mADC Operating Temperature Range 0°C to +70°C −65°C to +150°C Storage Temperature Range Lead Temperature PDIP Package Soldering Information (1) (2) (3) SOIC Package (Soldering, 10 sec.) 260°C Soldering (10 sec.) 260°C Vapor Phase (60 sec.) 215°C Infrared (15 sec.) 220°C “Absolute Maximum Ratings” indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is functional, but do not ensure specific performance limits. If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/ Distributors for availability and specifications. See Figure 22. Electrical Characteristics ISET(IN) = ISET(OUT) = 0.5 mA, Vsupply = 12V, TA = 25°C unless otherwise noted Parameter Conditions Open Loop Voltage Vsupply = 12V, RL = 1k, f = 100 Hz Gain TA = 125°C Bandwidth Unity Gain RIN = 1 kΩ, Ccomp = 10 pF Gain Bandwidth Product, Gain of 10 to 100 RIN = 50Ω to 200Ω Slew Rate Amplifier to Amplifier Coupling Mirror Gain (1) ΔMirror Gain (1) Input Bias Current Min Typ 62 72 Max Units dB 68 dB 15 30 MHz 200 400 MHz Unity Gain RIN = 1 kΩ, Ccomp = 10 pF 30 Gain of 10 to 100 RIN < 200Ω 60 V/μs −80 f = 100 Hz to 100 kHz, RL = 1k dB at 2 mA IIN(+), ISET = 5 μA, TA = 25°C 0.9 1.0 1.1 μA/μA at 0.2 mA IIN(+), ISET = 5 μA Over Temp. 0.9 1.0 1.1 μA/μA at 20 μA IIN(+), ISET = 5 μA Over Temp. 0.9 μA/μA 1.0 1.1 at 20 μA to 0.2 mA IIN(+) Over Temp, ISET = 5 μA 3 5 % Inverting Input, TA = 25°C 8 15 μA 30 μA Over Temp. Input Resistance (βre) Inverting Input 2.5 kΩ Output Resistance IOUT = 15 mA rms, f = 1 MHz 3.5 Ω (1) 2 Mirror gain is the current gain of the current mirror which is used as the non-inverting input. AI for two different mirror currents at any given temperature. Submit Documentation Feedback ΔMirror Gain is the % change in Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: LM359 LM359 www.ti.com SNOSBT4C – MAY 1999 – REVISED MARCH 2013 Electrical Characteristics (continued) ISET(IN) = ISET(OUT) = 0.5 mA, Vsupply = 12V, TA = 25°C unless otherwise noted Parameter Conditions Output Voltage Swing (RL VOUT High = 600Ω) VOUT Low Output Currents Supply Current IIN(−) and IIN(+) Grounded Typ 9.5 10.3 IIN(−) = 100 μA, IIN(+) = 0 Source IIN(−) and IIN(+) Grounded, RL = 100Ω Sink (Linear Region) Vcomp−0.5V = VOUT = 1V, IIN(+) = 0 Sink (Overdriven) IIN(−) = 100 μA, IIN(+) = 0, VOUT Force = 1V 2 16 Max Units 50 mV V 40 4.7 1.5 Non-Inverting Input Grounded, RL = ∞ Power Supply Rejection (2) f = 120 Hz, IIN(+) Grounded (2) Min 3 18.5 40 mA 22 50 mA dB See Figure 15 and Figure 16. Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: LM359 3 LM359 SNOSBT4C – MAY 1999 – REVISED MARCH 2013 www.ti.com Typical Performance Characteristics Open Loop Gain Figure 2. 4 Open Loop Gain Note: Shaded area refers to LM359 Figure 3. Open Loop Gain Gain Bandwidth Product Figure 4. Figure 5. Slew Rate Gain and Phase Feedback Gain = − 100 Figure 6. Figure 7. Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: LM359 LM359 www.ti.com SNOSBT4C – MAY 1999 – REVISED MARCH 2013 Typical Performance Characteristics (continued) Inverting Input Bias Current Figure 8. Inverting Input Bias Current Note: Shaded area refers to LM359 Figure 9. Mirror Gain Figure 10. Mirror Gain Note: Shaded area refers to LM359 Figure 11. Mirror Gain Figure 12. Mirror Current Note: Shaded area refers to LM359 Figure 13. Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: LM359 5 LM359 SNOSBT4C – MAY 1999 – REVISED MARCH 2013 www.ti.com Typical Performance Characteristics (continued) 6 Supply Current Supply Rejection Figure 14. Figure 15. Supply Rejection Output Sink Current Figure 16. Figure 17. Output Swing Output Impedance Figure 18. Figure 19. Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: LM359 LM359 www.ti.com SNOSBT4C – MAY 1999 – REVISED MARCH 2013 Typical Performance Characteristics (continued) Amplifier to Amplifier Coupling (Input Referred) Noise Voltage Figure 20. Figure 21. Maximum Power Dissipation Note: Shaded area refers to LM359J/LM359N Figure 22. Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: LM359 7 LM359 SNOSBT4C – MAY 1999 – REVISED MARCH 2013 www.ti.com APPLICATION HINTS The LM359 consists of two wide bandwidth, decompensated current differencing (Norton) amplifiers. Although similar in operation to the original LM3900, design emphasis for these amplifiers has been placed on obtaining much higher frequency performance as illustrated in Figure 23. This significant improvement in frequency response is the result of using a common-emitter/common-base (cascode) gain stage which is typical in many discrete and integrated video and RF circuit designs. Another versatile aspect of these amplifiers is the ability to externally program many internal amplifier parameters to suit the requirements of a wide variety of applications in which this type of amplifier can be used. Figure 23. DC BIASING The LM359 is intended for single supply voltage operation which requires DC biasing of the output. The current mirror circuitry which provides the non-inverting input for the amplifier also facilitates DC biasing the output. The basic operation of this current mirror is that the current (both DC and AC) flowing into the non-inverting input will force an equal amount of current to flow into the inverting input . The mirror gain (AI) specification is the measure of how closely these two currents match. For more details see TI Application Note AN-72 (Literature Number SNOA666). DC biasing of the output is accomplished by establishing a reference DC current into the (+) input, IIN(+), and requiring the output to provide the (−) input current. This forces the output DC level to be whatever value necessary (within the output voltage swing of the amplifier) to provide this DC reference current, Figure 24. Figure 24. 8 Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: LM359 LM359 www.ti.com SNOSBT4C – MAY 1999 – REVISED MARCH 2013 The DC input voltage at each input is a transistor VBE(≃ 0.6 VDC) and must be considered for DC biasing. For most applications, the supply voltage, V+, is suitable and convenient for establishing IIN(+). The inverting input bias current, Ib(−), is a direct function of the programmable input stage current (see OPERATING CURRENT PROGRAMMABILITY (ISET)) and to obtain predictable output DC biasing set IIN(+) ≥ 10Ib(−). The following figures illustrate typical biasing schemes for AC amplifiers using the LM359: Figure 25. Biasing an Inverting AC Amplifier Figure 26. Biasing a Non-Inverting AC Amplifier Figure 27. nVBE Biasing Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: LM359 9 LM359 SNOSBT4C – MAY 1999 – REVISED MARCH 2013 www.ti.com The nVBE biasing configuration is most useful for low noise applications where a reduced input impedance can be accommodated (see Typical Applications section). OPERATING CURRENT PROGRAMMABILITY (ISET) The input bias current, slew rate, gain bandwidth product, output drive capability and total device power consumption of both amplifiers can be simultaneously controlled and optimized via the two programming pins ISET(OUT) and ISET(IN). ISET(OUT) The output set current (ISET(OUT)) is equal to the amount of current sourced from pin 1 and establishes the class A biasing current for the Darlington emitter follower output stage. Using a single resistor from pin 1 to ground, as shown in Figure 28, this current is equal to: Figure 28. Establishing the Output Set Current The output set current can be adjusted to optimize the amount of current the output of the amplifier can sink to drive load capacitance and for loads connected to V+. The maximum output sinking current is approximately 10 times ISET(OUT). This set current is best used to reduce the total device supply current if the amplifiers are not required to drive small load impedances. ISET(IN) The input set current ISET(IN) is equal to the current flowing into pin 8. A resistor from pin 8 to V+ sets this current to be: Figure 29. Establishing the Input Set Current ISET(IN) is most significant in controlling the AC characteristics of the LM359 as it directly sets the total input stage current of the amplifiers which determines the maximum slew rate, the frequency of the open loop dominant pole, the input resistance of the (−) input and the biasing current Ib(−). All of these parameters are significant in wide band amplifier design. The input stage current is approximately 3 times ISET(IN) and by using this relationship the following first order approximations for these AC parameters are: 10 Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: LM359 LM359 www.ti.com SNOSBT4C – MAY 1999 – REVISED MARCH 2013 (1) where Ccomp is the total capacitance from the compensation pin (pin 3 or pin 13) to ground, AVOL is the low frequency open loop voltage gain in V/V and an ambient temperature of 25°C is assumed (KT/q = 26 mV and βtyp = 150). ISET(IN) also controls the DC input bias current by the expression: (2) which is important for DC biasing considerations. The total device supply current (for both amplifiers) is also a direct function of the set currents and can be approximated by: Isupply ≃ 27 × ISET(OUT) + 11 × ISET(IN) (3) with each set current programmed by individual resistors. PROGRAMMING WITH A SINGLE RESISTOR Operating current programming may also be accomplished using only one resistor by letting ISET(IN) equal ISET(OUT). The programming current is now referred to as ISET and it is created by connecting a resistor from pin 1 to pin 8 (Figure 30). (4) ISET(IN) = ISET(OUT) = ISET Figure 30. Single Resistor Programming of ISET This configuration does not affect any of the internal set current dependent parameters differently than previously discussed except the total supply current which is now equal to: Isupply ≃ 37 × ISET (5) Care must be taken when using resistors to program the set current to prevent significantly increasing the supply voltage above the value used to determine the set current. This would cause an increase in total supply current due to the resulting increase in set current and the maximum device power dissipation could be exceeded. The set resistor value(s) should be adjusted for the new supply voltage. One method to avoid this is to use an adjustable current source which has voltage compliance to generate the set current as shown in Figure 31. Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: LM359 11 LM359 SNOSBT4C – MAY 1999 – REVISED MARCH 2013 www.ti.com Figure 31. Current Source Programming of ISET This circuit allows ISET to remain constant over the entire supply voltage range of the LM359 which also improves power supply ripple rejection as illustrated in the Typical Performance Characteristics. It should be noted, however, that the current through the LM334 as shown will change linearly with temperature but this can be compensated for (see LM334 data sheet). Pin 1 must never be shorted to ground or pin 8 never shorted to V+ without limiting the current to 2 mA or less to prevent catastrophic device failure. CONSIDERATIONS FOR HIGH FREQUENCY OPERATION The LM359 is intended for use in relatively high frequency applications and many factors external to the amplifier itself must be considered. Minimization of stray capacitances and their effect on circuit operation are the primary requirements. The following list contains some general guidelines to help accomplish this end: 1. Keep the leads of all external components as short as possible. 2. Place components conducting signal current from the output of an amplifier away from that amplifier's noninverting input. 3. Use reasonably low value resistances for gain setting and biasing. 4. Use of a ground plane is helpful in providing a shielding effect between the inputs and from input to output. Avoid using vector boards. 5. Use a single-point ground and single-point supply distribution to minimize crosstalk. Always connect the two grounds (one from each amplifier) together. 6. Avoid use of long wires (> 2″) but if necessary, use shielded wire. 7. Bypass the supply close to the device with a low inductance, low value capacitor (typically a 0.01 μF ceramic) to create a good high frequency ground. If long supply leads are unavoidable, a small resistor (∼10Ω) in series with the bypass capacitor may be needed and using shielded wire for the supply leads is also recommended. COMPENSATION The LM359 is internally compensated for stability with closed loop inverting gains of 10 or more. For an inverting gain of less than 10 and all non-inverting amplifiers (the amplifier always has 100% negative current feedback regardless of the gain in the non-inverting configuration) some external frequency compensation is required because the stray capacitance to ground from the (−) input and the feedback resistor add additional lagging phase within the feedback loop. The value of the input capacitance will typically be in the range of 6 pF to 10 pF for a reasonably constructed circuit board. When using a feedback resistance of 30 kΩ or less, the best method of compensation, without sacrificing slew rate, is to add a lead capacitor in parallel with the feedback resistor with a value on the order of 1 pF to 5 pF as shown in Figure 32. 12 Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: LM359 LM359 www.ti.com SNOSBT4C – MAY 1999 – REVISED MARCH 2013 Cf = 1 pF to 5 pF for stability Figure 32. Best Method of Compensation Another method of compensation is to increase the effective value of the internal compensation capacitor by adding capacitance from the COMP pin of an amplifier to ground. An external 20 pF capacitor will generally compensate for all gain settings but will also reduce the gain bandwidth product and the slew rate. These same results can also be obtained by reducing ISET(IN) if the full capabilities of the amplifier are not required. This method is termed over-compensation. Another area of concern from a stability standpoint is that of capacitive loading. The amplifier will generally drive capacitive loads up to 100 pF without oscillation problems. Any larger C loads can be isolated from the output as shown in Figure 33. Over-compensation of the amplifier can also be used if the corresponding reduction of the GBW product can be afforded. Figure 33. Isolating Large Capacitive Loads In most applications using the LM359, the input signal will be AC coupled so as not to affect the DC biasing of the amplifier. This gives rise to another subtlety of high frequency circuits which is the effective series inductance (ESL) of the coupling capacitor which creates an increase in the impedance of the capacitor at high frequencies and can cause an unexpected gain reduction. Low ESL capacitors like solid tantalum for large values of C and ceramic for smaller values are recommended. A parallel combination of the two types is even better for gain accuracy over a wide frequency range. AMPLIFIER DESIGN EXAMPLES The ability of the LM359 to provide gain at frequencies higher than most monolithic amplifiers can provide makes it most useful as a basic broadband amplification stage. The design of standard inverting and non-inverting amplifiers, though different than standard op amp design due to the current differencing inputs, also entail subtle design differences between the two types of amplifiers. These differences will be best illustrated by design examples. For these examples a practical video amplifier with a passband of 8 Hz to 10 MHz and a gain of 20 dB will be used. It will be assumed that the input will come from a 75Ω source and proper signal termination will be considered. The supply voltage is 12 VDC and single resistor programming of the operating current, ISET, will be used for simplicity. Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: LM359 13 LM359 SNOSBT4C – MAY 1999 – REVISED MARCH 2013 www.ti.com AN INVERTING VIDEO AMPLIFIER 1. Basic circuit configuration: 2. Determine the required ISET from the characteristic curves for gain bandwidth product.GBWMIN= 10 × 10 MHz = 100 MHzFor a flat response to 10 MHz a closed loop response to two octaves above 10 MHz (40 MHz) will be sufficient. Actual GBW = 10 × 40 MHz = 400 MHz ISET required = 0.5 mA 3. Determine maximum value for Rf to provide stable DC biasing Optimum output DC level for maximum symmetrical swing without clipping is: Rf(MAX) can now be found: This value should not be exceeded for predictable DC biasing. 4. Select Rs to be large enough so as not to appreciably load the input termination resistance:Rs ≥ 750Ω; Let Rs = 750Ω 5. Select Rf for appropriate gain: predictability is insured. 6. Since Rf = 7.5k, for the output to be biased to 5.1 VDC, 7.5 kΩ is less than the calculated Rf(MAX) so DC the reference current IIN(+) must be: Now Rb can be found by: 7. Select Ci to provide the proper gain for the 8 Hz minimum input frequency: A larger value of Ci will allow a flat frequency response down to 8 Hz and a 0.01 μF ceramic capacitor in parallel with Ci will maintain high frequency gain accuracy. 8. Test for peaking of the frequency response and add a feedback “lead” capacitor to compensate if necessary. 14 Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: LM359 LM359 www.ti.com SNOSBT4C – MAY 1999 – REVISED MARCH 2013 Figure 34. Final Circuit Using Standard 5% Tolerance Resistor Values Vo(DC) = 5.1V Differential phase error < 1° for 3.58 MHz fIN Differential gain error < 0.5% for 3.58 MHz fIN f−3 dB low = 2.5 Hz Figure 35. Circuit Performance A NON-INVERTING VIDEO AMPLIFIER For this case several design considerations must be dealt with. • The output voltage (AC and DC) is strictly a function of the size of the feedback resistor and the sum of AC and DC “mirror current” flowing into the (+) input. • The amplifier always has 100% current feedback so external compensation is required. Add a small (1 pF–5 pF) feedback capacitance to leave the amplifier's open loop response and slew rate unaffected. • To prevent saturating the mirror stage the total AC and DC current flowing into the amplifier's (+) input should be less than 2 mA. • The output's maximum negative swing is one diode above ground due to the VBE diode clamp at the (−) input. Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: LM359 15 LM359 SNOSBT4C – MAY 1999 – REVISED MARCH 2013 www.ti.com DESIGN EXAMPLE eIN = 50 mV (MAX), fIN = 10 MHz (MAX), desired circuit BW = 20 MHz, AV = 20 dB, driving source impedance = 75Ω, V+ = 12V. 1. Basic circuit configuration: 2. Select ISET to provide adequate amplifier bandwidth so that the closed loop bandwidth will be determined by Rf and Cf. To do this, the set current should program an amplifier open loop gain of at least 20 dB at the desired closed loop bandwidth of the circuit. For this example, an ISET of 0.5 mA will provide 26 dB of open loop gain at 20 MHz which will be sufficient. Using single resistor programming for ISET: 3. Since the closed loop bandwidth will be determined by to obtain a 20 MHz bandwidth, both Rf and Cf should be kept small. It can be assumed that Cf can be in the range of 1 pF to 5 pF for carefully constructed circuit boards to insure stability and allow a flat frequency response. This will limit the value of Rf to be within the range of: Also, for a closed loop gain of +10, Rf must be 10 times Rs + re where re is the mirror diode resistance. 4. So as not to appreciably load the 75Ω input termination resistance the value of (Rs + re) is set to 750Ω. 5. For Av = 10; Rf is set to 7.5 kΩ. 6. The optimum output DC level for symmetrical AC swing is: 7. The DC feedback current must be: DC biasing predictability will be insured because 640 μA is greater than the minimum of ISET/5 or 100 μA. 8. For gain accuracy the total AC and DC mirror current should be less than 2 mA. For this example the maximum AC mirror current will be: μA to 706 μA which will insure gain accuracy. therefore the total mirror current range will be 574 9. Rb can now be found: 10. Since Rs + re will be 750Ω and re is fixed by the DC mirror current to be: Rs must be 750Ω–40Ω or 710Ω which can be a 680Ω resistor in series with a 30Ω resistor which are standard 5% tolerance resistor values. 16 Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: LM359 LM359 www.ti.com SNOSBT4C – MAY 1999 – REVISED MARCH 2013 11. As a final design step, Ci must be selected to pass the lower passband frequency corner of 8 Hz for this example. A larger value may be used and a 0.01 μF ceramic capacitor in parallel with Ci will maintain high frequency gain accuracy. Figure 36. Final Circuit Using Standard 5% Tolerance Resistor Values Vo(DC) = 5.4V Differential phase error < 0.5° Differential gain error < 2% f−3 dB low = 2.5 Hz Figure 37. Circuit Performance GENERAL PRECAUTIONS The LM359 is designed primarily for single supply operation but split supplies may be used if the negative supply voltage is well regulated as the amplifiers have no negative supply rejection. The total device power dissipation must always be kept in mind when selecting an operating supply voltage, the programming current, ISET, and the load resistance, particularly when DC coupling the output to a succeeding stage. To prevent damaging the current mirror input diode, the mirror current should always be limited to 10 mA, or less, which is important if the input is susceptible to high voltage transients. The voltage at any of the inputs must not be forced more negative than −0.7V without limiting the current to 10 mA. The supply voltage must never be reversed to the device; however, plugging the device into a socket backwards would then connect the positive supply voltage to the pin that has no internal connection (pin 5) which may prevent inadvertent device failure. Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: LM359 17 LM359 SNOSBT4C – MAY 1999 – REVISED MARCH 2013 www.ti.com Typical Applications DC Coupled Inputs Figure 38. Inverting Figure 39. Non-Inverting • • Eliminates the need for an input coupling capacitor Input DC level must be stable and can exceed the supply voltage of the LM359 provided that maximum input currents are not exceeded. Figure 40. Noise Reduction using nVBE Biasing 18 Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: LM359 LM359 www.ti.com • SNOSBT4C – MAY 1999 – REVISED MARCH 2013 R1 and C2 provide additional filtering of the negative biasing supply Figure 41. nVBE Biasing with a Negative Supply Figure 42. Typical Input Referred Noise Performance Submit Documentation Feedback Copyright © 1999–2013, Texas Instruments Incorporated Product Folder Links: LM359 19 LM359 SNOSBT4C – MAY 1999 – REVISED MARCH 2013 www.ti.com • FET input voltage mode op amp • For AV = +1; BW = 40 MHz, Sr = 60 V/μs; CC = 51 pF • For AV = +11; BW = 24 MHz, Sr = 130 V/μs; CC = 5 pF • For AV = +100; BW = 4.5 MHz, Sr = 150 V/μs; CC = 2 pF • VOS is typically