OPA2614
SBOS305D − JUNE 2004 − REVISED AUGUST 2008
Dual, High Gain Bandwidth, High Output Current,
Operational Amplifier with Current Limit
FEATURES
D
D
D
D
D
D
APPLICATIONS
D
D
D
D
D
LOW INPUT NOISE VOLTAGE: 1.8nV/√Hz
HIGH GAIN BANDWIDTH PRODUCT: 290MHz
HIGH OUTPUT CURRENT: 350mA
LOW INPUT OFFSET VOLTAGE: ±0.2mV
FLEXIBLE SUPPLY RANGE:
Single +5V to +12V Operation
Dual ±2.5V to ±6V Operation
LOW SUPPLY CURRENT: 6.0mA/ch
xDSL DIFFERENTIAL LINE DRIVERS
16-BIT ADC DRIVERS
TRANSIMPEDANCE AMPLIFIERS
PRECISION BASEBAND I/Q AMPLIFIERS
ACTIVE FILTERS
OPA2614 RELATED PRODUCTS
FEATURES
SINGLES
DUALS
Unity-Gain Stable
OPA2613
High Slew Rate VFB
OPA690
OPA2690
OPA3690
R/R Input/Output VFB
OPA353
OPA2353
Current-Feedback
OPA691
OPA2691
OPA3691
Current-Feedback
OPA2677
DESCRIPTION
The OPA2614 offers very low 1.8nV√Hz input noise in a
wideband, high gain bandwidth, voltage-feedback
architecture. Intended for xDSL driver applications, the
OPA2614 also supports this low input noise with
exceptionally low harmonic distortion, particularly in
differential configurations. Adequate output current is
provided to drive the potentially heavy load of a
twisted-pair line. Harmonic distortion for a 2VPP differential
output operating from +5V to +12V supplies is ≤ −80dBc
through 1MHz input frequencies. Operating on a low
6.0mA/ch supply current, the OPA2614 can satisfy most
xDSL driver requirements over a wide range of possible
supply voltagefrom a single +5 condition, to ±5V, on up
to a single +12V design.
General-purpose applications on a single +5V supply will
benefit from the high input and output voltage swing
available on this reduced supply voltage. Baseband I/Q
receiver channels can achieve almost perfect channel
match with noise and distortion to support signals through
5MHz with > 14-bit dynamic range.
TRIPLES
OPA2614
RO
n:1
xDSL Driver
RO
500Ω
1kΩ
500Ω
OP A2822
500Ω
1kΩ
xDSL Receiver
OP A2822
500Ω
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments
semiconductor products and disclaimers thereto appears at the end of this data sheet.
All trademarks are the property of their respective owners.
Copyright 2004-2008, Texas Instruments Incorporated
! !
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SBOS305D − JUNE 2004 − REVISED AUGUST 2008
ELECTROSTATIC
DISCHARGE SENSITIVITY
ABSOLUTE MAXIMUM RATINGS(1)
Supply Voltage (−40°C to +85°C) . . . . . . . . . . . . . . . . . . . . . . ±6.5V
Supply Voltage (0°C to +70°C) . . . . . . . . . . . . . . . . . . . . . . . ±6.65V
Internal Power Dissipation . . . . . . . . . See Thermal Characteristics
Differential Input Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±1.2V
Input Voltage Range . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±VS
Storage Temperature Range . . . . . . . . . . . . . . . . . . −65°C to +125°C
Lead Temperature (SO-8) . . . . . . . . . . . . . . . . . . . . . . . . . . . . +260°C
Junction Temperature (TJ) . . . . . . . . . . . . . . . . . . . . . . . . . . . +150°C
ESD Rating (Human Body Model) . . . . . . . . . . . . . . . . . . . . 2000V
(Machine Model) . . . . . . . . . . . . . . . . . . . . . . . . . . 200V
(Charge Device Model) . . . . . . . . . . . . . . . . . . . 1500V
This integrated circuit can be damaged by ESD. Texas Instruments
recommends that all integrated circuits be handled with appropriate
precautions. Failure to observe proper handling and installation
procedures can cause damage.
ESD damage can range from subtle performance degradation to
complete device failure. Precision integrated circuits may be more
susceptible to damage because very small parametric changes could
cause the device not to meet its published specifications.
(1) Stresses above these ratings may cause permanent damage.
Exposure to absolute maximum conditions for extended periods
may degrade device reliability. These are stress ratings only, and
functional operation of the device at these or any other conditions
beyond those specified is not supported.
PACKAGE/ORDERING INFORMATION(1)
PRODUCT
PACKAGE-LEAD
PACKAGE
DESIGNATOR
OPA2614
SO-8
D
SPECIFIED
TEMPERATURE
RANGE
PACKAGE
MARKING
−40°C to +85°C
OPA2614
ORDERING
NUMBER
TRANSPORT
MEDIA, QUANTITY
OPA2614ID
Rails, 100
OPA2614IDR
Tape and Reel, 2500
(1) For the most current package and ordering information, see the Package Option Addendum at the end of this document, or see the TI web site
at www.ti.com.
PIN CONFIGURATION
SO
TOP VIEW
OPA2614
2
Out A
1
8
+VS
−In A
2
7
Out B
+In A
3
6
−In B
−VS
4
5
+In B
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SBOS305D − JUNE 2004 − REVISED AUGUST 2008
ELECTRICAL CHARACTERISTICS: VS = ±6V
Boldface limits are tested at +25°C.
RF = 453Ω, RL = 100Ω, and G = +4, unless otherwise noted. See Figure 1 for AC performance only.
OPA2614ID
TYP
PARAMETER
AC Performance (see Figure 1)
Small-Signal Bandwidth
Gain-Bandwidth Product
Bandwidth for 0.1dB Gain Flatness
Peaking at a Gain of +2
Large-Signal Bandwidth
Slew Rate
Rise-and-Fall Time
Settling Time to 0.02%
0.1%
Harmonic Distortion
2nd-Harmonic
3rd-Harmonic
Input Voltage Noise
Input Current Noise
Channel-to-Channel Crosstalk
MIN/MAX OVER TEMPERATURE
+25°C(1)
0°C to
+70°C(2)
−40°C to
+85°C(2)
80
32
218
75
29
196
72
28
190
116
4.4
37
32
114
5.0
39
34
112
5.2
40
35
−65
−92
−87
−110
1.8
1.7
−68
−62
−90
−82
−104
2.0
2.1
−61
−88
−80
−102
2.1
2.2
97
±0.2
92
±1.0
−6
−12
±50
±300
TEST CONDITIONS
+25°C
G = +2, VO = 0.1VPP
G = +4, VO = 0.1VPP
G = +8, VO = 0.1VPP
G ≥ 20
G = +4, VO < 0.1VPP
VO < 0.1VPP
G = +4, VO = 2VPP
G = +4, 4V Step
G = +4, VO = 0.2V Step
G = +4, VO = 2V Step
G = +4, VO = 2V Step
G = +4, f = 1MHz, VO = 2VPP
RL = 20Ω
RL ≥ 500Ω
RL = 20Ω
RL ≥ 500Ω
f > 10kHz
f > 10kHz
f = 1MHz, Input-Referred
180
100
40
290
50
6
42
145
3.5
30
26
TEST
LEVEL
UNITS
MIN/
MAX
MHz
MHz
MHz
MHz
MHz
dB
MHz
V/µs
ns
ns
ns
typ
min
min
min
typ
typ
typ
min
typ
typ
typ
C
B
B
B
C
C
C
B
C
C
C
−60
−87
−78
−100
2.3
2.4
dBc
dBc
dBc
dBc
nV/√Hz
pA/√Hz
dBc
max
max
max
max
max
max
typ
B
B
B
B
B
B
C
92
±1.15
±3.3
−13
−30
±520
±5
91
±1.2
±3.3
−14.5
−35
±750
±7
dB
mV
µV/°C
µA
nA/°C
nA
nA/°C
min
max
max
max
max
max
max
A
A
B
A
B
A
B
(3)
DC Performance(4)
Open-Loop Gain (AOL)
Input Offset Voltage
Average Offset Voltage Drift
Input Bias Current
Average Bias Current Drift (Magnitude)
Input Offset Current
Average Offset Bias Current Drift
VO = 0V, RL = 100Ω
VCM = 0V
VCM = 0V
VCM = 0V
VCM = 0V
VCM = 0V
VCM = 0V
Input
±4.7
±4.5
±4.5
±4.4
V
min
A
VCM = ±1V
100
88
87
86
dB
min
A
VCM = 0
VCM = 0
18 0.6
7 1
kΩ pF
MΩ pF
typ
typ
C
C
No Load
100Ω
VO = 0, Linear Operation
VO = 0, Linear Operation
Output Shorted to Ground
G = +2, f = 100kHz
±5.0
±4.9
+350
−350
500
0.01
V
V
mA
mA
mA
Ω
min
min
min
min
typ
typ
A
A
A
A
C
C
Common-Mode Input Range (CMIR)(5)
Common-Mode Rejection Ratio (CMRR)
Input Impedance
Differential-Mode
Common-Mode
Output
Output Voltage Swing
Current Output, Sourcing
Current Output, Sinking
Short-Circuit Current
Closed-Loop Output Impedance
±4.8
±4.7
+280
−280
±4.8
±4.7
+240
−240
±4.7
±4.6
+220
−220
(1) Junction temperature = ambient for +25°C tested specifications.
(2) Junction temperature = ambient at low temperature limit; junction temperature = ambient +23°C at high temperature limit for over temperature
tested specifications.
(3) Test levels: (A) 100% tested at +25°C. Over temperature limits by characterization and simulation. (B) Limits set by characterization and
simulation. (C) Typical value only for information.
(4) Current is considered positive-out-of-node. VCM is the input common-mode voltage.
(5) Tested < 3dB below minimum CMRR specification at ± CMIR limits.
3
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SBOS305D − JUNE 2004 − REVISED AUGUST 2008
ELECTRICAL CHARACTERISTICS: VS = ±6V (continued)
Boldface limits are tested at +25°C.
RF = 453Ω, RL = 100Ω, and G = +4, unless otherwise noted. See Figure 1 for AC performance only.
OPA2614ID
TYP
PARAMETER
Power Supply
Specified Operating Voltage
Maximum Operating Voltage Range
Maximum Quiescent Current
Minimum Quiescent Current
Power-Supply Rejection Ratio (−PSRR)
TEST CONDITIONS
+25°C
MIN/MAX OVER TEMPERATURE
+25°C(1)
−40°C to
+85°C(2)
UNITS
MIN/
MAX
±6.3
12.4
11.6
90
±6.3
12.8
11.2
88
±6.3
13
11
87
V
V
mA
mA
dB
typ
max
max
min
min
C
A
A
A
A
°C
typ
C
°C/W
typ
C
±6
VS = ±6V, Both Channels
VS = ±6V, Both Channels
Input-Referred
12
12
95
TEST
LEVEL
0°C to
+70°C(2)
(3)
Thermal Characteristics
Specified Operating Range D Package
Thermal Resistance, qJA
D
SO-8
−40 to +85
Junction-to-Ambient
125
(1) Junction temperature = ambient for +25°C tested specifications.
(2) Junction temperature = ambient at low temperature limit; junction temperature = ambient +23°C at high temperature limit for over temperature
tested specifications.
(3) Test levels: (A) 100% tested at +25°C. Over temperature limits by characterization and simulation. (B) Limits set by characterization and
simulation. (C) Typical value only for information.
(4) Current is considered positive-out-of-node. VCM is the input common-mode voltage.
(5) Tested < 3dB below minimum CMRR specification at ± CMIR limits.
4
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SBOS305D − JUNE 2004 − REVISED AUGUST 2008
ELECTRICAL CHARACTERISTICS: VS = +5V
Boldface limits are tested at +25°C.
RF = 402Ω, RL = 100Ω, and G = +2, unless otherwise noted. See Figure 3 for AC performance only.
OPA2614ID
TYP
MIN/MAX OVER TEMPERATURE
+25°C(1)
0°C to
+70°C(2)
−40°C to
+85°C(2)
81
32
210
75
28
186
72
27
181
98
4.5
42
34
96
5.1
44
36
94
5.2
46
37
−64
−92
−85
−105
1.9
1.7
−68
−60
−89
−80
−100
2.1
2.1
−58
−87
−78
−98
2.2
2.2
95
±0.2
91
±1.0
−6
−11
±50
±300
Least Positive Input Voltage(5)
1.2
Most Positive Input Voltage(5)
3.8
VCM = ±1V
95
VCM = 0
VCM = 0
15 1
5 1.3
Most Positive Output Voltage
No Load
100Ω Load to 2.5V
4.0
3.95
3.85
3.8
3.8
3.75
Least Positive Output Voltage
No Load
100Ω Load to 2.5V
VO = 0, Linear Operation
VO = 0, Linear Operation
Output Shorted to Mid-Supply
G = +2, f = 100kHz
1.0
1.05
+300
−300
±400
0.01
1.15
1.20
1.2
1.25
PARAMETER
AC Performance (see Figure 3)
Small-Signal Bandwidth
Gain-Bandwidth Product
Bandwidth for 0.1dB Gain Flatness
Peaking at a Gain of +2
Large-Signal Bandwidth
Slew Rate
Rise-and-Fall Time
Settling Time to 0.02%
0.1%
Harmonic Distortion
2nd-Harmonic
3rd-Harmonic
Input Voltage Noise
Input Current Noise
Channel-to-Channel Crosstalk
TEST CONDITIONS
+25°C
G = +2, VO = 0.1VPP
G = +4, VO = 0.1VPP
G = +8, VO = 0.1VPP
G ≥ 20
G = +4, VO < 0.1VPP
VO < 0.1VPP
G = +4, VO = 2VPP
G = +4, 2V Step
G = +4, VO = 0.2V Step
G = +4, VO = 2V Step
G = +4, VO = 2V Step
G = +4, f = 1MHz, VO = 2VPP
RL = 20Ω to VS/2
RL ≥ 500Ω to VS/2
RL = 20Ω to VS/2
RL ≥ 500Ω to VS/2
f > 10kHz
f > 10kHz
f = 1MHz, Input-Referred
150
100
40
250
17
7.5
40
135
3.5
34
27
TEST
LEVELS
UNITS
MIN/
MAX
MHz
MHz
MHz
MHz
MHz
dB
MHz
V/µs
ns
ns
ns
typ
min
min
min
typ
typ
typ
min
typ
typ
typ
C
B
B
B
C
C
C
B
B
B
B
−57
−86
−76
−96
2.4
2.4
dBc
dBc
dBc
dBc
nV/√Hz
pA/√Hz
dBc
max
max
max
max
max
max
typ
B
B
B
B
B
B
C
89
±1.15
±3.3
−12
−35
±520
±5
88
±1.2
±3.3
−13.5
−35
±750
±7
dB
mV
µV/°C
µA
nA/°C
nA
nA/°C
min
max
max
max
max
max
max
A
A
B
A
B
A
B
1.4
1.4
1.5
V
max
A
3.6
3.6
3.5
V
min
A
85
84
83
dB
min
kΩ pF
MΩ pF
typ
typ
A
A
C
C
3.75
3.7
V
V
min
min
A
A
1.25
1.3
V
V
mA
mA
mA
Ω
min
min
typ
typ
typ
typ
A
A
C
C
C
C
(3)
DC Performance(4)
Open-Loop Gain (AOL)
Input Offset Voltage
Average Offset Voltage Drift
Input Bias Current
Average Bias Current Drift (Magnitude)
Input Offset Current
Average Offset Bias Current Drift
VO = 0V, RL = 100Ω
VCM = 0V
VCM = 0V
VCM = 0V
VCM = 0V
VCM = 0V
VCM = 0V
Input
Common-Mode Rejection Ratio (CMRR)
Input Impedance
Differential-Mode
Common-Mode
Output
Current Output, Sourcing
Current Output, Sinking
Short-Circuit Current
Closed-Loop Output Impedance
(1) Junction temperature = ambient for +25°C tested specifications.
(2) Junction temperature = ambient at low temperature limit; junction temperature = ambient +23°C at high temperature limit for over temperature
tested specifications.
(3) Test levels: (A) 100% tested at +25°C. Over temperature limits by characterization and simulation. (B) Limits set by characterization and
simulation. (C) Typical value only for information.
(4) Current is considered positive-out-of-node. VCM is the input common-mode voltage.
(5) Tested < 3dB below minimum CMRR specification at ± CMIR limits.
5
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SBOS305D − JUNE 2004 − REVISED AUGUST 2008
ELECTRICAL CHARACTERISTICS: VS = +5V (continued)
Boldface limits are tested at +25°C.
RF = 402Ω, RL = 100Ω, and G = +2, unless otherwise noted. See Figure 3 for AC performance only.
OPA2614ID
TYP
PARAMETER
TEST CONDITIONS
+25°C
MIN/MAX OVER TEMPERATURE
+25°C(1)
0°C to
+70°C(2)
−40°C to
+85°C(2)
12.6
11.0
9.4
12.6
11.3
9.4
12.6
11.5
9.1
TEST
LEVELS
UNITS
MIN/
MAX
V
V
mA
mA
dB
typ
max
max
min
typ
C
A
A
A
C
°C
typ
C
°C/W
typ
C
(3)
Power Supply
Specified Operating Voltage
Maximum Operating Voltage Range
Maximum Quiescent Current
Minimum Quiescent Current
Power-Supply Rejection Ratio (−PSRR)
Thermal Characteristics
Specified Operating Range D Package
Thermal Resistance, qJA
D
SO-8
5
VS = +5V, both channels
VS = +5V, both channels
Input-Referred
10.5
10.5
95
−40 to +85
Junction-to-Ambient
125
(1) Junction temperature = ambient for +25°C tested specifications.
(2) Junction temperature = ambient at low temperature limit; junction temperature = ambient +23°C at high temperature limit for over temperature
tested specifications.
(3) Test levels: (A) 100% tested at +25°C. Over temperature limits by characterization and simulation. (B) Limits set by characterization and
simulation. (C) Typical value only for information.
(4) Current is considered positive-out-of-node. VCM is the input common-mode voltage.
(5) Tested < 3dB below minimum CMRR specification at ± CMIR limits.
6
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SBOS305D − JUNE 2004 − REVISED AUGUST 2008
TYPICAL CHARACTERISTICS: VS = ±6V
At TA = +25°C, G = +4, RF = 453Ω, and RL = 100Ω, unless otherwise noted.
INVERTING SMALL−SIGNAL
FREQUENCY RESPONSE
NONINVERTING SMALL−SIGNAL
FREQUENCY RESPONSE
6
6
VO = 100mVPP
G = +2
3
G = −2
VO = 100mVPP
3
Normalized Gain (dB)
−3
−6
G = +8
−9
G = +12
−12
−15
−18
1
−6
100
−12
G = −16
See Figure 2
1
500
10
Frequency (MHz)
NONINVERTING LARGE−SIGNAL
FREQUENCY RESPONSE
INVERTING LARGE−SIGNAL
FREQUENCY RESPONSE
15
VO = 100mVPP
12
RL = 100Ω
VO = 500mVPP
G = −4V/V
VO = 100mVPP
RL = 100Ω
VO = 500mVPP
VO = 1VPP
VO = 2VPP
3
9
Gain (dB)
6
0
VO = 5VPP
−3
VO = 2VPP
3
VO = 5VPP
See
See Figure
Figure 44
−9
1
10
100
500
1
10
Frequency (MHz)
0
−1
Right Scale
0.1
0
−0.1
−2
−0.2
2
Output Voltage (1V/div)
0.2
Large Signal
Output Voltage (100mV/div)
Left Scale 4V
PP
200mVPP
Small Signal
500
INVERTING PULSE RESPONSE
3
0.3
1
100
Frequency (MHz)
NONINVERTING PULSE RESPONSE
G = +4V/V
RL = 100Ω
VO = 1VPP
0
−6
See Figure 1
−9
6
−3
−6
Output Voltage (1V/div)
500
12
9
2
100
Frequency (MHz)
G = +4V/V
3
G = −8
G = −12
−9
−18
10
15
Gain (dB)
G = −4
−3
−15
G = +16
See Figure 1
0
G = −4V/V
RL = 100Ω
0.3
Left Scale
4VPP
0.2
Large Signal
1
200mVPP
Small Signal
0
−1
0.1
0
−0.1
−2
See Figure 1
Right Scale
−0.2
Output Voltage (100mV/div)
Normalized Gain (dB)
G = +4
0
See Figure 2
−3
−0.3
Time (20ns/div)
−3
−0.3
Time (20ns/div)
7
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SBOS305D − JUNE 2004 − REVISED AUGUST 2008
TYPICAL CHARACTERISTICS: VS = ±6V (continued)
At TA = +25°C, G = +4, RF = 453Ω, and RL = 100Ω, unless otherwise noted.
HARMONIC DISTORTION vs FREQUENCY
G = +4
RL = 100Ω
−70
2nd−Harmonic
−80
−90
3rd−Harmonic
−100
0.1
1
G = +4
RL = 100Ω
f = 1MHz
−70
2nd−Harmonic
−80
−90
−100
3rd−Harmonic
Single Channel (see Figure 1)
−110
HARMONIC DISTORTION vs OUTPUT VOLTAGE
−60
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
−60
−110
0.1
10
1
HARMONIC DISTORTION vs NONINVERTING GAIN
VO = 2VPP
f = 1MHz
RL = 100Ω
−70
2nd−Harmonic
−80
−90
3rd−Harmonic
−100
Single Channel (see Figure 1)
−110
HARMONIC DISTORTION vs INVERTING GAIN
−60
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
−60
VO = 2VPP
f = 1MHz
RL = 100Ω
−70
10
2nd−Harmonic
−80
−90
3rd−Harmonic
−100
Single Channel (see Figure 2)
−110
1
20
1
10
Gain Magnitude (V/V)
Harmonic Distortion (dBc)
Gain Magnitude (V/V)
HARMONIC DISTORTION vs LOAD RESISTANCE
−60
VO = 2VPP
f = 1MHz
−70
2nd−Harmonic
−80
3rd−Harmonic
−90
−100
Single Channel
(see Figure 1)
−110
10
100
Load Resistance (Ω)
8
10
Output Voltage (VPP)
Frequency (MHz)
1000
20
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SBOS305D − JUNE 2004 − REVISED AUGUST 2008
TYPICAL CHARACTERISTICS: VS = ±6V (continued)
At TA = +25°C, G = +4, RF = 453Ω, and RL = 100Ω, unless otherwise noted.
MAXIMUM OUTPUT SWING
vs LOAD RESISTANCE
OUTPUT VOLTAGE AND CURRENT LIMITATIONS
6
5
6
RL = 100Ω
4
3
2
3
2
1
VO (V)
Output Voltage (V)
5
4
0
−1
−2
RL = 50Ω
1
0
−1
RL = 25Ω
−2
−3
−3
−4
−4
−5
−5
−6
See Figure 1
10
100
−6
−400
1000
1W Internal Power
Single Channel
−300
−200
−100
0
100
200
INPUT VOLTAGE AND CURRENT NOISE DENSITY
400
CHANNEL−TO−CHANNEL CROSSTALK
−30
10
Voltage Noise 1.8nV/√Hz
Crosstalk, Input Referred (dB)
Input-Referred
Voltage Noise (nV/√Hz)
Current Noise (pA/√Hz)
300
IO (mA)
Load Resistance (Ω)
G = +4V/V
RL = 100Ω
−40
−50
−60
−70
Current Noise 1.7pA/√Hz
−80
1
102
103
104
105
106
107
1
10
RECOMMENDED RS vs CAPACITIVE LOAD
FREQUENCY RESPONSE vs CAPACITIVE LOAD
Normalized Gain to Capacitive Load (dB)
70
Gain of +4V/V
0dB Peaking Targeted
60
RS (Ω)
50
40
30
20
10
0
1
10
100
Capacitive Load (pF)
100
Frequency (MHz)
Frequency (Hz)
1000
3
CL = 10pF
0
CL = 100pF
−3
CL = 47pF
CL = 22pF
−6
1/2
OPA2614
−9
−12
453Ω
−15
150Ω
RS
CL
1kΩ
1kΩ is optional.
−18
1
10
100
500
Frequency (MHz)
9
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SBOS305D − JUNE 2004 − REVISED AUGUST 2008
TYPICAL CHARACTERISTICS: VS = ±6V (continued)
At TA = +25°C, G = +4, RF = 453Ω, and RL = 100Ω, unless otherwise noted.
OPEN−LOOP GAIN AND PHASE
CMRR AND PSRR vs FREQUENCY
−PSRR
60
40
80
20
0
1k
10k
100k
1M
10M
100M
60
−90
40
−120
20
−150
0
−180
−20
100
−210
1k
10k
100k
1M
10M
100M
1G
Frequency (Hz)
Frequency (Hz)
CLOSED−LOOP OUTPUT IMPEDANCE
vs FREQUENCY
NONINVERTING OVERDRIVE RECOVERY
10
10
2.5
Input
8
1
Output Voltage (2V/div)
Output Impedance Magnitude (Ω)
−60
∠ AOL
0.1
0.01
0.001
6
1.0
2
0.5
0
0
−2
100k
1M
10M
−0.5
−4
−6
−10
10k
1.5
Output
4
−8
0.0001
2.0
−1.0
−1.5
G = +4V/V
RL = 100Ω
See Figure 1
Input Voltage (0.5V/div)
100
Open−Loop Phase (_ )
CMRR
−30
20 log (AOL)
100
100
80
0
120
+PSRR
Open−Loop Gain (dB)
Common−Mode Rejection Ratio (dB)
Power−Supply Rejection Ratio (dB)
120
−2.0
−2.5
100M
Time (100ns/div)
Frequency (Hz)
INVERTING OVERDRIVE RECOVERY
TYPICAL DC DRIFT OVER TEMPERATURE
1.5
4
1.0
2
0.5
0
0
−2
−0.5
−4
−1.0
−6
−1.5
−10
Output
See Figure 2
0.3
0.1
5
(10 Times Input Offset Current) 10 x IOS
0
−0.1
Input Offset Voltage (VIO)
−5
−0.3
−2.0
−2.5
Time (100ns/div)
10
2.0
6
−8
0.5
2.5
−0.5
−50
Input Bias Current (IB)
−25
0
25
50
75
Ambient Temperature (_C)
10
100
−10
125
Input Bias and Offset Current (µA)
Output Voltage (2V/div)
8
G = −4V/V
RL = 100Ω
Input Offset Voltage (mV)
Input
Input Voltage (0.5V/div)
10
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SBOS305D − JUNE 2004 − REVISED AUGUST 2008
TYPICAL CHARACTERISTICS: VS = ±6V (continued)
At TA = +25°C, G = +4, RF = 453Ω, and RL = 100Ω, unless otherwise noted.
COMMON−MODE INPUT RANGE AND OUTPUT SWING
vs SUPPLY VOLTAGE
SUPPLY AND OUTPUT CURRENT vs TEMPERATURE
12.3
300
6
12.2
12.1
280
Supply Current
Right Scale
12.0
270
11.9
260
5
Voltage Range (V)
290
Supply Current (0.1mA/div)
Output Current (10mA/div)
RL = 100Ω
Sourcing and Sinking Current
Left Scale
−V Input Voltage
4
3
−Output Voltage
2
+V Input Voltage
1
+Output Voltage
250
−50
−25
0
25
50
75
Ambient Temperature (_C)
100
11.8
125
0
2.5
3.0
3.5
4.0
4.5
5.0
5.5
6
Supply Voltage (±V)
11
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TYPICAL CHARACTERISTICS: VS = ±6V, Differential Configuration
At TA = +25°C, GD = 8, RF = 453Ω, and RL = 70Ω, unless otherwise noted. See Figure 5 for AC performance only.
DIFFERENTIAL SMALL−SIGNAL
FREQUENCY RESPONSE
3
21
GD = +2
RL = 70Ω
VO = 200mVPP
GD = +4
RL = 70Ω
GD = +8
18
VO = 0.5VPP
0
15
Gain (dB)
Normalized Gain (dB)
DIFFERENTIAL LARGE−SIGNAL
FREQUENCY RESPONSE
GD = +8
−3
GD = +16
VO = 2VPP
12
VO = 5VPP
9
−6
6
See Figure 5
−9
See Figure 5
3
10
1
100
200
1
10
Frequency (MHz)
−70
−60
GD = +8
f = 1MHz
VO = 2VPP
2nd−Harmonic
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
DIFFERENTIAL DISTORTION vs FREQUENCY
DIFFERENTIAL DISTORTION vs LOAD RESISTANCE
−60
−80
−90
3rd−Harmonic
−100
−110
See Figure 5
10
GD = +8
RL = 70Ω
VO = 2VPP
−70
−80
2nd−Harmonic
−90
−100
100
0.1
1k
1
Frequency (MHz)
DIFFERENTIAL DISTORTION
vs OUTPUT VOLTAGE
−80
Harmonic Distortion (dBc)
See Figure 5
3rd−Harmonic
−110
Load Resistance (Ω)
3rd−Harmonic
GD = +8V/V
RL = 70Ω
f = 1MHz
−85
−90
2nd−Harmonic
−95
See Figure 5
−100
0.1
1
Output Voltage Swing (VPP)
12
100
Frequency (MHz)
10
20
10
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TYPICAL CHARACTERISTICS: VS = +5V
At TA = +25°C, G = +4, RF = 453Ω, and RL = 100Ω, unless otherwise noted.
NONINVERTING SMALL−SIGNAL
FREQUENCY RESPONSE
NONINVERTING SMALL−SIGNAL
FREQUENCY RESPONSE
9
9
VO = 100mVPP
RL = 100Ω to VS/2
VO = 100mVPP
RL = 100Ω to VS/2
6
Normalized Gain (dB)
3
G = +4V/V
0
G = +8V/V
G = +12V/V
−3
G = +16V/V
−6
G = −2V/V
3
0
G = −4V/V
−3
G = −8V/V
G = −12V/V
−6
G = −16V/V
See Figure 4
See Figure 3
−9
−9
1
10
100
500
1
10
100
Frequency (MHz)
Frequency (MHz)
NONINVERTING LARGE−SIGNAL
FREQUENCY RESPONSE
INVERTING LARGE−SIGNAL
FREQUENCY RESPONSE
15
15
G = +4V/V
RL = 100Ω to VS/2
G = −4V/V
RL = 100Ω to VS/2
VO = 0.1VPP
VO = 0.1VPP
12
12
VO = 0.5VPP
9
Gain (dB)
Gain (dB)
VO = 0.5VPP
VO = 1VPP
6
9
VO = 1VPP
6
VO = 2VPP
VO = 2VPP
3
3
See Figure 3
0
1
See Figure 4
0
10
100
1
500
10
3.0
4.5
2.9
4.1
2.8
Large Signal
2.7
3.3
200mVPP
Small Signal
2.9
2.5
Right Scale
2.6
2.5
Output Voltage (500mV/div)
3.7
2VPP
G = +4V/V
RL = 100Ω to VS /2
Output Voltage (100mV/div)
Output Voltage (500mV/div)
Left Scale
3.7
2.9
3.0
G = −4V/V
RL = 100Ω to VS/2 2.9
2.8
Large Signal
2.7
200mVPP
Small Signal
Right Scale
2.6
2.5
2.1
2.4
1.7
2.3
1.3
2.2
2.1
1.7
2.3
1.3
2.2
0.9
2.1
0.9
2.0
0.5
Time (20ns/div)
2VPP
2.5
2.4
0.5
Left Scale
3.3
2.1
See Figure 3
500
INVERTING PULSE RESPONSE
NONINVERTING PULSE RESPONSE
4.5
100
Frequency (MHz)
Frequency (MHz)
4.1
500
Output Voltage (100mV/div)
Normalized Gain (dB)
6
G = +2V/V
2.0
Time (20ns/div)
13
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SBOS305D − JUNE 2004 − REVISED AUGUST 2008
TYPICAL CHARACTERISTICS: VS = +5V (continued)
At TA = +25°C, G = +4, RF = 453Ω, and RL = 100Ω, unless otherwise noted.
HARMONIC DISTORTION vs FREQUENCY
VO = 2VPP
G = +4
RL = 100Ω to VS/2
−70
2nd−Harmonic
−80
−90
3rd−Harmonic
−100
Single Channel
(see Figure 3)
−110
0.1
HARMONIC DISTORTION vs OUTPUT VOLTAGE
−60
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
−60
1
f = 1MHz
RL = 100Ω to VS/2
−70
2nd−Harmonic
−80
−90
3rd−Harmonic
−100
Single Channel
(see Figure 3)
−110
0.1
10
1
HARMONIC DISTORTION vs INVERTING GAIN
HARMONIC DISTORTION vs NONINVERTING GAIN
VO = 2VPP
f = 1MHz
RL = 100Ω to VS/2
−70
−60
2nd−Harmonic
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
−60
−80
−90
3rd−Harmonic
−100
VO = 2VPP
f = 1MHz
RL = 100Ω to VS/2
−70
2nd−Harmonic
−80
−90
3rd−Harmonic
−100
−110
−110
1
10
1
20
10
Gain Magnitude (V/V)
Gain Magnitude (V/V)
HARMONIC DISTORTION vs LOAD RESISTANCE
Harmonic Distortion (dBc)
−60
2nd−Harmonic
VO = 2VPP
f = 1MHz
G = +4V/V
RL to VS/2
−70
−80
−90
3rd−Harmonic
−100
−110
10
100
Load Resistance (Ω)
14
5
Output Voltage (VPP)
Frequency (MHz)
1000
20
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TYPICAL CHARACTERISTICS: VS = +5V, Differential Configuration
At TA = +25°C, GD = 8, RF = 453Ω, and RL = 70Ω, unless otherwise noted.
+5V
DIFFERENTIAL SMALL−SIGNAL
FREQUENCY RESPONSE
806Ω
6
RL = 70Ω
806Ω
Normalized Gain (dB)
1/2
OPA2614
RF
453Ω
0.01µF
RG
VI
RL
RF
453Ω
0.01µF
VI
G D = +2
3
GD = +4
0
−3
GD = +8
GD = +16
−6
0.01µF
806Ω
1/2
OPA2614
GD = 1 +
806Ω
−9
2RF
RG
0.1
10
100
200
Frequency (MHz)
DIFFERENTIAL LARGE−SIGNAL
FREQUENCY RESPONSE
DIFFERENTIAL DISTORTION vs LOAD RESISTANCE
−60
21
Harmonic Distortion (dBc)
RL = 70Ω
GD = 8V/V
15
Gain (dB)
GD = +8
VO = 2VPP
f = 1MHz
VO = 0.1VPP
18
VO = 2VPP
12
9
VO = 5VPP
6
−70
2nd−Harmonic
−80
−90
3rd−Harmonic
−100
3
−110
0
1
10
100
200
10
100
DIFFERENTIAL DISTORTION vs FREQUENCY
−70
GD = 8V/V
RL = 70Ω
VO = 2VPP
−80
2nd−Harmonic
−90
−100
3rd−Harmonic
−110
0.1
DIFFERENTIAL DISTORTION vs OUTPUT VOLTAGE
−80
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
−60
1k
Load Resistance (Ω)
Frequency (MHz)
GD = +8V/V
RL = 70Ω
f = 1MHz
−85
2nd−Harmonic
−90
3rd−Harmonic
−95
−100
1
Frequency (MHz)
10
0.1
1
4
Output Voltage Swing (VPP)
15
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SBOS305D − JUNE 2004 − REVISED AUGUST 2008
APPLICATION INFORMATION
WIDEBAND VOLTAGE-FEEDBACK OPERATION
The OPA2614 gives the exceptional AC performance of a
wideband voltage-feedback op amp with a highly linear,
high-power output stage. Requiring only 6mA/ch
quiescent current, the OPA2614 swings to within 1.0V of
either supply rail and delivers in excess of 280mA at room
temperature. This low-output headroom requirement,
along with supply voltage independent biasing, gives
remarkable single (+5V) supply operation. The OPA2614
delivers greater than 40MHz bandwidth driving a 2VPP
output into 100Ω on a single +5V supply. Previous boosted
output stage amplifiers typically suffer from very poor
crossover distortion as the output current goes through
zero. The OPA2614 achieves exceptional power gain with
much better linearity. Figure 1 shows the DC-coupled,
gain of +4, dual power-supply circuit configuration used as
the basis of the ±6V Electrical and Typical Characteristics.
For test purposes, the input impedance is set to 50Ω with
a resistor to ground, and the output impedance is set to
50Ω with a series output resistor. Voltage swings reported
in the electrical characteristics are taken directly at the
input and output pins, whereas load powers (dBm) are
defined at a matched 50Ω load. For the circuit of Figure 1,
the total effective load is 100Ω || 603Ω = 86Ω.
0.1µF
+6V
+VS
6.8µF
+
50Ω Source
VI
50Ω Load
50Ω
VO
1/2
OPA2614
50Ω
RF
453Ω
RG
150Ω
6.8µF
0.1µF
+
−VS
−6V
Figure 1. DC-Coupled, G = +4, Bipolar Supply,
Specification and Test Circuit
16
Figure 2 shows the DC-coupled, bipolar supply circuit
configuration used as the basis for the Inverting Gain
−4V/V Typical Characteristics. Key design considerations
of the inverting configuration are developed in the Inverting
Amplifier Operation section.
+5V
208Ω
50Ω
Source
RG
113Ω
Power−supply
decoupling
not shown.
1/2
OPA2614
VO
50Ω
50ΩLoad
−5V R
F
453Ω
VI
RM
89Ω
Figure 2. DC-Coupled, G = −4, Bipolar Supply,
Specification and Test Circuit
Figure 3 shows the AC-coupled, gain of +4, single-supply
circuit configuration used as the basis of the +5V Electrical
and Typical Characteristics. Though not a rail-to-rail
design, the OPA2614 requires minimal input and output
voltage headroom compared to other very wideband
voltage-feedback op amps. It will deliver a 2.6VPP output
swing on a single +5V supply with greater than 20MHz
bandwidth. The key requirement of broadband singlesupply operation is to maintain input and output signal
swings within the usable voltage ranges at both the input
and the output. The circuit of Figure 3 establishes an input
midpoint bias using a simple resistive divider from the +5V
supply (two 906Ω resistors). The input signal is then
AC-coupled into this midpoint voltage bias. The input
voltage can swing to within 1.4V of either supply pin, giving
a 2.2VPP input signal range centered between the supply
pins. The input impedance matching resistor (56.2Ω) used
for testing is adjusted to give a 50Ω input match when the
parallel combination of the biasing divider network is
included. The gain resistor (RG) is AC-coupled, giving the
circuit a DC gain of +1which puts the input DC bias
voltage (2.5V) on the output as well. Again, on a single +5V
supply, the output voltage can swing to within 1.1V of either
supply pin while delivering more than 100mA output
current. A demanding 100Ω load to a midpoint bias is used
in this characterization circuit. The new output stage used
in the OPA2614 can deliver large bipolar output currents
into this midpoint load with minimal crossover distortion,
as shown by the +5V supply, harmonic distortion plots.
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SBOS305D − JUNE 2004 − REVISED AUGUST 2008
the OPA2614. Each has its advantages and disadvantages. Figure 5 shows a basic starting point for
noninverting input differential I/O applications.
+5V
+VS
+
0.1µF
6.8µF
906Ω
+VCC
Power−supply
decoupling not
shown.
0.1µF
VI
56.2Ω
906Ω
1/2
OPA2614
VO
100Ω
1/2
OPA2614
VS /2
RF
453Ω
RF
453Ω
RG
150Ω
VI
RG
301Ω
RF
453Ω
VO
0.1µF
1/2
OPA2614
Figure 3. AC-Coupled, G = +4, Single-Supply,
Specification and Test Circuit
The last configuration used as the basis of the +5V
Electrical and Typical Characteristics is shown in Figure 4.
Design considerations for this inverting, bipolar supply
configuration are covered either in single-supply
configuration (as shown in Figure 3) or in the Inverting
Amplifier Operation section.
+5V
0.1µF
906Ω
+
−VCC
Figure 5. Noninverting Differential I/O Amplifier
This approach provides for a source termination
impedance that is independent of the signal gain. For
instance, simple differential filters may be included in the
signal path right up to the noninverting inputs without
interacting with the gain setting. The differential signal gain
for the circuit of Figure 5 is:
6.8µF
AD + 1 ) 2
0.1µF
RG
0.1µF 113Ω
906Ω
1/2
OPA2614
VO
100Ω
VS /2
RF
453Ω
VI
RM
89Ω
Figure 4. AC-Coupled, G = −4, Single-Supply,
Specification and Test Circuit
DIFFERENTIAL INTERFACE APPLICATIONS
Dual op amps are particularly suitable to differential input
to differential output applications. Typically, these fall into
either Analog-to-Digital Converter (ADC) input interface or
line driver applications. Two basic approaches to
differential I/O are noninverting or inverting configurations.
Since the output is differential, the signal polarity is
somewhat meaningless—the noninverting and inverting
terminology applies here to where the input is brought into
RF
RG
(1)
Since the OPA2614 is a voltage-feedback (VFB) amplifier,
its bandwidth is principally controlled by the noise gain.
The equivalent noise gain for Figure 5 is:
1)2
453W + 4VńV
301W
(2)
Various combinations of single-supply or AC-coupled gain
can also be delivered using the basic circuit of Figure 5.
Common-mode bias voltages on the two noninverting
inputs pass on to the output with a gain of 1 since an equal
DC voltage at each inverting node creates no current
through RG. This circuit does show a common-mode gain
of 1 from input to output. The source connection should
either remove this common-mode signal if undesired
(using an input transformer can provide this function), or
the common-mode voltage at the inputs can be used to set
the output common-mode bias. If the low common-mode
rejection of this circuit is a problem, the output interface
may also be used to reject that common-mode. For
instance, most modern differential input ADCs reject
common-mode signals very well, while a line driver
application through a transformer will also remove the
common-mode signal through to the line.
17
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The OPA2614 is a de-compensated version of the unity
gain stable OPA2613. This decompensation gives a flat
response at a gain of +4, higher gain bandwidth product,
and twice the slew rate of the OPA2613. The OPA2614
should not be used for integrator-based active filters as
unity gain stability is required for the correct operation of
that filter type. It can be used for Sallen-Key type filters
where the filter is implemented using a simple gain
stage—as long as that gain is ≥ 2 when using the
OPA2614.
The higher slew rate of the OPA2614 (145V/µs vs 70V/µs
for the OPA2613) will give a higher full-power bandwidth
and lower distortion to higher output swings. For example,
comparing the ±6V differential plots for the OPA2613 to
those of the OPA2614, we see about twice the large signal
bandwidth for the OPA2614. This is also operating at twice
the signal gain, but since the gain bandwidth for the
OPA2614 is approximately twice that of the OPA2613, this
is as expected.
The increased slew rate of the OPA2614 over the
OPA2613 will also give lower distortion at higher output
swings and/or frequency. Figure 8 and Figure 9 show the
differential test data for the OPA2613 and OPA2614,
respectively.
DIFFERENTIAL DISTORTION
vs OUTPUT VOLTAGE
−70
GD = 4
RL = 70Ω
f = 1MHz
−75
Harmonic Distortion (dBc)
OPA2614 vs OPA2613 PERFORMANCE
−80
3rd−Harmonic
−85
−90
−95
2nd−Harmonic
−100
−105
0.1
1
DIFFERENTIAL LARGE−SIGNAL
FREQUENCY RESPONSE
10
20
Output Voltage Swing (VPP)
15
RL = 70Ω
GD = +4
Figure 8. OPA2613 Differential Gain of +4
Distortion vs Output at 1MHz
VO = 0.2VPP
VO = 1VPP
VO = 2VPP
VO = 5VPP
0
1
10
100
Frequency (MHz)
Figure 6. OPA2613 Differential Gain of +4
Large-Signal Bandwidth
21
−85
−90
2nd−Harmonic
−95
0.1
1
10
20
Output Voltage Swing (VPP)
RL = 70Ω
GD = +8
18
3rd−Harmonic
GD = +8V/V
RL = 70Ω
f = 1MHz
−100
DIFFERENTIAL LARGE−SIGNAL
FREQUENCY RESPONSE
VO = 0.5VPP
Figure 9. OPA2614 Differential Gain of +8
Distortion vs Output at 1MHz
15
Gain (dB)
−80
6
3
VO = 2VPP
12
Notice how much lower the 3rd-harmonic is above 10VPP
for the OPA2614 vs the OPA2613. These test conditions
were set up to have the same loop gain so the difference
in high output 3rd-harmonics can be attributed principally
to the high slew rate for the OPA2614.
VO = 5VPP
9
6
3
1
10
100
Frequency (MHz)
Figure 7. OPA2614 Differential Gain of +8
Large-Signal Bandwidth
18
DIFFERENTIAL DISTORTION
vs OUTPUT VOLTAGE
9
Harmonic Distortion (dBc)
Gain (dB)
12
These differences show that the OPA2614 would be
preferred for higher gains, higher frequency applications
over the OPA2613 while the OPA2613 would be preferred
where unity gain stability is required in the application.
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SBOS305D − JUNE 2004 − REVISED AUGUST 2008
SINGLE-SUPPLY ADSL UPSTREAM DRIVER
Figure 10 shows an example of a single-supply ADSL
upstream driver. The dual OPA2614 is configured as a
differential gain stage to provide signal drive to the primary
of the transformer (here, a step-up transformer with a turns
ratio of 1:2). The main advantage of this configuration is
the cancellation of all even harmonic distortion products.
Another important advantage for ADSL applications is that
each amplifier needs only to swing half of the total output
required driving the load.
The two back-termination resistors (12.5Ω each) added at
each input of the transformer make the impedance of the
modem match the impedance of the phone line, and also
provide a means of detecting the received signal for the
receiver. The value of these resistors (RM) is a function of
the line impedance and the transformer turns ratio (n),
given by the following equation:
Z LINE
2n2
RM +
(4)
LINE DRIVER HEADROOM MODEL
+12V
The first step in a transformer-coupled, twisted-pair driver
design is to compute the peak-to-peak output voltage from
the target specifications. This is done using the following
equations:
20Ω
1/2
OPA2614
0.1µF
AFE
2VPP
Max
Assumed
+6.3V
0.1µF
IP = 150mA
RM
12.5Ω
1:2
RF
1kΩ
1kΩ
1kΩ
RG
308Ω
ZLINE
15VPP
100Ω
RM
12.5Ω
V RMS +
Figure 10. Single-Supply ADSL Upstream Driver
The analog front-end (AFE) signal is AC-coupled to the
driver, and the noninverting input of each amplifier is
biased slightly above the mid-supply voltage (+6.3V in this
case). In addition to providing the proper biasing to the
amplifier, this approach also provides a high-pass filtering
with a corner frequency, set here at 1.6kHz. As the
upstream signal bandwidth starts at 26kHz, this high-pass
filter does not generate any problems and has the
advantage of filtering out unwanted lower frequencies.
The input signal is amplified with a gain set by the following
equation:
2
(5)
RL
PL
10 10
V RMS + CF
(6)
V RMS (7)
with VP equal to the peak voltage at the load and CF as the
Crest Factor.
V LPP + 2
GD + 1 )
VRMS
(1mW) RL
Ǹ(1mW)
V P + Crest Factor
IP = 150mA
1/2
OPA2614
log
With PL power and VRMS voltage at the load, and RL line
impedance, this gives the following:
1µF
RF
1kΩ
20Ω
2
P L + 10
RF
RG
(3)
With RF = 1kΩ and RG = 308Ω, the gain for this differential
amplifier is 7.5. This gain boosts the AFE signal, assumed
to be a maximum of 2VPP, to a maximum of 15VPP.
CF
VRMS
(8)
with VLPP as the peak-to-peak voltage at the load.
Consolidating Equations 4 through 7 allows expressing
the required peak-to-peak voltage at the load as a function
of the crest factor, the load impedance, and the power at
the load. Thus:
V LPP + 2
CF
Ǹ(1mW)
RL
PL
10 10 (9)
This VLPP is usually computed for a nominal line
impedance and may be taken as a fixed design target.
The next step for the driver is to compute the individual
amplifier output voltage and currents as a function of VPP
on the line and transformer turns ratio. As the turns ratio
changes, the minimum allowed supply voltage changes
along with it. The peak current (IP) in the amplifier output
is given by:
"I P + 1
2
2
V LPP
n
1
4R M
(10)
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With VLPP as defined in Equation 8, and RM as defined in
Equation 4 and shown in Figure 11.
RM
Vpp =
2VLpp
n
Table 1. Line Driver Headroom Model Values
V1
R1
V2
R2
+5V
1.0V
2Ω
1.0V
5.5Ω
+12V
1.0V
2Ω
1.0V
5.5Ω
1:n
VLpp
n
RL
VLpp
RM
Figure 11. Driver Peak Output Voltage
With the previous information available, it is now possible
to select a supply voltage and the turns ratio desired for the
transformer as well as calculate the headroom for the
OPA2614.
TOTAL DRIVER POWER FOR xDSL
APPLICATIONS
The total internal power dissipation for the OPA2614 in an
xDSL line driver application will be the sum of the
quiescent power and the output stage power. The
OPA2614 holds a relatively constant quiescent current
versus supply voltage—giving a power contribution that is
simply the quiescent current times the supply voltage used
(the supply voltage will be greater than the solution given
in Equation 12). The total output stage power may be
computed with reference to Figure 13.
The model (shown in Figure 12) can be described with the
following set of equations:
1.
IAVG =
IP
CF
First, as available output swing:
V PP + VCC * (V1 ) V2) * I P
2.
+VCC
(R 1 ) R 2) (11)
RT
Or as required supply voltage:
V CC + VPP ) (V1 ) V2) ) I P
(R 1 ) R 2) (12)
The minimum supply voltage for a power and load
requirement is given by Equation 11.
Figure 13. Output Stage Power Model
+VCC
R1
V1
VO
IP
V2
R2
The two output stages used to drive the load of Figure 11
can be seen as an H-Bridge in Figure 13. The average
current drawn from the supply into this H-Bridge and load
will be the peak current in the load given by Equation 10
divided by the crest factor (CF) for the xDSL modulation.
This total power from the supply is then reduced by the
power in RT to leave the power dissipated internal to the
drivers in the four output stage transistors. That power is
simply the target line power used in Equation 5 plus the
power lost in the matching elements (RM). In the examples
here, a perfect match is targeted, giving the same power
in the matching elements as in the load. The output stage
power is then set by Equation 13.
P OUT +
Figure 12. Line Driver Headroom Model
V1, V2, R1, and R2 are given in Table 1 for both +12V and
+5V operation.
20
IP
CF
V CC * 2P L
(13)
The total amplifier power is then:
P TOT + I q
VCC )
IP
CF
V CC * 2P L
(14)
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For the ADSL CPE upstream driver design of Figure 10,
the peak current is 150mA for a signal that requires a crest
factor of 5.33 with a target line power of 13dBm into 100Ω
(20mW). With a typical quiescent current of 12mA and a
nominal supply voltage of +12V, the total internal power
dissipation for the solution of Figure 10 will be:
OPA2614. Figure 14 shows a typical inverting
configuration where the I/O impedances and signal gain
from Figure 1 are retained in an inverting circuit
configuration.
PTOT + 12mA(12V) ) 150mA (12V) * 2(20mW) + 400mW
5.33
+6V
(15)
Power−supply
decoupling not
shown.
DESIGN-IN TOOLS
DEMONSTRATION FIXTURE
0.01µF
A printed circuit board (PCB) is available to assist in the
initial evaluation of circuit performance using the
OPA2614. The fixture is offered free of charge as an
unpopulated PCB, delivered with user’s guide. The
summary information for this fixture is shown in Table 2.
Table 2. Demonstration Fixture
PRODUCT
PACKAGE
ORDERING
NUMBER
LITERATURE
NUMBER
OPA2614ID
SO-8
DEM-OPA-SO−2A
SBOU003
The demonstration fixture can be requested at the Texas
Instruments web site (www.ti.com) through the OPA2614
product folder.
MACROMODELS AND APPLICATIONS
SUPPORT
Computer simulation of circuit performance using SPICE
is often useful when analyzing the performance of analog
circuits and systems. This is particularly true for video and
RF amplifier circuits where parasitic capacitance and
inductance can have a major effect on circuit performance.
A SPICE model for the OPA2614 is available through the
TI web site (www.ti.com). This model does a good job of
predicting small-signal AC and transient performance
under a wide variety of operating conditions, but does not
do as well in predicting the harmonic distortion or video
dG/dP characteristics. This model does not attempt to
distinguish between the package types in small-signal AC
performance, nor does it attempt to simulate channel-tochannel coupling.
INVERTING AMPLIFIER OPERATION
As the OPA2614 is a general-purpose, wideband
voltage-feedback op amp, most of the familiar op amp
application circuits are available to the designer.
Wideband inverting operation is particularly suited to the
50Ω
Source
110Ω
1/2
OPA2614
RG
113Ω
VO
50Ω Load
50Ω
RF
453Ω
VI
VO
RM
89Ω
VI
=−
RF
RG
= −4
− 6V
Figure 14. Inverting Gain of −4 with Impedance
Matching
In the inverting configuration, two key design
considerations must be noted. The first is that the gain
resistor (RG) becomes part of the input impedance. If input
impedance matching is desired (which is beneficial
whenever the signal is coupled through a cable, twistedpair, long PCB trace, or other transmission line conductor),
it is normally necessary to add an additional matching
resistor to ground. RG, by itself, is not normally set to the
required input impedance since its value, along with the
desired gain, will determine an RF, which may be
non-optimal from a frequency response standpoint. The
total input impedance for the source becomes the parallel
combination of RG and RM.
The second major consideration, touched on in the
previous paragraph, is that the signal source impedance
becomes part of the noise gain equation and has an effect
on the bandwidth. In the example of Figure 14, the RM
value combines in parallel with the external 50Ω source
impedance, yielding an effective driving impedance of
50Ω || 89Ω = 32Ω. This impedance is added in series with
RG for calculating the noise gainwhich gives NG = 4.12.
Note that the noninverting input in this bipolar supply
inverting application is connected to ground through a
110Ω resistor. It is often suggested that an additional
resistor be connected to ground on the noninverting input
to achieve bias current error cancellation at the output.
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OUTPUT CURRENT AND VOLTAGE
The OPA2614 provides output voltage and current
capabilities that are unsurpassed in a low-cost dual
monolithic op amp. Under no-load conditions at +25°C, the
output voltage typically swings closer than 1V to either
supply rail; tested at +25°C, swing limit is within 1.1V of
either rail. Into a 12Ω load (the minimum tested load), it
delivers more than ±280mA continuous output current.
The specifications described previously, though familiar in
the industry, consider voltage and current limits separately.
In many applications, it is the voltage times current (or V-I
product) that is more relevant to circuit operation. Refer to
the Output Voltage and Current Limitations plot in the
Typical Characteristics. The X and Y axes of this graph
show the zero-voltage output current limit and the
zero-current output voltage limit, respectively. The four
quadrants give a more detailed view of the OPA2614
output drive capabilities, noting that the graph is bounded
by a safe operating area of 1W maximum internal power
dissipation (in this case, for one channel only).
Superimposing resistor load lines onto the plot shows that
the OPA2614 can drive +4.8 and −4.1 into 25Ω without
exceeding the output capabilities or the 1W dissipation
limit. A 100Ω load line (the standard test circuit load)
shows the full ±4.9V output swing capability, as shown in
the Electrical Characteristics tables. The minimum
specified output voltage and current over temperature are
set by worst-case simulations at the cold temperature
extreme. Only at cold startup will the output current and
voltage decrease to the numbers shown in the Electrical
Characteristics tables. As the output transistors deliver
power, the junction temperatures increase, decreasing the
VBEs (increasing the available output voltage swing), and
increasing the current gains (increasing the available
output current). In steady-state operation, the available
output voltage and current will always be greater than that
shown in the over-temperature specifications, since the
output stage junction temperatures will be higher than the
minimum specified operating ambient.
DRIVING CAPACITIVE LOADS
One of the most demanding and yet very common load
conditions for an op amp is capacitive loading. Often, the
capacitive load is the input of an ADCincluding
additional external capacitance that may be recommended to improve the ADC linearity. A high-speed, high
open-loop gain amplifier like the OPA2614 can be very
susceptible to decreased stability and closed-loop
22
response peaking when a capacitive load is placed directly
on the output pin. When the amplifier open-loop output
resistance is considered, this capacitive load introduces
an additional pole in the signal path that can decrease the
phase margin. Several external solutions to this problem
have been suggested.
When the primary considerations are frequency response
flatness, pulse response fidelity, and/or distortion, the
simplest and most effective solution is to isolate the
capacitive load from the feedback loop by inserting a
series isolation resistor between the amplifier output and
the capacitive load. This does not eliminate the pole from
the loop response, but rather shifts it and adds a zero at a
higher frequency. The additional zero acts to cancel the
phase lag from the capacitive load pole, thus increasing
the phase margin and improving stability. The Typical
Characteristics show the Recommended RS vs Capacitive
Load and the resulting frequency response at the load.
Parasitic capacitive loads greater than 2pF can begin to
degrade the performance of the OPA2614. Long PCB
traces, unmatched cables, and connections to multiple
devices can easily cause this value to be exceeded.
Always consider this effect carefully, and add the
recommended series resistor as close as possible to the
OPA2614 output pin (see the Board Layout Guidelines
section).
The very high output current and low gain stability for the
OPA2614 can be used to drive large capacitive loads with
moderate slew rates. An example is shown in Figure 15,
where a 2000pF load cap is driven with a 2MHz square
wave to give a ±5V swing. The supplies were slightly
increased to give more headroom for the charging current
through the 2Ω isolation resistor.
+6.2V
VI
±1.25V
2MHz Square
Wave
Input
1/2
OPA2614
Supply decoupling
not shown.
2Ω
VO
2000pF
453Ω
− 6.2V
113Ω
Figure 15. Large Capacitive Load Driver
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Figure 16 shows a comparison of 4 × Input voltage to the
capacitor voltage. The transition time is set by the 145V/µs
slew rate for the OPA2614. For this controlled dV/dT, the
charging current into the 2000pF load will be given by:
Slew Rate = IP/C
Solving for IP gives:
I P + 2000pF
145Vńms + 290mA peak current (16)
Input and Output Voltage
LARGE−SIGNAL CAPACITIVE LOAD DRIVE
6
5
4
3
2
1
0
−1
−2
−3
−4
−5
−6
Capacitor Voltage
4X Input Voltage
NOISE PERFORMANCE
Wideband voltage-feedback op amps generally have a
lower output noise than comparable current-feedback op
amps. The OPA2614 offers an excellent balance between
voltage and current noise terms to achieve low output
noise. The input voltage noise (1.8nV/√Hz) is lower than
most low-gain stable, wideband voltage-feedback op
amps. The op amp input voltage noise and the two input
current noise terms combine to give low output noise
under a wide variety of operating conditions. Figure 17
shows the op amp noise analysis model with all the noise
terms included. In this model, all noise terms are taken to
be noise voltage or current density terms in either nV/√Hz
or pA/√Hz.
145V/µs Slew Rate
ENI
1/2
OPA2614
RS
Time (50ns/div)
ERS
Figure 16. Large-Signal Capacitive Load Drive
At these larger capacitive loads, very low series R will
maintain stabilitybut some R is always required.
The OPA2614 provides good distortion performance into
a 100Ω load on ±6V supplies. Generally, until the
fundamental signal reaches high frequency or power
levels, the 2nd-harmonic dominates the distortion with a
negligible 3rd-harmonic component. Focusing then on the
2nd-harmonic, increasing the load impedance improves
distortion directly. Remember that the total load includes
the feedback networkin the noninverting configuration
(see Figure 1), this is the sum of RF + RG, whereas in the
inverting configuration, it is just RF. Also, providing an
additional supply decoupling capacitor (0.01µF) between
the supply pins (for bipolar operation) improves the
2nd-order distortion slightly (3dB to 6dB).
In most op amps, increasing the output voltage swing
increases harmonic distortion directly. The Typical
Characteristics show the 2nd-harmonic increasing at a
little less than the expected 2x rate whereas the
3rd-harmonic increases at a little less than the expected 3x
rate. Where the test power doubles, the difference
between it and the 2nd-harmonic decreases less than the
expected 6dB, whereas the difference between it and the
3rd-harmonic decreases by less than the expected 12dB.
Operating differentially will suppress the 2nd-order
harmonics below the 3rd.
Operating as a differential I/O stage will also suppress the
2nd-harmonic distortion.
RF
√4kTRS
√4kTRF
IBI
RG
4kT
RG
DISTORTION PERFORMANCE
EO
IBN
4kT = 1.6E −20J
at 290_K
Figure 17. Op Amp Noise Analysis Model
The total output spot noise voltage can be computed as the
square root of the sum of all squared output noise voltage
contributors. Equation 17 shows the general form for the
output noise voltage using the terms given in Figure 17.
EO +
Ǹǒ
E NI ) ǒI BN
RSǓ ) 4kTRS
2
2
Ǔ
NG 2 ) ǒI BI
2
RFǓ ) 4kTRFNG
(17)
Dividing this expression by the noise gain (NG = (1 + RF/RG))
gives the equivalent input-referred spot noise voltage at the
noninverting input, as shown in Equation 18.
EN +
Ǹ
ǒ
2
E NI ) I BN
R
Ǔ
S
2
) 4kTR )
S
ǒ
I BI
Ǔ
RF
NG
2
)
4kTR F
NG
(18)
Evaluating these two equations for the OPA2614 circuit
and component values (see Figure 1) gives a total output
spot noise voltage of 6.34nV/√Hz and a total equivalent
input spot noise voltage of 3.2nV/√Hz. This total input
referred spot noise voltage is higher than the 1.8nV/√Hz
specification for the op amp voltage noise alone. This
reflects the noise added to the output by the inverting
current noise times the feedback resistor.
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DIFFERENTIAL NOISE PERFORMANCE
Because the OPA2614 is used as a differential driver in
xDSL applications, it is important to analyze the noise in
such a configuration. Figure 18 shows the op amp noise
model for the differential configuration.
IN
Evaluating these equations for the OPA2614 ADSL circuit
and component values of Figure 10 gives a total output
spot noise voltage of 23.3nV/√Hz and a total equivalent
input spot noise voltage of 3.2nV/√Hz.
In order to minimize the output noise due to the
noninverting input bias current noise, it is recommended to
keep the noninverting source impedance as low as
possible.
DC ACCURACY AND OFFSET CONTROL
Driver
EN
RS
IN
ERS
√4kTRF
RF
√4kTRS
RG
EO2
√4kTRG
√4kTRF
RF
IN
EN
RS
IN
ERS
√4kTRS
Figure 18. Differential Op Amp Noise Analysis
Model
The OPA2614 can provide excellent DC signal accuracy
due to its high open-loop gain, high common-mode
rejection, high power-supply rejection, and low input offset
voltage and bias current offset errors. To take full
advantage of the low input offset voltage (±1.0mV
maximum at 25°C), careful attention to input bias current
cancellation is also required. The high-speed input stage
for the OPA2614 has relatively high input bias current (6µA
typical into the pins) but with a very close match between
the two input currents, typically 50nA input offset current.
The total output offset voltage may be reduced
considerably by matching the source impedances looking
out of the two inputs. For example, one way to add bias
current cancellation to the circuit of Figure 1 would be to
insert a 88Ω series resistor into the noninverting input from
the 50Ω terminating resistor. If the 50Ω source resistor is
DC-coupled, this will increase the source impedance for
the noninverting input bias current to 113Ω. Since this is
now equal to the impedance looking out of the inverting
input (RF || RG), the circuit will cancel the bias current
effects, leaving only the offset current times the feedback
resistor as a residual DC error term at the output.
Evaluating the configuration of Figure 1 adding a 88Ω in
series with the noninverting input pin, using worst-case
+25°C input offset voltage and the two input bias currents,
gives a worst-case output offset range equal to:
VOFF = ± (NG × VOS(MAX)) ± (IOS × RF)
As a reminder, the differential gain is expressed as:
GD + 1 )
2
where NG = noninverting signal gain
= ± (4 × 1.0mV) ± (453Ω × 300nA)
RF
RG
(19)
VOFF = ±4.14mV
The output noise can be expressed as shown below:
(20)
e
O
+
Ǹ
2
G
2
D
ǒ
ǒ
e 2) i
N
N
R
S
Ǔ
2
) 4kTR
S
Ǔ
2
) 2ǒi R Ǔ ) 2ǒ4kTR G Ǔ
I F
F D
Dividing this expression by the differential noise gain
(G D = (1 + 2R F/R G)) gives the equivalent input-referred
spot noise voltage at the noninverting input, as shown in
Equation 21.
ei +
24
Ǹ
2
(21)
ǒ
eN 2 ) ǒi N
Ǔ ǒ Ǔ ǒ
R SǓ ) 4kTR S ) 2
2
i IR F
GD
2
)2
= ±4.0mV ± 0.14mV
Ǔ
4kTR F
GD
THERMAL ANALYSIS
Due to the high output power capability of the OPA2614,
heat-sinking or forced airflow may be required under
extreme operating conditions. Maximum desired junction
temperature sets the maximum allowed internal power
dissipation as described below. In no case should the
maximum junction temperature be allowed to exceed
150°C. Operating junction temperature (TJ) is given by
TA + PD × qJA. The total internal power dissipation (PD) is
the sum of quiescent power (PDQ) and additional power
dissipation in the output stage (PDL) to deliver load power.
Quiescent power is the specified no-load supply current
times the total supply voltage across the part. PDL depends
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on the required output signal and load, but for a grounded
resistive load, PDL is at a maximum when the output is fixed
at a voltage equal to 1/2 of either supply voltage (for equal
bipolar supplies). Under this condition, PDL = VS2/(4 × RL)
where RL includes feedback network loading. Note that it
is the power in the output stage and not into the load that
determines internal power dissipation. As a worst-case
example, compute the maximum TJ using an OPA2614
SO-8 in the circuit of Figure 1 operating at the maximum
specified ambient temperature of +85°C with both outputs
driving a grounded 20Ω load to +3.0V.
PD = 12V × 13.0mA + 2 × [62/ (4 × (20Ω 804Ω))] = 1. 08W
Maximum TJ = +85°C + (1.08W × 125°C/W) = 220°C
This absolute worst-case condition exceeds specified
maximum junction temperature. This extreme case is not
normally encountered. Where high internal power dissipation is anticipated, consider the thermal slug package
version. Under the same worst-case conditions the
junction temperature will drop to 139°C with the 50°C/W
thermal impedance available using the PSO-8 package.
BOARD LAYOUT GUIDELINES
Achieving optimum performance with a high-frequency
amplifier like the OPA2614 requires careful attention to
board layout parasitic and external component types.
Recommendations that optimize performance include:
a) Minimize parasitic capacitance to any AC ground for
all of the signal I/O pins. Parasitic capacitance on the
output and inverting input pins can cause instability; on the
noninverting input, it can react with the source impedance
to cause unintentional band limiting. To reduce unwanted
capacitance, a window around the signal I/O pins should
be opened in all of the ground and power planes around
those pins. Otherwise, ground and power planes should
be unbroken elsewhere on the board.
b) Minimize the distance (< 0.25″) from the power-supply
pins to high-frequency 0.1µF decoupling capacitors. At the
device pins, the ground and power plane layout should not
be in close proximity to the signal I/O pins. Avoid narrow
power and ground traces to minimize inductance between
the pins and the decoupling capacitors. The power-supply
connections (on pins 4 and 7) should always be decoupled
with these capacitors. An optional supply decoupling
capacitor across the two power supplies (for bipolar
operation) improves 2nd-harmonic distortion performance.
Larger (2.2µF to 6.8µF) decoupling capacitors, effective at
a lower frequency, should also be used on the main supply
pins. These can be placed somewhat farther from the
device and may be shared among several devices in the
same area of the PCB.
c) Careful selection and placement of external
components preserve the high-frequency performance
of the OPA2614. Resistors should be of a very low
reactance type. Surface-mount resistors work best and
allow a tighter overall layout. Metal film and carbon
composition axially leaded resistors can also provide good
high-frequency performance. Again, keep the leads and
PCB trace length as short as possible. Never use
wire-wound type resistors in a high-frequency application.
Although the output pin and inverting input pin are the most
sensitive to parasitic capacitance, always position the
feedback and series output resistor, if any, as close as
possible to the output pin. Other network components,
such as noninverting input termination resistors, should
also be placed close to the package. Where double-side
component mounting is allowed, place the feedback
resistor directly under the package on the other side of the
board between the output and inverting input pins. The
453Ω feedback resistor used in the Typical Characteristics
at a gain of +4 on ±6V supplies is a good starting point for
design.
d) Connections to other wideband devices on the board
may be made with short direct traces or through onboard
transmission lines. For short connections, consider the
trace and the input to the next device as a lumped
capacitive load. Relatively wide traces (50mils to 100mils)
should be used, preferably with ground and power planes
opened up around them. Estimate the total capacitive load
and set RS from the plot of Recommended RS vs
Capacitive Load. Low parasitic capacitive loads (< 5pF)
may not need an RS because the OPA2614 is nominally
compensated to operate with a 2pF parasitic load. If a long
trace is required, and the 6dB signal loss intrinsic to a
doubly-terminated transmission line is acceptable,
implement a matched impedance transmission line using
microstrip or stripline techniques (consult an ECL design
handbook for microstrip and stripline layout techniques). A
50Ω environment is normally not necessary on board; in
fact, a higher impedance environment improves distortion
(see the distortion versus load plots). With a characteristic
board trace impedance defined based on board material
and trace dimensions, a matching series resistor into the
trace from the output of the OPA2614 is used, as well as
a terminating shunt resistor at the input of the destination
device. Remember also that the terminating impedance is
the parallel combination of the shunt resistor and the input
impedance of the destination device.
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SBOS305D − JUNE 2004 − REVISED AUGUST 2008
This total effective impedance should be set to match the
trace impedance. The high output voltage and current
capability of the OPA2614 allows multiple destination
devices to be handled as separate transmission lines,
each with their own series and shunt terminations. If the
6dB attenuation of a doubly-terminated transmission line
is unacceptable, a long trace can be series-terminated at
the source end only. Treat the trace as a capacitive load in
this case and set the series resistor value as shown in the
plot of RS vs Capacitive Load. However, this does not
preserve signal integrity as well as a doubly-terminated
line. If the input impedance of the destination device is low,
there is some signal attenuation due to the voltage divider
formed by the series output into the terminating
impedance.
e) Socketing a high-speed part like the OPA2614 is not
recommended. The additional lead length and pin-to-pin
capacitance introduced by the socket can create an
extremely troublesome parasitic network, which can make
it almost impossible to achieve a smooth, stable frequency
response. Best results are obtained by soldering the
OPA2614 onto the board.
INPUT AND ESD PROTECTION
The OPA2614 is built using a high-speed complementary
bipolar process. The internal junction breakdown voltages
are relatively low for these very small geometry devices
and are reflected in the absolute maximum ratings table.
All device pins have limited ESD protection using internal
diodes to the power supplies, as shown in Figure 19.
These diodes provide moderate protection to input
overdrive voltages above the supplies as well. The
protection diodes can typically support 30mA continuous
current. Where higher currents are possible (for example,
in systems with ±15V supply parts driving into the
OPA2614), current-limiting series resistors should be
added into the two inputs. Keep these resistor values as
low as possible, because high values degrade both noise
performance and frequency response.
+VCC
External
Pin
Internal
Circuitry
−VCC
Figure 19. Internal ESD Protection
26
www.ti.com
SBOS305D − JUNE 2004 − REVISED AUGUST 2008
Revision History
DATE
REV
PAGE
SECTION
8/08
D
2
Absolute Maximum Ratings
9/07
C
DESCRIPTION
Changed Storage Temperature minimum value from −40°C to −65°C.
Deleted all references to PSO−8 (OPA2614H) package.
1
Description
Deleted last paragraph.
25
Board Layout Guidelines
Deleted paragraph (F).
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
27
PACKAGE OPTION ADDENDUM
www.ti.com
13-Aug-2021
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
OPA2614ID
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
OPA
2614
OPA2614IDG4
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
OPA
2614
OPA2614IDR
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
OPA
2614
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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