OPA
OPA692
69 2
OPA6
92
SBOS236E – MARCH 2002 – REVISED DECEMBER 2008
www.ti.com
Wideband, Fixed Gain
Video BUFFER AMPLIFIER With Disable
FEATURES
DESCRIPTION
● FLEXIBLE SUPPLY RANGE:
+5V to +12V Single Supply
±2.5V to ±6V Dual Supplies
● INTERNALLY FIXED GAIN: +2 or ±1
● HIGH BANDWIDTH (G = +2): 225MHz
● LOW SUPPLY CURRENT: 5.1mA
● LOW DISABLED CURRENT: 150µA
● HIGH OUTPUT CURRENT: 190mA
● OUTPUT VOLTAGE SWING: ±4.0V
● SOT23-6 AVAILABLE
The OPA692 provides an easy to use, broadband fixed gain
video buffer amplifier. Depending on the external connections, the internal resistor network may be used to provide
either a fixed gain of +2 video buffer or a gain of +1 or –1
voltage buffer. Operating on a very low 5.1mA supply current, the OPA692 offers a slew rate and output power
normally associated with a much higher supply current. A
new output stage architecture delivers high output current
with minimal headroom and crossover distortion. This gives
exceptional single-supply operation. Using a single +5V
supply, the OPA692 can deliver a 1V to 4V output swing with
over 120mA drive current and > 200MHz bandwidth. This
combination of features makes the OPA692 an ideal RGB
line driver or single-supply Analog-to-Digital Converter (ADC)
input driver.
APPLICATIONS
●
●
●
●
●
BROADBAND VIDEO LINE DRIVERS
MULTIPLE LINE VIDEO DA
PORTABLE INSTRUMENTS
ADC BUFFERS
ACTIVE FILTERS
The low 5.1mA supply current for the OPA692 is precisely
trimmed at +25°C. This trim, along with low drift over temperature, ensures a lower maximum supply current than
competing products that report only a room temperature
nominal supply current. System power may be further reduced by using the optional disable control pin. Leaving this
disable pin open, or holding it HIGH, gives normal operation.
If pulled LOW, the OPA692 supply current drops to less than
150µA while the I/O pins go into a high-impedance state.
OPA692 RELATED PRODUCTS
Voltage-Feedback
SINGLES
DUALS
TRIPLES
OPA690
OPA2690
OPA3690
Current-Feedback
OPA691
OPA2691
OPA3691
Fixed Gain
OPA682
OPA2682
OPA3692
75Ω
Video
Out
OPA692
RG-59
75Ω
1
8
2
7
3
6
4
5
DIS
75Ω
Video
In
+5V
RG-59
75Ω
75Ω
75Ω
–5V
RG-59
75Ω
SO-8
G = +2
75Ω
RG-59
75Ω
225MHz, 4-Output Component Video DA
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
All trademarks are the property of their respective owners.
Copyright © 2002-2008, Texas Instruments Incorporated
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of Texas Instruments
standard warranty. Production processing does not necessarily include
testing of all parameters.
www.ti.com
ELECTROSTATIC
DISCHARGE SENSITIVITY
ABSOLUTE MAXIMUM RATINGS(1)
Power Supply ............................................................................... ±6.5VDC
Internal Power Dissipation(2) ............................ See Thermal Information
Differential Input Voltage(3) ............................................................... ±1.2V
Input Voltage Range ............................................................................ ±VS
Storage Temperature Range: D, DVB ........................... –65°C to +125°C
Lead Temperature (soldering, 10s) .............................................. +300°C
Junction Temperature (TJ ) ........................................................... +175°C
ESD Resistance: HBM ........................................................................ 2kV
MM ........................................................................ 200V
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling
and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits
may be more susceptible to damage because very small
parametric changes could cause the device not to meet its
published specifications.
NOTES: (1) Stresses above these ratings may cause permanent damage.
Exposure to absolute maximum conditions for extended periods may degrade
device reliability. These are stress ratings only, and functional operation of the
device at these or any other conditions beyond those specified is not implied.
(2) Packages must be derated based on specified θJA. Maximum TJ must be
observed. (3) Noninverting input to internal inverting node.
PACKAGE/ORDERING INFORMATION(1)
PRODUCT
PACKAGE-LEAD
PACKAGE
DESIGNATOR
OPA692ID
SPECIFIED
TEMPERATURE
RANGE
PACKAGE
MARKING
ORDERING
NUMBER
TRANSPORT
MEDIA, QUANTITY
OPA692ID
OPA692IDR
OPA692IDBVT
OPA692IDBVR
Rails, 100
Tape and Reel, 2500
Tape and Reel, 250
Tape and Reel, 3000
SO-8 Surface-Mount
D
–40°C to +85°C
OPA692
"
"
"
"
"
OPA692IDBV
SOT23-6
DBV
–40°C to +85°C
OAGI
"
"
"
"
"
NOTE: (1) For the most current package and ordering information, see the Package Option Addendum at the end of this document, or see the TI website at www.ti.com.
PIN CONFIGURATION
Top View
SO
Top View
SOT
Output
1
6
+VS
5
DIS
4
–IN
RF
402Ω
NC
1
RG
402Ω
RF
402Ω
8
–VS
2
+IN
3
DIS
–IN
2
7
+VS
+IN
3
6
Output
6
–VS
4
5
RG
402Ω
5
4
NC
OAGI
NC: No Connection
1
2
3
Pin Orientation/Package Marking
2
OPA692
www.ti.com
SBOS236E
ELECTRICAL CHARACTERISTICS: VS = ±5V
Boldface limits are tested at +25°C.
G = +2 (–IN grounded) and RL = 100Ω (see Figure 1 for AC performance only), unless otherwise noted.
OPA692ID, IDBV
TYP
PARAMETER
CONDITIONS
+25°C
MIN/MAX OVER TEMPERATURE
+25°C(1)
0°C to
70°C
–40°C to
+85°C
UNITS
MHz
typ
185
180
170
MHz
min
B
MHz
typ
C
MIN/ TEST
MAX LEVEL(2 )
AC PERFORMANCE (see Figure 1)
Small-Signal Bandwidth (VO < 0.5VPP)
Bandwidth for 0.1dB Gain Flatness
G = +1
280
G = +2
225
G = –1
220
C
G = +2, VO < 0.5VPP
120
40
35
30
MHz
min
B
Peaking at a Gain of +1
VO < 0.5VPP
0.2
1
1.5
2
dB
max
B
Large-Signal Bandwidth
G = +2, VO = 5VPP
220
MHz
typ
C
G = +2, 4V Step
2000
1400
1375
1350
V/µs
min
B
G = +2, VO = 0.5V Step
1.6
ns
typ
C
G = +2, VO = 5V Step
1.9
ns
typ
C
G = +2, VO = 2V Step
12
ns
typ
C
G = +2, VO = 2V Step
8
ns
typ
C
B
Slew Rate
Rise-and-Fall Time
Settling Time to 0.02%
0.1%
Harmonic Distortion
2nd-Harmonic
3rd-Harmonic
G = +2, f = 5MHz, VO = 2VPP
RL = 100Ω
–69
–62
–59
–57
dBc
max
RL ≥ 500Ω
–79
–70
–67
–65
dBc
max
B
RL = 100Ω
–76
–72
–70
–68
dBc
max
B
RL ≥ 500Ω
–94
–87
–82
–78
dBc
max
B
Input Voltage Noise
f > 1MHz
1.7
2.5
2.9
3.1
nV/√Hz
max
B
Noninverting Input Current Noise
f > 1MHz
12
14
15
15
pA/√Hz
max
B
Inverting Input Current Noise
f > 1MHz
15
17
18
19
pA/√Hz
max
B
NTSC, RL = 150Ω
0.07
%
typ
C
NTSC, RL = 37.5Ω
0.17
%
typ
C
NTSC, RL = 150Ω
0.02
deg
typ
C
NTSC, RL = 37.5Ω
0.07
deg
typ
C
G = +1
±0.2
%
typ
C
G = +2
±0.3
±1.5
±1.6
±1.7
%
max
A
G = –1
±0.2
±1.5
±1.6
±1.7
%
max
B
Maximum
402
457
462
464
Ω
max
A
Minimum
402
347
342
340
Ω
min
A
0.13
0.13
0.13
%/C°
max
B
±2.5
±3.2
±3.9
mV
max
A
±12
±20
µV/°C
max
B
Differential Gain
Differential Phase
DC PERFORMANCE(3)
Gain Error
Internal RF and RG
Average Drift
Input Offset Voltage
VCM = 0V
Average Offset Voltage Drift
VCM = 0V
Noninverting Input Bias Current
VCM = 0V
Average Noninverting Input Bias Current Drift
VCM = 0V
Inverting Input Bias Current
VCM = 0V
Average Inverting Input Bias Current Drift
VCM = 0V
±0.5
+15
±5
+35
±25
+43
+45
µA
max
A
–300
–300
nA/°C
max
B
±30
±40
µA
max
A
±90
±200
nA°C
max
B
±3.3
±3.2
INPUT
±3.5
Common-Mode Input Range
Noninverting Input Impedance
±3.4
100 || 2
V
min
B
kΩ || pF
typ
C
OUTPUT
Voltage Output Swing
No Load
±4.0
±3.6
V
min
A
±3.9
±3.8
±3.7
±3.7
100Ω Load
±3.6
±3.3
V
min
A
+190
+160
+140
+100
mA
min
A
–190
–160
–140
–100
mA
min
A
Current Output, Sourcing
Sinking
Short-Circuit Current
Closed-Loop Output Impedance
VO = 0
±250
mA
typ
C
G = +2, f = 100kHz
0.12
Ω
typ
C
NOTES: (1) Junction temperature = ambient temperature for low temperature limit and +25°C specifications. Junction temperature = ambient temperature +10°C
at high temperature limit specifications. (2) Test Levels: (A) 100% tested at +25°C. Over-temperature limits by characterization and simulation. (B) Limits set by
characterization and simulation. (C) Typical value only for information. (3) Current is considered positive out-of-node. VCM is the input common-mode voltage.
OPA692
SBOS236E
www.ti.com
3
ELECTRICAL CHARACTERISTICS: VS = ±5V (Cont.)
Boldface limits are tested at +25°C.
G = +2 (–IN grounded) and RL = 100Ω (see Figure 1 for AC performance only), unless otherwise noted.
OPA692ID, IDBV
TYP
PARAMETER
CONDITIONS
MIN/MAX OVER TEMPERATURE
+25°C
+25°C(1)
0°C to
70°C
–40°C to
+85°C
UNITS
–300
–350
–400
µA
max
A
µs
typ
C
MIN/ TEST
MAX LEVEL(2 )
DISABLE/POWER DOWN (DIS Pin)
Power-Down Supply Current (+VS)
VDIS = 0
–150
VIN = +1VDC
1
Enable Time
VIN = +1VDC
25
ns
typ
C
Off Isolation
G = +2, 5MHz
70
dB
typ
C
4
pF
typ
C
Turn-On Glitch
G = +2, RL = 150Ω
±50
mV
typ
C
Turn-Off Glitch
G = +2, RL= 150Ω
±20
C
Disable Time
Output Capacitance in Disable
mV
typ
Enable Voltage
3.3
3.5
3.6
3.7
V
min
A
Disable Voltage
1.8
1.7
1.6
1.5
V
max
A
75
130
150
160
µA
max
A
V
typ
C
±6
±6
±6
V
max
A
Control Pin Input Bias Current
VDIS = 0
POWER SUPPLY
±5
Specified Operating Voltage
Maximum Operating Voltage Range
Maximum Quiescent Current
VS = ±5V
5.1
5.3
5.5
5.8
mA
max
A
Minimum Quiescent Current
VS = ±5V
5.1
4.9
4.5
4.25
mA
min
A
Input Referred
58
52
50
49
dB
min
A
–40 to +85
°C
typ
C
Power-Supply Rejection Ratio (–PSRR)
TEMPERATURE RANGE
Specification: D, DBV
Thermal Resistance, θJA
D
SO-8
125
°C/W
typ
C
DBV
SOT23-6
150
°C/W
typ
C
NOTES: (1) Junction temperature = ambient temperature for low temperature limit and +25°C specifications. Junction temperature = ambient temperature +10°C
at high temperature limit specifications. (2) Test Levels: (A) 100% tested at +25°C. Over-temperature limits by characterization and simulation. (B) Limits set by
characterization and simulation. (C) Typical value only for information. (3) Current is considered positive out-of-node. VCM is the input common-mode voltage.
4
OPA692
www.ti.com
SBOS236E
ELECTRICAL CHARACTERISTICS: VS = +5V
Boldface limits are tested at +25°C.
G = +2 (–IN grounded though 0.1µF) and RL = 100Ω to VS/2 (see Figure 2 for AC performance only), unless otherwise noted.
OPA692ID, IDBV
TYP
PARAMETER
CONDITIONS
+25°C
MIN/MAX OVER TEMPERATURE
+25°C(1)
0°C to
70°C
–40°C to
+85°C
UNITS
MHz
typ
168
160
140
MHz
min
B
MHz
typ
C
MIN/ TEST
MAX LEVEL(2 )
AC PERFORMANCE (see Figure 2)
Small-Signal Bandwidth (VO < 0.5VPP)
Bandwidth for 0.1dB Gain Flatness
G = +1
240
G = +2
190
G = –1
195
C
G = +2, VO < 0.5VPP
90
40
30
25
MHz
min
B
Peaking at a Gain of +1
VO < 0.5VPP
0.2
1
2.5
3
dB
max
B
Large-Signal Bandwidth
G = +2, VO = 2VPP
210
MHz
typ
C
G = +2, 2V Step
830
600
575
550
V/µs
min
B
G = +2, VO = 0.5V Step
2.0
ns
typ
C
G = +2, VO = 2V Step
2.3
ns
typ
C
G = +2, VO = 2V Step
14
ns
typ
C
G = +2, VO = 2V Step
10
ns
typ
C
Slew Rate
Rise-and-Fall Time
Settling Time to 0.02%
0.1%
Harmonic Distortion
2nd-Harmonic
3rd-Harmonic
G = +2, f = 5MHz, VO = 2VPP
RL = 100Ω to VS /2
–66
–58
–57
–56
dBc
max
B
RL ≥ 500Ω to VS /2
–73
–65
–63
–62
dBc
max
B
RL = 100Ω to VS /2
–72
–68
–67
–65
dBc
max
B
RL ≥ 500Ω to VS /2
–77
–72
–70
–69
dBc
max
B
Input Voltage Noise
f > 1MHz
1.7
2.5
2.9
3.1
nV/√Hz
max
B
Noninverting Input Current Noise
f > 1MHz
12
14
15
15
pA/√Hz
max
B
Inverting Input Current Noise
f > 1MHz
15
17
18
19
pA/√Hz
max
B
G = +1
±0.2
%
typ
C
G = +2
±0.3
±1.5
±1.6
±1.7
%
max
A
G = –1
±0.2
±1.5
±1.6
±1.7
%
max
B
Maximum
402
457
462
464
Ω
max
B
Minimum
402
347
342
340
Ω
min
B
0.13
0.13
0.13
%/C°
max
B
±3
±3.6
±4.3
mV
max
A
±12
±20
µV/°C
max
B
DC PERFORMANCE(3)
Gain Error
Internal RF and RG
Average Drift
Input Offset Voltage
VCM = 2.5V
Average Offset Voltage Drift
VCM = 2.5V
Noninverting Input Bias Current
VCM = 2.5V
Average Noninverting Input Bias Current Drift
VCM = 2.5V
Inverting Input Bias Current
VCM = 2.5V
Average Inverting Input Bias Current Drift
VCM = 2.5V
±0.5
+20
±5
+40
±25
+46
+56
µA
max
A
–250
–250
nA/°C
max
B
±30
±40
µA
max
A
±112
±200
nA°C
max
B
INPUT
Least Positive Input Voltage
1.5
1.6
1.7
1.8
V
max
B
Most Positive Input Voltage
3.5
3.4
3.3
3.2
V
min
B
kΩ || pF
typ
C
Noninverting Input Impedance
100 || 2
OUTPUT
Most Positive Output Voltage
Least Positive Output Voltage
No Load
4.0
3.8
3.7
3.5
V
min
A
RL = 100Ω
3.9
3.7
3.6
3.4
V
min
A
No Load
1.0
1.2
1.3
1.5
V
max
A
RL = 100Ω
1.1
1.3
1.4
1.6
V
max
A
Current Output, Sourcing
+160
+120
+100
+80
mA
min
A
Sinking
–160
–120
–100
–80
mA
min
A
Short-Circuit Current
Output Impedance
VO = VS/2
±250
mA
typ
C
G = +2, f = 100kHz
0.12
Ω
typ
C
NOTES: (1) Junction temperature = ambient temperature for low temperature limit and +25°C specifications. Junction temperature = ambient temperature +10°C
at high temperature limit specifications. (2) Test Levels: (A) 100% tested at +25°C. Over-temperature limits by characterization and simulation. (B) Limits set by
characterization and simulation. (C) Typical value only for information. (3) Current is considered positive out-of-node. VCM is the input common-mode voltage.
OPA692
SBOS236E
www.ti.com
5
ELECTRICAL CHARACTERISTICS: VS = +5V (Cont.)
Boldface limits are tested at +25°C.
G = +2 (–IN grounded though 0.1µF) and RL = 100Ω to VS/2 (see Figure 2 for AC performance only), unless otherwise noted.
OPA692ID, IDBV
TYP
PARAMETER
MIN/MAX OVER TEMPERATURE
CONDITIONS
+25°C(1)
+25°C
0°C to
70°C
–40°C to
+85°C
–300
–350
–400
UNITS
MIN/ TEST
MAX LEVEL(2 )
DISABLE/POWER DOWN (DIS Pin)
Power-Down Supply Current (+VS)
µA
typ
C
dB
typ
C
pF
typ
C
mV
typ
B
VDIS = 0
–150
G = +2, 5MHz
65
4
Turn-On Glitch
G = +2, RL = 150Ω, VIN = 2.5V
±50
Turn-Off Glitch
G = +2, RL = 150Ω, VIN = 2.5V
±20
mV
typ
B
V
min
B
Off Isolation
Output Capacitance in Disable
Enable Voltage
3.3
Disable Voltage
Control Pin Input Bias Current (DIS )
VDIS = 0
3.5
3.6
3.7
1.8
1.7
1.6
1.5
V
max
B
75
130
150
160
µA
typ
C
V
typ
C
12
12
12
V
max
A
POWER SUPPLY
Specified Single-Supply Operating Voltage
5
Maximum Single-Supply Operating Voltage
Maximum Quiescent Current
VS = +5V
4.5
4.8
5.0
5.2
mA
max
A
Minimum Quiescent Current
VS = +5V
4.5
4.1
3.8
3.7
mA
min
A
Input Referred
55
dB
typ
C
–40 to +85
°C
typ
C
125
°C/W
typ
C
150
°C/W
typ
C
Power-Supply Rejection Ratio (+PSRR)
TEMPERATURE RANGE
Specification: D, DBV
Thermal Resistance, θJA
D
SO-8
DBV SOT23-6
NOTES: (1) Junction temperature = ambient temperature for low temperature limit and +25°C specifications. Junction temperature = ambient temperature +10°C
at high temperature limit specifications. (2) Test Levels: (A) 100% tested at +25°C. Over-temperature limits by characterization and simulation. (B) Limits set by
characterization and simulation. (C) Typical value only for information. (3) Current is considered positive out-of-node. VCM is the input common-mode voltage.
6
OPA692
www.ti.com
SBOS236E
TYPICAL CHARACTERISTICS: VS = ±5V
TA = +25°C, G = +2, and RL = 100Ω (see Figure 1 for DC performance only), unless otherwise noted.
SMALL-SIGNAL FREQUENCY RESPONSE
LARGE-SIGNAL FREQUENCY RESPONSE
1
7
VO = 1Vp-p
6
G = +1
VO = 2Vp-p
–1
G = –1
–2
5
Gain (1dB/div)
Normalized Gain (1dB/div)
0
–3
–4
–5
VO = 4Vp-p
3
VO = 7Vp-p
2
G = +2
–6
4
1
–7
–8
0
0
250MHz
500MHz
0
125MHz
Frequency (50MHz/div)
LARGE-SIGNAL PULSE RESPONSE
SMALL-SIGNAL PULSE RESPONSE
4
VO = 0.5Vp-p
G = +2
300
VO = 5Vp-p
G = +2
3
Output Voltage (1V/div)
Output Voltage (100mV/div)
400
200
100
0
–100
–200
2
1
0
–1
–2
–3
–300
–4
–400
Time (5ns/div)
Time (5ns/div)
COMPOSITE VIDEO dG/dP
0.20
+5V
0.18
Video In
0.16
DIS
Video Loads
DISABLED FEEDTHROUGH vs FREQUENCY
–50
No Pull-Down
With 1.3kΩ Pull-Down
0.12
Feedthrough (5dB/div)
–5V
dG
Optional
1.3kΩ
Pull-Down
dG
0.10
0.08
dP
0.06
0.04
VDIS = 0
–55
OPA692
0.14
dG/dP (%/°)
250MHz
Frequency (25MHz/div)
dP
–60
–65
–70
–75
Reverse
–80
Forward
–85
–90
0.02
0
–95
1
2
3
4
0.5
10
100
Frequency (MHz)
Number of 150Ω Loads
OPA692
SBOS236E
1
www.ti.com
7
TYPICAL CHARACTERISTICS: VS = ±5V (Cont.)
TA = +25°C, G = +2, and RL = 100Ω (see Figure 1 for DC performance only), unless otherwise noted.
HARMONIC DISTORTION vs LOAD RESISTANCE
5MHz HARMONIC DISTORTION vs SUPPLY VOLTAGE
–60
–50
VO = 2Vp-p
f = 5MHz
VO = 2Vp-p
RL = 100Ω
f = 5MHz
–55
–70
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
–65
2nd-Harmonic
–75
–80
–85
3rd-Harmonic
–90
–95
–100
–60
2nd-Harmonic
–65
–70
–75
–80
3rd-Harmonic
–85
–105
–110
–90
100
1000
2
2.5
3
Load Resistance (Ω)
4.5
5
5.5
6
–65
VO = 2Vp-p
RL = 100Ω
dBc = dB Below Carrier
RL = 100Ω
f = 5MHz
–60
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
4
HARMONIC DISTORTION vs OUTPUT VOLTAGE
HARMONIC DISTORTION vs FREQUENCY (G = +2)
–50
2nd-Harmonic
–70
–80
3rd-Harmonic
–90
–100
2nd-Harmonic
–70
–75
3rd-Harmonic
–80
–85
0.1
1
10
0.1
20
HARMONIC DISTORTION vs FREQUENCY (G = –1)
–50
5
HARMONIC DISTORTION vs FREQUENCY (G = +1)
–50
VO = 2Vp-p
RL = 100Ω
dBc = dB Below Carrier
1
Output Voltage Swing (Vp-p)
Frequency (MHz)
VO = 2Vp-p
RL = 100Ω
dBc = dB Below Carrier
–60
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
3.5
Supply Voltage (±VS)
2nd-Harmonic
–70
–80
3rd-Harmonic
–90
–60
–70
3rd-Harmonic
–80
–90
2nd-Harmonic
–100
–100
0.1
1
10
20
0.1
Frequency (MHz)
8
1
10
20
Frequency (MHz)
OPA692
www.ti.com
SBOS236E
TYPICAL CHARACTERISTICS: VS = ±5V (Cont.)
TA = +25°C, G = +2, and RL = 100Ω (see Figure 1 for DC performance only), unless otherwise noted.
2-TONE, 3RD-ORDER
INTERMODULATION SPURIOUS
INPUT VOLTAGE AND CURRENT NOISE DENSITY
–30
3rd-Order Spurious Level (dBc)
Inverting Input Current Noise (15pA/√Hz)
10
Noninverting Current Noise (12pA/√Hz)
Voltage Noise (1.7nV/√Hz)
1
dBc = dB below carriers
50MHz
–40
–50
–60
20MHz
–70
10MHz
–80
Load Power at Matched 50Ω Load
–90
100
1k
10k
100k
1M
10M
–8
–6
–4
Frequency (Hz)
RECOMMENDED RS vs CAPACITIVE LOAD
Normalized Gain to Capacitive Load (dB)
50
RS (Ω)
40
30
20
10
0
10
2
4
6
8
10
100
9
CL = 10pF
6
3
CL = 47pF
0
CL = 22pF
–3
VIN
RS
402Ω
–6
CL = 100pF
VO
OPA692
CL
1kΩ
402Ω
1kΩ is optional.
–9
1k
0
125MHz
Capacitive Load (pF)
250MHz
Frequency (25MHz/div)
PSRR vs FREQUENCY
SUPPLY AND OUTPUT CURRENT vs TEMPERATURE
65
10
250
Sourcing Output Current
60
+PSRR
55
Supply Current (2mA)
Power-Supply Rejection Ratio (dB)
0
FREQUENCY RESPONSE vs CAPACITIVE LOAD
60
1
–2
Single-Tone Load Power (dBm)
–PSRR
50
45
40
35
30
8
200
Sinking Output Current
6
150
4
100
Quiescent Supply Current
2
50
Output Current (50mA/div)
Current Noise (pA/√Hz)
Voltage Noise (nV/√Hz)
100
25
20
0
1k
10k
100k
1M
10M
100M
Frequency (Hz)
–25
0
25
50
75
100
125
Ambient Temperature (°C)
OPA692
SBOS236E
0
–50
www.ti.com
9
TYPICAL CHARACTERISTICS: VS = ±5V (Cont.)
TA = +25°C, G = +2, and RL = 100Ω (see Figure 1 for DC performance only), unless otherwise noted.
TYPICAL DC DRIFT OVER TEMPERATURE
OUTPUT VOLTAGE AND CURRENT LIMITATIONS
Output Current Limited
1W Internal
Power Limit
Input Offset Voltage (mV)
3
VO (V)
2
1
25Ω
Load Line
0
–1
50Ω Load Line
–2
100Ω Load Line
–3
–4
1W Internal
Power Limit
Output Current Limit
40
Noninverting Input Bias Current
1.5
30
1
20
0.5
10
Inverting Input Bias Current
0
0
–0.5
–10
Input Offset Voltage
–1
–20
–1.5
–30
–2
–5
–300 –250 –200 –150 –100 –50
0
50
–40
–50
100 150 200 250 300
–25
0
25
50
LARGE-SIGNAL DISABLE/ENABLE RESPONSE
0
2.0
Output Voltage (10mV/div)
Output Voltage (400mV/div)
2.0
6.0
VDIS
VDIS (2V/div)
4.0
VDIS
1.6
Output Voltage
0.8
0
125
DISABLE/ENABLE GLITCH
6.0
0.4
100
Ambient Temperature (°C)
IO (mA)
1.2
75
VIN = +1V
4.0
2.0
0
30
20
Output Voltage
(0V Input)
10
0
–10
–20
Time (200ns/div)
Time (20ns/div)
CLOSED-LOOP OUTPUT IMPEDANCE
10
Output Impedance (Ω)
+5V
50Ω
OPA692
ZO
–5V 402Ω
1
402Ω
0.1
10k
100k
1M
10M
100M
Frequency (Hz)
10
OPA692
www.ti.com
SBOS236E
VDIS (2V/div)
4
2
Input Bias Currents (µA)
5
TYPICAL CHARACTERISTICS: VS = +5V
TA = +25°C, G = +2, and RL = 100Ω (see Figure 2 for AC performance only), unless otherwise noted.
LARGE-SIGNAL FREQUENCY RESPONSE
SMALL-SIGNAL FREQUENCY RESPONSE
1
7
VO = 0.5Vp-p
6
G = +1
–1
5
–2
Gain (1dB/div)
Normalized Gain (1dB/div)
0
G = –1
–3
–4
–5
3
2
–6
G = +2
1
–7
RL = 100Ω to 2.5V
–8
0
0
250M
0
500M
125M
SMALL-SIGNAL PULSE RESPONSE
LARGE-SIGNAL PULSE RESPONSE
2.9
4.1
G = +2
VO = 0.5Vp-p
2.8
Output Voltage (400mV/div)
Output Voltage (100mV/div)
250M
Frequency (Hz)
Frequency (Hz)
2.7
2.6
2.5
2.4
2.3
2.2
G = +2
VO = 2Vp-p
3.7
3.3
2.9
2.5
2.1
1.7
1.3
2.1
0.9
Time (5ns/div)
Time (5ns/div)
RECOMMENDED RS vs CAPACITIVE LOAD
FREQUENCY RESPONSE vs CAPACITIVE LOAD
9
60
6
Normalized Gain to
Capacitive Load (dB)
70
50
RS (Ω)
VO = 1Vp-p
VO = 2Vp-p
4
40
30
20
CL = 10pF
CL = 22pF
3
CL = 47pF
0
+5V
–3
806Ω
VIN
0.1µF
RS
57.6Ω
10
–6
0
–9
806Ω OPA692
CL
CL = 100pF
VO
1kΩ
402Ω
402Ω
(1kΩ is optional)
0.1µF
1
10
100
1k
0
Capacitive Load (pF)
OPA692
SBOS236E
125MHz
250MHz
Frequency (25MHz/div)
www.ti.com
11
TYPICAL CHARACTERISTICS: VS = +5V (Cont.)
TA = +25°C, G = +2, and RL = 100Ω (see Figure 2 for AC performance only), unless otherwise noted.
HARMONIC DISTORTION vs FREQUENCY
HARMONIC DISTORTION vs LOAD RESISTANCE
–60
–50
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
VO = 2Vp-p
f = 5MHz
–65
2nd-Harmonic
–70
3rd-Harmonic
–75
–80
2nd-Harmonic
–70
–80
3rd-Harmonic
0.1
1k
1
Frequency (MHz)
HARMONIC DISTORTION vs OUTPUT VOLTAGE
2-TONE, 3RD-ORDER
INTERMODULATION SPURIOUS
–30
3rd-Order Spurious Level (dBc)
RL = 100Ω to 2.5V
f = 5MHz
–65
2nd-Harmonic
–70
–75
3rd-Harmonic
1
2
50MHz
–40
–45
–50
–55
20MHz
–60
–65
10MHz
–70
–75
Load Power at Matched 50Ω Load
–14
3
20
dBc = dB Below Carriers
–35
–80
–80
0.1
10
Load Resistance (Ω)
–60
Harmonic Distortion (dBc)
–60
–90
100
–12
–10
–8
–6
–4
–2
0
2
Single-Tone Load Power (dBm)
Output Voltage Swing (Vp-p)
12
VO = 2Vp-p
RL = 100Ω to 2.5V
OPA692
www.ti.com
SBOS236E
APPLICATIONS INFORMATION
WIDEBAND BUFFER OPERATION
The OPA692 gives the exceptional AC performance of a
wideband current-feedback op amp with a highly linear, highpower output stage. It features internal RF and RG resistors
that make it easy to select a gain of +2, +1, or –1 without any
external resistors. Requiring only 5.1mA quiescent current, the
OPA692 will swing to within 1V of either supply rail and deliver
in excess of 160mA at room temperature. This low output
headroom requirement, along with supply voltage independent biasing, gives remarkable single (+5V) supply operation.
The OPA692 will deliver greater than 200MHz bandwidth
driving a 2VPP output into 100Ω on a single +5V supply.
Previous boosted output stage amplifiers have typically suffered from very poor crossover distortion as the output current
goes through zero. The OPA692 achieves a comparable
power gain with much better linearity. The primary advantage
of a current-feedback op amp over a voltage-feedback op amp
is that AC performance (bandwidth and distortion) is relatively
independent of signal gain.
Figure 1 shows the DC-coupled, gain of +2, dual power-supply
circuit configuration used as the basis of the ±5V Electrical and
Typical Characteristics. For test purposes, the input impedance is set to 50Ω with a resistor to ground and the output
impedance is set to 50Ω with a series output resistor. Voltage
swings reported in the specifications are taken directly at the
input and output pins while load powers (dBm) are defined at
a matched 50Ω load. For the circuit of Figure 1, the total
effective load will be 100Ω || 804Ω = 89Ω. The disable control
line (DIS) is typically left open to ensure normal amplifier
operation. In addition to the usual power-supply decoupling
capacitors to ground, a 0.1µF capacitor can be included
between the two power-supply pins. This optional added
capacitor will typically improve the 2nd-harmonic distortion
performance by 3dB to 6dB.
Figure 2 shows the AC-coupled, gain of +2, single-supply
circuit configuration used as the basis of the +5V Electrical
and Typical Characteristics. Though not a rail-to-rail design,
the OPA692 requires minimal input and output voltage headroom compared to other very wideband current-feedback op
amps. It will deliver a 3VPP output swing on a single +5V
supply with greater than 150MHz bandwidth. The key requirement of broadband single-supply operation is to maintain input and output signal swings within the usable voltage
ranges at both the input and the output. The circuit of Figure
2 establishes an input midpoint bias using a simple resistive
divider from the +5V supply (two 806Ω resistors). The input
signal is then AC-coupled into this midpoint voltage bias. The
input voltage can swing to within 1.5V of either supply pin,
giving a 2VPP input signal range centered between the supply
pins. The input impedance matching resistor (57.6Ω) used
for testing is adjusted to give a 50Ω input match when the
parallel combination of the biasing divider network is included. The gain resistor (RG) is AC-coupled, giving the
circuit a DC gain of +1—which puts the input DC bias voltage
(2.5V) on the output as well. Again, on a single +5V supply,
the output voltage can swing to within 1V of either supply pin
while delivering more than 120mA output current. A demanding 100Ω load to a midpoint bias is used in this characterization circuit. The new output stage used in the OPA692 can
deliver large bipolar output currents into this midpoint load
with minimal crossover distortion, as shown by the +5V
supply, 3rd-harmonic distortion typical characteristics.
+VS
+5V
0.1µF
50Ω Source
0.1µF
57.6Ω
6.8µF
806Ω
DIS
VIN
+5V
+
VO
806Ω
OPA692
100Ω
VS/2
DIS
RF
402Ω
+
0.1µF
6.8µF
50Ω Source
VIN
50Ω
50Ω
RG
402Ω
50Ω Load
0.1µF
OPA692
RF
402Ω
FIGURE 2. AC-Coupled, G = +2, Single-Supply Specification
and Test Circuit.
RG
402Ω
SINGLE-SUPPLY ADC INTERFACE
0.1µF
+
6.8µF
–5V
FIGURE 1. DC-Coupled, G = +2, Bipolar Supply, Specification and Test Circuit.
Most modern, high-performance ADCs (such as the Texas
Instruments ADS8xx and ADS9xx series) operate on a single
+5V (or lower) power supply. It has been a considerable
challenge for single-supply op amps to deliver a low-distortion input signal at the ADC input for signal frequencies
OPA692
SBOS236E
www.ti.com
13
the output, centering the output voltage swing as well. Tested
performance at a 20MHz analog input frequency and a 60MSPS
clock rate on the converter gives > 58dBc SFDR.
exceeding 5MHz. The high slew rate, exceptional output
swing, and high linearity of the OPA692 make it an ideal
single-supply ADC driver. Figure 3 shows an example input
interface to a very high performance 10-bit, 60MSPS CMOS
converter.
WIDEBAND VIDEO MULTIPLEXING
The OPA692 in the circuit of Figure 3 provides 190MHz
bandwidth operating at a signal gain of +2 with a 2VPP output
swing. The noninverting input bias voltage is referenced to the
midpoint of the ADC signal range by dividing off the top and
bottom of the internal ADC reference ladder. With the gain
resistor (RG) AC-coupled, this bias voltage has a gain of +1 to
One common application for video speed amplifiers that
include a disable pin is to wire multiple amplifier outputs
together, then select which one of several possible video
inputs to source onto a single line. This simple wired-OR
video multiplexer can be easily implemented using the
OPA692, as shown in Figure 4.
+5V
+5V
RF
402Ω
RG
402Ω
0.1µF
ADS826
10-Bit
60MSPS
Clock
50Ω
Input
OPA692
2Vp-p
1Vp-p
22pF
Input
0.1µF
CM
2kΩ
DIS
+3.5V
REFT
0.1µF
+2.5V DC Bias
2kΩ
+1.5V
REFB
0.1µF
FIGURE 3. Wideband, AC-Coupled, Single-Supply ADC Driver.
+5V
2kΩ
|VOUT| < 2.6V
VDIS
+5V
Video 1
DIS
OPA692
75Ω
402Ω
–5V
402Ω
68.1Ω
75Ω Cable
VOUT
402Ω
RG-59
402Ω
+5V
68.1Ω
OPA692
Video 2
DIS
75Ω
2kΩ
–5V
FIGURE 4. 2-Channel Video Multiplexer.
14
OPA692
www.ti.com
SBOS236E
Typically, channel switching is performed either on sync or
retrace time in the video signal. The two inputs are approximately equal at this time. The make-before-break disable
characteristic of the OPA692 ensures that there is always
one amplifier controlling the line when using a wired-OR
circuit (see Figure 4). Since both inputs may be on for a short
period during the transition between channels, the outputs
are combined through the output impedance matching resistors (68.1Ω in this case). When one channel is disabled, its
feedback network forms part of the output impedance and
slightly attenuates the signal in getting out onto the cable.
The matching resistors have been set to get a signal gain of
+1 at the load while providing > 20dB return loss at the load.
The video multiplexer connection (see Figure 4) also insures
that the maximum differential voltage across the inputs of the
unselected channel do not exceed the rated ±1.2V maximum
for standard video signal levels. In any case, VOUT must be
< ±2.6VPP in order to not exceed the absolute maximum
differential input voltage (±1.2V) on the disabled channel.
The Disable Operation section shows the turn-on and turn-off
switching glitches using a grounded input for a single channel is typically less than ±50mV. Where two outputs are
switched (see Figure 4), the output line is always under the
control of one amplifier or the other due to the make-beforebreak disable timing. In this case, the switching glitches for
two 0V inputs drops to < 20mV.
4-CHANNEL FREQUENCY CHANNELIZER
The circuit of Figure 5 is a 4-channel multiplexer. In this
circuit the OPA691 provides the drive for all four channels.
Each channel includes a bandpass filter and each bandpass
filter is set for a different frequency band. This allows the
channelizing part of this circuit. The role of the OPA692 is to
provide impedance isolation. This is done through the use of
four matching resistances (59Ω in this case). These matching resistors ensure that the signals will combine during the
transition between channels. They have been used to get a
gain of +1 at the load.
This circuit may be used with a different number of channels.
Its limitation comes from the drive requirement for each
channel, as well as the minimum acceptable return loss.
The output resistor value (RO) to keep a gain of +1 at the
load, depends on the number of channels. For the OPA692,
Equation 1 gives:
(1)
RO =
[75Ω • (n – 2) + 804Ω] •
2
1+
[
241200Ω
75Ω • (n − 2) + 804Ω
]
2
– 1
Where n = number of devices in multiplexer.
+5V
75Ω
DIS 1
RO
59Ω
#1
OPA692
75Ω
–5V
+5V
75Ω
DIS 2
RO
59Ω
#2
OPA692
+5V
75Ω
75Ω Cable
–5V
OPA691
+5V
75Ω
RO
59Ω
#3
–5V
VOUT
DIS 3
RG-59
75Ω Load
OPA692
75Ω
–5V
+5V
75Ω
DIS 4
RO
59Ω
#4
OPA692
75Ω
–5V
G = +2 Stages
FIGURE 5. 4-Channel Frequency Channelizer.
OPA692
SBOS236E
www.ti.com
15
DELAY-EQUALIZED LOW-PASS FILTER
The circuit in Figure 6 realizes a 5th-order Butterworth lowpass filter with a –3dB bandwidth of 20MHz and group delay
equalization. This filter is based on the KRC active filter
topology using amplifiers with a fixed positive gain ≥ 1.
VIN
The component values have been predistorted to compensate
for the op amps parasitic effects. The low-Q pole section was
placed last to minimize noise peaking in the passband, while
maintaining good dynamic range performance.
OPA692
+5V
200Ω
The OPA692 makes a good amplifier for this type of filter. The
first stage is the group delay equalizer, which is based on a
gain of –1. The second stage has a high-Q pole, uses a gain
of +2 for minimum component sensitivity, and also produces a
real pole. The last stage has a low-Q pole, and uses a gain of
+1 for minimum component sensitivity.
80.6kΩ
2.7nF
402Ω
VOUT
402Ω
OPA227
200Ω
–5V
2.7nF
FIGURE 7. Precision Wideband, Unity-Gain Buffer.
PRECISION VOLTAGE BUFFER
DESIGN-IN TOOLS
The precision buffer in Figure 7 combines the DC precision
and low 1/f noise of the OPA227 with the high-speed performance of the OPA692. The 80.6kΩ resistor makes the highfrequency and low-frequency nominal gains equal. The
OPA692 takes over from the OPA227 at approximately 32kHz.
DEMONSTRATION FIXTURES
Two printed circuit boards (PCBs) are available to assist in
the initial evaluation of circuit performance using the OPA692
in its two package options. Both of these are offered free of
56pF
402Ω
402Ω
49.9Ω
105Ω
226Ω
VIN
220pF
OPA692
27pF
115Ω
OPA692
402Ω
100pF
402Ω
68pF
95.3Ω
226Ω
OPA692
39pF
VOUT
402Ω
402Ω
(Open)
FIGURE 6. Butterworth LP Filter with Delay Equalization.
16
OPA692
www.ti.com
SBOS236E
charge as unpopulated PCBs, delivered with a user's guide.
The summary information for these fixtures is shown in
the table below.
PRODUCT
OPA692ID
OPA692IDBV
PACKAGE
ORDERING
NUMBER
LITERATURE
NUMBER
SO-8
SOT23-6
DEM-OPA-SO-1A
DEM-OPA-SOT-1A
SBOU009
SBOU010
over-temperature specifications because the output stage
junction temperatures are higher than the minimum specified
operating ambient.
DRIVING CAPACITIVE LOADS
The demonstration fixtures can be requested at the Texas
Instruments web site (www.ti.com) through the OPA692
product folder.
OPERATING SUGGESTIONS
GAIN SETTING
Setting the gain with the OPA692 is very easy. For a gain of
+2, ground the –IN pin and drive the +IN pin with the signal.
For a gain of +1, leave the –IN pin open and drive the +IN pin
with the signal. For a gain of –1, ground the +IN pin and drive
the –IN pin with the signal. As the internal resistor values (not
their ratio) change over temperature and process, external
resistors should not be used to modify the gain.
OUTPUT CURRENT AND VOLTAGE
The OPA692 provides output voltage and current capabilities
that are unsurpassed in a low-cost monolithic op amp. Under
no-load conditions at +25°C, the output voltage typically
swings closer than 1V to either supply rail; the tested swing
limit is within 1.2V of either rail. Into a 15Ω load (the minimum
tested load), it is specified to deliver more than ±160mA.
The specifications described previously, though familiar in
the industry, consider voltage and current limits separately. In
many applications, it is the voltage times current, or V-I
product, which is more relevant to circuit operation. Refer to
the “Output Voltage and Current Limitations” plot in the
Typical Characteristics. The X- and Y-axes of this graph show
the zero-voltage output current limit and the zero-current
output voltage limit, respectively. The four quadrants give a
more detailed view of the OPA692 output drive capabilities,
noting that the graph is bounded by a safe operating area of
1W maximum internal power dissipation. Superimposing
resistor load lines onto the plot shows that the OPA692 can
drive ±2.5V into 25Ω, or ±3.5V into 50Ω without exceeding
the output capabilities or the 1W dissipation limit. A 100Ω
load line (the standard test circuit load) shows the full ±3.9V
output swing capability (see the Electrical Characteristics).
The minimum specified output voltage and current over
temperature are set by worst-case simulations at the cold
temperature extreme. Only at cold startup will the output
current and voltage decrease to the numbers shown in the
Electrical Characteristics. As the output transistors deliver
power, their junction temperatures increase, decreasing their
VBEs (increasing the available output voltage swing), and
increasing their current gains (increasing the available output
current). In steady-state operation, the available output voltage and current is always greater than that shown in the
One of the most demanding and yet very common load
conditions for an op amp is capacitive loading. Often, the
capacitive load is the input of an ADC—including additional
external capacitance which may be recommended to improve
ADC linearity. A high-speed amplifier like the OPA692 can be
very susceptible to decreased stability and frequency response peaking when a capacitive load is placed directly on
the output pin. When the amplifier’s open-loop output resistance is considered, this capacitive load introduces an additional pole in the signal path that can decrease the phase
margin. Several external solutions to this problem have been
suggested. When the primary considerations are frequency
response flatness, pulse response fidelity, and/or distortion,
the simplest and most effective solution is to isolate the
capacitive load from the feedback loop by inserting a series
isolation resistor between the amplifier output and the capacitive load. This does not eliminate the pole from the loop
response, but rather shifts it and adds a zero at a higher
frequency. The additional zero acts to cancel the phase lag
from the capacitive load pole, thus increasing the phase
margin and improving stability.
The Typical Characteristics show the recommended “RS vs
Capacitive Load” and the resulting frequency response at the
load. Parasitic capacitive loads greater than 2pF can begin to
degrade the performance of the OPA692. Long PC board
traces, unmatched cables, and connections to multiple devices can easily cause this value to be exceeded. Always
consider this effect carefully, and add the recommended
series resistor as close as possible to the OPA692 output pin
(see the Board Layout Guidelines section).
DISTORTION PERFORMANCE
The OPA692 provides good distortion performance into a
100Ω load on ±5V supplies. Relative to alternative solutions, it
provides exceptional performance into lighter loads and/or
operating on a single +5V supply. Generally, until the fundamental signal reaches very high-frequency or power levels, the
2nd-harmonic will dominate the distortion with a negligible 3rdharmonic component. Focusing then on the 2nd-harmonic,
increasing the load impedance improves distortion directly.
Remember that the total load includes the feedback network—
in the noninverting configuration (see Figure 1) this is the sum
of RF + RG, while in the inverting configuration, it is just RF.
Also, providing an additional supply decoupling capacitor
(0.1µF) between the supply pins (for bipolar operation) improves the 2nd-order distortion slightly (3dB to 6dB).
In most op amps, increasing the output voltage swing increases
harmonic distortion directly. The Typical Characteristics show the
2nd-harmonic increasing at a little less than the expected 2x rate
while the 3rd-harmonic increases at a much lower rate than the
expected 3x. Where the test power doubles, the difference
between it and the 2nd-harmonic decreases less than the
OPA692
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17
expected 6dB, while the difference between it and the 3rd
decreases by less than the expected 12dB. This also shows up
in the 2-tone, 3rd-order intermodulation spurious (IM3) response
curves. The 3rd-order spurious levels are extremely low at low
output power levels. The output stage continues to hold them low
even as the fundamental power reaches very high levels. As the
Typical Characteristics show, the spurious intermodulation powers do not increase as predicted by a traditional intercept model.
As the fundamental power level increases, the dynamic range
does not decrease significantly. For two tones centered at
20MHz, with 10dBm/tone into a matched 50Ω load (i.e., 2VPP for
each tone at the load, which requires 8VPP for the overall
2-tone envelope at the output pin), the Typical Characteristics
show 58dBc difference between the test-tone power and the 3rdorder intermodulation spurious levels. This exceptional performance improves further when operating at lower frequencies.
NOISE PERFORMANCE
The OPA692 offers an excellent balance between voltage and
current noise terms to achieve low output noise. The inverting
current noise (15pA/√Hz) is significantly lower than earlier
solutions while the input voltage noise (1.7nV√Hz) is lower
than most unity-gain stable, wideband, voltage-feedback op
amps. This low input voltage noise was achieved at the price
of higher noninverting input current noise (12pA/√Hz). As long
as the AC source impedance looking out of the noninverting
node is less than 100Ω, this current noise will not contribute
significantly to the total output noise. The op amp input voltage
noise and the two input current noise terms combine to give
low output noise for the gain settings, available using the
OPA692. Figure 8 shows the op amp noise analysis model
with all the noise terms included. In this model, all noise terms
are taken to be noise voltage or current density terms in either
nV/√Hz or pA/√Hz.
ENI
(2)
2
2
EO = ENI2 + (IBNR S ) + 4kTRS NG2 + (IBIRF ) + 4kTRFNG
Dividing this expression by the noise gain (NG = (1 + RF/RG))
will give the equivalent input-referred spot noise voltage at the
noninverting input, as shown in Equation 3.
(3)
2
4kTRF
2
I R
EN = ENI2 + (IBNR S ) + 4kTRS + BI F +
NG
NG
Evaluating these two equations for the OPA692 circuit and
component values (see Figure 1) will give a total output spot
noise voltage of 8.2nV/√Hz and a total equivalent input spot
noise voltage of 4.1nV/√Hz. This total input-referred spot
noise voltage is higher than the 1.7nV/√Hz specification for
the op amp voltage noise alone. This reflects the noise added
to the output by the inverting current noise times the feedback resistor.
DC ACCURACY
The OPA692 provides exceptional bandwidth in high gains,
giving fast pulse settling but only moderate DC accuracy. The
Electrical Characteristics show an input offset voltage comparable to high-speed voltage-feedback amplifiers. However,
the two input bias currents are somewhat higher and are
unmatched. Bias current cancellation techniques will not
reduce the output DC offset for OPA692. As the two input
bias currents are unrelated in both magnitude and polarity,
matching the source impedance looking out of each input to
reduce their error contribution to the output is ineffective.
Evaluating the configuration of Figure 1, using worst-case
+25°C input offset voltage and the two input bias currents,
gives a worst-case output offset range equal to:
±(NG • VOS(max)) + (IBN • RS/2 • NG) ± (IBI • RF)
EO
OPA692
RS
The total output spot noise voltage can be computed as the
square root of the sum of all squared output noise voltage
contributors. Equation 2 shows the general form for the output
noise voltage using the terms shown in Figure 8.
where NG = noninverting signal gain
= ±(2 • 2.5mV) + (35µA • 25Ω • 2) ± (402Ω • 25µA)
IBN
= ±5mV + 1.75mV ± 10.05mV
= –13.3mV → +16.80mV
ERS
RF
√4kTRS
4kT
RG
RG
IBI
√4kTRF
4kT = 1.6E –20J
at 290°K
Minimizing the resistance seen by the noninverting input will
give the best DC offset performance.
For significantly improved DC accuracy, consider the precision buffer circuit (see Figure 7).
DISABLE OPERATION
FIGURE 8. Noise Model.
18
The OPA692 provides an optional disable feature that may be
used either to reduce system power or to implement a simple
channel multiplexing operation. If the DIS control pin is left
OPA692
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SBOS236E
unconnected, the OPA692 will operate normally. To disable,
the control pin must be asserted LOW. Figure 9 shows a
simplified internal circuit for the disable control feature.
+VS
Q1
110kΩ
25kΩ
IS
Control
Due to the high output power capability of the OPA692,
heatsinking or forced airflow may be required under extreme
operating conditions. Maximum desired junction temperature
will set the maximum allowed internal power dissipation, as
described below. In no case should the maximum junction
temperature be allowed to exceed 175°C.
Operating junction temperature (TJ) is given by TA + PD • θJA.
The total internal power dissipation (PD) is the sum of
quiescent power (PDQ) and additional power dissipated in the
output stage (PDL) to deliver load power. Quiescent power is
simply the specified no-load supply current times the total
supply voltage across the part. PDL depends on the required
output signal and load but would, for a grounded resistive
load, be at a maximum when the output is fixed at a voltage
equal to 1/2 either supply voltage (for equal bipolar supplies).
Under this condition PDL = VS2/(4 • RL), where RL includes
feedback network loading.
15kΩ
VDIS
THERMAL ANALYSIS
–VS
FIGURE 9. Simplified Disable Control Circuit.
Note that it is the power in the output stage and not in the
load that determines internal power dissipation.
In normal operation, base current to Q1 is provided through
the 110kΩ resistor while the emitter current through the 15kΩ
resistor sets up a voltage drop that is inadequate to turn on
the two diodes in Q1’s emitter. As V DIS is pulled LOW,
additional current is pulled through the 15kΩ resistor eventually turning on these two diodes (≈ 75µA). At this point, any
further current pulled out of V DIS goes through those diodes
holding the emitter-base voltage of Q1 at approximately 0V.
This shuts off the collector current out of Q1, turning the
amplifier off. The supply current in the disable mode is only
that required to operate the circuit of Figure 8. Additional
circuitry ensures that turn-on time occurs faster than turn-off
time (make-before-break).
As a worst-case example, compute the maximum TJ using an
OPA692IDBV (SOT23-6 package) in the circuit of Figure 1
operating at the maximum specified ambient temperature of
+85°C and driving a grounded 20Ω load to +2.5VDC:
When disabled, the output and input nodes go to a highimpedance state. If the OPA692 is operating in a gain of +1,
this will show a very high impedance (4pF || 1MΩ) at the
output and exceptional signal isolation. If operating at a gain
of +2, the total feedback network resistance (RF + RG) will
appear as the impedance looking back into the output, but
the circuit will still show very high forward and reverse
isolation. If configured at a gain of –1, the input and output
will be connected through the feedback network resistance
(RF + RG) giving relatively poor input to output isolation.
One key parameter in disable operation is the output glitch
when switching in and out of the disabled mode. The Typical
Characteristics show these glitches for the circuit of Figure 1
with the input signal set to 0V. The glitch waveform at the
output pin is plotted along with the DIS pin voltage.
The transition edge rate (dV/dt) of the DIS control line will
influence this glitch. Slowing this edge can be achieved by
adding a simple RC filter into the V DIS pin from a higher
speed logic line. If extremely fast transition logic is used, a
2kΩ series resistor between the logic gate and the DIS input
pin will provide adequate bandlimiting using just the parasitic
input capacitance on the DIS pin while still ensuring an
adequate logic level swing.
PD = 10V • 5.8mA + 52/(4 • (20Ω || 800Ω)) = 378mW
Maximum TJ = +85°C + (0.39W • 150°C/W) = 142°C
Although this is still well below the specified maximum
junction temperature, system reliability considerations may
require lower junction temperatures. Remember, this is a
worst-case internal power dissipation—use your actual signal and load to compute PDL. The highest possible internal
dissipation occurs if the load requires current to be forced
into the output for positive output voltages or sourced from
the output for negative output voltages. This puts a high
current through a large internal voltage drop in the output
transistors. The “Output Voltage and Current Limitations” plot
shown in the Typical Characteristics include a boundary for
1W maximum internal power dissipation under these conditions.
BOARD LAYOUT GUIDELINES
Achieving optimum performance with a high-frequency amplifier like the OPA692 requires careful attention to board
layout parasitics and external component types. Recommendations that will optimize performance include:
a) Minimize parasitic capacitance to any AC ground for
all of the signal I/O pins. Parasitic capacitance on the
output pin can cause instability: on the noninverting input, it
can react with the source impedance to cause unintentional
bandlimiting. To reduce unwanted capacitance, a window
around the signal I/O pins should be opened in all of the
ground and power planes around those pins. Otherwise,
ground and power planes should be unbroken elsewhere on
the board.
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19
b) Minimize the distance (< 0.25") from the power-supply
pins to high-frequency 0.1µF decoupling capacitors. At
the device pins, the ground and power-plane layout should
not be in close proximity to the signal I/O pins. Avoid narrow
power and ground traces to minimize inductance between
the pins and the decoupling capacitors. The power-supply
connections (on pins 4 and 7) should always be decoupled
with these capacitors. An optional supply decoupling capacitor across the two power supplies (for bipolar operation) will
improve 2nd-harmonic distortion performance. Larger (2.2µF
to 6.8µF) decoupling capacitors, effective at lower frequencies, should also be used on the main supply pins. These
may be placed somewhat further from the device and may be
shared among several devices in the same area of the PC
board.
shunt resistor and the input impedance of the destination
device; this total effective impedance should be set to match
the trace impedance. The high output voltage and current
capability of the OPA692 allows multiple destination devices
to be handled as separate transmission lines, each with their
own series and shunt terminations. If the 6dB attenuation of
a doubly-terminated transmission line is unacceptable, a
long trace can be series-terminated at the source end only.
Treat the trace as a capacitive load in this case and set the
series resistor value as shown in the plot of “RS vs Capacitive
Load.” This will not preserve signal integrity as well as a
doubly-terminated line. If the input impedance of the destination device is low, there will be some signal attenuation due
to the voltage divider formed by the series output into the
terminating impedance.
c) Careful selection and placement of external components will preserve the high-frequency performance of
the OPA692. Any external resistors should be a very low
reactance type. Surface-mount resistors work best and allow
a tighter overall layout. Metal-film and carbon composition,
axially-leaded resistors can also provide good high-frequency
performance. Again, keep their leads and PC-board trace
length as short as possible. Never use wirewound type
resistors in a high-frequency application. All external components should also be placed close to the package.
e) Socketing a high-speed part like the OPA692 is not
recommended. The additional lead length and pin-to-pin
capacitance introduced by the socket can create an extremely troublesome parasitic network which can make it
almost impossible to achieve a smooth, stable frequency
response. Best results are obtained by soldering the OPA692
onto the board.
d) Connections to other wideband devices on the board
may be made with short direct traces or through onboard transmission lines. For short connections, consider
the trace and the input to the next device as a lumped
capacitive load. Relatively wide traces (50mils to 100mils)
should be used, preferably with ground and power planes
opened up around them. Estimate the total capacitive load
and set RS from the plot of recommended “RS vs Capacitive
Load.” Low parasitic capacitive loads (< 5pF) may not need
an RS because the OPA692 is nominally compensated to
operate with a 2pF parasitic load. If a long trace is required,
and the 6dB signal loss intrinsic to a doubly-terminated
transmission line is acceptable, implement a matched impedance transmission line using microstrip or stripline techniques (consult an ECL design handbook for microstrip and
stripline layout techniques). A 50Ω environment is normally
not necessary on board, and in fact, a higher impedance
environment will improve distortion as shown in the “Distortion vs Load” plots. With a characteristic board trace impedance defined based on board material and trace dimensions,
a matching series resistor into the trace from the output of the
OPA692 is used as well as a terminating shunt resistor at the
input of the destination device. Remember also that the
terminating impedance will be the parallel combination of the
20
INPUT AND ESD PROTECTION
The OPA692 is built using a very high-speed complementary
bipolar process. The internal junction breakdown voltages
are relatively low for these very small geometry devices.
These breakdowns are reflected in the Absolute Maximum
Ratings table. All device pins have limited ESD protection
using internal diodes to the power supplies, as shown in
+V CC
External
Pin
Internal
Circuitry
–V CC
FIGURE 10. Internal ESD Protection.
Figure 10.
These diodes provide moderate protection to input overdrive
voltages above the supplies as well. The protection diodes
can typically support 30mA continuous current. Where higher
currents are possible (e.g., in systems with ±15V supply parts
driving into the OPA692), current-limiting series resistors
should be added into the two inputs. Keep these resistor
values as low as possible since high values degrade both
noise performance and frequency response.
OPA692
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Revision History
DATE
REVISION
PAGE
12/08
E
2
3/06
D
16-17
SECTION
DESCRIPTION
Absolute Maximum Ratings Changed minimum Storage Temperature Range from −40°C to −65°C.
Design-In Tools
Board part number changed.
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
OPA692
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21
PACKAGE OPTION ADDENDUM
www.ti.com
13-Aug-2021
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
OPA692ID
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
OPA
692
OPA692IDBVR
ACTIVE
SOT-23
DBV
6
3000
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
OAGI
OPA692IDBVT
ACTIVE
SOT-23
DBV
6
250
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
OAGI
OPA692IDBVTG4
ACTIVE
SOT-23
DBV
6
250
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
OAGI
OPA692IDR
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
OPA
692
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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