OPA830IDBVRG4

OPA830 SBOS263F − AUGUST 2004 − REVISED AUGUST 2008 Low-Power, Single-Supply, Wideband Operational Amplifier FEATURES DESCRIPTION D HIGH BANDWIDTH: D D D D D D D 250MHz (G = +1) 110MHz (G = +2) LOW SUPPLY CURRENT: 3.9mA (VS = +5V) FLEXIBLE SUPPLY RANGE: ±1.4V to ±5.5V Dual Supply +2.8V to +11V Single Supply INPUT RANGE INCLUDES GROUND ON SINGLE SUPPLY 4.88V OUTPUT SWING ON +5V SUPPLY HIGH SLEW RATE: 550V/µs LOW INPUT VOLTAGE NOISE: 9.2nV/√Hz Pb-FREE SOT23 PACKAGE APPLICATIONS D SINGLE-SUPPLY ANALOG-TO-DIGITAL D D D D D RELATED PRODUCTS DESCRIPTION +3V +3V 374Ω VIN 100Ω THS1040 10−Bit 40MSPS OPA830 Low distortion operation is ensured by the high gain bandwidth product (110MHz) and slew rate (550V/µs), making the OPA830 an ideal input buffer stage to 3V and 5V CMOS ADCs. Unlike other low-power, single-supply amplifiers, distortion performance improves as the signal swing is decreased. A low 9.2nV/√Hz input voltage noise supports wide dynamic range operation. The OPA830 is available in an industry-standard SO-8 package. The OPA830 is also available in an ultra-small SOT23-5 package. For fixed-gain line driver applications, consider the OPA832. CONVERTER (ADC) INPUT BUFFERS SINGLE-SUPPLY VIDEO LINE DRIVERS CCD IMAGING CHANNELS LOW-POWER ULTRASOUND PLL INTEGRATORS PORTABLE CONSUMER ELECTRONICS 2.26kΩ The OPA830 is a low-power, single-supply, wideband, voltage-feedback amplifier designed to operate on a single +3V or +5V supply. Operation on ±5V or +10V supplies is also supported. The input range extends below the negative supply and to within 1.7V of the positive supply. Using complementary common-emitter outputs provides an output swing to within 25mV of either supply while driving 150Ω. High output drive current (±80mA) and low differential gain and phase errors also make them ideal for single-supply consumer video products. Rail-to-Rail Rail-to-Rail Fixed Gain General-Purpose (1800V/µs slew rate) Low-Noise, High DC Precision SINGLES DUALS TRIPLES QUADS — OPA832 OPA2830 OPA2832 — OPA3832 OPA4830 — OPA690 OPA2690 OPA3690 — OPA820 OPA2822 — OPA4820 22pF 562Ω 750Ω DC-Coupled, +3V ADC Driver Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. All trademarks are the property of their respective owners. Copyright  2004−2008, Texas Instruments Incorporated
          
             
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"#$ www.ti.com SBOS263F − AUGUST 2004 − REVISED AUGUST 2008 This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with appropriate precautions. Failure to observe proper handling and installation procedures can cause damage. ABSOLUTE MAXIMUM RATINGS(1) Power Supply . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12VDC Internal Power Dissipation . . . . . . . . . . . . . . See Thermal Analysis Differential Input Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±2.5V Input Voltage Range (Single Supply) . . . . . . . −0.5V to +VS + 0.3V Storage Temperature Range: D, DBV . . . . . . . . . −65°C to +125°C Lead Temperature (soldering, 10s) . . . . . . . . . . . . . . . . . . . . +300°C Junction Temperature (TJ) . . . . . . . . . . . . . . . . . . . . . . . . . . . +150°C ESD Rating: Human Body Model (HBM) . . . . . . . . . . . . . . . . . . . . . . . 2000V Charge Device Model (CDM) . . . . . . . . . . . . . . . . . . . . . 1500V Machine Model (MM) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 200V (1) Stresses above these ratings may cause permanent damage. Exposure to absolute maximum conditions for extended periods may degrade device reliability. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those specified is not supported. ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more susceptible to damage because very small parametric changes could cause the device not to meet its published specifications. PACKAGE/ORDERING INFORMATION(1) PRODUCT PACKAGE-LEAD PACKAGE DESIGNATOR SPECIFIED TEMPERATURE RANGE PACKAGE MARKING OPA830 SO-8 Surface-Mount D −40°C to +85°C OPA830 OPA830ID Rails, 100 ″ ″ ″ ″ ″ OPA830IDR Tape and Reel, 2500 OPA830 SOT23-5 DBV −40°C to +85°C A72 OPA830IDBVT Tape and Reel, 250 ″ ″ ″ ″ ″ OPA830IDBVR Tape and Reel, 3000 ORDERING NUMBER TRANSPORT MEDIA, QUANTITY (1) For the most current package and ordering information, see the Package Option Addendum at the end of this document, or see the TI website at www.ti.com. PIN CONFIGURATIONS NC 1 8 NC Inverting Input 2 7 +VS Noninverting Input 3 6 Output −VS 4 5 NC Output 1 −VS 2 Noninverting Input 3 5 +VS 4 Inverting Input 3 1 A72 2 SO−8 NC = No Connection 4 5 SOT23−5 Pin Orientation/Package Marking 2 
"#$ www.ti.com SBOS263F − AUGUST 2004 − REVISED AUGUST 2008 ELECTRICAL CHARACTERISTICS: VS = ±5V Boldface limits are tested at +25°C. At TA = 25°C, G = +2, RF = 750Ω, and RL = 150Ω to GND, unless otherwise noted (see Figure 3). OPA830ID, IDBV PARAMETER AC PERFORMANCE (see Figure 3) Small-Signal Bandwidth Gain-Bandwidth Product Peaking at a Gain of +1 Slew Rate Rise Time Fall Time Settling Time to 0.1% Harmonic Distortion 2nd-Harmonic 3rd-Harmonic Input Voltage Noise Input Current Noise NTSC Differential Gain NTSC Differential Phase DC PERFORMANCE(4) Open-Loop Voltage Gain Input Offset Voltage Average Offset Voltage Drift Input Bias Current Input Bias Current Drift Input Offset Current Input Offset Current Drift INPUT Negative Input Voltage(5) Positive Input Voltage(5) Common-Mode Rejection Ratio (CMRR) Input Impedance Differential Mode Common-Mode OUTPUT Output Voltage Swing Current Output, Sinking and Sourcing Short-Circuit Current Closed-Loop Output Impedance POWER SUPPLY Minimum Operating Voltage Maximum Operating Voltage Maximum Quiescent Current Minimum Quiescent Current Power-Supply Rejection Ratio (+PSRR) THERMAL CHARACTERISTICS Specification: ID, IDBV Thermal Resistance, qJA D SO-8 DBV SOT23-5 TYP MIN/MAX OVER TEMPERATURE CONDITIONS +25°C +25°C(1) 0°C to 70°C(2) −40°C to +85°C(2) G = +1, VO ≤ 0.2VPP G = +2, VO ≤ 0.2VPP G = +5, VO ≤ 0.2VPP G = +10, VO ≤ 0.2VPP G ≥ +10 VO ≤ 0.2VPP G = +2, 2V Step 0.5V Step 0.5V Step G = +2, 1V Step VO = 2VPP, f = 5MHz RL = 150Ω RL ≥ 500Ω RL = 150Ω RL ≥ 500Ω f > 1MHz f > 1MHz 310 120 25 11 110 6 600 3.3 3.5 42 70 18 8 85 68 16 7 82 65 15 6 80 280 5.8 5.9 63 270 5.85 5.95 65 260 5.9 6.0 66 −67 −71 −60 −77 9.5 3.7 0.07 0.17 −59 −62 −50 −65 10.5 4.7 −57 −61 −49 −62 11.0 5.2 74 ±1.5 — +5 66 ±7 UNITS MIN/ MAX TEST LEVEL (3) MHz MHz MHz MHz MHz dB V/µs ns ns ns typ min min min min typ min max max max C B B B B C B B B B −56 −60 −48 −59 11.5 5.7 dBc dBc dBc dBc nV/√Hz pA/√Hz % ° max max max max max max typ typ B B B B B B C C 65 ±8.1 ±25 +12 ±12 ±1.2 ±5 64 ±8.6 ±25 +13 ±12 ±1.4 ±5 dB mV µV/°C µA nA/°C µA nA/°C min max max max max max max A A B A B A B −5.3 3.0 74 −5.2 2.9 72 V V dB max min min A A A kΩ  pF kΩ  pF typ typ C C V V mA mA Ω min min min typ typ A A A C C V V mA mA dB typ max max min min C A A A A −40 to +85 °C typ C 125 150 °C/W °C/W typ typ C C RL = 150Ω VCM = 0V VCM = 0V Input-Referred +10 ±0.1 — ±1 −5.5 3.2 80 −5.4 3.1 76 10 2.1 400 1.2 G = +2, RL = 1kΩ to GND G = +2, RL = 150Ω to GND Output Shorted to Ground G = +2, f ≤ 100kHz ±4.88 ±4.64 ±85 150 0.06 ±4.86 ±4.60 ±65 ±4.85 ±4.58 ±60 ±4.84 ±4.56 ±55 ±5.5 4.7 4.0 61 ±5.5 5.3 3.6 60 ±5.5 5.9 3.3 59 ±1.4 VS = ±5V VS = ±5V Input-Referred 4.25 4.25 66 (1) Junction temperature = ambient for +25°C specifications. (2) Junction temperature = ambient at low temperature limits; junction temperature = ambient +5°C at high temperature limit for over temperature specifications. (3) Test levels: (A) 100% tested at +25°C. Over temperature limits by characterization and simulation. (B) Limits set by characterization and simulation. (C) Typical value only for information. (4) Current is considered positive out of pin. (5) Tested < 3dB below minimum specified CMRR at ± CMIR limits. 3 
"#$ www.ti.com SBOS263F − AUGUST 2004 − REVISED AUGUST 2008 ELECTRICAL CHARACTERISTICS: VS = +5V Boldface limits are tested at +25°C. At TA = 25°C, G = +2, RF = 750Ω, and RL = 150Ω to VS/2, unless otherwise noted (see Figure 1). OPA830ID, IDBV PARAMETER AC PERFORMANCE (see Figure 1) Small-Signal Bandwidth Gain-Bandwidth Product Peaking at a Gain of +1 Slew Rate Rise Time Fall Time Settling Time to 0.1% Harmonic Distortion 2nd-Harmonic 3rd-Harmonic Input Voltage Noise Input Current Noise NTSC Differential Gain NTSC Differential Phase DC PERFORMANCE(4) Open-Loop Voltage Gain Input Offset Voltage Average Offset Voltage Drift Input Bias Current Input Bias Current Drift Input Offset Current Input Offset Current Drift INPUT Least Positive Input Voltage(5) Most Positive Input Voltage(5) Common-Mode Rejection Ratio (CMRR) Input Impedance Differential-Mode Common-Mode OUTPUT Least Positive Output Voltage Most Positive Output Voltage Current Output, Sourcing and Sinking Short-Circuit Output Current Closed-Loop Output Impedance POWER SUPPLY Minimum Operating Voltage Maximum Operating Voltage Maximum Quiescent Current Minimum Quiescent Current Power-Supply Rejection Ratio (PSRR) THERMAL CHARACTERISTICS Specification: ID, IDBV Thermal Resistance, qJA D SO-8 DBV SOT23-5 (1) TYP MIN/MAX OVER TEMPERATURE CONDITIONS +25°C +25°C(1) 0°C to 70°C(2) −40°C to +85°C(2) G = +1, VO ≤ 0.2VPP G = +2, VO ≤ 0.2VPP G = +5, VO ≤ 0.2VPP G = +10, VO ≤ 0.2VPP G ≥ +10 VO ≤ 0.2VPP G = +2, 2V Step 0.5V Step 0.5V Step G = +2, 1V Step VO = 2VPP, f = 5MHz RL = 150Ω RL ≥ 500Ω RL = 150Ω RL ≥ 500Ω f > 1MHz f > 1MHz 250 110 24 11 110 5 550 3.3 3.3 43 72 17 8 84 70 16 7 80 68 15 6 79 280 5.7 5.7 64 270 5.8 5.8 66 260 5.9 5.9 67 −62 −64 −58 −84 9.2 3.5 0.08 0.09 −55 −58 −50 −66 10.2 4.5 −54 −57 −49 −63 10.7 5.0 72 ±0.5 — +5 66 ±5.0 ±0.1 — ±0.8 −0.5 3.2 80 −0.4 3.1 76 TEST LEVEL UNITS MIN/ MAX MHz MHz MHz MHz MHz dB V/µs ns ns ns typ min min min min typ min max max max C B B B B C B B B B −53 −56 −48 −60 11.2 5.5 dBc dBc dBc dBc nV/√Hz pA/√Hz % ° max max max max max max typ typ B B B B B B C C 65 ±6.0 ±20 +12 ±12 ±1 ±5 64 ±6.5 ±20 +13 ±12 ±1.2 ±5 dB mV µV/°C µA nA/°C µA nA/°C min max max max max max max A A B A B A B −0.3 3.0 74 −0.2 2.9 72 V V dB max min min A A A kΩ  pF kΩ  pF typ typ C C V V V V mA mA Ω max max min min min typ typ A A A A A C C V V mA mA dB typ max max min min C A A A A −40 to +85 °C typ C 125 150 °C/W °C/W typ typ C C (3) RL = 150Ω VCM = 2.5V VCM = 2.5V Input-Referred +10 10 2.1 4001.2 G = +5, RL = 1kΩ to 2.5V G = +5, RL = 150Ω to 2.5V G = +5, RL = 1kΩ to 2.5V G = +5, RL = 150Ω to 2.5V Output Shorted to Either Supply G = +2, f ≤ 100kHz 0.09 0.21 4.91 4.78 ±80 140 0.06 0.11 0.24 4.89 4.75 ±60 0.12 0.25 4.88 4.73 ±55 0.13 0.26 4.87 4.72 ±52 +11 4.1 3.7 61 +11 4.8 3.4 60 +11 5.5 3.1 59 +2.8 VS = +5V VS = +5V Input-Referred 3.9 3.9 66 Junction temperature = ambient for +25°C specifications. Junction temperature = ambient at low temperature limits; junction temperature = ambient +5°C at high temperature limit for over temperature. (3) Test levels: (A) 100% tested at +25°C. Over temperature limits by characterization and simulation. (B) Limits set by characterization and simulation. (C) Typical value only for information. (4) Current considered positive out of pin. (5) Tested < 3dB below minimum specified CMRR at ± CMIR limits. (2) 4 
"#$ www.ti.com SBOS263F − AUGUST 2004 − REVISED AUGUST 2008 ELECTRICAL CHARACTERISTICS: VS = +3V Boldface limits are tested at +25°C. At TA = 25°C, G = +2, and RL = 150Ω to VS/3, unless otherwise noted (see Figure 2). OPA830ID, IDBV TYP PARAMETER AC PERFORMANCE (see Figure 2) Small-Signal Bandwidth Gain-Bandwidth Product Slew Rate Rise Time Fall Time Settling Time to 0.1% Harmonic Distortion 2nd-Harmonic 3rd-Harmonic Input Voltage Noise Input Current Noise DC PERFORMANCE(4) Open-Loop Voltage Gain Input Offset Voltage Average Offset Voltage Drift Input Bias Current Input Bias Current Drift Input Offset Current Input Offset Current Drift OUTPUT Least Positive Output Voltage Most Positive Output Voltage Current Output, Sourcing Current Output, Sinking Short-Circuit Output Current Closed-Loop Output Impedance POWER SUPPLY Minimum Operating Voltage Maximum Operating Voltage Maximum Quiescent Current Minimum Quiescent Current Power-Supply Rejection Ratio (PSRR) UNITS MIN/ MAX TEST LEVEL (3) CONDITIONS +25°C +25°C(1) 0°C to 70°C(2) G = +2, VO ≤ 0.2VPP G = +5, VO ≤ 0.2VPP G = +10, VO ≤ 0.2VPP G ≥ +10 1V Step 0.5V Step 0.5V Step 1V Step VO = 1VPP, f = 5MHz RL = 150Ω RL ≥ 500Ω RL = 150Ω RL ≥ 500Ω f > 1MHz f > 1MHz 100 22 10 100 225 3.3 3.3 45 72 17 8 80 140 5.5 5.5 72 68 16 7 76 110 5.6 5.6 87 MHz MHz MHz MHz V/µs ns ns ns min min min min min max max max B B B B B B B B −67 −67 −66 −77 9.2 3.5 −61 −61 −59 −59 10.2 4.5 −59 −59 −58 −58 10.7 5.0 dBc dBc dBc dBc max max max max nV/√Hz pA/√Hz max max B B B B B B 72 ±1.5 — +5 66 ±7 65 ±8.1 ±25 +12 ±12 ±1.2 ±5 dB mV µV/°C µA nA/°C µA nA/°C min max max max max max max A A B A B A B −0.27 1.0 73 V V dB max min min A A A kΩ  pF kΩ  pF typ typ C C V V V V mA mA mA Ω max max min min min min typ typ A A A A A A C C V V mA mA dB min max max min min B A A A A −40 to +85 °C typ C 125 150 °C/W °C/W typ typ C C VCM = 1.0V VCM = 1.0V INPUT Least Positive Input Voltage(5) Most Positive Input Voltage(5) Common-Mode Rejection Ratio (CMRR) Input Impedance Differential-Mode Common-Mode MIN/MAX OVER TEMPERATURE Input-Referred +10 ±0.1 — ±1 −0.45 1.2 80 −0.4 1.1 75 10 2.1 4001.2 G = +5, RL = 1kΩ to 1.5V G = +5, RL = 150Ω to 1.5V G = +5, RL = 1kΩ to 1.5V G = +5, RL = 150Ω to 1.5V Output Shorted to Either Supply See Figure 2, f < 100kHz 0.08 0.17 2.91 2.82 30 30 45 0.06 0.11 0.39 2.88 2.74 20 20 0.125 0.40 2.85 2.70 18 18 +11 4.0 3.3 60 +11 4.7 3.1 58 +2.8 VS = +3V VS = +3V Input-Referred THERMAL CHARACTERISTICS Specification: ID, IDBV Thermal Resistance, qJA D SO-8 DBV SOT23-5 3.7 3.7 64 (1) Junction temperature = ambient for +25°C specifications. (2) Junction temperature = ambient at low temperature limits; junction temperature = ambient +5°C at high temperature limit for over temperature. (3) Test levels: (A) 100% tested at +25°C. Over temperature limits by characterization and simulation. (B) Limits set by characterization and simulation. (C) Typical value only for information. (4) Current considered positive out of pin. (5) Tested < 3dB below minimum specified CMRR at ± CMIR limits. 5 
"#$ www.ti.com SBOS263F − AUGUST 2004 − REVISED AUGUST 2008 TYPICAL CHARACTERISTICS: VS = ±5V At TA = 25°C, G = +2, RF = 750Ω, and RL = 150Ω to GND, unless otherwise noted (see Figure 3). INVERTING SMALL−SIGNAL FREQUENCY RESPONSE NONINVERTING SMALL−SIGNAL FREQUENCY RESPONSE 3 6 0 0 Normalized Gain (dB) Normalized Gain (dB) G = −2 G = +1 RF = 0Ω 3 G = +2 −3 G = +5 −6 −9 G = +10 −12 VO = 0.2VPP RL = 150Ω See Figure 3 −15 −18 1 G = −1 −3 −6 G = −5 −9 G = −10 −12 VO = 0.2VPP RL = 150Ω −15 −18 10 100 600 1 10 100 400 Frequency (MHz) Frequency (MHz) NONINVERTING LARGE−SIGNAL FREQUENCY RESPONSE INVERTING LARGE−SIGNAL FREQUENCY RESPONSE 9 3 6 0 3 −3 VO = 2VPP Gain (dB) 0 VO = 1VPP −3 VO = 4VPP −6 G = +2V/V RL = 150Ω See Figure 3 −9 −18 100 500 10 Frequency (MHz) 1.0 0.1 0 0.5 Small−Signal ± 100mV Left Scale 0 −0.1 −0.5 −0.2 −1.0 −0.3 −1.5 −0.4 −2.0 Time (10ns/div) 6 1.5 Small−Signal Output Voltage (100mV/div) 0.2 Large−Signal Output Voltage (500mV/div) Small−Signal Output Voltage (100mV/div) 0.3 Large−Signal ± 1V Right Scale 400 INVERTING PULSE RESPONSE 2.0 G = +2V/V See Figure 3 100 Frequency (MHz) NONINVERTING PULSE RESPONSE 0.4 VO = 4VPP G = −1V/V RL = 150Ω −15 VO = 0.5VPP 10 −9 −12 VO = 2VPP −12 VO = 0.5VPP −6 0.4 2.0 G = −1V/V 0.3 1.5 0.2 1.0 0.1 0 0.5 Small−Signal ± 100mV Left Scale −0.1 −0.2 −0.3 0 −0.5 Large−Signal ± 1V Right Scale −0.4 −1.0 −1.5 −2.0 Time (10ns/div) Large−Signal Output Voltage (500mV/div) Gain (dB) VO = 1VPP 
"#$ www.ti.com SBOS263F − AUGUST 2004 − REVISED AUGUST 2008 TYPICAL CHARACTERISTICS: VS = ±5V (continued) At TA = 25°C, G = +2, RF = 750Ω, and RL = 150Ω to GND, unless otherwise noted (see Figure 3). 5MHz HARMONIC DISTORTION vs SUPPLY VOLTAGE HARMONIC DISTORTION vs LOAD RESISTANCE −40 −60 3rd−Harmonic −65 −70 2nd−Harmonic −75 −80 −50 −55 Input Limited for VCM = 0V −60 −65 2nd−Harmonic −70 −75 −80 3rd−Harmonic −85 −85 100 −90 2.0 1000 2.5 3.0 Resistance (Ω ) −55 −70 −75 2nd−Harmonic −80 3rd−Harmonic −85 −90 5.0 5.5 −60 VO = 2VPP G = +2V/V See Figure 3 −65 2nd−Harmonic RL = 500Ω −70 −75 −80 −85 −90 3rd−Harmonic RL = 150Ω 2nd−Harmonic RL = 150Ω 3rd−Harmonic RL = 500Ω −95 −100 −95 −105 0.1 1 10 0.1 1 Output Voltage Swing (VPP) TWO−TONE, 3RD−ORDER INTERMODULATION SPURIOUS SUPPLY AND OUTPUT CURRENT vs TEMPERATURE 95 6.0 90 5.5 50Ω PO OPA830 20MHz 500Ω −50 750Ω −55 −60 750Ω 10MHz −65 −70 −75 5MHz −80 85 80 4.5 75 4.0 Supply Current Right Scale 50 −85 −90 −26 −20 −14 −8 −2 Single−Tone Load Power (2dBm/div) 6 5.0 Source/Sink Output Current Left Scale 25 −50 3.5 Supply Current (4mA/div) PI Output Current (50mA/div) −45 10 Frequency (MHz) −40 3rd−Order Spurious Level (dBc) 4.5 HARMONIC DISTORTION vs FREQUENCY Harmonic Distortion (dBc) Harmonic Distortion (dBc) −65 4.0 −50 f = 5MHz RL = 500Ω G = +2V/V See Figure 3 −60 3.5 Supply Voltage (±VS) HARMONIC DISTORTION vs OUTPUT VOLTAGE −55 VO = 2VPP RL = 500Ω G = +2V/V See Figure 3 −45 f = 5MHz VO = 2VPP G = +2V/V See Figure 3 −55 Harmonic Distortion (dBc) Harmonic Distortion (dBc) −50 3.0 −25 0 25 50 75 100 125 Ambient Temperature (_ C) 7 
"#$ www.ti.com SBOS263F − AUGUST 2004 − REVISED AUGUST 2008 TYPICAL CHARACTERISTICS: VS = ±5V (continued) At TA = 25°C, G = +2, RF = 750Ω, and RL = 150Ω to GND, unless otherwise noted (see Figure 3). RECOMMENDED RS vs CAPACITIVE LOAD 120 CL = 10pF 7 6 5 100 90 CL = 1000pF 4 80 3 2 1 50Ω VO OPA830 CL −1 750Ω 60 40 1kΩ(1) 750Ω −2 70 50 RS VI 0 0dB Peaking Targeted 110 CL = 100pF RS (Ω) Normalized Gain to Capacitive Load (dB) FREQUENCY RESPONSE vs CAPACITIVE LOAD 8 30 NOTE: (1) 1kΩ is optional. 20 −3 10 1 10 100 200 1 10 Frequency (MHz) 5 5 4 4 3 3 2 2 1 G = +5V/V VS = ±5V −2 −3 −3 −4 −4 −5 −5 10 100 Resistance (Ω ) 1k 1W Internal Power Lim it Output Current Lim it RL = 500Ω RL = 50Ω RL = 100Ω 0 −1 −2 −6 8 VO (V) Output Voltage (V) 6 0 −1 1k OUTPUT VOLTAGE AND CURRENT LIMITATIONS OUTPUT SWING vs LOAD RESISTANCE 6 1 100 Capacitive Load (pF) Output 1W Internal Current Limit P ower Limit −6 −160 −120 −80 −40 0 IO (mA) 40 80 120 160 
"#$ www.ti.com SBOS263F − AUGUST 2004 − REVISED AUGUST 2008 TYPICAL CHARACTERISTICS: VS = ±5V, Differential Configuration At TA = 25°C, GD = +2, RF = 604Ω, and RL = 500Ω, unless otherwise noted. DIFFERENTIAL SMALL−SIGNAL FREQUENCY RESPONSE +5V 3 GD = 1 OPA830 Normalized Gain (dB) 0 −5V 6 0 4Ω RG 6 0 4Ω VI RL 50 0 Ω +5V VO RG −6 GD = 5 −9 G D = 10 −12 OPA830 −5V GD = 2 −3 GD = −15 604Ω RG VO = 200mVPP RL = 500Ω 1 10 100 200 Frequency (MHz) DIFFERENTIAL LARGE−SIGNAL FREQUENCY RESPONSE DIFFERENTIAL DISTORTION vs LOAD RESISTANCE −45 9 VO = 5VPP Gain (dB) 3 0 VO = 2VPP −3 VO = 1VPP −6 GD = 2 RL = 500Ω −9 Harmonic Distortion (dBc) −50 6 VO = 200mVPP 1 10 100 −55 3rd−Harmonic −60 −65 −70 VO = 4VPP GD = 2 f = 5MHz −75 −80 −85 −90 2nd−Harmonic −95 −100 100 200 150 300 350 400 450 500 −55 GD = 2 VO = 4VPP RL = 500Ω 3rd−Harmonic −70 −80 −90 −100 GD = 2 RL = 500Ω f = 5MHz −60 Harmonic Distrtion (dBc) Harmonic Distortion (dBc) −60 250 DIFFERENTIAL DISTORTION vs OUTPUT VOLTAGE DIFFERENTIAL DISTORTION vs FREQUENCY −40 −50 200 Resistance (Ω) Frequency (MHz) −65 −70 3rd−Harmonic −75 −80 −85 2nd−Harmonic −90 −95 −100 2nd−Harmonic −110 −105 0.1 1 10 Frequency (MHz) 100 1 10 Output Voltage Swing (VPP) 9 
"#$ www.ti.com SBOS263F − AUGUST 2004 − REVISED AUGUST 2008 TYPICAL CHARACTERISTICS: VS = +5V At TA = 25°C, G = +2, RF = 750Ω, RL = 150Ω to VS/2, and input VCM = 2.5V, unless otherwise noted (see Figure 1). INVERTING SMALL−SIGNAL FREQUENCY RESPONSE NONINVERTING SMALL−SIGNAL FREQUENCY RESPONSE 3 6 G = +1 RF = 0Ω G = −2 0 Normalized Gain (dB) 0 G = +2 −3 G = +5 −6 −9 G = +10 −12 VO = 0.2VPP RL = 150Ω See Figure 1 −15 −18 1 G = −1 −3 −6 G = −5 −9 G = −10 −12 VO = 0.2VPP RL = 150Ω See Figure 9 −15 −18 10 100 500 1 10 NONINVERTING LARGE−SIGNAL FREQUENCY RESPONSE 3 6 0 Gain (dB) Gain (dB) −3 VO = 1VPP 0 VO = 0.5VPP −3 −6 VO = 0.5VPP −12 −9 100 500 10 Frequency (MHz) 2.5 3.5 Small−Signal ± 100mV Left Scale 3.0 2.5 2.4 2.0 2.3 1.5 2.2 1.0 2.1 0.5 Time (10ns/div) 10 4.0 Small−Signal Output Voltage (100mV/div) 2.7 Large−Signal Output Voltage (500mV/div) Small−Signal Output Voltage (100mV/div) Large−Signal ± 1V Right Scale 500 INVERTING PULSE RESPONSE 4.5 G = +2V/V See Figure 1 100 Frequency (MHz) NONINVERTING PULSE RESPONSE 2.9 VO = 2VPP G = −1V/V RL = 150Ω See Figure 9 −15 −18 10 2.6 VO = 1VPP −6 −12 VO = 2VPP G = +2V/V RL = 150Ω See Figure 1 −9 2.8 400 INVERTING LARGE−SIGNAL FREQUENCY RESPONSE 9 3 100 Frequency (MHz) Frequency (MHz) 2.9 4.5 G = −1V/V 2.8 4.0 2.7 3.5 2.6 3.0 2.5 Small−Signal ± 100mV Left Scale 2.4 2.3 2.2 2.5 2.0 Large−Signal ± 1V Right Scale 2.1 1.5 1.0 0.5 Time (10ns/div) Large−Signal Output Voltage (500mV/div) Normalized Gain (dB) 3 
"#$ www.ti.com SBOS263F − AUGUST 2004 − REVISED AUGUST 2008 TYPICAL CHARACTERISTICS: VS = +5V (continued) At TA = 25°C, G = +2, RF = 750Ω, RL = 150Ω to VS/2, and input VCM = 2.5V, unless otherwise noted (see Figure 1). HARMONIC DISTORTION vs FREQUENCY HARMONIC DISTORTION vs LOAD RESISTANCE −50 −50 −55 −55 −60 2nd−Harmonic −65 −70 −75 f = 5MHz VO = 2VPP G = +2V/V See Figure 1 −80 −85 −60 Harmonic Distortion (dBc) Harmonic Distortion (dBc) G = +2V/V VO = 2VPP See Figure 1 3rd−Harmonic 3rd−Harmonic R L = 150Ω 2nd−Harmonic RL = 500Ω −65 −70 −75 2nd−Harmonic RL = 150Ω −80 −85 −90 −95 3rd−Harmonic RL = 500Ω −100 −105 −90 100 1000 0.1 1 HARMONIC DISTORTION vs OUTPUT VOLTAGE HARMONIC DISTORTION vs NONINVERTING GAIN −45 −60 Input Limited Harmonic Distortion (dBc) −55 Harmonic Distortion (dBc) −55 f = 5MHz RL = 500Ω G = +2V/V See Figure 1 −50 −65 −70 −75 2nd−Harmonic −80 −85 −90 −60 2nd−Harmonic −65 −70 −75 −80 f = 5MHz RL = 500Ω VO = 2VPP See Figure 1 3rd−Harmonic −85 3rd−Harmonic −95 −100 −90 0.1 1 10 1 10 Output Voltage Swing (VPP) Gain (V/V) TWO−TONE, 3RD−ORDER INTERMODULATION SPURIOUS HARMONIC DISTORTION vs INVERTING GAIN −45 3rd−Order Spurious Level (dBc) −55 Harmonic Distortion (dBc) 10 Frequency (MHz) Load Resistance (Ω ) −60 2nd−Harmonic −65 −70 3rd−Harmonic −75 f = 5MHz RL = 500Ω VO = 2VPP −80 −85 1 10 Gain ( V/V ) −50 −55 PI 50Ω PO OPA830 500Ω 20MHz 750Ω −60 −65 750Ω 10MHz −70 −75 −80 5MHz −85 −90 −95 −26 −24 −22 −20 −18 −16 −14 −12 −10 −8 −6 −4 −2 Single−Tone Load Power (dBm) 11 
"#$ www.ti.com SBOS263F − AUGUST 2004 − REVISED AUGUST 2008 TYPICAL CHARACTERISTICS: VS = +5V (continued) At TA = 25°C, G = +2, RF = 750Ω, RL = 150Ω to VS/2, and input VCM = 2.5V, unless otherwise noted (see Figure 1). INPUT VOLTAGE AND CURRENT NOISE DENSITY CLOSED−LOOP OUTPUT IMPEDANCE vs FREQUENCY 100 Output Impedance (Ω) Voltage Noise (nV/√Hz) Current Noise (pA/√Hz) 100 Voltage Noise (9.2nV/√Hz) 10 Current Noise (3.5pA/√Hz) 10 1 0.1 0.01 1 10 100 1k 10k 100k 1M 10M 1k 10k 100k Normalized Gain to Capacitive Load (dB) 0dB Peaking Targeted 110 100 RS (Ω) 90 80 70 60 50 40 30 20 10 1 10 100 8 6 CL = 100pF 5 CL = 1000pF 4 3 2 1 RS VI 50Ω 0 VO O P A830 CL −1 750Ω NOTE: (1) 1kΩis optional. −3 1k 1 10 −20 4.5 −40 4.0 20 log (AOL) −60 −80 40 ∠(AOL) −120 10 −140 0 −160 −10 −180 1k 10k 100k 1M Frequency (Hz) 12 −100 10M 100M −200 1G Voltage Range (V) 70 Open−Loop Phase (_ ) Open−Loop Gain (dB) 5.0 −20 100 300 VOLTAGE RANGES vs TEMPERATURE 0 20 100 Frequency (MHz) OPEN−LOOP GAIN AND PHASE 30 1kΩ(1) 750Ω −2 80 50 100M CL = 10pF 7 Capacitive Load (pF) 60 10M FREQUENCY RESPONSE vs CAPACITIVE LOAD RECOMMENDED RS vs CAPACITIVE LOAD 130 120 1M Frequency (Hz) Frequency (Hz) Most Positive Output Voltage 3.5 3.0 2.5 Most Positive Input Voltage 2.0 RL = 150Ω 1.5 1.0 Least Positive Output Voltage 0.5 0 −0.5 −1.0 −50 Least Positive Input Voltage 0 50 Ambient Temperature (10_ C/div) 110 
"#$ www.ti.com SBOS263F − AUGUST 2004 − REVISED AUGUST 2008 TYPICAL CHARACTERISTICS: VS = +5V (continued) At TA = 25°C, G = +2, RF = 750Ω, RL = 150Ω to VS/2, and input VCM = 2.5V, unless otherwise noted (see Figure 1). TYPICAL DC DRIFT OVER TEMPERATURE SUPPLY AND OUTPUT CURRENT vs TEMPERATURE 6 2 4 1 2 10 × Input Offset Current (IOS) 0 0 −1 −2 −2 −4 Input Offset Voltage (VOS) −3 −8 −50 −25 −6 25 50 75 100 90 85 5.0 4.5 Output Current, Sinking 80 4.0 75 3.5 70 3.0 Output Current, Sourcing 60 −50 125 2.5 −25 2.0 0 25 50 75 100 125 Ambient Temperature (_C) Ambient Temperature (_ C) CMRR AND PSRR vs FREQUENCY OUTPUT SWING vs LOAD RESISTANCE 90 5.5 80 5.0 4.5 70 CMRR Output Voltage (V) Common−Mode Rejection Ratio (dB) Power−Supply Rejection Ratio (dB) 5.5 Quiescent Current 65 −8 0 6.0 95 Output Current (5mA/div) Input Offset Voltage (mV) 3 100 Supply Current (0.5mA/div) 8 Input Bias Current (IB) Input Bias and Offset Current (µV) 4 60 50 40 30 PSRR 4.0 3.5 G = +5V/V 3.0 2.5 2.0 1.5 1.0 20 0.5 10 0 0 1k 10k 100k 1M Frequency (Hz) 10M 100M −0.5 10 100 1k Load Resistance (Ω ) 13 
"#$ www.ti.com SBOS263F − AUGUST 2004 − REVISED AUGUST 2008 TYPICAL CHARACTERISTICS: VS = +5V, Differential Configuration At TA = 25°C, G = +2, RF = 604Ω, and RL = 500Ω differential, unless otherwise noted. DIFFERENTIAL SMALL−SIGNAL FREQUENCY RESPONSE +5V 3 1.2kΩ GD = 1 2.5V 0 R 0.1µF OPA830 Normalized Gain (dB) 1.2kΩ 60 4Ω G 60 4Ω VI R VO L +5V R GD = 2 −3 −6 GD = 5 −9 GD = 10 −12 G 1.2kΩ 1.2kΩ VO = 200mVPP RL = 500Ω OPA830 2.5V 0.1µF GD = −15 604Ω RG 1 10 100 200 Frequency (MHz) DIFFERENTIAL LARGE−SIGNAL FREQUENCY RESPONSE DIFFERENTIAL DISTORTION vs LOAD RESISTANCE −40 9 VO = 3VPP Gain (dB) 3 VO = 2VPP 0 −3 −6 −9 VO = 1VPP GD = 2 RL = 500Ω 1 Harmonic Distortion (dBc) −45 6 VO = 0.2VPP 10 −50 3rd−Harmonic −55 −60 −65 VO = 4VPP GD = 2 f = 5MHz −70 −75 −80 2nd−Harmonic −85 100 −90 100 200 150 −50 3rd−Harmonic −60 −70 2nd−Harmonic −80 −90 400 450 500 −80 −85 2nd−Harmonic −90 −95 100 3rd−Harmonic −75 −100 10 350 −70 −110 Frequency (MHz) 14 −65 −100 1 300 GD = 2 RL = 500Ω f = 5MHz −60 Harmonic Distrtion (dBc) Harmonic Distrtion (dBc) −55 VO = 4VPP GD = 2 RL = 500Ω −40 250 DIFFERENTIAL DISTORTION vs OUTPUT VOLTAGE DIFFERENTIAL DISTORTION vs FREQUENCY −30 200 Resistance (Ω) Frequency (MHz) 1 10 Output Voltage Swing (VPP) 
"#$ www.ti.com SBOS263F − AUGUST 2004 − REVISED AUGUST 2008 TYPICAL CHARACTERISTICS: VS = +3V At TA = 25°C, G = +2, and RL = 150Ω to VS/3, unless otherwise noted (see Figure 2). INVERTING SMALL−SIGNAL FREQUENCY RESPONSE 3 3 0 Normalized Gain (dB) Normalized Gain (dB) NONINVERTING SMALL−SIGNAL FREQUENCY RESPONSE 6 0 G = +2 −3 G = +5 −6 −9 G = +10 −12 RL = 150Ω VO = 0.2VPP See Figure 2 −15 −18 1 G = −1 −3 G = −2 −6 G = −5 −9 G = −10 −12 RL = 150Ω VO = 0.2VPP −15 −18 10 100 400 1 10 100 300 Frequency (MHz) Frequency (MHz) NONINVERTING LARGE−SIGNAL FREQUENCY RESPONSE INVERTING LARGE−SIGNAL FREQUENCY RESPONSE 9 3 6 0 3 −3 VO = 0.5VPP VO = 1VPP VO = 0.5VPP 0 Gain (dB) VO = 1.5VPP −3 −6 −6 VO = 1.5VPP −9 −12 RL = 150Ω G = +2V/V See Figure 2 −9 −12 −15 RL = 150Ω G = −1V/V −18 10 100 400 10 Frequency (MHz) Large−Signal ± 0.5V Right Scale 1.10 1.05 1.00 1.75 1.50 Small−Signal ± 100mV Left Scale 1.25 1.00 0.95 0.75 0.90 0.50 0.85 0.25 0.80 0 Time (10ns/div) Small−Signal Output Voltage (90mV/div) 1.15 G = +2V/V See Figure 2 300 INVERTING PULSE RESPONSE 2.00 Large−Signal Output Voltage (250mV/div) Small−Signal Output Voltage (90mV/div) NONINVERTING PULSE RESPONSE 1.20 100 Frequency (MHz) 1.20 1.15 2.00 G = −1V/V Large−Signal ± 0.5V Right Scale 1.10 1.05 1.00 1.75 1.50 Small−Signal ± 100mV Left Scale 1.25 1.00 0.95 0.75 0.90 0.50 0.85 0.25 0.80 0 Large−Signal Output Voltage (250mV/div) Gain (dB) VO = 1VPP Time (10ns/div) 15 
"#$ www.ti.com SBOS263F − AUGUST 2004 − REVISED AUGUST 2008 TYPICAL CHARACTERISTICS: VS = +3V (continued) At TA = 25°C, G = +2, and RL = 150Ω to VS/3, unless otherwise noted (see Figure 2). HARMONIC DISTORTION vs LOAD RESISTANCE −50 Harmonic Distortion (dBc) −60 −65 2nd−Harmonic −70 −75 −80 3rd−Harmonic f = 5MHz RL = 500Ω G = +2V/V See Figure 2 −50 Harmonic Distortion (dBc) f = 5MHz VO = 1VPP G = +2V/V See Figure 2 −55 HARMONIC DISTORTION vs OUTPUT VOLTAGE −40 −60 Input Limited −70 −90 −85 −100 −90 100 0.1 1000 1 TWO−TONE, 3RD−ORDER INTERMODULATION SPURIOUS HARMONIC DISTORTION vs FREQUENCY −55 VO = 1VPP G = +2V/V See Figure 2 Harmonic Distortion (dBc) −65 2nd−Harmonic RL = 500Ω −70 −75 3rd−Order Spurious Level (dBc) −40 −60 2nd−Harmonic RL = 150Ω −80 −85 3rd−Harmonic RL = 150Ω −90 3rd−Harmonic RL = 500Ω −95 −100 0.1 1 10 −45 PI 50Ω −50 500Ω 750Ω −55 −60 −65 −70 −75 10MHz −80 −85 5MHz −90 −95 −28 −26 −24 −22 −20 −18 −16 −14 −12 −10 150 130 110 90 70 50 30 10 Capacitive Load (pF) 1k Normalized Gain to Capacitive Load (dB) 0dB Peaking Targeted 170 100 −8 FREQUENCY RESPONSE vs CAPACITIVE LOAD RECOMMENDED RS vs CAPACITIVE LOAD 10 20MHz 750Ω Single−Tone Load Power (dBm) 190 1 PO OPA830 Frequency (MHz) RS (Ω) 10 Output Voltage Swing (VPP ) Resistance (Ω ) 16 3rd−Harmonic 2nd−Harmonic −80 8 CL = 10pF 7 CL = 100pF 6 5 CL = 1000pF 4 3 2 1 RS VI 50Ω 0 VO O P A830 CL 1kΩ(1) 750Ω −1 −2 750Ω NOTE: (1) 1kΩ is optional. −3 1 10 Frequency (MHz) 100 200 
"#$ www.ti.com SBOS263F − AUGUST 2004 − REVISED AUGUST 2008 TYPICAL CHARACTERISTICS: VS = +3V (continued) At TA = 25°C, G = +2, and RL = 150Ω to VS/3, unless otherwise noted (see Figure 2). OUTPUT SWING vs LOAD RESISTANCE 3.5 Output Voltage (V) 3.0 2.5 2.0 G = +5V/V 1.5 1.0 0.5 0 −0.5 10 100 1k Load Resistance (Ω ) 17 
"#$ www.ti.com SBOS263F − AUGUST 2004 − REVISED AUGUST 2008 TYPICAL CHARACTERISTICS: VS = +3V, Differential Configuration At TA = 25°C, G = +2, RF = 604Ω, and RL = 500Ω differential, unless otherwise noted. +3V DIFFERENTIAL SMALL−SIGNAL FREQUENCY RESPONSE 3 2kΩ 1V 0 OPA830 Normalized Gain (dB) 0.1µF 1kΩ 60 4Ω RG 60 4Ω VI RL VO +3V RG GD = 1 −3 GD = 5 −9 GD = 10 −12 2kΩ OPA830 1V −15 0.1µ F 1kΩ GD = GD = 2 −6 604Ω VO = 200mVPP RL = 500Ω 1 10 RG 100 200 Frequency (MHz) DIFFERENTIAL LARGE−SIGNAL FREQUENCY RESPONSE DIFFERENTIAL DISTORTION vs LOAD RESISTANCE −40 9 −45 Gain (dB) 3 Harmonic Distortion (dBc) 6 VO = 2VPP 0 VO = 1VPP −3 VO = 200mVPP −6 −55 −60 −65 1 10 100 −70 −75 −80 2nd−Harmonic −90 100 200 150 GD = 2 VO = 2VPP RL = 500Ω −65 3rd−Harmonic −75 −85 −95 2nd−Harmonic −105 −115 0.1 1 10 Frequency (MHz) 18 300 350 400 450 500 −75 Harmonic Distortion (dBc) Harmonic Distortion (dBc) −55 250 DIFFERENTIAL DISTORTION vs OUTPUT VOLTAGE DIFFERENTIAL DISTORTION vs FREQUENCY −45 200 Resistance (Ω) Frequency (MHz) −35 3rd−Harmonic VO = 4VPP GD = 2 f = 5MHz −85 GD = 2 −9 −50 100 −80 GD = 2 RL = 500Ω f = 5MHz 3rd−Harmonic −85 2nd−Harmonic −90 −95 −100 0.50 0.75 1.00 1.25 1.50 Output Voltage Swing (VPP) 1.75 2.00 
"#$ www.ti.com SBOS263F − AUGUST 2004 − REVISED AUGUST 2008 APPLICATIONS INFORMATION WIDEBAND VOLTAGE-FEEDBACK OPERATION The OPA830 is a unity-gain stable, very high-speed voltage-feedback op amp designed for single-supply operation (+3V to +10V). The input stage supports input voltages below ground and to within 1.7V of the positive supply. The complementary common-emitter output stage provides an output swing to within 25mV of ground and the positive supply. The OPA830 is compensated to provide stable operation with a wide range of resistive loads. Figure 1 shows the AC-coupled, gain of +2 configuration used for the +5V Specifications and Typical Characteristic Curves. For test purposes, the input impedance is set to 50Ω with a resistor to ground. Voltage swings reported in the Electrical Characteristics are taken directly at the input and output pins. For the circuit of Figure 1, the total effective load on the output at high frequencies is 150Ω || 1500Ω. The 1.5kΩ resistors at the noninverting input provide the common-mode bias voltage. Their parallel combination equals the DC resistance at the inverting input (RF), reducing the DC output offset due to input bias current. the Electrical Characteristics are taken directly at the input and output pins. For the circuit of Figure 2, the total effective load on the output at high frequencies is 150Ω || 1500Ω. The 1.13kΩ and 2.26kΩ resistors at the noninverting input provide the common-mode bias voltage. Their parallel combination equals the DC resistance at the inverting input (RF), reducing the DC output offset due to input bias current. VS = +3V 6.8µF + 2.26kΩ 0.1µF 53.6Ω 0.1µF VIN 53.6Ω 0.1µF 2.5V 1.50kΩ VOUT OPA830 RL 150Ω RG 750Ω +VS/2 RF 750Ω 1.13kΩ VOUT OPA830 RL 150Ω RG 750Ω RF 750Ω +VS 3 Figure 2. AC-Coupled, G = +2, +3V Single-Supply Specification and Test Circuit 6.8µF + 1.50kΩ +1V VIN +VS/3 VS = +5V 0.1µF +VS 2 Figure 1. AC-Coupled, G = +2, +5V Single-Supply Specification and Test Circuit Figure 2 shows the AC-coupled, gain of +2 configuration used for the +3V Specifications and Typical Characteristic Curves. For test purposes, the input impedance is set to 50Ω with a resistor to ground. Voltage swings reported in Figure 3 shows the DC-coupled, gain of +2, dual power-supply circuit configuration used as the basis of the ±5V Electrical Characteristics and Typical Characteristics. For test purposes, the input impedance is set to 50Ω with a resistor to ground and the output impedance is set to 150Ω with a series output resistor. Voltage swings reported in the specifications are taken directly at the input and output pins. For the circuit of Figure 3, the total effective load will be 150Ω || 1.5kΩ. Two optional components are included in Figure 3. An additional resistor (348Ω) is included in series with the noninverting input. Combined with the 25Ω DC source resistance looking back towards the signal generator, this gives an input bias current cancelling resistance that matches the 375Ω source resistance seen at the inverting input (see the DC Accuracy and Offset Control section). In addition to the usual power-supply decoupling capacitors to ground, a 0.01µF capacitor is included between the two power-supply pins. In practical PC board layouts, this optional capacitor will typically improve the 2nd-harmonic distortion performance by 3dB to 6dB. 19 
"#$ www.ti.com SBOS263F − AUGUST 2004 − REVISED AUGUST 2008 +5V 0.1µF +VS 6.8µF + R2 50Ω Source R1 348Ω VIN VIN VO 50Ω 150Ω 0.01µF RF 750Ω RG 750Ω + R3 6.8µF 0.1µF −5V Figure 3. DC-Coupled, G = +2, Bipolar Supply Specification and Test Circuit SINGLE-SUPPLY ADC INTERFACE The ADC interface on the front page shows a DC-coupled, single-supply ADC driver circuit. Many systems are now requiring +3V supply capability of both the ADC and its driver. The OPA830 provides excellent performance in this demanding application. Its large input and output voltage ranges and low distortion support converters such as the THS1040 shown in the figure on page 1. The input level-shifting circuitry was designed so that VIN can be between 0V and 0.5V, while delivering an output voltage of 1V to 2V for the THS1040. DC LEVEL-SHIFTING Figure 4 shows a DC-coupled noninverting amplifier that level-shifts the input up to accommodate the desired output voltage range. Given the desired signal gain (G), and the amount VOUT needs to be shifted up (∆VOUT) when VIN is at the center of its range, the following equations give the resistor values that produce the desired performance. Assume that R4 is between 200Ω and 1.5kΩ. NG = G + VOUT/VS R1 = R4/G R2 = R4/(NG − G) R3 = R4/(NG −1) where: NG = 1 + R4/R3 VOUT = (G)VIN + (NG − G)VS Make sure that VIN and VOUT stay within the specified input and output voltage ranges. 20 OPA830 OPA830 VOUT R4 Figure 4. DC Level-Shifting The circuit on the front page is a good example of this type of application. It was designed to take VIN between 0V and 0.5V and produce VOUT between 1V and 2V when using a +3V supply. This means G = 2.00, and ∆VOUT = 1.50V − G × 0.25V = 1.00V. Plugging these values into the above equations (with R4 = 750Ω) gives: NG = 2.33, R1 = 375Ω, R2 = 2.25kΩ, and R3 = 563Ω. The resistors were changed to the nearest standard values for the front page circuit. AC-COUPLED OUTPUT VIDEO LINE DRIVER Low-power and low-cost video line drivers often buffer digital-to-analog converter (DAC) outputs with a gain of 2 into a doubly-terminated line. Those interfaces typically require a DC blocking capacitor. For a simple solution, that interface often has used a very large value blocking capacitor (220µF) to limit tilt, or SAG, across the frames. One approach to creating a very low high-pass pole location using much lower capacitor values is shown in Figure 5. This circuit gives a voltage gain of 2 at the output pin with a high-pass pole at 8Hz. Given the 150Ω load, a simple blocking capacitor approach would require a 133µF value. The two much lower valued capacitors give this same low-pass pole using this simple SAG correction circuit of Figure 5. 
"#$ www.ti.com SBOS263F − AUGUST 2004 − REVISED AUGUST 2008 +5V 1.87kΩ Video DAC 47µF 75Ω VO OPA830 78.7Ω 75Ω Load 22µF 845Ω 325Ω 528Ω 650Ω Figure 5. Video Line Driver with SAG Correction The input is shifted slightly positive in Figure 5 using the voltage divider from the positive supply. This gives about a 200mV input DC offset that will show up at the output pin as a 400mV DC offset when the DAC output is at zero current during the sync tip portion of the video signal. This acts to hold the output in its linear operating region. This will pass on any power-supply noise to the output with a gain of approximately −20dB, so good supply decoupling is recommended on the power-supply pin. Figure 6 shows the frequency response for the circuit of Figure 5. This plot shows the 8Hz low-frequency high-pass pole and a high-end cutoff at approximately 100MHz. impedance source, such as an op amp. The resistor values are low to reduce noise. Using both RT and RF helps minimize the impact of parasitic impedances. +5V RT VIN RC OPA830 RG VOUT RF 3 Normalized Gain (dB) 0 Figure 7. Compensated Noninverting Amplifier −3 −6 The Noise Gain can be calculated as follows: −9 G1 + 1 ) −12 −15 RF RG (1) RF −18 G2 + 1 ) −21 1 10 102 103 104 105 106 107 108 RT ) G 109 Frequency (Hz) Figure 6. Video Line Driver Response to Matched Load NONINVERTING AMPLIFIER WITH REDUCED PEAKING Figure 7 shows a noninverting amplifier that reduces peaking at low gains. The resistor RC compensates the OPA830 to have higher Noise Gain (NG), which reduces the AC response peaking (typically 5dB at G = +1 without RC) without changing the DC gain. VIN needs to be a low NG + G 1 RC G2 1 (2) (3) A unity-gain buffer can be designed by selecting RT = RF = 20.0Ω and RC = 40.2Ω (do not use RG). This gives a noise gain of 2, so the response will be similar to the Characteristics Plots with G = +2. Decreasing RC to 20.0Ω will increase the noise gain to 3, which typically gives a flat frequency response, but with less bandwidth. The circuit in Figure 1 can be redesigned to have less peaking by increasing the noise gain to 3. This is accomplished by adding RC = 2.55kΩ across the op amp inputs. 21 
"#$ www.ti.com SBOS263F − AUGUST 2004 − REVISED AUGUST 2008 SINGLE-SUPPLY ACTIVE FILTER The OPA830, while operating on a single +3V or +5V supply, lends itself well to high-frequency active filter designs. Again, the key additional requirement is to establish the DC operating point of the signal near the supply midpoint for highest dynamic range. Figure 8 shows an example design of a 1MHz low-pass Butterworth filter using the Sallen-Key topology. Both the input signal and the gain setting resistor are AC-coupled using 0.1µF blocking capacitors (actually giving bandpass response with the low-frequency pole set to 32kHz for the component values shown). As discussed for Figure 1, this allows the midpoint bias formed by the two 1.87kΩ resistors to appear at both the input and output pins. The midband signal gain is set to +4 (12dB) in this case. The capacitor to ground on the noninverting input is intentionally set larger to dominate input parasitic terms. At a gain of +4, the OPA830 on a single supply will show 30MHz small- and large-signal bandwidth. The resistor values have been slightly adjusted to account for this limited bandwidth in the amplifier stage. Tests of this circuit show a precise 1MHz, −3dB point with a maximally-flat passband (above the 32kHz AC-coupling corner), and a maximum stop band attenuation of 36dB at the amplifier’s −3dB bandwidth of 30MHz. DESIGN-IN TOOLS DEMONSTRATION BOARDS Two printed circuit boards (PCBs) are available to assist in the initial evaluation of circuit performance using the OPA830 in its two package options. Both of these are offered free of charge as unpopulated PCBs, delivered with a user’s guide. The summary information for these fixtures is shown in Table 1. Table 1. Demonstration Fixtures by Package ORDERING PACKAGE NUMBER NUMBER OPA830ID SO-8 DEM-OPA-SO-1A SBOU009 OPA830IDBV SOT23-5 DEM-OPA-SOT-1A SBOU010 The demonstration fixtures can be requested at the Texas Instruments web site (www.ti.com) through the OPA830 product folder. +5V 100pF 1.87kΩ 0.1µF 137Ω 432Ω VI 1.87kΩ 150pF OPA830 4V I 1MHz, 2nd−Order Butterworth Filter 1.5kΩ 500Ω 0.1µF Figure 8. Single-Supply, High-Frequency Active Filter 22 LITERATURE PRODUCT 
"#$ www.ti.com SBOS263F − AUGUST 2004 − REVISED AUGUST 2008 MACROMODEL AND APPLICATIONS SUPPORT Computer simulation of circuit performance using SPICE is often a quick way to analyze the performance of the OPA830 and its circuit designs. This is particularly true for video and RF amplifier circuits where parasitic capacitance and inductance can play a major role on circuit performance. A SPICE model for the OPA830 is available through the TI web page (www.ti.com). The applications department is also available for design assistance. These models predict typical small signal AC, transient steps, DC performance, and noise under a wide variety of operating conditions. The models include the noise terms found in the electrical specifications of the data sheet. These models do not attempt to distinguish between the package types in their small-signal AC performance. OPERATING SUGGESTIONS OPTIMIZING RESISTOR VALUES Since the OPA830 is a unity-gain stable, voltage-feedback op amp, a wide range of resistor values may be used for the feedback and gain setting resistors. The primary limits on these values are set by dynamic range (noise and distortion) and parasitic capacitance considerations. For a noninverting unity-gain follower application, the feedback connection should be made with a direct short. Below 200Ω, the feedback network will present additional output loading which can degrade the harmonic distortion performance of the OPA830. Above 1kΩ, the typical parasitic capacitance (approximately 0.2pF) across the feedback resistor may cause unintentional band limiting in the amplifier response. A good rule of thumb is to target the parallel combination of RF and RG (see Figure 3) to be less than about 400Ω. The combined impedance RF || RG interacts with the inverting input capacitance, placing an additional pole in the feedback network, and thus a zero in the forward response. Assuming a 2pF total parasitic on the inverting node, holding RF || RG < 400Ω will keep this pole above 200MHz. By itself, this constraint implies that the feedback resistor RF can increase to several kΩ at high gains. This is acceptable as long as the pole formed by RF and any parasitic capacitance appearing in parallel is kept out of the frequency range of interest. In the inverting configuration, an additional design consideration must be noted. RG becomes the input resistor and therefore the load impedance to the driving source. If impedance matching is desired, RG may be set equal to the required termination value. However, at low inverting gains, the resultant feedback resistor value can present a significant load to the amplifier output. For example, an inverting gain of 2 with a 50Ω input matching resistor (= RG) would require a 100Ω feedback resistor, which would contribute to output loading in parallel with the external load. In such a case, it would be preferable to increase both the RF and RG values, and then achieve the input matching impedance with a third resistor to ground (see Figure 9). The total input impedance becomes the parallel combination of RG and the additional shunt resistor. BANDWIDTH vs GAIN: NONINVERTING OPERATION Voltage-feedback op amps exhibit decreasing closed-loop bandwidth as the signal gain is increased. In theory, this relationship is described by the Gain Bandwidth Product (GBP) shown in the specifications. Ideally, dividing GBP by the noninverting signal gain (also called the Noise Gain, or NG) will predict the closed-loop bandwidth. In practice, this only holds true when the phase margin approaches 90°, as it does in high-gain configurations. At low gains (increased feedback factors), most amplifiers will exhibit a more complex response with lower phase margin. The OPA830 is compensated to give a slightly peaked response in a noninverting gain of 2 (see Figure 3). This results in a typical gain of +2 bandwidth of 110MHz, far exceeding that predicted by dividing the 110MHz GBP by 2. Increasing the gain will cause the phase margin to approach 90° and the bandwidth to more closely approach the predicted value of (GBP/NG). At a gain of +10, the 11MHz bandwidth shown in the Electrical Characteristics agrees with that predicted using the simple formula and the typical GBP of 110MHz. Frequency response in a gain of +2 may be modified to achieve exceptional flatness simply by increasing the noise gain to 3. One way to do this, without affecting the +2 signal gain, is to add an 2.55kΩ resistor across the two inputs, as shown in Figure 7. A similar technique may be used to reduce peaking in unity-gain (voltage follower) applications. For example, by using a 750Ω feedback resistor along with a 750Ω resistor across the two op amp inputs, the voltage follower response will be similar to the gain of +2 response of Figure 2. Further reducing the value of the resistor across the op amp inputs will further dampen the frequency response due to increased noise gain. The OPA830 exhibits minimal bandwidth reduction going to single-supply (+5V) operation as compared with ±5V. This minimal reduction is because the internal bias control circuitry retains nearly constant quiescent current as the total supply voltage between the supply pins is changed. INVERTING AMPLIFIER OPERATION All of the familiar op amp application circuits are available with the OPA830 to the designer. See Figure 9 for a typical inverting configuration where the I/O impedances and signal gain from Figure 1 are retained in an inverting circuit configuration. Inverting operation is one of the more common requirements and offers several performance benefits. It also allows the input to be biased at VS/2 without any headroom issues. The output voltage can be independently moved to be within the output voltage range with coupling capacitors, or bias adjustment resistors. 23 
"#$ www.ti.com SBOS263F − AUGUST 2004 − REVISED AUGUST 2008 +5V + 0.1µF 6.8µF 2RT 1.5kΩ 150Ω 0.1µF 50Ω Source 0.1µF 2RT 1.5kΩ RG 374Ω OPA830 +VS 2 RF 750Ω RM 57.6Ω Figure 9. AC-Coupled, G = −2 Example Circuit In the inverting configuration, three key design considerations must be noted. The first consideration is that the gain resistor (RG) becomes part of the signal channel input impedance. If input impedance matching is desired (which is beneficial whenever the signal is coupled through a cable, twisted pair, long PC board trace, or other transmission line conductor), RG may be set equal to the required termination value and RF adjusted to give the desired gain. This is the simplest approach and results in optimum bandwidth and noise performance. However, at low inverting gains, the resulting feedback resistor value can present a significant load to the amplifier output. For an inverting gain of 2, setting RG to 50Ω for input matching eliminates the need for RM but requires a 100Ω feedback resistor. This configuration has the interesting advantage of the noise gain becoming equal to 2 for a 50Ω source impedance—the same as the noninverting circuits considered above. The amplifier output will now see the 100Ω feedback resistor in parallel with the external load. In general, the feedback resistor should be limited to the 200Ω to 1.5kΩ range. In this case, it is preferable to increase both the RF and RG values, as shown in Figure 9, and then achieve the input matching impedance with a third resistor (RM) to ground. The total input impedance becomes the parallel combination of RG and RM. The second major consideration, touched on in the previous paragraph, is that the signal source impedance becomes part of the noise gain equation and hence influences the bandwidth. For the example in Figure 9, the RM value combines in parallel with the external 50Ω source impedance (at high frequencies), yielding an effective driving impedance of 50Ω || 57.6Ω = 26.8Ω. This impedance is added in series with RG for calculating the noise gain. The resulting noise gain is 2.87 for Figure 9, as opposed to only 2 if RM could be eliminated as discussed above. The bandwidth will therefore be lower for the gain 24 of −2 circuit of Figure 9 (NG = +2.87) than for the gain of +2 circuit of Figure 1. The third important consideration in inverting amplifier design is setting the bias current cancellation resistors on the noninverting input (a parallel combination of RT = 750Ω). If this resistor is set equal to the total DC resistance looking out of the inverting node, the output DC error, due to the input bias currents, will be reduced to (Input Offset Current) times RF. With the DC blocking capacitor in series with RG, the DC source impedance looking out of the inverting mode is simply RF = 750Ω for Figure 9. To reduce the additional high-frequency noise introduced by this resistor and power-supply feed-through, RT is bypassed with a capacitor. OUTPUT CURRENT AND VOLTAGES The OPA830 provides outstanding output voltage capability. For the +5V supply, under no-load conditions at +25°C, the output voltage typically swings closer than 90mV to either supply rail. The minimum specified output voltage and current specifications over temperature are set by worst-case simulations at the cold temperature extreme. Only at cold startup will the output current and voltage decrease to the numbers shown in the ensured tables. As the output transistors deliver power, their junction temperatures will increase, decreasing their VBEs (increasing the available output voltage swing) and increasing their current gains (increasing the available output current). In steady-state operation, the available output voltage and current will always be greater than that shown in the over-temperature specifications, since the output stage junction temperatures will be higher than the minimum specified operating ambient. To maintain maximum output stage linearity, no output short-circuit protection is provided. This will not normally be a problem, since most applications include a series matching resistor at the output that will limit the internal power dissipation if the output side of this resistor is shorted to ground. However, shorting the output pin directly to the adjacent positive power-supply pin (8-pin packages) will, in most cases, destroy the amplifier. If additional short-circuit protection is required, consider a small series resistor in the power-supply leads. This will reduce the available output voltage swing under heavy output loads. DRIVING CAPACITIVE LOADS One of the most demanding and yet very common load conditions for an op amp is capacitive loading. Often, the capacitive load is the input of an ADC—including additional external capacitance which may be recommended to improve ADC linearity. A high-speed, high open-loop gain amplifier like the OPA830 can be very susceptible to decreased stability and closed-loop response peaking when a capacitive load is placed directly on the output pin. When the primary considerations are frequency response flatness, pulse response fidelity, and/or distortion, the simplest and most effective solution 
"#$ www.ti.com SBOS263F − AUGUST 2004 − REVISED AUGUST 2008 is to isolate the capacitive load from the feedback loop by inserting a series isolation resistor between the amplifier output and the capacitive load. The Typical Characteristic curves show the recommended RS versus capacitive load and the resulting frequency response at the load. Parasitic capacitive loads greater than 2pF can begin to degrade the performance of the OPA830. Long PC board traces, unmatched cables, and connections to multiple devices can easily exceed this value. Always consider this effect carefully, and add the recommended series resistor as close as possible to the output pin (see the Board Layout Guidelines section). The criterion for setting this RS resistor is a maximum bandwidth, flat frequency response at the load. For a gain of +2, the frequency response at the output pin is already slightly peaked without the capacitive load, requiring relatively high values of RS to flatten the response at the load. Increasing the noise gain will also reduce the peaking (see Figure 7). DISTORTION PERFORMANCE The OPA830 provides good distortion performance into a 150Ω load. Relative to alternative solutions, it provides exceptional performance into lighter loads and/or operating on a single +3V supply. Generally, until the fundamental signal reaches very high frequency or power levels, the 2nd-harmonic will dominate the distortion with a negligible 3rd-harmonic component. Focusing then on the 2nd-harmonic, increasing the load impedance improves distortion directly. Remember that the total load includes the feedback network; in the noninverting configuration (see Figure 3) this is sum of RF + RG, while in the inverting configuration, only RF needs to be included in parallel with the actual load. Running differential suppresses the 2nd-harmonic, as shown in the differential typical characteristic curves. NOISE PERFORMANCE High slew rate, unity-gain stable, voltage-feedback op amps usually achieve their slew rate at the expense of a higher input noise voltage. The 9.2nV/√Hz input voltage noise for the OPA830 however, is much lower than comparable amplifiers. The input-referred voltage noise and the two input-referred current noise terms (2.8pA/√Hz) combine to give low output noise under a wide variety of operating conditions. Figure 10 shows the op amp noise analysis model with all the noise terms included. In this model, all noise terms are taken to be noise voltage or current density terms in either nV/√Hz or pA/√Hz. ENI EO OPA830 RS IBN ERS RF √ 4kTRS RG 4kT RG √ 4kTRF I BI 4kT = 1.6E − 20J at 290_K Figure 10. Noise Analysis Model The total output spot noise voltage can be computed as the square root of the sum of all squared output noise voltage contributors. Equation 4 shows the general form for the output noise voltage using the terms shown in Figure 10: EO + Ǹǒ Ǔ E NI ) ǒI BNRSǓ ) 4kTRS NG 2 ) ǒI BIR FǓ ) 4kTRFNG 2 2 2 (4) Dividing this expression by the noise gain (NG = (1 + RF/RG)) will give the equivalent input-referred spot noise voltage at the noninverting input, as shown in Equation 5: EN + Ǹ ENI ) ǒIBNR SǓ ) 4kTRS ) 2 2 ǒ Ǔ IBIRF NG 2 ) 4kTRF NG (5) Evaluating these two equations for the circuit and component values shown in Figure 1 will give a total output spot noise voltage of 19.3nV/√Hz and a total equivalent input spot noise voltage of 9.65nV/√Hz. This is including the noise added by the resistors. This total input-referred spot noise voltage is not much higher than the 9.2nV/√Hz specification for the op amp voltage noise alone. DC ACCURACY AND OFFSET CONTROL The balanced input stage of a wideband voltage-feedback op amp allows good output DC accuracy in a wide variety of applications. The power-supply current trim for the OPA830 gives even tighter control than comparable products. Although the high-speed input stage does require relatively high input bias current (typically 5µA out of each input terminal), the close matching between them may be used to reduce the output DC error caused by this current. This is done by matching the DC source resistances appearing at the two inputs. Evaluating the configuration of Figure 3 (which has matched DC input 25 
"#$ www.ti.com SBOS263F − AUGUST 2004 − REVISED AUGUST 2008 resistances), using worst-case +25°C input offset voltage and current specifications, gives a worst-case output offset voltage equal to: (NG = noninverting signal gain at DC) ±(NG × VOS(MAX)) + (RF × IOS(MAX)) = ±(2 × 7mV) × (375Ω × 1µA) = ±14.38mV A fine-scale output offset null, or DC operating point adjustment, is often required. Numerous techniques are available for introducing DC offset control into an op amp circuit. Most of these techniques are based on adding a DC current through the feedback resistor. In selecting an offset trim method, one key consideration is the impact on the desired signal path frequency response. If the signal path is intended to be noninverting, the offset control is best applied as an inverting summing signal to avoid interaction with the signal source. If the signal path is intended to be inverting, applying the offset control to the noninverting input may be considered. Bring the DC offsetting current into the inverting input node through resistor values that are much larger than the signal path resistors. This will insure that the adjustment circuit has minimal effect on the loop gain and hence the frequency response. THERMAL ANALYSIS Maximum desired junction temperature will set the maximum allowed internal power dissipation, as described below. In no case should the maximum junction temperature be allowed to exceed 150°C. Operating junction temperature (TJ) is given by TA + P D × q JA. The total internal power dissipation (P D) is the sum of quiescent power (P DQ ) and additional power dissipated in the output stage (P DL ) to deliver load power. Quiescent power is simply the specified no-load supply current times the total supply voltage across the part. PDL will depend on the required output signal and load; though, for resistive loads connected to mid-supply (V S/2), PDL is at a maximum when the output is fixed at a voltage equal to VS/4 or 3V S/4. Under this condition, PDL = V S2 /(16 × R L ), where RL includes feedback network loading. Note that it is the power in the output stage, and not into the load, that determines internal power dissipation. As a worst-case example, compute the maximum TJ using an OPA830 (SOT23-5 package) in the circuit of Figure 1 operating at the maximum specified ambient temperature of +85°C and driving a 150Ω load at mid-supply. PD = 10V × 3.9mA + 52/(16 × (150Ω || 750Ω)) = 51.5mW Maximum TJ = +85°C + (0.051W × 150°C/W) = 93°C. Although this is still well below the specified maximum junction temperature, system reliability considerations may require lower ensured junction temperatures. The 26 highest possible internal dissipation will occur if the load requires current to be forced into the output at high output voltages or sourced from the output at low output voltages. This puts a high current through a large internal voltage drop in the output transistors. BOARD LAYOUT GUIDELINES Achieving optimum performance with a high-frequency amplifier like the OPA830 requires careful attention to board layout parasitics and external component types. Recommendations that will optimize performance include: a) Minimize parasitic capacitance to any AC ground for all of the signal I/O pins. Parasitic capacitance on the output and inverting input pins can cause instability: on the noninverting input, it can react with the source impedance to cause unintentional bandlimiting. To reduce unwanted capacitance, a window around the signal I/O pins should be opened in all of the ground and power planes around those pins. Otherwise, ground and power planes should be unbroken elsewhere on the board. b) Minimize the distance ( < 0.25”) from the power-supply pins to high-frequency 0.1µF decoupling capacitors. At the device pins, the ground and power-plane layout should not be in close proximity to the signal I/O pins. Avoid narrow power and ground traces to minimize inductance between the pins and the decoupling capacitors. Each powersupply connection should always be decoupled with one of these capacitors. An optional supply decoupling capacitor (0.1µF) across the two power supplies (for bipolar operation) will improve 2nd-harmonic distortion performance. Larger (2.2µF to 6.8µF) decoupling capacitors, effective at lower frequency, should also be used on the main supply pins. These may be placed somewhat farther from the device and may be shared among several devices in the same area of the PC board. c) Careful selection and placement of external components will preserve the high-frequency performance. Resistors should be a very low reactance type. Surface-mount resistors work best and allow a tighter overall layout. Metal film or carbon composition axially-leaded resistors can also provide good highfrequency performance. Again, keep their leads and PC board traces as short as possible. Never use wire-wound type resistors in a high-frequency application. Since the output pin and inverting input pin are the most sensitive to parasitic capacitance, always position the feedback and series output resistor, if any, as close as possible to the output pin. Other network components, such as noninverting input termination resistors, should also be placed close to the package. Where double-side component mounting is allowed, place the feedback resistor directly under the package on the other side of the board between the output and inverting input pins. Even with a low parasitic capacitance shunting the external resistors, excessively high resistor values can create 
"#$ www.ti.com SBOS263F − AUGUST 2004 − REVISED AUGUST 2008 significant time constants that can degrade performance. Good axial metal film or surface-mount resistors have approximately 0.2pF in shunt with the resistor. For resistor values > 1.5kΩ, this parasitic capacitance can add a pole and/or zero below 500MHz that can effect circuit operation. Keep resistor values as low as possible consistent with load driving considerations. The 750Ω feedback used in the Typical Characteristics is a good starting point for design. d) Connections to other wideband devices on the board may be made with short direct traces or through onboard transmission lines. For short connections, consider the trace and the input to the next device as a lumped capacitive load. Relatively wide traces (50mils to 100mils) should be used, preferably with ground and power planes opened up around them. Estimate the total capacitive load and set RS from the typical characteristic curve Recommended RS vs Capacitive Load. Low parasitic capacitive loads (< 5pF) may not need an RS since the OPA830 is nominally compensated to operate with a 2pF parasitic load. Higher parasitic capacitive loads without an RS are allowed as the signal gain increases (increasing the unloaded phase margin). If a long trace is required, and the 6dB signal loss intrinsic to a doubly-terminated transmission line is acceptable, implement a matched impedance transmission line using microstrip or stripline techniques (consult an ECL design handbook for microstrip and stripline layout techniques). A 50Ω environment is normally not necessary onboard, and in fact, a higher impedance environment will improve distortion as shown in the distortion versus load plots. With a characteristic board trace impedance defined (based on board material and trace dimensions), a matching series resistor into the trace from the output of the OPA830 is used as well as a terminating shunt resistor at the input of the destination device. Remember also that the terminating impedance will be the parallel combination of the shunt resistor and the input impedance of the destination device; this total effective impedance should be set to match the trace impedance. If the 6dB attenuation of a doubly-terminated transmission line is unacceptable, a long trace can be series-terminated at the source end only. Treat the trace as a capacitive load in this case and set the series resistor value as shown in the typical characteristic curve Recommended RS vs Capacitive Load. This will not preserve signal integrity as well as a doubly-terminated line. If the input impedance of the destination device is low, there will be some signal attenuation due to the voltage divider formed by the series output into the terminating impedance. e) Socketing a high-speed part is not recommended. The additional lead length and pin-to-pin capacitance introduced by the socket can create an extremely troublesome parasitic network which can make it almost impossible to achieve a smooth, stable frequency response. Best results are obtained by soldering the OPA830 onto the board. INPUT AND ESD PROTECTION The OPA830 is built using a very high-speed complementary bipolar process. The internal junction breakdown voltages are relatively low for these very small geometry devices. These breakdowns are reflected in the Absolute Maximum Ratings table. All device pins are protected with internal ESD protection diodes to the power supplies, as shown in Figure 11. +VCC External Pin Internal Circuitry − VCC Figure 11. Internal ESD Protection These diodes provide moderate protection to input overdrive voltages above the supplies as well. The protection diodes can typically support 30mA continuous current. Where higher currents are possible (that is, in systems with ±15V supply parts driving into the OPA830), current-limiting series resistors should be added into the two inputs. Keep these resistor values as low as possible, since high values degrade both noise performance and frequency response. 27 www.ti.com SBOS263F − AUGUST 2004 − REVISED AUGUST 2008 Revision History DATE REV PAGE SECTION 8/08 F 2 Absolute Maximum Ratings 8/07 E 1 Features DESCRIPTION Changed Storage Temperature minimum value from −40°C to −65°C. Changed 550V/ns to 550V/µs. NOTE: Page numbers for previous revisions may differ from page numbers in the current version. 28 PACKAGE OPTION ADDENDUM www.ti.com 10-Dec-2020 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Pins Package Drawing Qty Eco Plan (2) Lead finish/ Ball material MSL Peak Temp Op Temp (°C) Device Marking (3) (4/5) (6) OPA830ID ACTIVE SOIC D 8 75 RoHS & Green NIPDAU Level-2-260C-1 YEAR -40 to 85 OPA 830 OPA830IDBVR ACTIVE SOT-23 DBV 5 3000 RoHS & Green NIPDAU Level-2-260C-1 YEAR -40 to 85 A72 OPA830IDBVT ACTIVE SOT-23 DBV 5 250 RoHS & Green NIPDAU Level-2-260C-1 YEAR -40 to 85 A72 OPA830IDR ACTIVE SOIC D 8 2500 RoHS & Green NIPDAU Level-2-260C-1 YEAR -40 to 85 OPA 830 (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may reference these types of products as "Pb-Free". RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption. Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of