OPA861
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SBOS338G – AUGUST 2005 – REVISED MAY 2013
Wide Bandwidth
Operational Transconductance
Amplifier (OTA)
Check for Samples: OPA861
FEATURES
1
•
•
•
•
•
Wide Bandwidth (80MHz, Open-Loop, G = +5)
High Slew Rate (900V/µs)
High Transconductance (95mA/V)
External IQ-Control
Low Quiescent Current (5.4mA)
APPLICATIONS
•
•
•
•
•
•
•
Video/Broadcast Equipment
Communications Equipment
High-Speed Data Acquisition
Wideband LED Drivers
Control Loop Amplifiers
Wideband Active Filters
Line Drivers
The OTA or voltage-controlled current source can be
viewed as an ideal transistor. Like a transistor, it has
three terminals—a high impedance input (base), a
low-impedance input/output (emitter), and the current
output (collector). The OPA861, however, is selfbiased and bipolar. The output collector current is
zero for a zero base-emitter voltage. AC inputs
centered about zero produce an output current, which
is bipolar and centered about zero. The
transconductance of the OPA861 can be adjusted
with an external resistor, allowing bandwidth,
quiescent current, and gain trade-offs to be
optimized.
Used as a basic building block, the OPA861
simplifies the design of AGC amplifiers, LED driver
circuits for fiber optic transmission, integrators for fast
pulses, fast control loop amplifiers and control
amplifiers for capacitive sensors, and active filters.
The OPA861 is available in SO-8 and SOT23-6
surface-mount packages.
DESCRIPTION
The OPA861 is a versatile monolithic component
designed for wide-bandwidth systems, including high
performance video, RF and IF circuitry. The OPA861
is a wideband, bipolar operational transconductance
amplifier (OTA).
0
−10
R
C1
R
V IN
V OUT
C2
Gain (dB)
−20
−30
10MHz
Low−Pass Filter
−40
20kHz
Low−Pass Filter
−50
−60
−70
−80
1k
10k
100k
1M
10M
100M
1G
Frequency (Hz)
Low−Pass Negative Impedance Converter (NIC) Filter
Frequency Response of 20kHz and 10MHz
Low−Pass NIC Filters
1
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2005–2013, Texas Instruments Incorporated
OPA861
SBOS338G – AUGUST 2005 – REVISED MAY 2013
www.ti.com
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more
susceptible to damage because very small parametric changes could cause the device not to meet its published specifications.
ORDERING INFORMATION (1)
(1)
SPECIFIED
TEMPERATURE
RANGE
PACKAGE
MARKING
PRODUCT
PACKAGE
PACKAGE
DESIGNATOR
OPA861
SO-8
D
–45°C to +85°C
OPA861
OPA861
SOT23-6
DBV
–45°C to +85°C
N5R
ORDERING
NUMBER
TRANSPORT MEDIA,
QUANTITY
OPA861ID
Rails, 75
OPA861IDR
Tape and Reel, 2500
OPA861IDBVT
Tape and Reel, 250
OPA861IDBVR
Tape and Reel, 3000
For the most current package and ordering information, see the Package Option Addendum at the end of this document, or see the TI
web site at www.ti.com.
ABSOLUTE MAXIMUM RATINGS (1)
Power Supply
±6.5VDC
Internal Power Dissipation
See Thermal Information
Differential Input Voltage
±1.2V
Input Common-Mode Voltage Range
±VS
Storage Temperature Range: D
–65°C to +125°C
Lead Temperature (soldering, 10s)
+260°C
Junction Temperature (TJ)
+150°C
ESD Rating:
(1)
(2)
Human Body Model (HBM) (2)
1500V
Charge Device Model (CDM)
1000V
Stresses above these ratings may cause permanent damage. Exposure to absolute maximum conditions for extended periods may
degrade device reliability. These are stress ratings only, and functional operations of the device at these and any other conditions
beyond those specified is not supported.
Pin 2 for the SO-8 package > 500V HBM. Pin 4 for the SOT23-6 package > 500V HBM.
Figure 1. PIN CONFIGURATION
Top View
I Q Adjust
1
8
C
E
2
7
V+ = +5V
B
3
6
NC
V− = −5V
4
5
NC
I Q Adjust
1
6
+VS
−VS
2
5
C
B
3
4
E
SOT23−6
SO−8
2
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SBOS338G – AUGUST 2005 – REVISED MAY 2013
ELECTRICAL CHARACTERISTICS: VS = ±5V
RL = 500Ω and RADJ = 250Ω, unless otherwise noted.
OPA861ID, IDBV
TYP
PARAMETER
MIN/MAX OVER TEMPERATURE
CONDITIONS
+25°C
+25°C (2)
0°C to
70°C (3)
–40°C to
+85°C (3)
UNITS
MIN/
MAX
TEST
LEVEL (1)
G = +5, VO = 200mVPP,
RL = 500Ω
80
77
75
74
MHz
min
B
G = +5, VO = 1VPP
80
MHz
typ
C
G = +5, VO = 5VPP
80
MHz
typ
C
G = +5, VO = 5V Step
900
V/µs
min
B
VO = 1V Step
4.4
ns
typ
C
OTA—Open-Loop (see Figure 33)
AC PERFORMANCE
Bandwidth
Slew Rate
Rise Time and Fall Time
Harmonic Distortion
860
850
840
G = +5, VO = 2VPP, 5MHz
2nd-Harmonic
RL = 500Ω
–68
–55
–54
–53
dB
max
B
3rd-Harmonic
RL = 500Ω
–57
–52
–51
–49
dB
max
B
Base Input Voltage Noise
f > 100kHz
2.4
3.0
3.3
3.4
nV/√Hz
max
B
Base Input Current Noise
f > 100kHz
1.7
2.4
2.45
2.5
pA/√Hz
max
B
Emitter Input Current Noise
f > 100kHz
5.2
15.3
16.6
17.5
pA/√Hz
max
B
Minimum OTA Transconductance (gm)
VO = ±10mV, RC = 50Ω, RE = 0Ω
95
80
77
75
mA/V
min
A
Maximum OTA Transconductance (gm)
VO = ±10mV, RC = 50Ω, RE = 0Ω
95
150
155
160
mA/V
max
A
VB = 0V, RC = 0Ω, RE = 100Ω
±3
±12
±15
±20
mV
max
A
±67
±120
μV/°C
max
B
±6
±6.6
μA
max
A
±20
±25
nA/°C
max
B
±125
±140
μA
max
A
±500
±600
nA/°C
max
B
±30
±38
μA
max
A
±250
±300
nA/°C
max
B
±3.6
±3.6
OTA DC PERFORMANCE (4) (see Figure 33)
B-Input Offset Voltage
Average B-Input Offset Voltage Drift
B-Input Bias Current
Average B-Input Bias Current Drift
E-Input Bias Current
Average E-Input Bias Current Drift
C-Output Bias Current
Average C-Output Bias Current Drift
VB = 0V, RC = 0Ω, RE = 100Ω
VB = 0V, RC = 0Ω, RE = 100Ω
±1
±5
VB = 0V, RC = 0Ω, RE = 100Ω
VB = 0V, VC = 0V
±30
±100
VB = 0V, VC = 0V
VB = 0V, VC = 0V
±5
±18
VB = 0V, VC = 0V
OTA INPUT (see Figure 33)
B-Input Voltage Range
±4.2
B-Input Impedance
±3.7
455 || 2.1
V
min
B
kΩ || pF
typ
C
Min E-Input Resistance
10.5
12.5
13.0
13.3
Ω
max
B
Max E-Input Resistance
10.5
6.7
6.5
6.3
Ω
min
B
IE = ±1mA
±4.2
±3.7
±3.6
±3.6
V
min
A
VE = 0
±15
±10
±9
±9
mA
min
A
IC = ±1mA
±4.7
±4.0
±3.9
±3.9
V
min
A
VC = 0
±15
±10
±9
±9
mA
min
A
kΩ || pF
typ
C
OTA OUTPUT
E-Output Voltage Compliance
E-Output Current, Sinking/Sourcing
C-Output Voltage Compliance
C-Output Current, Sinking/Sourcing
C-Output Impedance
(1)
(2)
(3)
(4)
54 || 2
Test levels: (A) 100% tested at +25°C. Over temperature limits set by characterization and simulation. (B) Limits set by characterization
and simulation. (C) Typical value only for information.
Junction temperature = ambient for +25°C specifications.
Junction temperature = ambient at low temperature limit; junction temperature = ambient + 7°C at high temperature limit for over
temperature specifications.
Current is considered positive out of node.
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OPA861
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ELECTRICAL CHARACTERISTICS: VS = ±5V (continued)
RL = 500Ω and RADJ = 250Ω, unless otherwise noted.
OPA861ID, IDBV
TYP
MIN/MAX OVER TEMPERATURE
+25°C (2)
0°C to
70°C (3)
–40°C to
+85°C (3)
Maximum Operating Voltage
±6.3
±6.3
±6.3
Minimum Operating Voltage
±2.0
±2.0
PARAMETER
CONDITIONS
+25°C
MIN/
MAX
TEST
LEVEL (1)
V
typ
C
V
max
A
±2.0
V
min
B
UNITS
POWER SUPPLY
Specified Operating Voltage
±5
Maximum Quiescent Current
RADJ = 250Ω
5.4
5.9
7.0
7.4
mA
max
A
Minimum Quiescent Current
RADJ = 250Ω
5.4
4.9
4.3
3.4
mA
min
A
ΔIC/ΔVS
±20
±50
±60
±65
µA/V
max
A
–40 to +85
°C
typ
C
OTA Power-Supply Rejection Ratio (+PSRR)
THERMAL CHARACTERISTICS
Specification: ID, IDBV
Thermal Resistance θ JA
D
SO-8
Junction-to-Ambient
+125
°C/W
typ
C
DBV
SOT23-6
Junction-to-Ambient
+150
°C/W
typ
C
4
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OPA861
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SBOS338G – AUGUST 2005 – REVISED MAY 2013
ELECTRICAL CHARACTERISTICS: VS = +5V
RL = 500Ω to VS/2 and RADJ = 250Ω, unless otherwise noted.
OPA861ID, IDBV
TYP
MIN/MAX OVER TEMPERATURE
CONDITIONS
+25°C
+25°C (2)
0°C to
70°C (3)
–40°C to
+85°C (3)
UNITS
MIN/
MAX
TEST
LEVEL (1)
Bandwidth
G = +5, VO = 200mVPP,
RL = 500Ω
73
72
72
70
MHz
min
B
G = +5, VO = 1VPP
73
MHz
typ
C
Slew Rate
G = +5, VO = 2.5V Step
410
395
390
390
V/µs
min
B
VO = 1V Step
4.4
ns
typ
C
PARAMETER
OTA—Open-Loop (see Figure 33)
AC PERFORMANCE
Rise Time and Fall Time
Harmonic Distortion
G = +5, VO = 2VPP, 5MHz
2nd-Harmonic
RL = 500Ω
–67
–55
–54
–54
dB
max
B
3rd-Harmonic
RL = 500Ω
–57
–50
–49
–48
dB
max
B
Base Input Voltage Noise
f > 100kHz
2.4
3.0
3.3
3.4
nV/√Hz
max
B
Base Input Current Noise
f > 100kHz
1.7
2.4
2.45
2.5
pA/√Hz
max
B
Emitter Input Current Noise
f > 100kHz
5.2
15.3
16.6
17.5
pA/√Hz
max
B
Minimum OTA Transconductance (gm)
VO = ±10mV, RC = 50Ω, RE = 0Ω
85
70
67
65
mA/V
min
A
Maximum OTA Transconductance (gm)
VO = ±10mV, RC = 50Ω, RE = 0Ω
85
140
145
150
mA/V
max
A
VB = 0V, RC = 0Ω, RE = 100Ω
±3
±12
±15
±20
mV
max
A
±67
±120
μV/°C
max
B
±6
±6.6
μA
max
A
±20
±25
nA/°C
max
B
±125
±140
μA
max
A
±500
±600
nA/°C
max
B
μA
typ
C
B
OTA DC PERFORMANCE (4) (see Figure 33)
B-Input Offset Voltage
Average B-Input Offset Voltage Drift
B-Input Bias Current
Average B-Input Bias Current Drift
E-Input Bias Current
Average E-Input Bias Current Drift
C-Output Bias Current
VB = 0V, RC = 0Ω, RE = 100Ω
VB = 0V, RC = 0Ω, RE = 100Ω
±1
±5
VB = 0V, RC = 0Ω, RE = 100Ω
VB = 0V, VC = 0V
±30
±100
VB = 0V, VC = 0V
VB = 0V, VC = 0V
±15
OTA INPUT (see Figure 33)
Most Positive B-Input Voltage
4.2
3.7
3.6
3.6
V
min
Least Positive B-Input Voltage
0.8
1.3
1.4
1.4
V
max
B
kΩ || pF
typ
C
B-Input Impedance
455 || 2.1
Min E-Input Resistance
11.8
14.4
14.9
15.4
Ω
max
B
Max E-Input Resistance
11.8
7.1
6.9
6.7
Ω
min
B
OTA OUTPUT
Maximum E-Output Voltage Compliance
IE = ±1mA
4.2
3.7
3.6
3.6
V
min
A
Minimum E-Output Voltage Compliance
IE = ±1mA
0.8
1.3
1.4
1.4
V
max
A
VE = 0
±8
±7
±6.5
±6.5
mA
min
A
Maximum C-Output Voltage Compliance
IC = ±1mA
4.7
4.0
3.9
3.9
V
min
A
Minimum C-Output Voltage Compliance
IC = ±1mA
0.3
1.0
1.1
1.1
V
max
A
VC = 0
±8
±7
±6.5
±6.5
mA
min
A
kΩ || pF
typ
C
E-Output Current, Sinking/Sourcing
C-Output Current, Sinking/Sourcing
C-Output Impedance
(1)
(2)
(3)
(4)
54 || 2
Test levels: (A) 100% tested at +25°C. Over temperature limits set by characterization and simulation. (B) Limits set by characterization
and simulation. (C) Typical value only for information.
Junction temperature = ambient for +25°C specifications.
Junction temperature = ambient at low temperature limit; junction temperature = ambient + 3°C at high temperature limit for over
temperature specifications.
Current is considered positive out of node.
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OPA861
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ELECTRICAL CHARACTERISTICS: VS = +5V (continued)
RL = 500Ω to VS/2 and RADJ = 250Ω, unless otherwise noted.
OPA861ID, IDBV
TYP
MIN/MAX OVER TEMPERATURE
+25°C (2)
0°C to
70°C (3)
–40°C to
+85°C (3)
Maximum Operating Voltage
12.6
12.6
12.6
Minimum Operating Voltage
4
4
PARAMETER
CONDITIONS
+25°C
MIN/
MAX
TEST
LEVEL (1)
V
typ
C
V
max
A
4
V
min
B
UNITS
POWER SUPPLY
Specified Operating Voltage
5
Maximum Quiescent Current
RADJ = 250Ω
4.7
5.2
6.0
6.4
mA
max
A
Minimum Quiescent Current
RADJ = 250Ω
4.7
4.2
3.4
3.0
mA
min
A
ΔIC/ΔVS
±20
±50
±60
±65
µA/V
max
A
–40 to +85
°C
typ
C
OTA Power-Supply Rejection Ratio (+PSRR)
THERMAL CHARACTERISTICS
Specification: ID, IDBV
Thermal Resistance θ JA
D
SO-8
Junction-to-Ambient
+125
°C/W
typ
C
DBV
SOT23-6
Junction-to-Ambient
+150
°C/W
typ
C
6
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SBOS338G – AUGUST 2005 – REVISED MAY 2013
TYPICAL CHARACTERISTICS: VS = ±5V
At TA = +25°C, IQ = 5.4mA, and RL = 500Ω, unless otherwise noted.
OTA TRANSCONDUCTANCE vs FREQUENCY
OTA TRANSCONDUCTANCE vs QUIESCENT CURRENT
150
1000
IO UT
VIN = 100mVPP
RL = 50Ω
VIN = 10mVPP
50Ω
Transconductance (mA/V)
Transconductance (mA/V)
VI N
50Ω
I Q = 5.4mA (102mA/V)
IQ = 6.5mA (117mA/V)
100
IQ = 1.9mA (51mA/V)
120
2
gm = -0.8265.IQ + 24.197.IQ - 1.466
90
IOUT
60
VIN
50W
30
50W
IQ = 3.4mA (79mA/V)
10
0
1M
10M
100M
1G
0
1
2
Figure 2.
140
6
IQ = 6.5mA
100
IQ = 3.4mA
80
60
IQ = 1.9mA
40
Small signal around input voltage.
−40
−30
−20
−10
0
2
IQ = 3.4mA
0
IQ = 1.9mA
−2
VIN
−4
50Ω
50Ω
20
30
−70 −60 −50 −40 −30 −20 −10
40
10
20
30
40
50
60
70
Figure 5.
OTA LARGE-SIGNAL PULSE RESPONSE
3
0.6
0.2
0
G = +5V/V
RL = 500Ω
VIN = 0.25VPP
f IN = 20MHz
See Figure 48
Output Voltage (V)
2
0.4
−0.8
0
OTA Input Voltage (mV)
OTA SMALL-SIGNAL PULSE RESPONSE
Output Voltage (V)
IOUT
−8
10
0.8
−0.6
8
IQ = 5.4mA
Figure 4.
−0.4
7
4
Input Voltage (mV)
−0.2
6
IQ = 6.5mA
−6
20
0
5
OTA TRANSFER CHARACTERISTICS
8
OTA Output Current (mA)
Transconductance (mA/V)
OTA TRANSCONDUCTANCE vs INPUT VOLTAGE
IQ = 5.4mA
4
Figure 3.
160
120
3
Quiescent Current (mA)
Frequency (Hz)
1
0
−1
−2
G = +5V/V
RL = 500Ω
VIN = 1VPP
fIN = 20MHz
See Figure 48
−3
Time (10ns/div)
Time (10ns/div)
Figure 6.
Figure 7.
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TYPICAL CHARACTERISTICS: VS = ±5V (continued)
At TA = +25°C, IQ = 5.4mA, and RL = 500Ω, unless otherwise noted.
B-INPUT RESISTANCE vs QUIESCENT CURRENT
C-OUTPUT RESISTANCE vs QUIESCENT CURRENT
120
OTA C- Output Resistance (kW )
OTA B- Input Resistance (kW )
500
490
480
470
460
450
440
430
110
100
90
80
70
60
50
40
0
1
2
3
4
5
6
7
0
8
1
2
Figure 8.
E-OUTPUT RESISTANCE vs QUIESCENT CURRENT
6
7
8
INPUT VOLTAGE AND CURRENT NOISE DENSITY
Input Voltage Noise Density (nV/√Hz)
Input Current Noise Density (pA/√Hz)
OTA E- Output Resistance (W )
5
100
50
40
30
20
10
0
E−Input Current Noise (5.2pA/√Hz)
10
B−Input Voltage Noise (2.4nV/√Hz)
B−Input Current Noise (1.65pA/√Hz)
1
0
1
2
3
4
5
6
7
100
8
Quiescent Current (mA)
1k
10k
100k
1M
10M
Frequency (Hz)
Figure 10.
Figure 11.
QUIESCENT CURRENT vs RADJ
1MHz OTA VOLTAGE AND CURRENT NOISE DENSITY
vs QUIESCENT CURRENT ADJUST RESISTOR
16
Input Voltage Noise Density (nV/√Hz)
Input Current Noise Density (pA/√Hz)
8
7
Quiescent Current (mA)
4
Figure 9.
60
6
5
4
3
2
1
E−Input Current Noise (pA/√Hz)
14
12
10
8
B−Input Voltage Noise (nV/√Hz)
6
B−Input Current Noise (pA/√Hz)
4
2
0
0
0.1
1
10
100
1k
10k
100k
0
200 400 600 800 1000 1200 1400 1600 1800 2000
Quiescent Current Adjust Resistor (Ω )
Quiescent Current Adjust Resistor (Ω)
Figure 12.
8
3
Quiescent Current (mA)
Quiescent Current (mA)
Figure 13.
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SBOS338G – AUGUST 2005 – REVISED MAY 2013
TYPICAL CHARACTERISTICS: VS = ±5V (continued)
At TA = +25°C, IQ = 5.4mA, and RL = 500Ω, unless otherwise noted.
QUIESCENT CURRENT vs TEMPERATURE
3
9
4
2
8
2
1
B−Input Offset Voltage
0
0
−2
−1
−4
B−Input Bias Current
−6
−40
−20
0
20
40
60
80
100
Quiescent Current (mA)
6
Input Bias Current (µA)
Offset Voltage (mV)
B-INPUT OFFSET VOLTAGE AND BIAS CURRENT
vs TEMPERATURE
7
6
5
−2
4
−3
3
−40
120
−20
0
Ambient Temperature (_ C)
20
Figure 14.
Five Representative Units
IQ
300
20
100
120
IADJ
4
3
2
= -5E-18 x RADJ + 1E-12 x RADJ - 7E-08 x RADJ + 0.0046 x RADJ + 37.8
250
IQ/IADJ Ratio
OTA C−Output Bias Current (µA)
80
IQ/IADS Ratio vs RADJ
350
30
10
0
−10
200
150
−20
100
−30
50
−40
−40
60
Figure 15.
C-OUTPUT BIAS CURRENT vs TEMPERATURE
40
40
Ambient Temperature (_ C)
−20
0
20
40
60
80
100
IQ = Quiescent Current.
IADJ = Current flowing out of IQ adjust pin.
0
0.01
120
0.1
1
10
100
1k
10k
100k
Quiescent Current Adjust Resistor (W )
Ambient Temperature (_ C)
Figure 16.
Figure 17.
QUIESCENT CURRENT vs ADJUST PIN BIAS CURRENT
IQ Adjust Pin Bias Current (mA)
250
200
150
100
50
0
0.01
0.1
1
10
100
1k
10k
100k
Quiescent Current Adjust Resistor (W)
Figure 18.
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TYPICAL CHARACTERISTICS: VS = +5V
At TA = +25°C, IQ = 4.7mA, and RL = 500Ω to VS/2, unless otherwise noted.
OTA TRANSCONDUCTANCE vs FREQUENCY
100
OTA TRANSCONDUCTANCE vs IQ
150
OTA Transconductance (mA/V)
IQ = 5.8mA
(93mA/V)
Transconductance (mA/V)
IQ = 4.7mA (80mA/V)
IQ = 3.1mA (60mA/V)
IQ = 1.65mA (37mA/V)
I OU T
VIN
50Ω
50Ω
IOUT
VIN
120
50Ω
50Ω
90
60
30
RL = 50Ω
VIN = 10mVPP
VIN = 100mVPP
10
0
1
10
100
1k
0
1
Frequency (Hz)
OTA TRANSCONDUCTANCE vs INPUT VOLTAGE
5
6
7
OTA TRANSFER CHARACTERISTICS
6
IQ = 5.8mA
IQ = 5.8mA
IQ = 4.7mA
100
80
OTA Output Current (mA)
Transconductance (mA/V)
4
Figure 20.
120
IQ = 3.1mA
60
IQ = 1.65mA
40
20
Small−signal around input voltage.
0
−30
−20
−10
0
10
20
4
IQ = 3.1mA
2
IQ = 4.7mA
I OUT
−2
V IN
−4
−6
−50 −40 −30 −20 −10
30
1.5
0.10
1.0
−0.15
−0.20
Output Voltage (V)
0.15
−0.10
20
30
40
50
OTA LARGE-SIGNAL PULSE RESPONSE
2.0
G = +5V/V
R L = 500Ω
VIN = 0.07VPP
f IN = 20MHz
10
Figure 22.
OTA SMALL-SIGNAL PULSE RESPONSE
−0.05
0
OTA Input Voltage (mV)
0.20
0
50 Ω
50 Ω
Figure 21.
0.05
IQ = 1.65mA
0
Input Voltage (mV)
Output Voltage (V)
3
Quiescent Current (mA)
Figure 19.
0.5
0
−0.5
−1.0
−1.5
−2.0
Time (10ns/div)
G = +5V/V
R L = 500Ω
VIN = 0.7VPP
fIN = 20MHz
Time (10ns/div)
Figure 23.
10
2
Figure 24.
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TYPICAL CHARACTERISTICS: VS = +5V (continued)
At TA = +25°C, IQ = 4.7mA, and RL = 500Ω to VS/2, unless otherwise noted.
C-OUTPUT RESISTANCE vs QUIESCENT CURRENT
120
490
110
OTA C−Output Resistance (kΩ )
OTA B−Input Resistance (kΩ )
B-INPUT RESISTANCE vs QUIESCENT CURRENT
500
480
470
460
450
440
430
100
90
80
70
60
50
40
420
0
1
2
3
4
5
6
7
0
1
2
3
Figure 25.
5
6
7
Figure 26.
E-OUTPUT RESISTANCE vs QUIESCENT CURRENT
QUIESCENT CURRENT vs RADJ
60
7
50
6
Quiescent Current (mA)
OTA E−Output Resistance (Ω )
4
Quiescent Current (mA)
Quiescent Current (mA)
40
30
20
10
5
4
3
2
1
0
0
0
1
2
3
4
5
6
7
0.1
Quiescent Current (mA)
1
10
100
1k
10k
100k
Quiescent Current Adjust Resistor (Ω)
Figure 27.
Figure 28.
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APPLICATION INFORMATION
The
OPA861
is
a
versatile
monolithic
transconductance amplifier designed for widebandwidth systems, including high-performance
video, RF, and IF circuitry. The operation of the
OPA861 is discussed in the OTA (Operational
Transconductance Amplifier) section of this data
sheet. Over the years and depending on the writer,
the OTA section of an op amp has been referred to
as a Diamond Transistor, Voltage-Controlled Current
source, Transconductor, Macro Transistor, or positive
second-generation
current
conveyor
(CCII+).
Corresponding symbols for these terms are shown in
Figure 29.
C
3
VIN1
B
1
IOUT
2
VIN2
E
Diamond
Transistor
Transconductor
(used here)
Voltage−Controlled
Current Source
C
VIN1
Z
CCII+
VIN2
B
IOUT
E
Current Conveyor II+
TRANSCONDUCTANCE (OTA) SECTION—AN
OVERVIEW
The symbol for the OTA section is similar to a
transistor (see Figure 29). Applications circuits for the
OTA look and operate much like transistor
circuits—the transistor is also a voltage-controlled
current source. Not only does this characteristic
simplify the understanding of application circuits, it
aids the circuit optimization process as well. Many of
the same intuitive techniques used with transistor
designs apply to OTA circuits. The three terminals of
the OTA are labeled B, E, and C. This labeling calls
attention to its similarity to a transistor, yet draws
distinction for clarity. While the OTA is similar to a
transistor, one essential difference is the sense of the
C-output current: it flows out the C terminal for
positive B-to-E input voltage and in the C terminal for
negative B-to-E input voltage. The OTA offers many
advantages over a discrete transistor. The OTA is
self-biased, simplifying the design process and
reducing component count. In addition, the OTA is far
more linear than a transistor. Transconductance of
the OTA is constant over a wide range of collector
currents—this feature implies a fundamental
improvement of linearity.
BASIC CONNECTIONS
Macro Transistor
Figure 29. Symbols and Terms
Regardless of its depiction, the OTA section has a
high-input impedance (B-input), a low-input/output
impedance (E-input), and a high-impedance current
source output (C-output).
Figure 30 shows basic connections required for
operation. These connections are not shown in
subsequent circuit diagrams. Power-supply bypass
capacitors should be located as close as possible to
the device pins. Solid tantalum capacitors are
generally best.
RQ = 250W, roughly sets IQ = 5.4mA.
RC
1
8
RE
RS
(25W to 200W)
RADJ
250W
3
6
4
5
(1)
0.1mF
2.2mF
Solid Tantalum
-VS
(1)
7
+
VIN
-5V
2
+5V
+VS
0.1mF
+
2.2mF
Solid
Tantalum
NOTE: (1) VS = ±6.5V absolute maximum.
Figure 30. Basic Connections
12
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QUIESCENT CURRENT CONTROL PIN
The quiescent current of the transconductance
portion of the OPA861 is set with a resistor, RADJ,
connected from pin 1 to –VS. The maximum
quiescent current is 6mA. RADJ should be set
between 50Ω and 1kΩ for optimal performance of the
OTA section. This range corresponds to the 5mA
quiescent current for RADJ = 50Ω, and 1mA for RADJ =
1kΩ. If the IQ adjust pin is connected to the negative
supply, the quiescent current will be set by the 250Ω
internal resistor.
Reducing or increasing the quiescent current for the
OTA section controls the bandwidth and AC behavior
as well as the transconductance. With RADJ = 250Ω,
this sets approximately 5.4mA total quiescent current
at 25°C. It may be appropriate in some applications to
trim this resistor to achieve the desired quiescent
current or AC performance.
Applications circuits generally do not show the
resistor RQ, but it is required for proper operation.
With a fixed RADJ resistor, quiescent current
increases with temperature (see Figure 12 in the
Typical Characteristics section). This variation of
current with temperature holds the transconductance,
gm, of the OTA relatively constant with temperature
(another advantage over a transistor).
It is also possible to vary the quiescent current with a
control signal. The control loop in Figure 31 shows
1/2 of a REF200 current source used to develop
100mV on R1. The loop forces 125mV to appear on
R2. Total quiescent current of the OPA861 is
approximately 37 × I1, where I1 is the current made to
flow out of pin 1.
OPA861
R1
1.25kΩ
BASIC APPLICATIONS CIRCUITS
Most applications circuits for the OTA section consist
of a few basic types, which are best understood by
analogy to a transistor. Used in voltage-mode, the
OTA section can operate in three basic operating
states—common emitter, common base, and
common collector. In the current-mode, the OTA can
be useful for analog computation such as current
amplifier, current differentiator, current integrator, and
current summer.
Common-E Amplifier or Forward Amplifier
Figure
32
compares
the
common-emitter
configuration for a BJT with the common-E amplifier
for the OTA section. There are several advantages in
using the OTA section in place of a BJT in this
configuration. Notably, the OTA does not require any
biasing, and the transconductance gain remains
constant over temperature. The output offset voltage
is close to 0, compared with several volts for the
common-emitter amplifier.
The gain is set in a similar manner as for the BJT
equivalent with Equation 1:
R
G+ 1 L
gm ) R E
V+
1/2 REF200
100µA
With this control loop, quiescent current will be nearly
constant with temperature. Since this method differs
from the temperature-dependent behavior of the
internal current source, other temperature-dependent
behavior may differ from that shown in the Typical
Characteristics. The circuit of Figure 31 will control
the IQ of the OPA861 somewhat more accurately than
with a fixed external resistor, RQ. Otherwise, there is
no fundamental advantage to using this more
complex biasing circuitry. It does, however,
demonstrate the possibility of signal-controlled
quiescent current. This capability may suggest other
possibilities such as AGC, dynamic control of AC
behavior, or VCO.
IQ Adjust
(1)
1 I1
R2
425Ω
TLV2262
Figure 31. Optional Control Loop for Setting
Quiescent Current
Just as transistor circuits often use emitter
degeneration, OTA circuits may also use
degeneration. This option can be used to reduce the
effects that offset voltage and offset current might
otherwise have on the DC operating point of the OTA.
The E-degeneration resistor may be bypassed with a
large capacitor to maintain high AC gain. Other
circumstances may suggest a smaller value capacitor
used to extend or optimize high-frequency
performance.
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The forward amplifier shown in Figure 33 and
Figure 34 corresponds to one of the basic circuits
used to characterize the OPA861. Extended
characterization of this topology appears in the
Typical Characteristics section of this datasheet.
V+
RL
RS
VO
VO
VI
Inverting Gain
VOS = Several Volts
R1
160Ω
RE
RS
VI
8
C
3 B
OPA861
E
2
V−
VI
8
C
3 B
VO
OPA861
Figure 33. Forward Amplifier Configuration and
Test Circuit
RL
E
2
RE
G = 5V/V
IQ = 5.4mA
RE
78Ω
(a) Transistor Common−Emitter Amplifier
Transconductance varies over temperature.
100Ω
RC
500Ω
RL1
Noninverting Gain
VOS = 0V
VO
Network
Analyzer
8
3
(b) OTA Common−E Amplifier
Transconductance remains constant over temperature.
R1
100W
Figure 32. Common-Emitter vs Common-E
Amplifier
(2)
A positive voltage at the B-input, pin 3, causes a
positive current to flow out of the C-input, pin 8. This
gives a noninverting gain where the circuit of
Figure 32a is inverting. Figure 32b shows an amplifier
connection of the OPA861, the equivalent of a
common-emitter transistor amplifier. Input and output
can be ground-referenced without any biasing. The
amplifier is non-inverting because of the sense of the
output current.
14
RL2
rE
2
VI
The transconductance of the OTA with degeneration
can be calculated by Equation 2:
g m_deg + 1 1
gm ) R E
RIN
50W
OTA
RL = RL1 + RL2 || RIN
RE
G =
RL
RE + rE
At IQ = 5.4mA
G=
RL
RE + 10.5W
rE =
rE =
1
gm
1
95mA/V
= 10.5W
at IQ = 5.4mA
Figure 34. Forward Amplifier Design Equations
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Common-C Amplifier
Figure 35b shows the OPA861 connected as an Efollower—a voltage buffer. It is interesting to notice
that the larger the RE resistor, the closer to unity gain
the buffer will be. If the OPA861 is to be used as a
buffer, use RE ≥ 500Ω for best results. For the
OPA861 used as a buffer, the gain is given by
Equation 3:
1
G+
[1
1 ) g 1R
m
(3)
E
This low impedance can be converted to a high
impedance by inserting the buffer amplifier in series.
Current-Mode Analog Computations
As mentioned earlier, the OPA861 can be used
advantageously for analog computation. Among the
application possibilities are functionality as a current
amplifier, current differentiator, current integrator,
current summer, and weighted current summer.
Table 1 lists these different uses with the associated
transfer functions.
These functions can easily be combined to form
active filters. Some examples using these currentmode functions are shown later in this document.
V+
G=1
VOS = 0.7V
VI
V+
VO
RL
RE
VO
Noninverting Gain
VOS = Several Volts
V−
(a) Transistor Common−Collector Amplifier
(Emitter Follower)
G+
100Ω
VI
OPA861
1)g
R1
+1
1
VIN
RE
m
ǒ
R O + g1 ø R E
m
8
C
3 B
1
RE
Ǔ
V(a) Transistor Common-Base Amplifier
G=1
VOS = 0V
E
2
RE
RL
G=
RE +
VO
(b) OTA Common−C Amplifier
(Buffer)
100W
8
C
3 B
=-
RL
RE
VO
OPA861
E
2
Figure 35. Common-Collector vs Common-C
Amplifier
1
gm
Inverting Gain
VOS = 0V
RL
RE
A low value resistor in series with the B-input is
recommended. This resistor helps isolate trace
parasitic from the inputs, reduces any tendency to
oscillate, and controls frequency response peaking.
Typical resistor values are from 25Ω to 200Ω.
VIN
(b) OTA Common-B Amplifier
Figure 36. Common-Base Transistor vs
Common-B OTA
Common-B Amplifier
Figure 36 shows the Common-B amplifier. This
configuration produces an inverting gain and a low
impedance input. Equation 4 shows the gain for this
configuration.
RL
R
G+
[* L
1
RE
RE ) g
m
(4)
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Table 1. Current-Mode Analog Computation Using the OTA Section
FUNCTIONAL ELEMENT
TRANSFER FUNCTION
IMPLEMENTATION WITH THE OTA SECTION
IOUT
IIN
Current Amplifier
I OUT +
R1
R2
R1
I IN
R2
IOUT
Current Integrator
IIN
1
I OUT +
C
C
ŕ I dt
R
R
IN
IOUT
Current Summer
n
I OUT + 1 S I j
j+1
I1
I2
In
I OUT
Weighted Current Summer
n
I OUT + 1 S I j
j+1
Rj
R
R
R1
I1
R
Rn
In
OPA861 APPLICATIONS
Control-Loop Amplifier
DC-Restore Circuit
A new type of control loop amplifier for fast and
precise control circuits can be designed with the
OPA861. The circuit of Figure 37 illustrates a series
connection of two voltage control current sources that
have an integral (and at higher frequencies, a
proportional) behavior versus frequency. The control
loop amplifiers show an integrator behavior from DC
to the frequency represented by the RC time constant
of the network from the C-output to GND. Above this
frequency, they operate as an amp with constant
gain. The series connection increases the overall gain
to about 110dB and thus minimizes the control loop
deviation. The differential configuration at the inputs
enables one to apply the measured output signal and
the reference voltage to two identical high-impedance
inputs. The output buffer decouples the C-output of
the second OTA in order to insure the AC
performance and to drive subsequent output stages.
The OPA861 can be used advantageously with an
operational amplifier, here the OPA656, as a DCrestore circuit. Figure 38 illustrates this design.
Depending on the collector current of the
transconductance amplifier (OTA) of the OPA861, a
switching function is realized with the diodes D1 and
D2.
16
When the C-output is sourcing current, the capacitor
C1 is being charged. When the C-output is sinking
current, D1 is turned off and D2 is turned on, letting
the voltage across C1 be discharged through R2.
The condition to charge C1 is set by the voltage
difference between VREF and VOUT. For the OTA Coutput to source current, VREF has to be greater than
VOUT. The rate of charge of C1 is set by both R1 and
C1. The discharge rate is given by R2 and C1.
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6
5
8
BUF602
VOUT
3
8
180Ω
2
10pF
VREF
10pF
3
2
10Ω
180Ω
33Ω
10Ω
33Ω
6
VIN
Figure 37. Control-Loop Amplifier Using Three OPA861s
C1
100pF
20Ω
JFET−Input, Wideband
VIN
D 1, D2 = 1N4148
RQ = 1kΩ
OPA656
R2
100kΩ
D1
VOUT
20Ω
D2
CCII
8 C
The OTA amplifier works as a current conveyor (CCII) in this circuit, with a current gain of 1.
R1 and C1 set the DC restoration time constant.
D1 adds a propagation delay to the DC restoration.
R2 and C1 set the decay time constant.
E 2
R1
40.2Ω
B
3
R2
100Ω
VREF
Figure 38. DC Restorer Circuit
Negative Impedance Converter Filter: Low-Pass
Filter
The OPA861 can be used as a negative impedance
converter to realize the low-pass filer shown in
Figure 39.
The transfer function is shown in Equation 5:
VOUT
1
=
VIN
1 + sR(C1 + C2) + s2C1C2R2
(5)
with:
w0 +
R
C1
R
VIN
1
ǸC1C 2 R
C 1C 2
VOUT
Q=
C2
C1 + C2
Figure 39. Low-Pass Negative Impedance
Converter Filter
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Differential Line Driver/Receiver
The input impedance is shown in Equation 6:
Z IN + 1 ) R 1 ) sRC
2sC
1 ) 2sRC
(6)
Figure 40 shows the frequency responses for lowpass, Butterworth filters set at 20kHz and 10MHz.
For the 20kHz filter, set R to 1kΩ and
C 1 + 1 C 2 + 5.6mF
2
. For the 10MHz filter, the
parasitic capacitance at the output pin needs to be
taken into consideration. In the example of Figure 40,
the parasitic is 3pF, which gives us the settings of R
= 1.13kΩ, C1 = 10pF, and C2 = 17pF.
The wide bandwidth and high slew rate of the
OPA861 current-mode amplifier make it an ideal line
driver. The circuit in Figure 42 makes use of two
OPA861s to realize a single-ended to differential
conversion. The high-impedance current source
output of the OPA861 allows it to drive lowimpedance or capacitive loads without series
resistances and avoids any attenuation that would
have otherwise occured in the resistive network.
The OPA861 used as a differential receiver exhibits
excellent common-mode rejection ratio, as can be
seen in Figure 41.
Common−Mode Rejection Ratio (dBc)
0
−10
Gain (dB)
−20
−30
−40
−50
−60
−70
−80
1k
10k
100k
1M
10M
100M
1G
0
−10
−20
−30
−40
−50
−60
−70
−80
−90
−100
0.001
0.01
Frequency (Hz)
0.1
1
10
100
Frequency (MHz)
Figure 40. Small-Signal Frequency Response for
a Low-Pass Negative Impedance Converter Filter
Figure 41. Differential Driver Common-Mode
Rejection Ratio for 2VPP Input Signals
To 50Ω Load
50Ω
VIN
50Ω
10Ω
50Ω
100Ω
50Ω
10Ω
50Ω
Figure 42. Twisted-Pair Differential Driver and Receiver with the OPA861
18
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ACTIVE FILTERS USING THE OPA861 IN
CURRENT CONVEYOR STRUCTURE
One further example of the versatility of the Diamond
Transistor and Buffer is the construction of highfrequency (> 10MHz) active filters. Here, the Current
Conveyor structure, shown in Figure 43, is used with
the Diamond Transistor as a Current Conveyor.
VOUT
+1
C
IOUT
E
CCII− B
C
VIN
C
R
R
R
R
of the operational amplifier becomes a negative
second type of Current Conveyor (CCII), as shown in
Figure 43. Both arrangements have identical transfer
functions and the same level of sensitivity to
deviations. The most recent implementation of active
filters in a Current-Conveyor structure produced a
second-order Bi-Quad filter. The value of the
resistance in the emitter of the Diamond Transistor
controls the filter characteristic. For more information,
refer to application note SBOS047, New Ultra HighSpeed Circuit Techniques with Analog ICs.
IIN
C/2
C/2
Reciprocal Networks
T(s) =
VOUT
VIN
=
IOUT
IIN
=
+
4KQ2/R2C2
s2 + 2/RC[2Q(1 − K) + 1]s + 4KQ2/R2C2
VIN
VOUT
N
I OUT
N
I IN
NA
I IN
−
Figure 43. Current Conveyor
VOUT
=
I OUT
VIN
IIN
Interreciprocal Networks
The method of converting RC circuit loops with
operational amplifiers in Current Conveyor structures
is based upon the adjoint network concept. A network
is reversible or reciprocal when the transfer function
does not change even when the input and output
have been exchanged. Most networks, of course, are
nonreciprocal. The networks of Figure 44, perform
interreciprocally when the input and output are
exchanged, while the original network, N, is
exchanged for a new network NA. In this case, the
transfer function remains the same, and NA is the
adjoing network. It is easy to construct an adjoint
network for any given circuit, and these networks are
the base for circuits in Current-Conveyor structure.
Individual elements can be interchanged according to
the list in Figure 45. Voltage sources at the input
become short circuits, and the current flowing there
becomes the output variable. In contrast, the voltage
output becomes the input, which is excitated by a
current source. The following equation describes the
interreciprocal features of the circuit: VOUT/VIN =
IOUT/IIN. Resistances and capacitances remain
unchanged. In the final step, the operational amplifier
with infinite input impedance and 0Ω output
impedance is transformed into a current amplifier with
0Ω input impedance and infinite output impedance. A
Diamond Transistor with the base at ground comes
quite close to an ideal current amplifier. The wellknown Sallen-Key low-pass filter with positive
feedback, is an example of conversion into CurrentConveyor structure, see Figure 46. The positive gain
+
VIN
VOUT
N
I OUT
−
Figure 44. Networks
Element
1
− VOUT +
R
1
Passive
Elements
Controlled
Sources
C
1
1
1
Adjoint
VIN
1
Signal
Sources
+
V
−
2
1
2
1
2
1
2
1
IOUT
2
IIN
R
C
2
2
3
3
µV
2
µI
I
4
4
Figure 45. Individual Elements in the Current
Conveyor
R3
R2
VIN
BUF602
C1
R1
RB1
C2
R1M
RB2
R1S
R2S
VOUT
R2M
RB3
R3S
Figure 46. Universal Active Filter
Transfer Function
Filter Characteristics
The transfer function of the universal active filter of
Figure 46 is shown in Equation 7.
Five filter types can be made with this structure:
• For a low-pass filter, set R2 = R3 = ∞,
• For a high-pass filter, set R1 = R2 = ∞,
• For a bandpass filter, set R1 = R3 = ∞,
• For a band rejection filter, set R2 = ∞; R1 = R3,
• For an all-pass filter, set R1 = R1S; R2 = R2S; and
R3 = R3S.
R
R
1M R
1R
s 2C1C 2R 1M R2M ) sC 1 R1M ) R1
V OUT
2
1
3
F(p) +
+
R
R
VIN
s2C C R 2M ) sC 1M ) 1
1
2
3S
2S
R
1S
(7)
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A few designs for a low-pass filter are shown in
Figure 47 and Table 2.
Gain (dB)
Table 2. Component Values for Filters Shown In
Figure 47
fO
R
RO
CO
1MHz
150
100
2nF
20MHz
150
100
112.5pF
50MHz
150
100
25pF
3
0
-3
-6
-9
-12
-15
-18
-21
-24
-27
-30
-33
-36
-39
-42
-45
-48
50MHz Filter
High-CMRR, Moderate Precision, Differential
I/O ADC Driver
The circuit shown in Figure 48 depicts an ADC driver
implemented with two OPA861s. Since the gain is set
here by the ratio of the internal 600Ω resistors and
RE, its accuracy will only be as good as the input
resistor of the ADS5272. The small-signal frequency
response for this circuit has 150MHz at –3dB
bandwidth for a gain of approximately 5.6dB, as
shown in Figure 49. The advantage of this circuit lies
in its high CMRR to 100kHz; see Figure 50. This
circuit also has more than 10 bits of linearity.
VIN1
20MHz Filter
100k
1M
RE
600Ω
100M
1G
VIN2
The advantages of building active filters using a
Current Conveyor structure are:
• The increase in output resistance of operational
amplifiers at high frequencies makes it difficult to
construct feedback filter structures (decrease in
stop-band attenuation).
• All filter coefficients are represented by
resistances, making it possible to adjust the filter
frequency response without affecting the filter
coefficients.
The capacitors which determine the frequency are
located between the ground and the current
source outputs and are thus grounded on one
side. Therefore, all parasitic capacitances can be
viewed as part of these capacitors, making them
easier to comprehend.
The features which determine the frequency
characteristics are currents, which charge the
integration capacitors. This situation is similar to
the transfer characteristic of the Diamond
Transistor.
600Ω
Figure 48. High CMRR, Moderate Precision,
Differential I/O ADC Driver
6
5.6dB
3
Gain (dB)
Figure 47. Butterworth Low-Pass Filter with the
Universal Active Filter
20
VCM
OPA861
10M
Frequency (Hz)
•
600Ω
For All Filters:
R2 = R3 = ¥
R1 = R1S = R2S = 1/2 R3S = R
R1M = R2M = R0
C1 = C2 = C0
10k
•
ADS5272
OPA861
1MHz Filter
0
−3
−6
−9
1M
10M
100M
1G
Frequency (Hz)
Figure 49. ADC Driver, Small-Signal Frequency
Response
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NOISE PERFORMANCE
Common−Mode Rejection Ratio(dB)
75
Input−Referred
70
65
60
55
50
45
The OTA noise model consists of three elements: a
voltage noise on the B-input; a current noise on the
B-input; and a current noise on the E-input. Figure 51
shows the OTA noise analysis model with all the
noise terms included. In this model, all noise terms
are taken to be noise voltage or current density terms
in either nV/√Hz or pA/√Hz.
40
35
en
30
VO
25
RL
20
1k
10k
100k
1M
10M
100M
1G
RS
Frequency (Hz)
√4kTRS
Figure 50. CMRR of the ADC Driver
ibn
RG
ibi
√4kTRS
DESIGN-IN TOOLS
Figure 51. OTA Noise Analysis Model
DEMONSTRATION BOARDS
A printed circuit board (PCB) is available to assist in
the initial evaluation of circuit performance using the
OPA861. This module is available free, as an
unpopulated PCB delivered with descriptive
documentation. The summary information for the
board is shown below:
The total output spot noise voltage can be computed
as the square root of the sum of all squared output
noise voltage contributors. Equation 8 shows the
general form for the output noise voltage using the
terms shown in Figure 51.
eO =
PRODUCT
PACKAGE
BOARD PART
NUMBER
OPA861ID
SO-8
DEM-OTA-SO-1A
The board can be requested
Instruments web site (www.ti.com).
on
LITERATURE
REQUEST
NUMBER
SBOU035
the
Texas
MACROMODELS AND APPLICATIONS
SUPPORT
Computer simulation of circuit performance using
SPICE is often useful when analyzing the
performance of analog circuits and systems. This
principle is particularly true for Video and RF amplifier
circuits where parasitic capacitance and inductance
can have a major effect on circuit performance. A
SPICE model for the OPA861 is available through the
Texas Instruments web page (www.ti.com). These
models do a good job of predicting small-signal AC
and transient performance under a wide variety of
operating conditions. They do not do as well in
predicting the harmonic distortion. These models do
not attempt to distinguish between the package types
in their small-signal AC performance.
[eN2 + (RSiBN)2 + 4kTRS + (RGiBI)2 + 4kTRG]
2
RL
RG +
1
gm
(8)
THERMAL ANALYSIS
Maximum desired junction temperature will set the
maximum allowed internal power dissipation as
described below. In no case should the maximum
junction temperature be allowed to exceed 150°C.
Operating junction temperature (TJ) is given by
TA + PD × θ JA. The total internal power dissipation
(PD) is the sum of quiescent power (PDQ) and
additional power dissipated in the output stage (PDL)
to deliver output current. Quiescent power is simply
the specified no-load supply current times the total
supply voltage across the part. PDL will depend on the
required output signal and load but would, for the
OPA861 be at a maximum when the maximum IO is
being driven into a voltage source that puts the
maximum voltage across the output stage. Maximum
IO is 15mA times a 9V maximum across the output.
Note that it is the power in the output stage and not
into the load that determines internal power
dissipation.
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As a worst-case example, compute the maximum TJ
using an OPA861IDBV in the circuit of Figure 32b
operating at the maximum specified ambient
temperature of +85°C and driving a –1V voltage
reference.
PD = 10V × 5.4mA + (15mA × 9V) = 185mW
Maximum TJ = +85°C + (0.19W × 150°C/W) = 114°C.
Although this is still well below the specified
maximum junction temperature, system reliability
considerations may require lower tested junction
temperatures. The highest possible internal
dissipation will occur if the load requires current to be
forced into the output for positive output voltages or
sourced from the output for negative output voltages.
This puts a high current through a large internal
voltage drop in the output transistors.
BOARD LAYOUT GUIDELINES
Achieving optimum performance with a highfrequency amplifier like the OPA861 requires careful
attention to board layout parasitics and external
component types. Recommendations that will
optimize performance include:
a) Minimize parasitic capacitance to any AC ground
for all of the signal I/O pins. Parasitic capacitance on
the inverting input pin can cause instability: on the
noninverting input, it can react with the source
impedance to cause unintentional bandlimiting. To
reduce unwanted capacitance, a window around the
signal I/O pins should be opened in all of the ground
and power planes around those pins. Otherwise,
ground and power planes should be unbroken
elsewhere on the board.
b) Minimize the distance (< 0.25") from the powersupply pins to high-frequency 0.1µF decoupling
capacitors. At the device pins, the ground and powerplane layout should not be in close proximity to the
signal I/O pins. Avoid narrow power and ground
traces to minimize inductance between the pins and
the decoupling capacitors. The power-supply
connections should always be decoupled with these
capacitors. An optional supply decoupling capacitor
(0.1µF) across the two power supplies (for bipolar
operation) will improve 2nd-harmonic distortion
performance. Larger (2.2µF to 6.8µF) decoupling
capacitors, effective at lower frequency, should also
be used on the main supply pins. These may be
placed somewhat farther from the device and may be
shared among several devices in the same area of
the PC board.
22
c) Careful selection and placement of external
components will preserve the high-frequency
performance of the OPA861. Resistors should be a
very low reactance type. Surface-mount resistors
work best and allow a tighter overall layout. Metal film
or carbon composition, axially-leaded resistors can
also provide good high-frequency performance.
Again, keep their leads and PC board traces as short
as possible. Never use wirewound type resistors in a
high-frequency application.
d) Connections to other wideband devices on the
board may be made with short, direct traces or
through onboard transmission lines. For short
connections, consider the trace and the input to the
next device as a lumped capacitive load. Relatively
wide traces (50mils to 100mils) should be used,
preferably with ground and power planes opened up
around them.
e) Socketing a high-speed part like the OPA861 is
not recommended. The additional lead length and
pin-to-pin capacitance introduced by the socket can
create an extremely troublesome parasitic network
that makes it almost impossible to achieve a smooth,
stable frequency response. Best results are obtained
by soldering the OPA861 onto the board.
INPUT AND ESD PROTECTION
The OPA861 is built using a very high-speed
complementary bipolar process. The internal junction
breakdown voltages are relatively low for these very
small geometry devices. These breakdowns are
reflected in the Absolute Maximum Ratings table. All
device pins are protected with internal ESD protection
diodes to the power supplies as shown in Figure 52.
+VCC
External
Pin
Internal
Circuitry
−VCC
Figure 52. Internal ESD Protection
These diodes provide moderate protection to input
overdrive voltages above the supplies as well. The
protection diodes can typically support 30mA
continuous current. Where higher currents are
possible (for example, in systems with ±15V supply
parts driving into the OPA861), current-limiting series
resistors should be added into the two inputs. Keep
these resistor values as low as possible since high
values degrade both noise performance and
frequency response.
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SBOS338G – AUGUST 2005 – REVISED MAY 2013
REVISION HISTORY
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision F (May 2011) to Revision G
•
Page
Changed transfer function equations in Negative Impedance Converter Filter: Low-Pass Filter section .......................... 17
Changes from Revision E (August 2008) to Revision F
Page
•
Updated Figure 30 .............................................................................................................................................................. 12
•
Updated Equation 8 ............................................................................................................................................................ 21
Changes from Revision D (August 2006) to Revision E
•
Page
Changed storage temperature range rating in Absolute Maximum Ratings table from –40°C to +125°C to –65°C to
+125°C .................................................................................................................................................................................. 2
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PACKAGE OPTION ADDENDUM
www.ti.com
10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
OPA861ID
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
OPA
861
OPA861IDBVT
ACTIVE
SOT-23
DBV
6
250
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
NSR
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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