SM73302
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SNOSB93A – AUGUST 2011 – REVISED APRIL 2013
SM73302 88 MHz, Precision, Low Noise, 1.8V CMOS Input, Decompensated Operational
Amplifier
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FEATURES
DESCRIPTION
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The SM73302 low noise, CMOS input operational
amplifier offers a low input voltage noise density of
5.8 nV/√Hz while consuming only 1.15 mA of
quiescent current. The SM73302 is stable at a gain of
10 and has a gain bandwidth (GBW) product of 88
MHz. The SM73302 has a supply voltage range of
1.8V to 5.5V and can operate from a single supply.
The SM73302 features a rail-to-rail output stage, and
is part of the precision amplifier family and is ideal for
a variety of instrumentation applications.
1
2
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(Typical 5V Supply, Unless Otherwise Noted)
Renewable Energy Grade
Input Offset Voltage ±150 µV (max)
Input Referred Voltage Noise 5.8 nV/√Hz
Input Bias Current 100 fA
Gain Bandwidth Product 88 MHz
Supply Voltage Range 1.8V to 5.5V
Supply Current 1.15mA
Rail-to-Rail Output Swing
– @ 10 kΩ Load 25 mV from Rail
– @ 2 kΩ Load 45 mV from Rail
2.5V and 5.0V Performance
Total Harmonic Distortion 0.04% @1 kHz, 600Ω
Temperature Range −40°C to 125°C
APPLICATIONS
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ADC Interface
Photodiode Amplifiers
Active Filters and Buffers
Low Noise Signal Processing
Medical Instrumentation
Sensor Interface Applications
The SM73302 provides optimal performance in low
voltage and low noise systems. A CMOS input stage,
with typical input bias currents in the range of a few
femtoamperes, and an input common mode voltage
range, which includes ground, makes the SM73302
ideal for low power sensor applications where high
speeds are needed.
The SM73302 is manufactured using an advanced
VIP50 process. The SM73302 is offered in a 5-Pin
SOT-23 package.
Typical Application
CF
RF
IIN
CCM
CD
VB
+
+
VOUT
CIN = CD + CCM
VOUT
= - RF
IIN
VOLTAGE NOISE (nV/ Hz)
1000
5V
100
2.5V
10
1
0.1
1
10
100
1k
10k
FREQUENCY (Hz)
Figure 1. Photodiode Transimpedance Amplifier
Figure 2. Input Referred Voltage Noise vs.
Frequency
1
2
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
All trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2011–2013, Texas Instruments Incorporated
SM73302
SNOSB93A – AUGUST 2011 – REVISED APRIL 2013
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These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
Absolute Maximum Ratings (1) (2)
ESD Tolerance (3)
Human Body Model
Machine Model
Charge-Device Model
VIN Differential
2000V
200V
1000V
±0.3V
Supply Voltage (V+ – V−)
6.0V
Input/Output Pin Voltage
V+ +0.3V, V− −0.3V
−65°C to 150°C
Storage Temperature Range
Junction Temperature (4)
+150°C
For soldering specifications: see http://www.ti.com/lit/SNOA549
(1)
(2)
(3)
(4)
Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for
which the device is intended to be functional. For specifications and the test conditions, see the Electrical Characteristics Tables.
If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/ Distributors for availability and
specifications.
Human Body Model, applicable std. MIL-STD-883, Method 3015.7. Machine Model, applicable std. JESD22-A115-A (ESD MM std. of
JEDEC)Field-Induced Charge-Device Model, applicable std. JESD22-C101-C (ESD FICDM std. of JEDEC).
The maximum power dissipation is a function of TJ(MAX), θJA. The maximum allowable power dissipation at any ambient temperature is
PD = (TJ(MAX) - TA)/θJA. All numbers apply for packages soldered directly onto a PC Board.
Operating Ratings (1)
Temperature Range (2)
Supply Voltage (V+ – V−)
Package Thermal Resistance (θJA (2))
(1)
(2)
2
−40°C to 125°C
−40°C ≤ TA ≤ 125°C
2.0V to 5.5V
0°C ≤ TA ≤ 125°C
1.8V to 5.5V
5-Pin SOT-23
180°C/W
Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for
which the device is intended to be functional. For specifications and the test conditions, see the Electrical Characteristics Tables.
The maximum power dissipation is a function of TJ(MAX), θJA. The maximum allowable power dissipation at any ambient temperature is
PD = (TJ(MAX) - TA)/θJA. All numbers apply for packages soldered directly onto a PC Board.
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2.5V Electrical Characteristics (1)
Unless otherwise specified, all limits are specified for TA = 25°C, V+ = 2.5V, V− = 0V, VCM = V+/2 = VO. Boldface limits apply
at the temperature extremes.
Symbol
Parameter
Conditions
Min (2)
Typ (3)
Max (2)
Units
VOS
Input Offset Voltage
±20
±180
±480
µV
TC VOS
Input Offset Voltage Temperature
Drift (4) (5)
−1.0
±4
μV/°C
IB
Input Bias Current
−40°C ≤ TA ≤ 85°C
0.05
1
25
−40°C ≤ TA ≤ 125°C
0.05
1
100
.006
0.5
50
VCM = 1.0V (6) (5)
IOS
Input Offset Current
VCM = 1.0V (5)
CMRR
Common Mode Rejection Ratio
0V ≤ VCM ≤ 1.4V
83
80
94
PSRR
Power Supply Rejection Ratio
2.0V ≤ V+ ≤ 5.5V, VCM = 0V
85
80
100
1.8V ≤ V+ ≤ 5.5V, VCM = 0V
85
98
Common Mode Voltage Range
CMRR ≥ 60 dB
CMRR ≥ 55 dB
AVOL
Open Loop Voltage Gain
VOUT = 0.15V to 2.2V,
RL = 2 kΩ to V+/2
88
82
98
VOUT = 0.15V to 2.2V,
RL = 10 kΩ to V+/2
92
88
110
Output Voltage Swing High
Output Voltage Swing Low
IOUT
Output Current
IS
Supply Current
SR
Slew Rate
pA
dB
dB
CMVR
VOUT
pA
−0.3
−0.3
1.5
1.5
dB
RL = 2 kΩ to V+/2
25
70
77
RL = 10 kΩ to V+/2
20
60
66
RL = 2 kΩ to V+/2
30
70
73
RL = 10 kΩ to V+/2
15
60
62
Sourcing to V−
VIN = 200 mV (7)
36
30
47
Sinking to V+
VIN = –200 mV (7)
7.5
5
15
0.95
AV = +10, Rising (10% to 90%)
32
AV = +10, Falling (90% to 10%)
24
V
mV from
either rail
mA
1.30
1.65
mA
V/μs
GBW
Gain Bandwidth
AV = +10, RL = 10 kΩ
88
MHz
en
Input Referred Voltage Noise Density
f = 1 kHz
6.2
nV/√Hz
in
Input Referred Current Noise Density
f = 1 kHz
0.01
pA/√Hz
THD+N
Total Harmonic Distortion + Noise
f = 1 kHz, AV = 1, RL = 600Ω
0.01
%
(1)
(2)
(3)
(4)
(5)
(6)
(7)
Electrical Table values apply only for factory testing conditions at the temperature indicated. Factory testing conditions result in very
limited self-heating of the device such that TJ = TA. No indication of parametric performance exists in the electrical tables under
conditions of internal self-heating where TJ > TA.
Limits are 100% production tested at 25°C. Limits over the operating temperature range are specified through correlations using the
statistical quality control (SQC) method.
Typical values represent the most likely parametric norm as determined at the time of characterization. Actual typical values may vary
over time and will also depend on the application and configuration. The typical values are not tested and are not specified on shipped
production material.
Offset voltage average drift is determined by dividing the change in VOS by temperature change.
Parameter is specified by design and/or characterization and is not test in production.
Positive current corresponds to current flowing into the device.
The short circuit test is a momentary test, the short circuit duration is 1.5 ms.
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5V Electrical Characteristics (1)
Unless otherwise specified, all limits are specified for TA = 25°C, V+ = 5V, V− = 0V, VCM = V+/2 = VO. Boldface limits apply at
the temperature extremes.
Symbo
l
VOS
Parameter
Min (2)
Typ (3)
Max (2)
Units
±10
±150
±450
µV
−1.0
±4
μV/°C
−40°C ≤ TA ≤ 85°C
0.1
1
25
−40°C ≤ TA ≤ 125°C
0.1
1
100
.01
0.5
50
Conditions
Input Offset Voltage
TC VOS Input Offset Voltage Temperature
Drift (4) (5)
IB
Input Bias Current
VCM = 2.0V (6) (5)
IOS
Input Offset Current
VCM = 2.0V (5)
CMRR
Common Mode Rejection Ratio
0V ≤ VCM ≤ 3.7V
85
80
100
PSRR
Power Supply Rejection Ratio
2.0V ≤ V+ ≤ 5.5V, VCM = 0V
85
80
100
1.8V ≤ V+ ≤ 5.5V, VCM = 0V
85
98
Common Mode Voltage Range
CMRR ≥ 60 dB
CMRR ≥ 55 dB
AVOL
Open Loop Voltage Gain
VOUT = 0.3V to
4.7V,
RL = 2 kΩ to V+/2
88
82
107
VOUT = 0.3V to
4.7V,
RL = 10 kΩ to V+/2
92
88
110
Output Voltage Swing High
Output Voltage Swing Low
(1)
(2)
(3)
(4)
(5)
(6)
4
pA
dB
dB
CMVR
VOUT
pA
−0.3
−0.3
4
4
V
dB
RL = 2 kΩ to V+/2
35
70
77
RL = 10 kΩ to V+/2
25
60
66
RL = 2 kΩ to V+/2
42
70
73
RL = 10 kΩ to V+/2
25
60
66
mV from
either rail
Electrical Table values apply only for factory testing conditions at the temperature indicated. Factory testing conditions result in very
limited self-heating of the device such that TJ = TA. No indication of parametric performance exists in the electrical tables under
conditions of internal self-heating where TJ > TA.
Limits are 100% production tested at 25°C. Limits over the operating temperature range are specified through correlations using the
statistical quality control (SQC) method.
Typical values represent the most likely parametric norm as determined at the time of characterization. Actual typical values may vary
over time and will also depend on the application and configuration. The typical values are not tested and are not specified on shipped
production material.
Offset voltage average drift is determined by dividing the change in VOS by temperature change.
Parameter is specified by design and/or characterization and is not test in production.
Positive current corresponds to current flowing into the device.
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5V Electrical Characteristics(1) (continued)
Unless otherwise specified, all limits are specified for TA = 25°C, V+ = 5V, V− = 0V, VCM = V+/2 = VO. Boldface limits apply at
the temperature extremes.
Symbo
l
IOUT
Min (2)
Typ (3)
Sourcing to V−
VIN = 200 mV (7)
46
38
60
Sinking to V+
VIN = –200 mV (7)
10.5
6.5
21
Parameter
Conditions
Output Short Circuit Current
IS
Supply Current
SR
Slew Rate
1.15
AV = +10, Rising (10% to 90%)
35
AV = +10, Falling (90% to 10%)
28
Max (2)
Units
mA
1.40
1.75
mA
V/μs
GBW
Gain Bandwidth
AV = +10, RL = 10 kΩ
88
MHz
en
Input Referred Voltage Noise Density
f = 1 kHz
5.8
nV/√Hz
in
Input Referred Current Noise Density
f = 1 kHz
0.01
pA/√Hz
f = 1 kHz, AV = 1, RL = 600Ω
0.01
%
THD+N Total Harmonic Distortion + Noise
(7)
The short circuit test is a momentary test, the short circuit duration is 1.5 ms.
Connection Diagram
OUTPUT
V
-
5
1
2
+
+IN
+
V
3
4
-IN
Figure 3. 5-Pin SOT-23, Top View
See DBV Package
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Typical Performance Characteristics
Unless otherwise specified, TA = 25°C, V– = 0, V+ = 5V, VS = V+ - V−, VCM = VS/2.
TCVOS Distribution
Offset Voltage Distribution
25
25
-40°C d TA d 125qC
VS = 2.5V, 5V
20
PERCENTAGE (%)
PERCENTAGE (%)
20 V
CM = VS/2
UNITS TESTED:
10,000
15
10
5
VS = 5V
VCM = VS/2
UNITS TESTED: 10,000
15
10
5
0
-4
-3
-2
-1
0
1
0
-200
2
100
Figure 4.
Figure 5.
200
Offset Voltage Distribution
25
-40°C d TA d 125°C
VS = 2.5V, 5V
20 VCM = VS/2
UNITS TESTED:
10,000
15
20
PERCENTAGE (%)
PERCENTAGE (%)
0
OFFSET VOLTAGE (PV)
TCVOS Distribution
25
-100
TCVOS (PV/°C)
10
5
VS = 2.5V
VCM = VS/2
UNITS TESTED:10,000
15
10
5
0
-4
-3
-2
-1
0
-200
0
-100
0
100
TCVOS (PV/°C)
OFFSET VOLTAGE (PV)
Figure 6.
Figure 7.
Supply Current
vs.
Supply Voltage
Offset Voltage
vs.
VCM
2
200
200
VS = 1.8V
1.6
125°C
OFFSET VOLTAGE (PV)
SUPPLY CURRENT (mA)
150
25°C
1.2
0.8
-40°C
0.4
-40°C
100
50
25°C
0
-50
125°C
-100
-150
0
1.5
2.5
3.5
4.5
5.5 6.0
+
0
0.3
0.6
0.9
1.2
1.5
VCM (V)
V (V)
Figure 8.
6
-200
-0.3
Figure 9.
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Typical Performance Characteristics (continued)
Unless otherwise specified, TA = 25°C, V– = 0, V+ = 5V, VS = V+ - V−, VCM = VS/2.
Offset Voltage
vs.
VCM
Offset Voltage
vs.
VCM
200
200
VS = 2.5V
VS = 5V
150
-40°C
OFFSET VOLTAGE (PV)
OFFSET VOLTAGE (PV)
150
100
25°C
50
0
125°C
-50
-100
-150
0
0.3
0.6
0.9
1.2
1.5
1.8
50
25°C
0
125°C
-50
-100
-200
-0.3
2.1
0.7
1.7
2.7
3.7
VCM (V)
VCM (V)
Figure 10.
Figure 11.
Offset Voltage
vs.
Temperature
Slew Rate
vs.
Supply Voltage
150
36
100
34
50
32
4.7
RISING EDGE
SLEW RATE (V/Ps)
OFFSET VOLTAGE (PV)
-40°C
-150
-200
-0.3
VS = 2.5V
0
SM73302
-50
-100
VS = 5V
-150
30
28
26
FALLING EDGE
24
-200
-40 -20
0
20
40
60
22
1.5
80 100 120 125
2.5
3.5
4.5
5.5
+
TEMPERATURE (°C)
V (V)
Figure 12.
Figure 13.
Input Bias Current Over Temperature
Input Bias Current
vs.
VCM
50
1000
VS = 5V
25°C
500
+
V = 5V
40
30
0
20
-500
IBIAS (pA)
INPUT BIAS CURRENT (fA)
100
-40°C
-1000
-1500
125°C
10
0
85°C
-10
-20
-2000
-30
-2500
-40
-50
-3000
0
1
2
3
4
0
1
2
3
4
VCM (V)
VCM (V)
Figure 14.
Figure 15.
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Typical Performance Characteristics (continued)
Unless otherwise specified, TA = 25°C, V– = 0, V+ = 5V, VS = V+ - V−, VCM = VS/2.
Sourcing Current
vs.
Supply Voltage
200
80
150
70
100
50
25°C
0
125°C
-50
125°C
60
-40°C
ISOURCE (mA)
OFFSET VOLTAGE (PV)
Offset Voltage
vs.
Supply Voltage
-40°C
50
25°C
40
30
-100
20
-150
10
0
-200
1.5
2.5
3.5
4.5
5.5
1
6
2
3
4
5
6
+
VS (V)
V (V)
Figure 16.
Figure 17.
Sinking Current
vs.
Supply Voltage
Sourcing Current
vs.
Output Voltage
35
70
30
60
125°C
125°C
50
ISOURCE (mA)
ISINK (mA)
25
25°C
20
15
-40°C
10
-40°C
40
25°C
30
20
5
10
0
0
1
2
3
4
5
6
0
1
2
+
3
4
V (V)
VOUT (V)
Figure 18.
Figure 19.
Sinking Current
vs.
Output Voltage
Positive Output Swing
vs.
Supply Voltage
30
5
40
RLOAD = 10 k:
125°C
35
VOUT FROM RAIL (mV)
25
ISINK (mA)
20
25°C
15
10
-40°C
30
125°C
25
25°C
20
-40°C
15
10
5
5
0
0
8
1
2
3
4
0
1.8
5
2.5
3.2
3.9
4.6
5.3
6
+
VOUT (V)
V (V)
Figure 20.
Figure 21.
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Typical Performance Characteristics (continued)
Unless otherwise specified, TA = 25°C, V– = 0, V+ = 5V, VS = V+ - V−, VCM = VS/2.
Negative Output Swing
vs.
Supply Voltage
Positive Output Swing
vs.
Supply Voltage
25
50
-40°C
45
20
VOUT FROM RAIL (mV)
VOUT FROM RAIL (mV)
25°C
15
125°C
10
5
125°C
40
25°C
35
30
25
20
-40°C
15
10
RLOAD = 10 k:
0
1.8
2.5
3.2
5
3.9
4.6
5.3
RLOAD = 2 k:
0
1.8
2.5
3.2
6
3.9
+
V (V)
Figure 23.
Negative Output Swing
vs.
Supply Voltage
Positive Output Swing
vs.
Supply Voltage
-40°C
90
25°C
40
VOUT FROM RAIL (mV)
VOUT FROM RAIL (mV)
6
100
50
35
125°C
30
25
20
15
125°C
70
50
40
10
5.3
-40°C
30
20
4.6
25°C
60
5
3.9
RLOAD = 600:
80
10
RLOAD = 2 k:
0
1.8
2.5
3.2
0
1.8
6
2.5
3.2
3.9
+
4.6
5.3
6
+
V (V)
V (V)
Figure 24.
Figure 25.
Negative Output Swing
vs.
Supply Voltage
Input Referred Voltage Noise
vs.
Frequency
1000
120
125°C
RLOAD = 600:
25°C
VOLTAGE NOISE (nV/ Hz)
100
VOUT FROM RAIL (mV)
5.3
Figure 22.
45
80
-40°C
60
40
20
0
1.8
4.6
+
V (V)
2.5
3.2
3.9
4.6
5.3
2.5V
10
1
0.1
6
5V
100
1
10
100
+
V (V)
FREQUENCY (Hz)
Figure 26.
Figure 27.
1k
10k
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Typical Performance Characteristics (continued)
Unless otherwise specified, TA = 25°C, V– = 0, V+ = 5V, VS = V+ - V−, VCM = VS/2.
Overshoot and Undershoot
vs.
CLOAD
Time Domain Voltage Noise
70
OVERSHOOT AND UNDERSHOOT %
VS = ±2.5V
400 nV/DIV
VCM = 0.0V
US%
60
50
OS%
40
30
20
10
0
1 s/DIV
20
0
40
80
60
100
120
CLOAD (pF)
Figure 28.
Figure 29.
THD+N
vs.
Frequency
THD+N
vs.
Frequency
0.04
0.05
RL = 600:
RL = 600:
0.04
THD+N (%)
THD+N (%)
0.03
+
0.03
V = 2.5V
0.02
AV = +10
VO = 1 VPP
0.02
VO = 4 VPP
0.01
0.01
+
V = 5V
AV = +10
RL = 100 k:
0
10
100
RL = 100 k:
1k
10k
0
10
100k
100
1k
10k
100k
FREQUENCY (Hz)
FREQUENCY (Hz)
Figure 30.
Figure 31.
THD+N
vs.
Peak-to-Peak Output Voltage (VOUT)
THD+N
vs.
Peak-to-Peak Output Voltage (VOUT)
-40
-40
-50
-60
RL = 600:
-70
V+ = 2.5V
f = 1 kHz
AV = +10
-80
0.01
10
THD+N (dB)
THD+N (dB)
-50
1
-70
-80 V+ = 5V
f = 1 kHz
AV = +10
-90
0.01
RL = 100 k:
0.1
RL = 600:
-60
10
RL = 100 k:
0.1
1
OUTPUT AMPLITUDE (VPP)
OUTPUT AMPLITUDE (VPP)
Figure 32.
Figure 33.
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Typical Performance Characteristics (continued)
Unless otherwise specified, TA = 25°C, V– = 0, V+ = 5V, VS = V+ - V−, VCM = VS/2.
Closed Loop Output Impedance
vs.
Frequency
Open Loop Gain and Phase
100
1000
100
80
60
60
GAIN
40
40
20
20
+
V = 5V
0
CL = 20 pF
RL = 2 k:, 10 k:
-20
100k
1M
OUTPUT IMPEDANCE (:)
80
PHASE (°)
GAIN (dB)
PHASE
100
10
1
0
0.1
10k
-20
100M
10M
100k
1M
10M
100M
FREQUENCY (Hz)
FREQUENCY (Hz)
Figure 34.
Figure 35.
Crosstalk Rejection
Small Signal Transient Response, AV = +10
140
120
10 mV/DIV
CROSSTALK REJECTION RATION (dB)
160
100
80
60
40
VIN = 2 mVPP
f = 1 MHz, AV = +10
20
+
V = 5V, CL = 10 pF
0
1k
100k
10k
1M
10M
100 ns/DIV
100M
FREQUENCY (Hz)
Figure 37.
Large Signal Transient Response, AV = +10
Small Signal Transient Response, AV = +10
10 mV/DIV
200 mV/DIV
Figure 36.
VIN = 100 mVPP
VIN = 2 mVPP
f = 1 MHz, AV = +10
f = 1 MHz, AV = +10
V = 2.5V, CL = 10 pF
V = 2.5V, CL = 10 pF
+
+
100 ns/DIV
100 ns/DIV
Figure 38.
Figure 39.
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Typical Performance Characteristics (continued)
Unless otherwise specified, TA = 25°C, V– = 0, V+ = 5V, VS = V+ - V−, VCM = VS/2.
PSRR
vs.
Frequency
Large Signal Transient Response, AV = +10
100
-PSRR, 5V
90
PSRR (dB)
200 mV/DIV
80
-PSRR, 2.5V
+PSRR, 5V
+PSRR, 2.5V
70
60
VIN = 100 mVPP
50
f = 1 MHz, AV = +10
+
V = 5V, CL = 10 pF
40
100
100 ns/DIV
1k
10k
100k
1M
FREQUENCY (Hz)
Figure 40.
Figure 41.
CMRR
vs.
Frequency
Input Common Mode Capacitance
vs.
VCM
120
25
+
V = 5V
100
20
+
80
CCM (pF)
CMRR (dB)
V = 2.5V
60
+
V = 5V
15
10
40
5
20
0
10
100
1k
10k
100k
1M
10M
FREQUENCY (Hz)
0
1
2
3
4
VCM (V)
Figure 42.
12
0
Figure 43.
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APPLICATION INFORMATION
ADVANTAGES OF THE SM73302
Wide Bandwidth at Low Supply Current
The SM73302 is a high performance op amp that provides a GBW of 88 MHz with a gain of 10 while drawing a
low supply current of 1.15 mA. This makes it ideal for providing wideband amplification in data acquisition
applications.
With the proper external compensation, the SM73302 can be operated at gains of ±1 and still maintain much
faster slew rates than comparable unity gain stable amplifiers. The increase in bandwidth and slew rate is
obtained without any additional power consumption over the LMP7715.
Low Input Referred Noise and Low Input Bias Current
The SM73302 has a very low input referred voltage noise density (5.8 nV/√Hz at 1 kHz). A CMOS input stage
ensures a small input bias current (100 fA) and low input referred current noise (0.01 pA/√Hz). This is very
helpful in maintaining signal integrity, and makes the SM73302 ideal for audio and sensor based applications.
Low Supply Voltage
The SM73302 operates with a 2.5V and 5V supply. This device operates at all supply voltages between 2.0V and
5.5V, for ambient temperatures ranging from −40°C to 125°C, thus utilizing the entire battery lifetime. The
SM73302 is also operational at 1.8V supply voltage, for temperatures between 0°C and 125°C optimizing their
usage in low-voltage applications.
RRO and Ground Sensing
Rail-to-Rail output (RRO) swing provides the maximum possible dynamic range. This is particularly important
when operating at low supply voltages. An innovative positive feedback scheme is used to boost the current drive
capability of the output stage. This allows the SM73302 to source more than 40 mA of current at 1.8V supply.
This also limits the performance of this part as a comparator, and hence the usage of the SM73302 in an openloop configuration is not recommended. The input common-mode range includes the negative supply rail which
allows direct sensing at ground in single supply operation.
Small Size
The small footprint of the SM73302 package saves space on printed circuit boards, and enables the design of
smaller electronic products, such as cellular phones, pagers, or other portable systems. Long traces between the
signal source and the op amp make the signal path more susceptible to noise pick up.
The physically smaller SM73302 allows the op amp to be placed closer to the signal source, thus reducing noise
pickup and maintaining signal integrity.
USING THE DECOMPENSATED SM73302
Advantages of Decompensated Op Amp
A unity gain stable op amp, which is fully compensated, is designed to operate with good stability down to gains
of ±1. The large amount of compensation does provide an op amp that is relatively easy to use; however, a
decompensated op amp is designed to maximize the bandwidth and slew rate without any additional power
consumption. This can be very advantageous.
The SM73302 requires a gain of ±10 to be stable. However, with an external compensation network (a simple
RC network) these parts can be stable with gains of ±1 and still maintain the higher slew rate. Looking at the
Bode plots for the SM73302 and its closest equivalent unity gain stable op amp, the LMP7715, one can clearly
see the increased bandwidth of the SM73302. Both plots are taken with a parallel combination of 20 pF and 10
kΩ for the output load.
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100
100
80
80
60
60
GAIN
40
40
PHASE (°)
GAIN (dB)
PHASE
20
20
0
0
-20
1k
10k
100k
1M
-20
100M
10M
FREQUENCY (Hz)
100
100
80
80
60
60
40
40
20
20
0
0
-20
1k
10k
100k
1M
PHASE (°)
GAIN (dB)
Figure 44. SM73302 AVOL vs. Frequency
-20
100M
10M
FREQUENCY (Hz)
Figure 45. LMP7715 AVOL vs. Frequency
Figure 44 shows the much larger 88 MHz bandwidth of the SM73302 as compared to the 17 MHz bandwidth of
the LMP7715 shown in Figure 45. The decompensated SM73302 has five times the bandwidth of the LMP7715.
What is a Decompensated Op Amp?
The differences between the unity gain stable op amp and the decompensated op amp are shown in Figure 46.
This Bode plot assumes an ideal two pole system. The dominant pole of the decompensated op amp is at a
higher frequency, f1, as compared to the unity gain stable op amp which is at fd as shown in Figure 46. This is
done in order to increase the speed capability of the op amp while maintaining the same power dissipation of the
unity gain stable op amp. The SM73302 has a dominant pole at 1.6 kHz. The unity gain stable
LMP7715/LMP7716 have their dominant pole at 300 Hz.
DECOMPENSATED OP AMP
AOL
UNITY-GAIN STABLE OP AMP
Gmin
fGBWP
fd
f1
fu
f2
fu'
Figure 46. Open Loop Gain for Unity Gain Stable Op Amp and Decompensated Op Amp
14
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Having a higher frequency for the dominate pole will result in:
1. The DC open loop gain (AVOL) extending to a higher frequency.
2. A wider closed loop bandwidth.
3. Better slew rate due to reduced compensation capacitance within the op amp.
The second open loop pole (f2) for the SM73302 occurs at 45 MHz. The unity gain (fu’) occurs after the second
pole at 51 MHz. An ideal two pole system would give a phase margin of 45° at the location of the second pole.
The SM73302 has parasitic poles close to the second pole, giving a phase margin closer to 0°. Therefore it is
necessary to operate the SM73302 at a closed loop gain of 10 or higher, or to add external compensation in
order to assure stability.
For the LMP7715, the gain bandwidth product occurs at 17 MHz. The curve is constant from fd to fu which occurs
before the second pole.
For the SM73302 the GBW = 88 MHz and is constant between f1 and f2. The second pole at f2 occurs before
AVOL =1. Therefore fu’ occurs at 51 MHz, well before the GBW frequency of 88 MHz. For decompensated op
amps the unity gain frequency and the GBW are no longer equal. Gmin is the minimum gain for stability and for
the SM73302 this is a gain of 18 to 20 dB.
Input Lead-Lag Compensation
The recommended technique which allows the user to compensate the SM73302 for stable operation at any gain
is lead-lag compensation. The compensation components added to the circuit allow the user to shape the
feedback function to make sure there is sufficient phase margin when the loop gain is as low as 0 dB and still
maintain the advantages over the unity gain op amp. Figure 47 shows the lead-lag configuration. Only RC and C
are added for the necessary compensation.
RF
RIN
RC
C
+
Figure 47. SM73302 with Lead-Lag Compensation for Inverting Configuration
To cover how to calculate the compensation network values it is necessary to introduce the term called the
feedback factor or F. The feedback factor F is the feedback voltage VA-VB across the op amp input terminals
relative to the op amp output voltage VOUT.
F=
VA - V B
VOUT
(1)
From feedback theory the classic form of the feedback equation for op amps is:
VOUT
VIN
=
A
1 + AF
(2)
A is the open loop gain of the amplifier and AF is the loop gain. Both are highly important in analyzing op amps.
Normally AF >>1 and so the above equation reduces to:
VOUT
VIN
=
1
F
(3)
Deriving the equations for the lead-lag compensation is beyond the scope of this datasheet. The derivation is
based on the feedback equations that have just been covered. The inverse of feedback factor for the circuit in
Figure 47 is:
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©
§
¨
¨
©
§
= ¨¨1 +
RF §¨1 + s(Rc + RIN || RF) C
1 + sRcC
RIN ¨
©
§
¨
¨
©
1
F
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(4)
where 1/F's pole is located at
fp =
1
2SRcC
(5)
1/F's zero is located at
fz =
1
2S Rc + RIN || RF)C
(6)
1
F
f=0
RF
=1+
RIN
(7)
The circuit gain for Figure 47 at low frequencies is −RF/RIN, but F, the feedback factor is not equal to the circuit
gain. The feedback factor is derived from feedback theory and is the same for both inverting and non-inverting
configurations. Yes, the feedback factor at low frequencies is equal to the gain for the non-inverting configuration.
f=f
©
§
¨
¨
©
F
= ¨¨1 +
RIN || RF
RF §¨
1+
RC
RIN ¨©
§
¨
¨
©
§
1
(8)
From this formula, we can see that
• 1/F's zero is located at a lower frequency compared with 1/F's pole.
• 1/F's value at low frequency is 1 + RF/RIN.
• This method creates one additional pole and one additional zero.
• This pole-zero pair will serve two purposes:
– To raise the 1/F value at higher frequencies prior to its intercept with A, the open loop gain curve, in order
to meet the Gmin = 10 requirement. For the SM73302 some overcompensation will be necessary for good
stability.
– To achieve the previous purpose above with no additional loop phase delay.
Please note the constraint 1/F ≥ Gmin needs to be satisfied only in the vicinity where the open loop gain A and
1/F intersect; 1/F can be shaped elsewhere as needed. The 1/F pole must occur before the intersection with the
open loop gain A.
In order to have adequate phase margin, it is desirable to follow these two rules:
Rule 1 1/F and the open loop gain A should intersect at the frequency where there is a minimum of 45° of phase
margin. When over-compensation is required the intersection point of A and 1/F is set at a frequency
where the phase margin is above 45°, therefore increasing the stability of the circuit.
Rule 2 1/F’s pole should be set at least one decade below the intersection with the open loop gain A in order to
take advantage of the full 90° of phase lead brought by 1/F’s pole which is F’s zero. This ensures that the
effect of the zero is fully neutralized when the 1/F and A plots intersect each other.
Calculating Lead-Lag Compensation for SM73302
Figure 48 is the same plot as Figure 44, but the AVOL and phase curves have been redrawn as smooth lines to
more readily show the concepts covered, and to clearly show the key parameters used in the calculations for
lead-lag compensation.
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100
PHASE
ADDITIONAL
COMPENSATION
GAIN (dB) and PHASE (°C)
80
AVOL
60
45° PHASE
MARGIN
40
20
1
with ADDITIONAL
F
COMPENSATION
1
F
0
-20
1k
2nd POLE
f2
10k
100k
1M
GBP
10M
100M
FREQUENCY (Hz)
Figure 48. SM73302 Simplified Bode Plot
To obtain stable operation with gains under 10 V/V the open loop gain margin must be reduced at high
frequencies to where there is a 45° phase margin when the gain margin of the circuit with the external
compensation is 0 dB. The pole and zero in F, the feedback factor, control the gain margin at the higher
frequencies. The distance between F and AVOL is the gain margin; therefore, the unity gain point (0 dB) is where
F crosses the AVOL curve.
For the example being used RIN = RF for a gain of −1. Therefore F = 6 dB at low frequencies. At the higher
frequencies the minimum value for F is 18 dB for 45° phase margin. From Equation 8 we have the following
relationship:
RF
RIN
§
¨
¨
©
+
§
¨
¨1
©
+
RIN || RF
RC
§
¨
¨
©
§
¨1
¨
©
= 18 dB = 7.9
(9)
Now set RF = RIN = R. With these values and solving for RC we have RC = R/5.9. Note that the value of C does
not affect the ratio between the resistors. Once the value of the resistors is set, then the position of the pole in F
must be set. A 2 kΩ resistor is used for RF and RIN in this design. Therefore the value for RC is set at 330Ω, the
closest standard value for 2 kΩ/5.9.
Rewriting Equation 5 to solve for the minimum capacitor value gives the following equation:
C = 1/(2πfpRC)
(10)
The feedback factor curve, F, intersects the AVOL curve at about 12 MHz. Therefore the pole of F should not be
any larger than 1.2 MHz. Using this value and RC = 330Ω the minimum value for C is 390 pF. Figure 49 shows
that there is too much overshoot, but the part is stable. Increasing C to 2.2 nF did not improve the ringing, as
shown in Figure 50.
Figure 49. First Try at Compensation, Gain = −1
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Figure 50. C Increased to 2.2 nF, Gain = −1
Some over-compensation appears to be needed for the desired overshoot characteristics. Instead of intersecting
the AVOL curve at 18 dB, 2 dB of over-compensation will be used, and the AVOL curve will be intersected at 20
dB. Using Equation 8 for 20 dB, or 10 V/V, the closest standard value of RC is 240Ω. The following two
waveforms show the new resistor value with C = 390 pF and 2.2 nF. Figure 52 shows the final compensation and
a very good response for the 1 MHz square wave.
Figure 51. RC = 240Ω and C = 390 pF, Gain = −1
Figure 52. RC = 240Ω and C = 2.2 nF, Gain = −1
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To summarize, the following steps were taken to compensate the SM73302 for a gain of −1:
1. Values for Rc and C were calculated from the Bode plot to give an expected phase margin of 45°. The values
were based on RIN = RF = 2 kΩ. These calculations gave Rc = 330Ω and C = 390 pF.
2. To reduce the ringing C was increased to 2.2 nF which moved the pole of F, the feedback factor, farther
away from the AVOL curve.
3. There was still too much ringing so 2 dB of over-compensation was added to F. This was done by
decreasing RC to 240Ω.
The LMP7715 is the fully compensated part (without the Renewable Energy Grade), which is comparable to the
SM73302 . Using the LMP7715 in the same setup, but removing the compensation network, provided the
response shown in Figure 53 .
Figure 53. LMP7715 Response
For large signal response the rise and fall times are dominated by the slew rate of the op amps. Even though
both parts are quite similar the SM73302 will give rise and fall times about 2.5 times faster than the LMP7715.
This is possible because the SM73302 is a decompensated op amp and even though it is being used at a gain of
−1, the speed is preserved by using a good technique for external compensation.
Non-Inverting Compensation
For the non-inverting amp the same theory applies for establishing the needed compensation. When setting the
inverting configuration for a gain of −1, F has a value of 2. For the non-inverting configuration both F and the
actual gain are the same, making the non-inverting configuration more difficult to compensate. Using the same
circuit as shown in Figure 47, but setting up the circuit for non-inverting operation (gain of +2) results in similar
performance as the inverting configuration with the inputs set to half the amplitude to compensate for the
additional gain. Figure 54 below shows the results.
Figure 54. RC = 240Ω and C = 2.2 nF, Gain = +2
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Figure 55. LMP7715 Response Gain = +2
The response shown in Figure 54 is close to the response shown in Figure 52. The part is actually slightly faster
in the non-inverting configuration. Decreasing the value of RC to around 200Ω can decrease the negative
overshoot but will have slightly longer rise and fall times. The other option is to add a small resistor in series with
the input signal. Figure 55 shows the performance of the LMP7715 with no compensation. Again the
decompensated parts are almost 2.5 times faster than the fully compensated op amp.
The most difficult op amp configuration to stabilize is the gain of +1. With proper compensation the SM73302 can
be used in this configuration and still maintain higher speeds than the fully compensated parts. Figure 56 shows
the gain = 1, or the buffer configuration, for these parts.
RF
RC
RP
C
+
Figure 56. SM73302 with Lead-Lag Compensation for Non-Inverting Configuration
Figure 56 is the result of using Equation 8 and additional experimentation in the lab. RP is not part of Equation 8,
but it is necessary to introduce another pole at the input stage for good performance at gain = +1. Equation 8 is
shown below with RIN = ∞.
+
RF
Rc
§
¨
¨
©
§
¨1
¨
©
= 18 dB = 7.9
(11)
Using 2 kΩ for RF and solving for RC gives RC = 2000/6.9 = 290Ω. The closest standard value for RC is 300Ω.
After some fine tuning in the lab RC = 330Ω and RP = 1.5 kΩ were chosen as the optimum values. RP together
with the input capacitance at the non-inverting pin inserts another pole into the compensation for the SM73302.
Adding this pole and slightly reducing the compensation for 1/F (using a slightly higher resistor value for RC)
gives the optimum response for a gain of +1. Figure 57 is the response of the circuit shown in Figure 56.
Figure 58 shows the response of the LMP7715 in the buffer configuration with no compensation and RP = RF = 0.
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Figure 57. RC = 330Ω and C = 10 nF, Gain = +1
Figure 58. LMP7715 Response Gain = +1
With no increase in power consumption the decompensated op amp offers faster speed than the compensated
equivalent part . These examples used RF = 2 kΩ. This value is high enough to be easily driven by the SM73302,
yet small enough to minimize the effects from the parasitic capacitance of both the PCB and the op amp.
Note: When using the SM73302, proper high frequency PCB layout must be followed. The GBW of these parts is
88 MHz, making the PCB layout significantly more critical than when using the compensated counterparts which
have a GBW of 17 MHz.
TRANSIMPEDANCE AMPLIFIER
An excellent application for the SM73302 is as a transimpedance amplifier. With a GBW product of 88 MHz this
part is ideal for high speed data transmission by light. The circuit shown on the front page of the datasheet is the
circuit used to test the SM73302 as a transimpedance amplifier. The only change is that VB is tied to the VCC of
the part, thus the direction of the diode is reversed from the circuit shown on the front page.
Very high speed components were used in testing to check the limits of the SM73302 in a transimpedance
configuration. The photodiode part number is PIN-HR040 from OSI Optoelectronics. The diode capacitance for
this part is only about 7 pF for the 2.5V bias used (VCC to virtual ground). The rise time for this diode is 1 nsec. A
laser diode was used for the light source. Laser diodes have on and off times under 5 nsec. The speed of the
selected optical components allowed an accurate evaluation of the SM73302 as a transimpedance amplifier. An
evaluation board for decompensated op amps, PN 551013271-001 A, was used and only minor modifications
were necessary and no traces had to be cut.
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CF
2.5V
2.5V
RF
DPHOTO
CD
VOUT
CCM
+
-2.5V
Figure 59. Transimpedance Amplifier
Figure 59 is the complete schematic for a transimpedance amplifier. Only the supply bypass capacitors are not
shown. CD represents the photodiode capacitance which is given on its datasheet. CCM is the input common
mode capacitance of the op amp and, for the SM73302 it is shown in Figure 43. In Figure 59 the inverting input
pin of the SM73302 is kept at virtual ground. Even though the diode is connected to the 2.5V line, a power
supply line is AC ground, thus CD is connected to ground.
Figure 60 shows the schematic needed to derive F, the feedback factor, for a transimpedance amplifier. In this
figure CD + CCM = CIN. Therefore it is critical that the designer knows the diode capacitance and the op amp input
capacitance. The photodiode is close to an ideal current source once its capacitance is included in the model.
What kind of circuit is this? Without CF there is only an input capacitor and a feedback resistor. This circuit is a
differentiator! Remember, differentiator circuits are inherently unstable and must be compensated. In this case CF
compensates the circuit.
CF
RF
VA
IDIODE
VOUT
CIN
+
Figure 60. Transimpedance Feedback Model
Using feedback theory, F = VA/VOUT, this becomes a voltage divider giving the following equation:
F=
1 + sCFRF
1 + sRF (CF + CIN)
(12)
The noise gain is 1/F. Because this is a differentiator circuit, a zero must be inserted. The location of the zero is
given by:
´
¶z
=
1
1 + sRF (CF + CIN)
(13)
CF has been added for stability. The addition of this part adds a pole to the circuit. The pole is located at:
´
¶p
=
1
1 + sCFRF
(14)
To attain maximum bandwidth and still have good stability the pole is to be located on the open loop gain curve
which is A. If additional compensation is required one can always increase the value of CF, but this will also
reduce the bandwidth of the circuit. Therefore A = 1/F, or AF = 1. For A the equation is:
A=
22
ZGBW
Z =
´
¶GBW
´
¶
(15)
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The expression fGBW is the gain bandwidth product of the part. For a unity gain stable part this is the frequency
where A = 1. For the SM73302 fGBW = 88 MHz. Multiplying A and F results in the following equation:
´
¶ GBW
´
¶
´
¶ GBW
x
´
¶
1 + sCFRF
1 + sRF (CF + CIN)
1+
x
§
¨
¨
©
1+
§ CFRF
¨
¨
© CFRF
=
2
=1
RF (CF + CIN)
CFRF
§
¨
¨
©
P
§
¨
¨
©
AF ´ =
¶
2
(16)
For the above equation s = jω. To find the actual amplitude of the equation the square root of the square of the
real and imaginary parts are calculated. At the intersection of F and A, we have:
Z=
1
CFRF
(17)
After a bit of algebraic manipulation the above equation reduces to:
§C + C
IN
¨ F
¨
CF
©
§
¨
¨
©
1+
2
2
= 8S2 ´
¶GBW RF CF
2
2
(18)
In the above equation the only unknown is CF. In trying to solve this equation the fourth power of CF must be
dealt with. An excel spread sheet with this equation can be used and all the known values entered. Then through
iteration, the value of CF when both sides are equal will be found. That is the correct value for CF and of course
the closest standard value is used for CF.
Before moving to the lab, the transfer function of the transimpedance amplifier must be found and the units must
be in Ohms.
VOUT =
-RF
1 + sCFRF
x IDIODE
(19)
The SM73302 was evaluated for RF = 10 kΩ and 100 kΩ, representing a somewhat lower gain configuration and
with the 100 kΩ feedback resistor a fairly high gain configuration. The RF = 10 kΩ is covered first. Looking at the
Input Common Mode Capacitance vs. VCM chart for CCM for the operating point selected CCM = 15 pF. Note that
for split supplies VCM = 2.5V, CIN = 22 pF and fGBW = 88 MHz. Solving for CF the calculated value is 1.75 pF, so
1.8 pF is selected for use. Checking the frequency of the pole finds that it is at 8.8 MHz, which is right at the
minimum gain recommended for this part. Some over compensation was necessary for stability and the final
selected value for CF is 2.7 pF. This moves the pole to 5.9 MHz. Figure 61 and Figure 62 show the rise and fall
times obtained in the lab with a 1V output swing. The laser diode was difficult to drive due to thermal effects
making the starting and ending point of the pulse quite different, therefore the two separate scope pictures.
Figure 61. Fall Time
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SM73302
SNOSB93A – AUGUST 2011 – REVISED APRIL 2013
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Figure 62. Rise Time
In Figure 61 the ringing and the hump during the on time is from the laser. The higher drive levels for the laser
gave ringing in the light source as well as light changing from the thermal characteristics. The hump is due to the
thermal characteristics.
Solving for CF using a 100 kΩ feedback resistor, the calculated value is 0.54 pF. One of the problems with more
gain is the very small value for CF. A 0.5 pF capacitor was used, its measured value being 0.64 pF. For the 0.64
pF location the pole is at 2.5 MHz. Figure 63 shows the response for a 1V output.
Figure 63. High Gain Response
A transimpedance amplifier is an excellent application for the SM73302. Even with the high gain using a 100 kΩ
feedback resistor, the bandwidth is still well over 1 MHz. Other than a little over compensation for the 10 kΩ
feedback resistor configuration using the SM73302 was quite easy. Of course a very good board layout was also
used for this test.
24
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SM73302
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SNOSB93A – AUGUST 2011 – REVISED APRIL 2013
REVISION HISTORY
Changes from Original (April 2013) to Revision A
•
Page
Changed layout of National Data Sheet to TI format .......................................................................................................... 24
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25
PACKAGE OPTION ADDENDUM
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10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
SM73302MF/NOPB
ACTIVE
SOT-23
DBV
5
1000
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 125
SC3B
SM73302MFE/NOPB
ACTIVE
SOT-23
DBV
5
250
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 125
SC3B
SM73302MFX/NOPB
ACTIVE
SOT-23
DBV
5
3000
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 125
SC3B
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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