TPA112DR

TPA112 www.ti.com SLOS212E – AUGUST 1998 – REVISED JUNE 2004 150-mW STEREO AUDIO POWER AMPLIFIER FEATURES • • • • • DESCRIPTION 150-mW Stereo Output Wide Range of Supply Voltages – Fully Specified for 3.3-V and 5-V Operation – Operational From 2.5 V to 5.5 V Thermal and Short-Circuit Protection Surface-Mount Packaging – PowerPAD™ MSOP – SOIC Standard Operational Amplifier Pinout D OR DGN PACKAGE (TOP VIEW) VO1 IN1– IN1+ GND 1 8 2 7 3 6 4 5 VDD VO2 IN2– IN2+ The TPA112 is a stereo audio power amplifier packaged in an 8-pin PowerPAD™ MSOP package capable of delivering 150 mW of continuous RMS power per channel into 8-Ω loads. Amplifier gain is externally configured by means of two resistors per input channel and does not require external compensation for settings of 1 to 10. THD+N when driving an 8-Ω load from 5 V is 0.1% at 1 kHz, and less than 2% across the audio band of 20 Hz to 20 kHz. For 32-Ω loads, the THD+N is reduced to less than 0.06% at 1 kHz, and is less than 1% across the audio band of 20 Hz to 20 kHz. For 10-kΩ loads, the THD+N performance is 0.01% at 1 kHz, and less than 0.02% across the audio band of 20 Hz to 20 kHz. FUNCTIONAL BLOCK DIAGRAM RF CI RI LIN− CI VDD 8 VO1 1 Short-Circuit Protection RI LIN+ 2 IN1− 3 IN1+ VDD CC RO RC RF To Headphone Jack (See TPA152) VDD/2 CI RIN− CI RI RF 6 IN2− 5 IN2+ VO2 RO RI RIN+ CC 7 Over-Temperature Protection RC 4 RF Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. PowerPAD is a trademark of Texas Instruments. PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of the Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. Copyright © 1998–2004, Texas Instruments Incorporated TPA112 www.ti.com SLOS212E – AUGUST 1998 – REVISED JUNE 2004 These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam during storage or handling to prevent electrostatic damage to the MOS gates. AVAILABLE OPTIONS PACKAGED DEVICES (1) MSOP SYMBOLIZATION TA SMALL OUTLINE (1) (D) MSOP (1) (DGN) –40°C to 85°C TPA112D TPA112DGN TI AAD The D and DGN packages are available in left-ended tape and reel only (e.g., TPA112DR, TPA112DGNR). Terminal Functions TERMINAL NAME I/O NO. DESCRIPTION GND 4 I GND is the ground connection. IN1- 2 I IN1- is the inverting input for channel 1. IN1+ 3 I IN1+ is the noninverting input for channel 1. IN2- 6 I IN2- is the inverting input for channel 2. IN2+ 5 I IN2+ is the noninverting input for channel 2. VDD 8 I VDD is the supply voltage terminal. VO1 1 O VO1 is the audio output for channel 1. VO2 7 O VO2 is the audio output for channel 2. ABSOLUTE MAXIMUM RATINGS over operating free-air temperature range (unless otherwise noted) (1) UNIT VDD Supply voltage VI Differential input voltage 6V II Input current ±2.5 µA IO Output current ±250 mA –0.3 V to VDD + 0.3 V Continuous total power dissipation Internally llimited TJ Operating junction temperature range –40°C to 150°C Tstg Storage temperature range –65°C to 150°C Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds (1) 260°C Stresses beyond those listed under "absolute maximum ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated under "recommended operating conditions” is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability. DISSIPATION RATING TABLE PACKAGE (1) 2 TA ≤ 25°C POWER RATING DERATING FACTOR ABOVE TA = 25°C TA = 70°C POWER RATING TA = 85°C POWER RATING D 725 mW 5.8 mW/°C 464 mW 377 mW DGN 2.14 W (1) 17.1 mW/°C 1.37 W 1.11 W See the Texas Instruments document, PowerPAD Thermally Enhanced Package Application Report (SLMA002), for more information on the PowerPAD package. The thermal data was measured on a PCB layout based on the information in the section entitled Texas Instruments Recommended Board for PowerPAD, of that document. TPA112 www.ti.com SLOS212E – AUGUST 1998 – REVISED JUNE 2004 RECOMMENDED OPERATING CONDITIONS MIN MAX VDD Supply voltage 2.5 5.5 UNIT V TA Operating free-air temperature –40 85 °C DC ELECTRICAL CHARACTERISTICS at TA = 25°C, VDD = 3.3 V PARAMETER TEST CONDITIONS MIN TYP MAX VOO Output offset voltage 10 PSRR Power supply rejection ratio IDD(q) Supply current 1.5 ZI Input impedance >1 VDD = 3.2 V to 3.4 V 83 UNIT mV dB 3 mA MΩ AC OPERATING CHARACTERISTICS VDD = 3.3 V, TA = 25°C, RL = 8 Ω PARAMETER TEST CONDITIONS MIN TYP MAX UNIT 70 (1) mW PO Output power (each channel) THD ≤ 0.1% THD+N Total harmonic distortion + noise PO = 70 mW, 20 Hz–20 kHz BOM Maximum output power BW G = 10, THD < 5% Phase margin Open loop 58° Supply ripple rejection f = 1 kHz 68 dB 86 dB SVRR Channel/channel output separation f = 1 kHz SNR Signal-to-noise ratio PO = 100 mW Vn Noise output voltage (1) 2% > 20 kHz 100 dB 9.5 µV(rms) Measured at 1 kHz DC ELECTRICAL CHARACTERISTICS at TA = 25°C, VDD = 5 V PARAMETER TEST CONDITIONS MIN TYP MAX VOO Output offset voltage PSRR Power supply rejection ratio IDD(q) Supply current 1.5 ZI Input impedance >1 VDD = 4.9 V to 5.1 V UNIT 10 mV 3 mA 76 dB MΩ AC OPERATING CHARACTERISTICS VDD = 5 V, TA = 25°C, RL = 8 Ω PARAMETER TEST CONDITIONS MIN TYP MAX UNIT 70 (1) mW PO Output power (each channel) THD ≤ 0.1% THD+N Total harmonic distortion + noise PO = 150 mW, 20 Hz–20 kHz BOM Maximum output power BW G = 10, THD < 5% Phase margin Open loop 56° Supply ripple rejection f = 1 kHz 68 dB Channel/channel output separation f = 1 kHz 86 dB SNR Signal-to-noise ratio PO = 150 mW Vn Noise output voltage SVRR (1) 2% > 20 kHz 100 dB 9.5 µV(rms) Measured at 1 kHz 3 TPA112 www.ti.com SLOS212E – AUGUST 1998 – REVISED JUNE 2004 AC OPERATING CHARACTERISTICS VDD = 3.3 V, TA = 25°C, RL = 32 Ω TYP MAX UNIT PO Output power (each channel) PARAMETER THD ≤ 0.1% 40 (1) mW THD+N Total harmonic distortion + noise PO = 30 mW, 20 Hz–20 kHz 0.5% BOM Maximum output power BW G = 10, THD < 2% > 20 Phase margin Open loop 58° Supply ripple rejection f = 1 kHz 68 dB 86 dB SVRR TEST CONDITIONS Channel/channel output separation f = 1 kHz SNR Signal-to-noise ratio PO = 100 mW Vn Noise output voltage (1) MIN kHz 100 dB 9.5 µV(rms) Measured at 1 kHz AC OPERATING CHARACTERISTICS VDD = 5 V, TA = 25°C, RL = 32 Ω PARAMETER TEST CONDITIONS MIN TYP MAX UNIT mW PO Output power (each channel) THD ≤ 0.1% 40 (1) THD+N Total harmonic distortion + noise PO = 60 mW, 20 Hz–20 kHz 0.4% BOM Maximum output power BW G = 10, THD < 2% > 20 Phase margin Open loop 56° Supply ripple rejection f = 1 kHz 68 dB Channel/channel output separation f = 1 kHz 86 dB SNR Signal-to-noise ratio PO = 150 mW 100 dB Vn Noise output voltage 9.5 µV(rms) SVRR (1) 4 Measured at 1 kHz kHz TPA112 www.ti.com SLOS212E – AUGUST 1998 – REVISED JUNE 2004 TYPICAL CHARACTERISTICS Table of Graphs FIGURE THD+N Total harmonic distortion plus noise 1, 2, 4, 5, 7, 8, 10, 11, 13, 14, 16, 17, 34, 36 vs Frequency vs Output power 3, 6, 9, 12, 15, 18 PSSR Power supply rejection ratio vs Frequency Vn Output noise voltage vs Frequency 19, 20 21, 22 Crosstalk vs Frequency 23-26, 37, 38 Mute attenuation vs Frequency 27, 28 Open-loop gain vs Frequency 29, 30 Phase margin vs Frequency 29, 30 Phase vs Frequency 39-44 Output power vs Load resistance 31, 32 ICC Supply current vs Supply voltage 33 SNR Signal-to-noise ratio vs Voltage gain 35 Closed-loop gain vs Frequency 39-44 Power dissipation/amplifier vs Output power 45, 46 TOTAL HARMONIC DISTORTION + NOISE vs FREQUENCY TOTAL HARMONIC DISTORTION + NOISE vs FREQUENCY 1 10 VDD = 3.3 V PO = 30 mW CB = 1 µ F RL = 32 Ω THD+N −Total Harmonic Distortion + Noise − % THD+N −Total Harmonic Distortion + Noise − % 10 AV = 5 AV = 10 0.1 AV = 1 0.01 0.001 20 100 1k f − Frequency − Hz Figure 1. 10k 20k 1 0.1 VDD = 3.3 V AV = 1 V/V RL = 32 Ω CB = 1 µ F PO = 15 mW PO = 10 mW 0.01 PO = 30 mW 0.001 20 100 1k 10k 20k f − Frequency − Hz Figure 2. 5 TPA112 www.ti.com SLOS212E – AUGUST 1998 – REVISED JUNE 2004 TOTAL HARMONIC DISTORTION + NOISE vs OUTPUT POWER TOTAL HARMONIC DISTORTION + NOISE vs FREQUENCY 10 THD+N −Total Harmonic Distortion + Noise − % THD+N −Total Harmonic Distortion + Noise − % 10 VDD = 3.3 V RL = 32 Ω AV = 1 V/V CB = 1 µF 20 kHz 10 kHz 1 0.1 1 kHz 20 Hz 10 AV = 10 mW 0.1 50 AV = 5 mW 0.01 AV = 1 mW 0.001 20 0.01 1 1 VDD = 5 V PO = 60 mW RL = 32 Ω CB = 1 µF 100 PO − Output Power − mW Figure 4. TOTAL HARMONIC DISTORTION + NOISE vs FREQUENCY TOTAL HARMONIC DISTORTION + NOISE vs OUTPUT POWER THD+N −Total Harmonic Distortion + Noise − % THD+N −Total Harmonic Distortion + Noise − % 10 VDD = 5 V RL = 32 Ω AV = 1 V/V CB = 1 µF 1 PO = 30 mW PO = 15 mW 0.01 PO = 60 mW 0.001 20 100 1k f − Frequency − Hz Figure 5. 6 10k 20k Figure 3. 10 0.1 1k f − Frequency − Hz 10k 20k VDD = 5 V AV = 1 V/V RL = 32 Ω CB = 1 µF 20 kHz 1 10 kHz 0.1 1 kHz 20 Hz 0.01 0.002 0.01 0.1 PO − Output Power − W Figure 6. 0.2 TPA112 www.ti.com SLOS212E – AUGUST 1998 – REVISED JUNE 2004 TOTAL HARMONIC DISTORTION + NOISE vs FREQUENCY TOTAL HARMONIC DISTORTION + NOISE vs FREQUENCY 10 VDD = 3.3 V RL = 10 kΩ PO = 100 µF CB = 1 µF THD+N −Total Harmonic Distortion + Noise − % THD+N −Total Harmonic Distortion + Noise − % 10 1 AV = 5 mW 0.1 0.01 AV = 2 mW 0.001 20 100 1k VDD = 3.3 V RL = 10 kΩ AV = 1 V/V CB = 1 µF 1 0.1 PO = 45 µW 0.01 0.001 20 10k 20k 100 f − Frequency − Hz 10k 20k Figure 7. Figure 8. TOTAL HARMONIC DISTORTION + NOISE vs OUTPUT POWER TOTAL HARMONIC DISTORTION + NOISE vs FREQUENCY 10 VDD = 3.3 V RL = 10 k Ω AV = 1 V/V CB = 1 µF 1 0.1 20 Hz 10 kHz 0.01 20 Hz 1 kHz 0.001 5 1k f − Frequency − Hz THD+N −Total Harmonic Distortion + Noise − % THD+N −Total Harmonic Distortion + Noise − % 10 PO = 90 µW PO = 130 µW 10 100 PO − Output Power − µW Figure 9. 200 1 VDD = 5 V RL = 10 kΩ PO = 300 µW CB = 1 µF 0.1 AV = 5 AV = 1 0.01 AV = 2 0.001 20 100 1k 10k 20k f − Frequency − Hz Figure 10. 7 TPA112 www.ti.com SLOS212E – AUGUST 1998 – REVISED JUNE 2004 TOTAL HARMONIC DISTORTION + NOISE vs FREQUENCY TOTAL HARMONIC DISTORTION + NOISE vs OUTPUT POWER 1 10 VDD = 5 V RL = 10 kΩ AV = 1 V/V CB = 1 µF PO = 300 µW 0.1 PO = 200 µW 0.01 PO = 100 µW 0.001 20 THD+N −Total Harmonic Distortion + Noise − % THD+N −Total Harmonic Distortion + Noise − % 10 VDD = 5 V RL = 10 kΩ AV = 1 V/V CB = 1 µ F 1 0.1 20 Hz 20 kHz 0.01 10 kHz 1 kHz 0.001 100 1k 10k 20k 5 500 Figure 12. TOTAL HARMONIC DISTORTION + NOISE vs FREQUENCY TOTAL HARMONIC DISTORTION + NOISE vs FREQUENCY 10 VDD = 3.3 V PO = 75 mW RL = 8 Ω CB = 1 µF 1 THD+N −Total Harmonic Distortion + Noise − % THD+N − Total Harmonic Distortion Plus Noise − % 100 Figure 11. 2 AV = 5 AV = 2 0.1 AV = 1 0.01 0.001 20 100 1k f − Frequency − Hz Figure 13. 8 10 PO − Output Power − µW f − Frequency − Hz 10k 20k VDD = 3.3 V RL = 8 Ω AV = 1 V/V PO = 30 mW 1 PO = 15 mW 0.1 0.01 PO = 75 mW 0.001 20 100 1k f − Frequency − Hz Figure 14. 10k 20k TPA112 www.ti.com SLOS212E – AUGUST 1998 – REVISED JUNE 2004 TOTAL HARMONIC DISTORTION + NOISE vs OUTPUT POWER TOTAL HARMONIC DISTORTION + NOISE vs FREQUENCY VDD = 3.3 V RL = 8 Ω AV = 1 V/V THD+N − Total Harmonic Distortion Plus Noise − % THD+N −Total Harmonic Distortion + Noise − % 10 20 kHz 10 kHz 1 1 kHz 0.1 20 Hz 0.01 10m 0.1 0.3 2 VDD = 5 V PO = 100 mW RL = 8 Ω CB = 1 µF 1 AV = 1 0.01 0.001 20 100 10k 20k Figure 15. Figure 16. TOTAL HARMONIC DISTORTION + NOISE vs FREQUENCY TOTAL HARMONIC DISTORTION + NOISE vs OUTPUT POWER 10 THD+N −Total Harmonic Distortion + Noise − % THD+N −Total Harmonic Distortion + Noise − % 1k f − Frequency − Hz 10 VDD = 5 V RL = 8 kΩ AV = 1 V/V PO = 30 mW 1 PO = 60 mW 0.01 PO = 10 mW 0.001 20 AV = 5 0.1 PO − Output Power − W 0.1 AV = 2 100 1k f − Frequency − Hz Figure 17. 10k 20k VDD = 5 V RL = 8 Ω AV = 1 V/V 20 kHz 1 10 kHz 1 kHz 0.1 20 Hz 0.01 10m 0.1 1 PO − Output Power − W Figure 18. 9 TPA112 www.ti.com SLOS212E – AUGUST 1998 – REVISED JUNE 2004 POWER SUPPLY REJECTION RATIO vs FREQUENCY POWER SUPPLY REJECTION RATIO vs FREQUENCY 0 VDD = 3.3 V RL = 8 Ω to 10 kΩ −10 −20 CB = 0.1 µF −30 CB = 1 µF −40 −50 −60 CB = 2 µF −70 Bypass = 1.65 V −80 −90 −100 20 100 1k PSRR − Power Supply Rejection Ratio − dB PSRR − Power Supply Rejection Ratio − dB 0 VDD = 5 V RL = 8 Ω to 10 kΩ −10 −20 CB = 0.1 µF −30 CB = 1 µF −40 −50 CB = 2 µF −60 −70 −80 −90 Bypass = 2.5 V −100 20 10k 20k 100 f − Frequency − Hz Figure 19. Figure 20. OUTPUT NOISE VOLTAGE vs FREQUENCY OUTPUT NOISE VOLTAGE vs FREQUENCY Vn − Output Noise Voltage − µV Vn − Output Noise Voltage − µV 10 VDD = 3.3 V BW = 10 Hz to 22 kHz AV = 1 V/V RL = 8 Ω to 10 kΩ 100 1k f − Frequency − Hz Figure 21. 10 10k 20k 20 20 1 20 1k f − Frequency − Hz 10k 20k 10 VDD = 5 V BW = 10 Hz to 22 kHz RL = 8 Ω to 10 kΩ AV = 1 V/V 1 20 100 1k f − Frequency − Hz Figure 22. 10k 20k TPA112 www.ti.com SLOS212E – AUGUST 1998 – REVISED JUNE 2004 CROSSTALK vs FREQUENCY CROSSTALK vs FREQUENCY −50 −60 −70 Crosstalk − dB −75 −60 −65 −80 IN2 TO OUT1 −85 −90 −95 −70 −75 IN2 TO OUT1 −80 −85 −100 IN1 TO OUT2 −90 IN1 TO OUT2 −105 −110 20 PO = 100 mW VDD = 3.3 V RL = 8 Ω CB = 1 µF AV = 1 V/V −55 Crosstalk − dB −65 PO = 25 mW VDD = 3.3 V RL = 32 Ω CB = 1 µF AV = 1 V/V −95 −100 100 1k 10k 20k 20 100 f − Frequency − Hz Figure 23. Figure 24. CROSSTALK vs FREQUENCY CROSSTALK vs FREQUENCY −60 10k 20k −50 VDD = 5 V PO = 25 mW CB = 1 µF RL = 32 Ω AV = 1 V/V −65 −75 −80 −85 −55 −60 −65 Crosstalk − dB −65 Crosstalk − dB 1k f − Frequency − Hz IN2 TO OUT1 −90 −95 VDD = 5 V PO = 100 mW CB = 1 µF RL = 8 Ω AV = 1 V/V −70 IN2 TO OUT1 −75 −80 −85 −100 −90 IN1 TO OUT2 IN1 TO OUT2 −105 −110 20 −95 100 1k 10k 20k −100 20 100 1k f − Frequency − Hz f − Frequency − Hz Figure 25. Figure 26. 10k 20k 11 TPA112 www.ti.com SLOS212E – AUGUST 1998 – REVISED JUNE 2004 MUTE ATTENUATION vs FREQUENCY MUTE ATTENUATION vs FREQUENCY 0 Mute Attenuation − dB −20 −20 −30 −40 −50 −60 −70 −30 −40 −50 −60 −70 −80 −80 −90 −90 −100 20 VDD = 5 V CB = 1 µF RL = 32 Ω −10 Mute Attenuation − dB −10 0 VDD = 3.3 V RL = 32 Ω CB = 1 µF 100 1k −100 20 10k 20k 100 f − Frequency − Hz 1k f − Frequency − Hz Figure 27. Figure 28. OPEN-LOOP GAIN AND PHASE MARGIN vs FREQUENCY 150° VDD = 3.3 V TA = 25°C No Load Open-Loop Gain − dB 80 Phase 60 40 90° 60° Gain 20 30° 0 0° −20 100 1k 10k 100k f − Frequency − Hz Figure 29. 12 120° 1M −30° 10M φ m − Phase Margin 100 10k 20k TPA112 www.ti.com SLOS212E – AUGUST 1998 – REVISED JUNE 2004 OPEN-LOOP GAIN AND PHASE MARGIN vs FREQUENCY 100 150° VDD = 5 V TA = 25°C No Load 120° Phase 60 40 90° 60° Gain 20 30° 0 0° −20 100 1k 10k 100k φ m − Phase Margin Open-Loop Gain − dB 80 −30° 10M 1M f − Frequency − Hz . . Figure 30. OUTPUT POWER vs LOAD RESISTANCE OUTPUT POWER vs LOAD RESISTANCE 300 120 THD+N = 1 % VDD = 3.3 V AV = 1 V/V 250 PO − Output Power − mW PO − Output Power − mW 100 THD+N = 1 % VDD = 5 V AV = 1 V/V 80 60 40 200 150 100 50 20 0 0 8 16 24 32 40 48 RL − Load Resistance − Ω Figure 31. 56 64 8 16 24 32 40 48 56 64 RL − Load Resistance − Ω Figure 32. 13 TPA112 www.ti.com SLOS212E – AUGUST 1998 – REVISED JUNE 2004 SUPPLY CURRENT vs SUPPLY VOLTAGE TOTAL HARMONIC DISTORTION + NOISE vs FREQUENCY THD+N − Total Harmonic Distortion Plus Noise − % 1.4 I DD − Supply Current − mA 1.2 1 0.8 0.6 0.4 0.2 0 2.5 3 3.5 4 4.5 5 1 VI = 1 V AV = 1 V/V RL = 10 kΩ CB = 1 µF 0.1 0.01 0.001 5.5 20 100 Figure 34. SIGNAL-TO-NOISE RATIO vs VOLTAGE GAIN TOTAL HARMONIC DISTORTION + NOISE vs FREQUENCY THD+N − Total Harmonic Distortion Plus Noise − % VI = 1 V SNR − Signal−to−Ratio − dB 102 100 98 96 94 92 2 3 4 5 6 7 AV − Voltage Gain − V/V Figure 35. 14 10k 20k Figure 33. 104 1 1k f − Frequency − Hz VDD − Supply Voltage − V 8 9 10 1 VDD = 5 V AV = 1 V/V RL = 10 kΩ CB = 1 µF 0.1 0.01 0.001 20 100 1k f − Frequency − Hz Figure 36. 10k 20k TPA112 www.ti.com SLOS212E – AUGUST 1998 – REVISED JUNE 2004 CROSSTALK vs FREQUENCY CROSSTALK vs FREQUENCY −60 −60 −70 −80 Crosstalk − dB −90 −100 IN2 to OUT1 −110 −90 −100 −120 IN2 to OUT1 −110 −120 −130 −130 IN1 to OUT2 −140 IN1 to OUT2 −140 −150 −150 100 1k 20 10k 20k 100 f − Frequency − Hz 1k 10k 20k f − Frequency − Hz Figure 37. Figure 38. CLOSED-LOOP GAIN AND PHASE vs FREQUENCY 200° 180° Phase 160° 140° Phase 20 120° Closed−Loop Gain − dB Crosstalk − dB −80 VDD = 5 V VO = 1 V RL = 10 kΩ CB = 1 µF −70 VDD = 3.3 V VO = 1 V RL = 10 kΩ CB = 1 µF VDD = 3.3 V RI = 20 kΩ RF = 20 kΩ RL = 32 Ω CI = 1 µF AV = −1 V/V 30 20 10 100° 80° Gain 0 −10 10 100 1k 10k 100k 1M f − Frequency − Hz Figure 39. 15 TPA112 www.ti.com SLOS212E – AUGUST 1998 – REVISED JUNE 2004 CLOSED-LOOP GAIN AND PHASE vs FREQUENCY 200° 180° 160° 140° Phase Phase Closed−Loop Gain − dB 120° VDD = 5 V RI = 20 kΩ RF = 20 kΩ RL = 32 Ω CI = 1 µF AV = −1 V/V 30 100° 80° 20 10 Gain 0 −10 10 100 1k 10k 100k 1M f − Frequency − Hz Figure 40. CLOSED-LOOP GAIN AND PHASE vs FREQUENCY 200° 180° Phase 140° Closed−Loop Gain − dB 120° VDD = 3.3 V RI = 20 kΩ RF = 20 kΩ RL = 8 Ω CI = 1 µF AV = −1 V/V 40 80° 60° Gain 20 0 −20 10 100 1k 10k f − Frequency − Hz Figure 41. 16 100° 100k 1M Phase 160° TPA112 www.ti.com SLOS212E – AUGUST 1998 – REVISED JUNE 2004 CLOSED-LOOP GAIN AND PHASE vs FREQUENCY 200° 160° 140° Phase 180° Phase Closed−Loop Gain − dB 120° VDD = 3.3 V RI = 20 kΩ RF = 20 kΩ RL = 10 kΩ CI = 1 µF AV = −1 V/V 30 20 10 100° 80° Gain 0 −10 10 100 1k 10k 100k 1M f − Frequency − Hz Figure 42. CLOSED-LOOP GAIN AND PHASE vs FREQUENCY 200° 180° Phase Closed−Loop Gain − dB 140° VDD = 5 V RI = 20 kΩ RF = 20 kΩ RL = 8 Ω CI = 1 µF AV = −1 V/V 120° Phase 160° 100° 80° 60° 40° Gain 20 0 −20 10 100 1k 10k 100k 1M f − Frequency − Hz Figure 43. 17 TPA112 www.ti.com SLOS212E – AUGUST 1998 – REVISED JUNE 2004 CLOSED-LOOP GAIN AND PHASE vs FREQUENCY 200° 180° 160° 140° 120° VDD = 5 V RI = 20 kΩ RF = 20 kΩ RL = 10 kΩ CI = 1 µF AV = −1 V/V 30 100° 80° Phase Closed−Loop Gain − dB Phase 20 10 Gain 0 −10 10 100 1k 100k 10k 1M f − Frequency − Hz Figure 44. POWER DISSIPATION/AMPLIFIER vs OUTPUT POWER POWER DISSIPATION/AMPLIFIER vs OUTPUT POWER 80 180 VDD = 3.3 V VDD = 5 V 8Ω 70 140 Amplifier Power − mW Amplifier Power − mW 60 50 40 16 Ω 30 32 Ω 20 120 100 16 Ω 80 60 32 Ω 40 64 Ω 10 64 Ω 20 0 0 0 20 40 60 80 100 120 140 160 180 Load Power − mW Figure 45. 18 8Ω 160 200 0 20 40 60 80 100 120 140 160 180 Load Power − mW Figure 46. 200 TPA112 www.ti.com SLOS212E – AUGUST 1998 – REVISED JUNE 2004 APPLICATION INFORMATION GAIN SETTING RESISTORS, RF and RI The gain for the TPA112 is set by resistors RF and RI according to Equation 1. Gain 
  RF RI (1) Given that the TPA112 is an MOS amplifier, the input impedance is high. Consequently, input leakage currents are not generally a concern, although noise in the circuit increases as the value of RF increases. In addition, a certain range of RF values is required for proper start-up operation of the amplifier. Taken together, it is recommended that the effective impedance seen by the inverting node of the amplifier be set between 5 kΩ and 20 kΩ. The effective impedance is calculated in Equation 2. R FR I Effective Impedance  RF
RI (2) As an example, consider an input resistance of 20 kΩ and a feedback resistor of 20 kΩ. The gain of the amplifier would be -1 and the effective impedance at the inverting terminal would be 10 kΩ, which is within the recommended range. For high-performance applications, metal film resistors are recommended because they tend to have lower noise levels than carbon resistors. For values of RF above 50 kΩ, the amplifier tends to become unstable due to a pole formed from RF and the inherent input capacitance of the MOS input structure. For this reason, a small compensation capacitor of approximately 5 pF should be placed in parallel with RF. In effect, this creates a low-pass filter network with the cutoff frequency defined in Equation 3. 1 f co(lowpass)
2
R F CF (3) For example, if RF is 100 kΩ and CF is 5 pF then fco(lowpass) is 318 kHz, which is well outside the audio range. INPUT CAPACITOR, CI In the typical application, input capacitor CI is required to allow the amplifier to bias the input signal to the proper dc level for optimum operation. In this case, CI and RI form a high-pass filter with the corner frequency determined in Equation 4. 1 f co(highpass)
2
R I CI (4) The value of CI is important to consider, as it directly affects the bass (low-frequency) performance of the circuit. Consider the example where RI is 20 kΩ and the specification calls for a flat bass response down to 20 Hz. Equation 4 is reconfigured as Equation 5. 1 CI
2
R I f co(highpass) (5) In this example, CI is 0.4 µF, so one would likely choose a value in the range of 0.47 µF to 1 µF. A further consideration for this capacitor is the leakage path from the input source through the input network (RI, CI) and the feedback resistor (RF) to the load. This leakage current creates a dc offset voltage at the input to the amplifier that reduces useful headroom, especially in high-gain applications (> 10). For this reason a low-leakage tantalum or ceramic capacitor is the best choice. When polarized capacitors are used, the positive side of the capacitor should face the amplifier input in most applications, as the dc level there is held at VDD/2, which is likely higher that the source dc level. It is important to confirm the capacitor polarity in the application. 19 TPA112 www.ti.com SLOS212E – AUGUST 1998 – REVISED JUNE 2004 APPLICATION INFORMATION (continued) POWER SUPPLY DECOUPLING, CS The TPA112 is a high-performance CMOS audio amplifier that requires adequate power supply decoupling to ensure that the output total harmonic distortion (THD) is as low as possible. Power supply decoupling also prevents oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling is achieved by using two capacitors of different types that target different types of noise on the power supply leads. For higher frequency transients, spikes, or digital hash on the line, a good low equivalent-series-resistance (ESR) ceramic capacitor; typically, 0.1 µF, placed as close as possible to the device VDD lead, works best. For filtering lower frequency noise signals, a larger aluminum electrolytic capacitor of 10 µF or greater placed near the power amplifier is recommended. MIDRAIL VOLTAGE The TPA112 is a single-supply amplifier; so, it must be properly biased to accommodate audio signals. Normally, the amplifier is biased at VDD/2, but it can actually be biased at any voltage between VDD and ground. However, biasing the amplifier at a point other than VDD/2 reduces the amplifier's maximum output swing. In some applications where the circuitry driving the TPA112 has a different midrail voltage, it might make sense to use the same midrail voltage for the TPA112, and possibly eliminate the use of the dc-blocking capacitors. The two concerns with the midrail voltage source are the amount of noise present and its output impedance. Any noise present on the midrail voltage source that is not present on the audio input signal will be input to the amplifier, and passed to the output (and increased by the gain of the circuit). Common-mode noise is cancelled out by the differential configuration of the circuit. The output impedance of the circuit used to generate the midrail voltage needs to be low enough so as not to be influenced by the audio signal path. A common method of generating the midrail voltage is to form a voltage divider from the supply to ground, with a bypass capacitor from the common node to ground. This capacitor improves the PSRR of the circuit. However, this circuit has a limited range of output impedances; so, to achieve low output impedances, the voltage generated by the voltage divider is fed into a unity-gain amplifier to lower the output impedance of the circuit. VDD VDD R R + _ Midrail CBYPASS R CBYPASS a) Midrail Voltage Generator Using a Simple Resistor-Divider TLV2460 Midrail R b) Buffered Midrail Voltage Generator to Provide Low Output Impedance Figure 47. Midrail Voltage Generator If a voltage step is applied to a speaker, it causes a noise pop. To reduce popping, the midrail voltage should rise at a subsonic rate. That is, a rate less than the rise time of a 20-Hz waveform. If the voltage rises faster than that, there is the possibility of a pop from the speaker. Pop can also be heard in the speaker if the midrail voltage rises faster than the charge of either the input coupling capacitor or the output coupling capacitor. If midrail rises first, the charging of the input and output capacitors is heard in the speaker. To keep this noise as low as possible, the relationship shown in Equation 6 should be maintained. 20 TPA112 www.ti.com SLOS212E – AUGUST 1998 – REVISED JUNE 2004 APPLICATION INFORMATION (continued) 1  1 CB
RSOURCE CI RI  1 R LC C (6) Where CBYPASS is the value of the bypass capacitor, and RSOURCE is the equivalent source impedance of the voltage divider (the parallel combination of the two resistors). For example, if the voltage divider is constructed using two 20-kΩ resistors, then RSOURCE is 10 kΩ. MIDRAIL BYPASS CAPACITOR, CB The midrail bypass capacitor CB serves several important functions. During start-up, CB determines the rate at which the amplifier starts up. This helps to push the start-up pop noise into the subaudible range (so slow it can not be heard). The second function is to reduce noise produced by the power supply caused by coupling into the output drive signal. This noise is from the midrail generation circuit internal to the amplifier. The capacitor is fed from the resistor divider with equivalent resistance of RSOURCE. To keep the start-up pop as low as possible, the relationship shown in Equation 7 should be maintained. 1  1 C
R  CI RI B SOURCE (7) As an example, consider a circuit where CB is 1 µF, RSOURCE = 160 kΩ, CI is 1 µF, and RI is 20 kΩ. Inserting these values into the Equation 8 results in: 6.25
50 (8) which satisfies the rule. Recommended values for bypass capacitor CB are 0.1 µF to 1 µF, ceramic or tantalum low-ESR, for the best THD and noise performance. OUTPUT COUPLING CAPACITOR, CC In the typical single-supply, single-ended (SE) configuration, an output coupling capacitor (CC) is required to block the dc bias at the output of the amplifier, thus preventing dc currents in the load. As with the input coupling capacitor, the output coupling capacitor and impedance of the load form a high-pass filter governed by Equation 9. 1 f (out high)
2
R L CC (9) The main disadvantage, from a performance standpoint, is that the typically small load impedances drive the low-frequency corner higher. Large values of CC are required to pass low frequencies into the load. Consider the example where a CC of 68 µF is chosen and loads vary from 32 Ω to 47 kΩ. Table 1 summarizes the frequency response characteristics of each configuration. Table 1. Common Load Impedances vs Low Frequency Output Characteristics in SE Mode RL CC LOWEST FREQUENCY 32 Ω 68 µF 73 Hz 10,000 Ω 68 µF 0.23 Hz 47,000 Ω 68 µF 0.05 Hz As Table 1 indicates, headphone response is adequate and drive into line level inputs (a home stereo for example) is good. The output coupling capacitor required in single-supply, SE mode also places additional constraints on the selection of other components in the amplifier circuit. With the rules described earlier still valid, add the following relationship: • Output Pulldown Resistor, RC + RO – Placing a 100-Ω resistor, RC, from the output side of the coupling capacitor to ground ensures the coupling capacitor, CC, is charged before a plug is inserted into the jack. Without this resistor, the coupling capacitor would charge rapidly upon insertion of a plug, leading to an audible pop in the headphones. 21 TPA112 SLOS212E – AUGUST 1998 – REVISED JUNE 2004 • www.ti.com – Placing a 20-kΩ resistor, RO, from the output of the IC to ground ensures that the coupling capacitor fully discharges at power down. If the supply is rapidly cycled without this capacitor, a small pop may be audible in 10-kΩ loads. Using Low-ESR Capacitors – Low-ESR capacitors are recommended throughout this application. A real capacitor can be modeled simply as a resistor in series with an ideal capacitor. The voltage drop across this resistor minimizes the beneficial effects of the capacitor in the circuit. The lower the equivalent value of this resistance, the more the real capacitor behaves like an ideal capacitor. 5-V VERSUS 3.3-V OPERATION The TPA112 is designed for operation over a supply range of 2.5 V to 5.5 V. This data sheet provides full specifications for 5-V and 3.3-V operation because these are considered to be the two most common standard voltages. There are no special considerations for 3.3-V versus 5-V operation as far as supply bypassing, gain setting, or stability. The most important consideration is that of output power. Each amplifier in the TPA112 can produce a maximum voltage swing of VDD – 1 V. This means, for 3.3-V operation, clipping starts to occur when VO(PP) = 2.3 V, as opposed to VO(PP) = 4 V for 5-V operation. The reduced voltage swing subsequently reduces maximum output power into the load before distortion begins to become significant. 22 PACKAGE OPTION ADDENDUM www.ti.com 10-Dec-2020 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Pins Package Drawing Qty Eco Plan (2) Lead finish/ Ball material MSL Peak Temp Op Temp (°C) Device Marking (3) (4/5) (6) TPA112D ACTIVE SOIC D 8 75 RoHS & Green NIPDAU Level-1-260C-UNLIM TPA112 TPA112DR ACTIVE SOIC D 8 2500 RoHS & Green NIPDAU Level-1-260C-UNLIM TPA112 (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may reference these types of products as "Pb-Free". RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption. Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of