TPA2000D4DAP

 
      
 
   SLOS337B − DECEMBER 2000 − REVISED NOVEMBER 2005 D Ideal for Notebook PCs and USB-Powered D D D D D D D D DAP PACKAGE (TOP VIEW) Speakers 2 W Into 4 Ω From 5-V Supply Integrated Class-AB Headphone Amplifier Second-Generation Modulation Technique − Filterless Operation − Improved Efficiency Low Supply Current . . . 9 mA typ at 5 V Shutdown Control . . . < 0.05 µA Typ Shutdown Pin Is TTL Compatible −40°C to 85°C Operating Temperature Range Space-Saving, Thermally-Enhanced PowerPAD Packaging LINN LINP HPLIN GAIN0 GAIN1 PVDDL LOUTP PGNDL PGNDL LOUTN PVDDL HPLGAIN HPLOUT MODE HPRGAIN HPROUT description 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15 16 32 31 30 29 28 27 26 25 24 23 22 21 20 19 18 17 RINN RINP HPRIN BYPASS SHUTDOWN PVDDR ROUTP PGNDR PGNDR ROUTN PVDDR NC VDD COSC ROSC AGND NC − No internal connection The TPA2000D4 is a 2-W stereo bridge-tied-load (BTL) class-D amplifier designed to drive speakers with as low as 4-Ω impedance. The amplifier uses TI’s second-generation modulation technique, which results in improved efficiency and SNR, and also allows the device to be connected directly to the speaker without the use of the LC output filter commonly associated with class-D amplifiers (this will result in an EMI which must be shielded at the system level). These features make the device ideal for use in notebook PCs where high-efficiency is needed to extend battery run-time. For speakers powered off the USB bus, the high-efficiency allows for higher output power levels without tripping the USB’s overcurrent circuitry. The gain of the amplifier is controlled by two input terminals, GAIN1, and GAIN0. This allows the amplifier to be configured for a gain of 6, 12, 18, and 23.5 dB. The differential input terminals are high-impedance CMOS inputs, and can be used as summing nodes. The headphone amplifier is a stereo single-ended (SE) class-AB amplifier which requires two external resistors per channel to set the gain. The MODE pin selects which amplifier is active; the unused amplifier is placed in shutdown to reduce supply current. Both the class-D BTL amplifier, and the class-AB SE amplifier include depop circuitry to reduce the amount of turnon pop at power up, when cycling SHUTDOWN, and when switching modes of operation. The TPA2000D4 is available in the 32-pin thermally-enhanced TSSOP package (DAP) which allows stereo 2-W continuous output power levels in 4-Ω loads when placed on a board with proper thermal board design. The TPA2000D4 operates over an ambient temperature range of −40°C to 85°C. These packages deliver levels of thermal performance that were previously only achievable in TO-220-type packages. Thermal impedances of less than 35°C/W are readily realized in multilayer PCB applications when using the DAP package. Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. PowerPAD is a trademark of Texas Instruments. Copyright  2000−2005, Texas Instruments Incorporated  
 
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   SLOS337B − DECEMBER 2000 − REVISED NOVEMBER 2005 AVAILABLE OPTIONS PACKAGED DEVICE TSSOP (DAP)† TA −40°C to 85°C TPA2000D4DAP † The DAP package is available taped and reeled. To order a taped and reeled part, add the suffix R to the part number (e.g., TPA2000D4DAPR). functional schematic VDD AGND PVDDR VDD Gain Adjust RINN Rs2 100 Ω + _ _ Deglitch Logic Gate Drive ROUTN + _ PGNDR + _ cmv + Gain Adjust RINP PVDDR _ Rs1 + _ 100 Ω + Deglitch Logic Gate Drive ROUTP SD−z MODE GAIN1 GAIN0 PGNDR Input Buffers Gain 2 Biases and References Startup Protection Logic Ramp Generator Thermal COSC ROSC Gain Adjust Rs1 100 Ω + _ _ Deglitch Logic Gate Drive Gain Adjust LOUTP + + _ cmv _ + LINN VDD ok PVDDL BYPASS LINP OC Detect OC Detect PGNDL PVDDL Rs2 + _ _ 100 Ω + _ HPLIN BYPASS + Deglitch Logic Gate Drive LOUTN PGNDR HPROUT HPLGAIN _ HPRIN BYPASS + HPRGAIN 2 WWW.TI.COM HPLOUT 
 
      
 
   SLOS337B − DECEMBER 2000 − REVISED NOVEMBER 2005 Terminal Functions TERMINAL NAME NO. I/O DESCRIPTION AGND 17 I Analog ground BYPASS 29 I Connect capacitor to ground for BYPASS voltage filtering COSC 19 I Connect capacitor to ground to set oscillation frequency GAIN0 4 I Bit 0 of gain control GAIN1 5 I Bit 1 of gain control HPLGAIN 12 I Place RF between pins 12 and 13 HPRGAIN 15 I Place RF between pins 14 and 15 HPLIN 3 I Left HP single-ended (SE) input HPLOUT 13 O Left headphone output HPRIN 30 I Right HP SE input HPROUT 16 O Right headphone output LINN 1 I Left class-D negative differential input LINP 2 I Left class-D positive differential input LOUTP 7 O Left positive bridge-tied load (BTL) output LOUTN 10 O Left negative BTL output MODE 14 I Mode = 1, then HP, Mode = 0, then BTL NC 21 — PGNDL 8, 9 I Left class-D high-current ground PGNDR 24, 25 I Right class-D high-current ground PVDDL 6, 11 I Left class-D high-current power supply PVDDR 22, 27 I Right class-D high power supply ROSC 18 I Connect resistor to ground to set oscillation frequency RINP 31 I Right class-D positive differential signal RINN 32 I Right class-D negative differential signal ROUTN 23 O Right negative BTL output ROUTP 26 O Right positive BTL output SHUTDOWN 28 I Shutdown terminal (negative logic) VDD 20 I Power supply No connection absolute maximum ratings over operating free-air temperature (unless otherwise noted)† Supply voltage, VDD, PVDDL,R . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −0.3 V to 5.5 V Input voltage, VI . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −0.3 V to VDD+0.3 V Continuous total power dissipation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . See Dissipation Rating Table Operating free-air temperature range, TA . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −40°C to 85°C Operating junction temperature range, TJ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −40°C to 150°C Storage temperature range, Tstg . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −65°C to 150°C Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 260°C † Stresses beyond those listed under “absolute maximum ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated under “recommended operating conditions” is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability. WWW.TI.COM 3 
 
      
 
   SLOS337B − DECEMBER 2000 − REVISED NOVEMBER 2005 DISSIPATION RATING TABLE PACKAGE TA ≤ 25°C POWER RATING DERATING FACTOR ABOVE TA = 25°C TA = 70°C POWER RATING TA = 85°C POWER RATING DAP 5.3 W 42.5 mW/°C 3.4 W 2.8 W recommended operating conditions MIN Supply voltage, VDD, PVDD, VCC GAIN0, GAIN1, SHUTDOWN High-level input voltage, VIH NOM MAX 3.7 5.5 2 MODE V V 0.8 VDD V GAIN0, GAIN1, SHUTDOWN Low-level input voltage, VIL UNIT 0.8 MODE 0.4 VDD V V Oscillator resistance, ROSC 120 kΩ Oscillator capacitance, COSC 220 pF PWM Frequency 200 300 kHz Operating free-air temperature, TA −40 85 °C electrical characteristics over recommended operating free-air temperature range, TA = 25°C, VDD = PVDD = 5 V (unless otherwise noted) PARAMETER |VOS| Output offset voltage (measured differentially) PSRR Power supply rejection ratio | IIH | High-level input current | IIL | Low-level input current IDD Supply current IDD(SD) TEST CONDITIONS VI = 0 V, AV = −2 V/V PVDD = 4.5 V to 5.5 V MIN Class D Headphone MODE = 5 V MAX 15 Class-D −70 PVDD = 4.5 V to 5.5 V Headphone PVDD = 5.5 V, VI = PVDD PVDD = 5.5 V, MODE = 0 V TYP Shutdown mode mV dB −75 VI = 0 V UNIT 1 µA 1 µA 9 12 7 11 0.05 1 mA µA A operating characteristics, class-D amplifier, TA = 25°C, VDD = PVDD = 5 V, RL = 4 Ω, Gain = all gains (unless otherwise noted) PARAMETER TEST CONDITIONS PO THD+N Output power THD = 0.1%, Total harmonic distortion plus noise BOM kSVR Maximum output power bandwidth PO = 1 W,f = 20 Hz to 20 kHz THD = 1% Supply ripple rejection ratio f = 1 kHz, SNR Signal-to-noise ratio Vn ZI Noise output voltage 4 f = 1 kHz CBYPASS = 1 µF CBYPASS = 1 µF, Input impedance f = 20 Hz to 20 kHz MIN TYP 2 UNIT W < 0.4% 20 kHz −71 dB 85 dB 20 µVRMS kΩ >15 WWW.TI.COM MAX 
 
      
 
   SLOS337B − DECEMBER 2000 − REVISED NOVEMBER 2005 operating characteristics, class-D amplifier, TA = 25°C, VDD = PVDD = 5 V, RL = 8 Ω, Gain = all gains (unless otherwise noted) PARAMETER TEST CONDITIONS MIN PO THD+N Output power THD = 0.1%, f = 1 kHz Total harmonic distortion plus noise Maximum output power bandwidth PO = 0.5 W, THD = 1% f = 20 Hz to 20 kHz BOM kSVR Supply ripple rejection ratio f = 1 kHz, CBYPASS = 1 µF SNR Signal-to-noise ratio Vn ZI Noise output voltage CBYPASS = 1 µF, TYP MAX 1.5 UNIT W 15 operating characteristics, headphone amplifier, TA = 25°C, VDD = PVDD = 5 V, RL = 32 Ω, Gain = 1 V/V (unless otherwise noted) PARAMETER TEST CONDITIONS MIN PO THD+N Output power THD = 0.1%, f = 1 kHz Total harmonic distortion plus noise f = 20 Hz to 20 kHz BOM kSVR Maximum output power bandwidth PO = 75 mW, THD = 1% Supply ripple rejection ratio f = 1 kHz, CBYPASS = 1 µF SNR Signal-to-noise ratio Vn Noise output voltage CBYPASS = 1 µF, TYP 90 f = 20 Hz to 20 kHz MAX UNIT mW 0 V 5V 0V Across Load −5 V Current Figure 28. The TPA2000D4 Output Voltage and Current Waveforms Into an Inductive Load efficiency: why you must use a filter with the traditional class-D modulation scheme The main reason that the traditional class-D amplifier needs an output filter is that the switching waveform results in maximum current flow. This causes more loss in the load, which causes lower efficiency. The ripple current is large for the traditional modulation scheme because the ripple current is proportional to voltage multiplied by the time at that voltage. The differential voltage swing is 2 × VDD and the time at each voltage is half the period for the traditional modulation scheme. An ideal LC filter is needed to store the ripple current from each half cycle for the next half cycle, while any resistance causes power dissipation. The speaker is both resistive and reactive, whereas an LC filter is almost purely reactive. The TPA2000D4 modulation scheme has very little loss in the load without a filter because the pulses are very short and the change in voltage is VDD instead of 2 × VDD. As the output power increases, the pulses widen, making the ripple current larger. Ripple current could be filtered with an LC filter for increased efficiency, but for most applications the filter is not needed. An LC filter with a cutoff frequency less than the class-D switching frequency allows the switching current to flow through the filter instead of the load. The filter has less resistance than the speaker that results in less power dissipated, which increases efficiency. 16 WWW.TI.COM 
 
      
 
   SLOS337B − DECEMBER 2000 − REVISED NOVEMBER 2005 APPLICATION INFORMATION effects of applying a square wave into a speaker Audio specialists have said for years not to apply a square wave to speakers. If the amplitude of the waveform is high enough and the frequency of the square wave is within the bandwidth of the speaker, the square wave could cause the voice coil to jump out of the air gap and/or scar the voice coil. A 250-kHz switching frequency, however, is not significant because the speaker cone movement is proportional to 1/f2 for frequencies beyond the audio band. Therefore, the amount of cone movement at the switching frequency is very small. However, damage could occur to the speaker if the voice coil is not designed to handle the additional power. To size the speaker for added power, the ripple current dissipated in the load needs to be calculated by subtracting the theoretical supplied power, PSUP THEORETICAL, from the actual supply power, PSUP, at maximum output power, POUT. The switching power dissipated in the speaker is the inverse of the measured efficiency, ηMEASURED, minus the theoretical efficiency, ηTHEORETICAL. PSPKR = PSUP – PSUP THEORETICAL (at max output power) (1) PSPKR = PSUP / POUT – PSUP THEORETICAL / POUT (at max output power) (2) PSPKR = 1/ηMEASURED – 1/ηTHEORETICAL (at max output power) (3) The maximum efficiency of the TPA2000D4 with an 8-Ω load is 85%. Using equation 3 with the efficiency at maximum power from Figure 2 (78%), we see that there is an additional 106 mW dissipated in the speaker. The added power dissipated in the speaker is not an issue as long as it is taken into account when choosing the speaker. when to use an output filter Design the TPA2000D4 without the filter if the traces from amplifier to speaker are short. The TPA2000D4 passed FCC and CE radiated emissions with no shielding with speaker wires 8 inches long or less. Notebook PCs and powered speakers where the speaker is in the same enclosure as the amplifier are good applications for class-D without a filter. A ferrite bead filter can often be used if the design is failing radiated emissions without a filter, and the frequency sensitive circuit is greater than 1 MHz. This is good for circuits that just have to pass FCC and CE because FCC and CE only test radiated emissions greater than 30 MHz. If choosing a ferrite bead, choose one with high impedance at high frequencies, but very low impedance at low frequencies. Use an output filter if there are low frequency (< 1 MHz) EMI sensitive circuits and/or there are long leads from amplifier to speaker. gain setting via GAIN0 and GAIN1 inputs The gain of the TPA2000D4 is set by two input terminals, GAIN0 and GAIN1. The gains listed in Table 1 are realized by changing the taps on the input resistors inside the amplifier. This causes the input impedance, ZI, to be dependent on the gain setting. The actual gain settings are controlled by ratios of resistors, so the actual gain distribution from part-to-part is quite good. However, the input impedance may shift by 30% due to shifts in the actual resistance of the input resistors. For design purposes, the input network (discussed in the next section) should be designed assuming an input impedance of 20 kΩ, which is the absolute minimum input impedance of the TPA2000D4. At the higher gain settings, the input impedance could increase as high as 115 kΩ. WWW.TI.COM 17 
 
      
 
   SLOS337B − DECEMBER 2000 − REVISED NOVEMBER 2005 APPLICATION INFORMATION Table 2. Gain Settings AMPLIFIER GAIN (dB) INPUT IMPEDANC (kΩ) TYP TYP 6 104 12 74 0 18 44 1 23.5 24 GAIN1 GAIN0 0 0 0 1 1 1 input resistance Each gain setting is achieved by varying the input resistance of the amplifier, which can range from its smallest value to over 6 times that value. ZF CI Input Signal IN ZI The −3 dB frequency can be calculated using equation 4: f *3 dB + 18 1 2p C I Z I (4) WWW.TI.COM 
 
      
 
   SLOS337B − DECEMBER 2000 − REVISED NOVEMBER 2005 APPLICATION INFORMATION input capacitor, CI In the typical application an input capacitor, CI, is required to allow the amplifier to bias the input signal to the proper dc level for optimum operation. In this case, CI and the input impedance of the amplifier, ZI, form a high-pass filter with the corner frequency determined in equation 5. −3 dB f c(highpass) + (5) 1 2 p ZI CI fc The value of CI is important as it directly affects the bass (low frequency) performance of the circuit. Consider the example where ZI is 20 kΩ and the specification calls for a flat bass response down to 80 Hz. Equation 5 is reconfigured as equation 6. CI + 1 2p Z I f c (6) In this example, CI is 0.1 µF, so one would likely choose a value in the range of 0.1 µF to 1 µF. If the gain is known and is constant, use ZI from Table 1 to calculate CI. A further consideration for this capacitor is the leakage path from the input source through the input network (CI) and the feedback network to the load. This leakage current creates a dc offset voltage at the input to the amplifier that reduces useful headroom, especially in high gain applications. For this reason a low-leakage tantalum or ceramic capacitor is the best choice. When polarized capacitors are used, the positive side of the capacitor should face the amplifier input in most applications as the dc level there is held at VDD/2, which is likely higher than the source dc level. Note that it is important to confirm the capacitor polarity in the application. CI must be 10 times smaller than the bypass capacitor to reduce clicking and popping noise from power on/off and entering and leaving shutdown. After sizing CI for a given cutoff frequency, size the bypass capacitor to 10 times that of the input capacitor. CI ≤ CBYP / 10 (7) switching frequency The switching frequency is determined using the values of the components connected to ROSC (pin 18) and COSC (pin 19) and is calculated with the following equation: fs + 6.6 R OSC C OSC (8) The switching frequency was chosen to be centered on 250 kHz. This frequency represents the optimization of audio fidelity due to oversampling and the maximization of efficiency by minimizing the switching losses of the amplifier. The recommended values are a resistance of 120 kΩ and a capacitance of 220 pF. Using these component values, the amplifier operates properly by using 5% tolerance resistors and 10% tolerance capacitors. The tolerance of the components can be changed, as long as the switching frequency remains between 200 kHz and 300 kHz. Within this range, the internal circuitry of the device provides stable operation. WWW.TI.COM 19 
 
      
 
   SLOS337B − DECEMBER 2000 − REVISED NOVEMBER 2005 APPLICATION INFORMATION gain setting resistors, RF and RI for HP amplifier The voltage gain for the TPA2000D4 headphone amplifier is set by resistors RF and RI according to equation 9. Gain + * ǒ Ǔ RF RI or Gain (dB) + 20 log ǒ Ǔ RF RI (9) Given that the TPA2000D4 is a MOS amplifier, the input impedance is very high. Consequently input leakage currents are not generally a concern, although noise in the circuit increases as the value of RF increases. In addition, a certain range of RF values is required for proper start-up operation of the amplifier. Taken together it is recommended that the effective impedance seen by the inverting node of the amplifier be set between 5 kΩ and 20 kΩ. The effective impedance is calculated in equation 10. Effective Impedance + R FR I (10) RF ) RI As an example, consider an input resistance of 20 kΩ and a feedback resistor of 20 kΩ. The gain of the amplifier would be − 1 and the effective impedance at the inverting terminal would be 10 kΩ, which is within the recommended range. For high performance applications, metal film resistors are recommended because they tend to have lower noise levels than carbon resistors. For values of RF above 50 kΩ, the amplifier tends to become unstable due to a pole formed from RF and the inherent input capacitance of the MOS input structure. For this reason, a small compensation capacitor of approximately 5 pF should be placed in parallel with RF. This, in effect, creates a low-pass filter network with the cutoff frequency defined in equation 11. fc + 1 2p R F C F (11) For example, if RF is 100 kΩ and CF is 5 pF then fc is 318 kHz, which is well outside the audio range. For maximum signal swing and output power at low supply voltages like 1.6 V to 3.3 V, BYPASS is biased to VDD/4. However, to allow the output to be biased at VDD/2, a resistor, R, equal to RF must be placed from the negative input to ground. input capacitor, CI for HP amplifier In the typical application, an input capacitor, CI, is required to allow the amplifier to bias the input signal to the proper dc level for optimum operation. In this case, CI and RI form a high-pass filter with the corner frequency determined in equation 12. fc + 1 2p R I C I (12) The value of CI is important to consider, as it directly affects the bass (low frequency) performance of the circuit. Consider the example where RI is 20 kΩ and the specification calls for a flat bass response down to 20 Hz. Equation 4 is reconfigured as equation 13. CI + 1 2p R I f c (13) In this example, CI is 0.40 µF, so one would likely choose a value in the range of 0.47 µF to 1 µF. A further consideration for this capacitor is the leakage path from the input source through the input network (RI, CI) and the feedback resistor (RF) to the load. This leakage current creates a dc offset voltage at the input to the amplifier that reduces useful headroom, especially in high-gain applications (>10). For this reason a low-leakage tantalum or ceramic capacitor is the best choice. When polarized capacitors are used, the positive side of the capacitor should face the amplifier input in most applications, as the dc level there is held at VDD/4, which is likely higher than the source dc level. It is important to confirm the capacitor polarity in the application. 20 WWW.TI.COM 
 
      
 
   SLOS337B − DECEMBER 2000 − REVISED NOVEMBER 2005 APPLICATION INFORMATION output coupling capacitor, CC for HP amplifier In the typical single-supply single-ended (SE) configuration, an output coupling capacitor (CC) is required to block the dc bias at the output of the amplifier, thus preventing dc currents in the load. As with the input coupling capacitor, the output coupling capacitor and impedance of the load form a high-pass filter governed by equation 14. fc + 1 2p R L C C (14) The main disadvantage, from a performance standpoint, is that the typically small load impedances drive the low-frequency corner higher. Large values of CC are required to pass low frequencies into the load. Consider the example where a CC of 68 µF is chosen and loads vary from 32 Ω to 47 kΩ. Table 3 summarizes the frequency response characteristics of each configuration. Table 3. Common Load Impedances vs Low Frequency Output Characteristics in SE Mode RL CC Lowest Frequency 32 Ω 68 µF Ą73 Hz 10,000 Ω 68 µF 0.23 Hz 47,000 Ω 68 µF 0.05 Hz As Table 3 indicates, headphone response is adequate and drive into line level inputs (a home stereo for example) is very good. The output coupling capacitor required in single-supply SE mode also places additional constraints on the selection of other components in the amplifier circuit. With the rules described earlier still valid, add the following relationship: ǒC B 1 v 1 Ơ 1 ǒCI RIǓ RLCC 55 kΩǓ (15) power supply decoupling, CS The TPA2000D4 is a high-performance CMOS audio amplifier that requires adequate power supply decoupling to ensure the output total harmonic distortion (THD) is as low as possible. Power supply decoupling also prevents oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling is achieved by using two capacitors of different types that target different types of noise on the power supply leads. For higher frequency transients, spikes, or digital hash on the line, a good low equivalent-series-resistance (ESR) ceramic capacitor, typically 0.1 µF, placed as close as possible to the device VDD lead works best. For filtering lower-frequency noise signals, a larger aluminum electrolytic capacitor of 10 µF or greater, placed near the audio power amplifier is recommended. midrail bypass capacitor, CBYP The midrail bypass capacitor, CBYP, is the most critical capacitor and serves several important functions. During start-up or recovery from shutdown mode, CBYP determines the rate at which the amplifier starts up. The second function is to reduce noise produced by the power supply caused by coupling into the output drive signal. This noise is from the midrail generation circuit internal to the amplifier, which appears as degraded PSRR and THD+N. Bypass capacitor (CBYP) values of 0.47-µF to 1-µF ceramic or tantalum, low-ESR capacitors are recommended for the best THD and noise performance. WWW.TI.COM 21 
 
      
 
   SLOS337B − DECEMBER 2000 − REVISED NOVEMBER 2005 APPLICATION INFORMATION midrail bypass capacitor, CBYP (continued) Increasing the bypass capacitor reduces clicking and popping noise from power on/off and entering and leaving shutdown. To have minimal pop, CBYP should be 10 times larger than CI. CBYP ≥ 10 × CI (16) differential input The differential input stage of the amplifier cancels any noise that appears on both input lines of a channel. To use the TPA2000D4 EVM with a differential source, connect the positive lead of the audio source to the RINP (LINP) input and the negative lead from the audio source to the RINN (LINN) input. To use the TPA2000D4 with a single-ended source, ac ground the RINN and LINN inputs through a capacitor and apply the audio single to the RINP and LINP inputs. In a single-ended input application, the RINN and LINN inputs should be ac-grounded at the audio source instead of at the device inputs for best noise performance. shutdown modes The TPA2000D4 employs a shutdown mode of operation designed to reduce supply current, IDD, to the absolute minimum level during periods of nonuse for battery-power conservation. The SHUTDOWN input terminal should be held high during normal operation when the amplifier is in use. Pulling SHUTDOWN low causes the outputs to mute and the amplifier to enter a low-current state, IDD(SD) = 0.05 µA. SHUTDOWN should never be left unconnected because amplifier operation would be unpredictable. using low-ESR capacitors Low-ESR capacitors are recommended throughout this application section. A real (as opposed to ideal) capacitor can be modeled simply as a resistor in series with an ideal capacitor. The voltage drop across this resistor minimizes the beneficial effects of the capacitor in the circuit. The lower the equivalent value of this resistance the more the real capacitor behaves like an ideal capacitor. 22 WWW.TI.COM 
 
      
 
   SLOS337B − DECEMBER 2000 − REVISED NOVEMBER 2005 APPLICATION INFORMATION evaluation circuit Right HP SE Input Right Audio Input+ Right Audio Input− C5 0.1 µF C1 0.22 µF C2 0.22 µF C3 0.22 µF 1 C4 0.22 µF 2 C5 0.1 µF R1 20 kΩ TPA2000D4 Left Audio Input− Left Audio Input+ Left HP SE Input R2 20 kΩ 3 4 GAIN SELECT 5 GAIN SELECT C15 10 µF C10 0.1 µF LOUTP 6 7 8 9 10 LOUTN PVDD 11 C9 1 µF R4 20 kΩ 12 13 14 15 16 C16 220 µF R3 20 kΩ RINN LINP RINP HPLIN HPRIN GAIN0 BYPASS GAIN1 32 31 30 29 C8 0.47 µF 28 To System Control SHUTDOWN PVDDL PVDDR LOUTP ROUTP PGNDL PGNDR PGNDL PGNDR LOUTN ROUTN PVDDL PVDDR HPLGAIN HPLOUT MODE NC VDD COSC HPRGAIN ROSC HPROUT AGND C17 + R8 120 kΩ R7 120 kΩ LINN 220 µF 27 26 C11 1 µF C14 10 µF ROUTP 25 24 23 ROUTN 22 C12 1 µF 21 20 C13 0.1 µF 19 220 pF C7 18 PVDD R9 120 kΩ 17 R12 1 kΩ VDD R11 1 kΩ WWW.TI.COM 23 PACKAGE OPTION ADDENDUM www.ti.com 10-Dec-2020 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Pins Package Drawing Qty Eco Plan (2) Lead finish/ Ball material MSL Peak Temp Op Temp (°C) Device Marking (3) (4/5) (6) TPA2000D4DAP ACTIVE HTSSOP DAP 32 46 RoHS & Green NIPDAU Level-3-260C-168 HR -40 to 85 TPA2000D4 TPA2000D4DAPR ACTIVE HTSSOP DAP 32 2000 RoHS & Green NIPDAU Level-3-260C-168 HR -40 to 85 TPA2000D4 (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may reference these types of products as "Pb-Free". RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption. Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of