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SLUS536C − AUGUST 2002 − REVISED AUGUST 2004
FEATURES
D
D
D
D
D
D
D
D
D
D
D
D
DESCRIPTION
Wide Input Supply Range: −20 V to −80 V
The TPS2392 and TPS2393 integrated circuits are hot
swap power managers optimized for use in nominal
−48-V systems. They operate with supply voltage
ranges from −20-V to −80-V, and are rated to withstand
spikes to −100 V. In conjunction with an external
N-channel FET and sense resistor, they can be used to
enable live insertion of plug-in cards and modules in
powered systems. Each device provides load current
slew rate control and peak magnitude limiting.
Undervoltage and overvoltage shutdown thresholds
are easily programmed via a three-resistor divider
network. In addition, two active-low, debounced inputs
provide plug-in insertion detection. A power good (PG)
output enables downstream converters. The TPS2392
and TPS2393 also provide the basic hot swap functions
of electrical isolation of faulty cards, filtered protection
against nuisance overcurrent trips, and single-line fault
reporting. The 44-pin part supports designs where
telecomm creepage and clearance requirements must
be followed.
Transient Rating to −100 V
Programmable Current Limit
Programmable Current Slew Rate
Programmable UV/OV Thresholds/Hysteresis
Debounced Insertion Detection Inputs
Open-Drain Power Good (PG) Output
Fault Timer to Eliminate Nuisance Trips
Open-Drain Fault Output (FAULT)
Enable Input (EN)
14-Pin TSSOP package
44-Pin TSSOP Package for
Creapage/Clearance
APPLICATIONS
D −48-V Distributed Power Systems
D Central Office Switching
D Wireless Base Station
R1
200 kΩ
1%
GND
The TPS2392 latches off in response to current faults,
while the TPS2393 periodically retries the load in the
event of a fault.
DC/DC
CONVERTER
R2
4.99 kΩ
1%
VIN+
C4
100 µF
100 V
R6
10 kΩ
R3
3.92 kΩ
1%
TPS2392/TPS2393
R5
100
kΩ
D2
5.6 V
C3
1500 pF
−48V
C2
0.1 µF
C1
3900 pF
UVLO
OVLO 14
2
INSA DRAINSNS 13
3
INSB
4
FAULT
5
EN
GATE 10
6
FLTTIME
ISENS
7
IRAMP
VOUT+
COUT
EN
VIN−
1
VOUT+
VDD
VOUT−
VOUT−
D1
BAS19
PG 12
RTN 11
Q1
IRF530
VUV
= 32.8 V
VUV
= 30.8 V
VOV
= 72.6 V
9
−VIN 8
R4
20 mΩ
1/4, 1%
UDG−02098
.
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments
semiconductor products and disclaimers thereto appears at the end of this data sheet.
!"# $ %& ' # "$ (&)*%"# +"#', +&%#$
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Copyright 2004, Texas Instruments Incorporated
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SLUS536C − AUGUST 2002 − REVISED AUGUST 2004
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam during
storage or handling to prevent electrostatic damage to the MOS gates.
ORDERING INFORMATION
TA
FAULT OPERATION
LATCH OFF
PERIODIC RETRY
−40°C to 85°C
LATCH OFF
PERIODIC RETRY
PACKAGE
TSSOP (PW)(1)
TSSOP (PW)(1)
PART NUMBER
TSSOP (PW)(1)
TSSOP (PW)(1)
TPS2392DBT
TPS2392PW
TPS2393PW
TPS2393DBT
(1) The PW and DBT package is also available taped and reeled. Add an R suffix to the device type (i.e., TPS2392PWR) for quantities of 2,500 per
reel.
ABSOLUTE MAXIMUM RATINGS
over operating free-air temperature range unless otherwise noted(1)
TPS2392
TPS2393
Input voltage range, VI
UVLO, INSA, INSB, FLTTIME, IRAMP, OVLO,
DRAINSNS, GATE, ISENS(2)
RTN(2)
EN(2)(3)
FAULT(2)(4)
Output voltage range, VO
UNIT
−0.3 to 15
V
−0.3 to 100
PG(2)(4)
FAULT
Continuous output current
PG
10
Operating junction temperature range, TJ
−55 to 125
Storage temperature, Tstg
−65 to 150
mA
°C
C
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds
260
(1) Stresses beyond those listed under “absolute maximum ratings” may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated under “recommended operating conditions” is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
(2) All voltages are with respect to −VIN (unless otherwise noted).
(3) With 100-kΩ minimum input series resistance.
(4) With 10-kΩ minimum series resistance.
2
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SLUS536C − AUGUST 2002 − REVISED AUGUST 2004
RECOMMENDED OPERATING CONDITIONS
MIN
NOM
MAX
UNIT
Input supply voltage, −VIN to RTN
−80
−20
V
Operating junction temperature, TJ
−40
85
°C
DISSIPATION RATINGS
PACKAGE
TA < 25°C
POWER RATING
DERATING FACTOR
ABOVE TA = 25°C
TA = 85°C
POWER RATING
TSSOP−14
750 mW
7.5 mW/°C
300 mW
DBT PACKAGE
(TOP VIEW)
PW PACKAGE
(TOP VIEW)
UVLO
INSA
INSB
FAULT
EN
FLTTIME
IRAMP
1
2
3
4
5
6
7
14
13
12
11
10
9
8
OVLO
DRAINSNS
PG
RTN
GATE
ISENS
−VIN
INSA
N/C
N/C
N/C
N/C
N/C
INSB
N/C
N/C
N/C
N/C
N/C
FAULT(BAR)
EN
FLTTIME
IRAMP
N/C
N/C
N/C
N/C
N/C
−VIN
1
44
2
43
3
42
4
41
5
40
6
39
7
38
8
37
9
36
10
35
11
34
12
33
13
32
14
31
15
30
16
29
17
28
18
27
19
26
20
25
21
24
22
23
UVLO
OVLO
N/C
DRAIN SENSE
PG(BAR)
N/C
N/C
N/C
N/C
N/C
RTN
N/C
N/C
N/C
N/C
N/C
GATE
ISENS
N/C
N/C
N/C
N/C
3
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SLUS536C − AUGUST 2002 − REVISED AUGUST 2004
ELECTRICAL CHARACTERISTICS
VI(−VIN) = −48 V with respect to RTN, VI(EN) = 2.8 V, VI(INSA) = 0 V, VI(INSB) = 0 V, VI(UVLO) = 2.5 V, VI(OVLO) = 0 V, VI(ISENS) = 0 V, all outputs
unloaded, TA = −40°C to 85°C (unless otherwise noted)(1)(2)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
1050
1500
1350
1700
−16
−13
UNIT
INPUT SUPPLY
ICC1
ICC2
Supply current, RTN
VUVLO_L Internal UVLO threshold, VIN rising
VHYS
Internal UVLO hysteresis
VI(RTN) = 48 V
VI(RTN) = 80 V
To GATE pull-up
−19
200
µA
A
V
mV
ENABLE INPUT (EN)
VTH
Threshold voltage, VIN rising
ISRC_EN EN pin switched pull-up current
To GATE pull-up
1.3
1.4
1.5
V
−12
−10
−8
µA
To GATE pull-up
1.36
1.40
1.44
V
VI(UVLO) = 2.5 V
VI(UVLO) = 1 V
−11.7
−10.0
−8.3
µA
UVLO low-level input current
1
µA
Threshold voltage, VIN rising, OVLO
To GATE pull-up
1.36
1.40
1.44
V
−11.7
−10.0
−8.3
µA
1
µA
UNDERVOLTAGE/OVERVOLTAGE COMPARATORS
VTH_UV Threshold voltage, VIN rising, UVLO
ISRC_UV UVLO pin switched pull-up current
IIL
VTH_OV
ISRC_OV OLVO pin switched pull-up current
IIL
OVLO low-level input current
VI(OVLO) = 2.5 V
VI(OVLO) = 1 V
−1
−1
INSERTION DETECTION
VTH
ISRC_INSx
Threshold voltage, VIN rising, INSA, INSB
To GATE pull-down
1.0
1.4
1.8
V
INSA, INSB pin pull-up current
VI(INSA) = 0 V, VI(INSB) = 0 V
To GATE pull-up
−14
−11
−8
µA
1.5
2.5
4.1
ms
11
14
17
V
5
10
tD_INS
Insertion delay time, VIN falling, INSA, INSB
LINEAR CURRENT AMPLIFIER (LCA)
VI(ISENS) = 0 V, IO(GATE) = −10 µA
VI(ISENS) = 80 mV, VO(GATE) = 5 V
VO(FLTTIME) = 2 V
VI(ISENS) = 80 mV, VO(GATE) = 5 V
VO(FLTTIME) > 4 V
VOH
High-level output voltage, GATE
ISINK
Output sink current, linear mode
IFAULT
Output sink current, fault shutdown
II
VREF_K
Input current, ISENS
0 V < VI(ISENS) < 0.2 V
−1
Reference clamp voltage
VO(IRAMP) = OPEN
VO(IRAMP) = 2 V
33
VO(IRAMP) = 0.25 V
VO(IRAMP) = 1 V
−850
−600
−400
−11
−10
−9
−11
−10
−9
VIO
Input offset voltage
RAMP GENERATOR
ISRC1
IRAMP source current, reduced rate turn-on
ISRC2
IRAMP source current, normal rate
VOL
AV
Low-level output voltage, IRAMP
VO(IRAMP) = 3 V
VI(EN) = 0 V
Voltage gain, relative to ISENS
mA
50
100
1
40
−7
µA
47
7
2
mV
nA
µA
A
mV
9.5
10.0
10.5
mV/V
80
100
120
mV
2
4
7
µs
OVERLOAD COMPARATOR
VTH_OL
tDLY
(1)
(2)
4
Current overload threshold, ISENS
Glitch filter delay time
VI(ISENS) = 200 mV
All voltages are with respect to the −VIN terminal, unless otherwise stated.
Currents are positive into and negative out of the specified terminals.
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SLUS536C − AUGUST 2002 − REVISED AUGUST 2004
ELECTRICAL CHARACTERISTICS (continued)
VI(−VIN) = −48 V with respect to RTN, VI(EN) = 2.8 V, VI(INSA) = 0 V, VI(INSB) = 0 V, VI(UVLO) = 2.5 V, VI(OVLO) = 0 V, VI(ISENS) = 0 V, all outputs
unloaded, TA = −40°C to 85°C (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
VOL
ICHG
Low-level output voltage, FLTTIME
VI(EN) = 0 V
VI(ISENS) = 80 mV, VO(FLTTIME) = 2 V
5
mV
−55
−50
−45
µA
VFLT
IDSG
Fault threshold voltage
3.75
4.00
4.25
V
Discharge current, retry mode
TPS2393
0.38
0.61
µA
D
Output duty cycle
TPS2393
1.0%
1.5%
FAULT TIMER
Charging current, current limit mode
IRST
Discharge current, timer reset mode
POWERGOOD SENSING
VTH
ISRC
DRAINSNS threshold voltage
IOH
High-level output leakage current, PG output
DRAINSNS pull-up current
RDS(on) Driver on-resistance, PG output
VI(ISENS) = 80 mV, VO(FLTTIME) = 2 V
VI(ISENS) = 80 mV
VO(FLTTIME) = 2 V, VI(ISENS) = 0 V
VI(DRAINSNS) = 0 V
VI(EN) = 0 V, VO(PG) = 65 V
1
mA
1.20
1.35
1.50
−14
−11
−8
VI(ISENS) = 0 V, VI(DRAINSNS) = 0 V
IO(PG) = 1 mA
10
50
V
µA
A
80
Ω
10
µA
80
Ω
FAULT OUTPUT
IOH
High-level output leakage current, FAULT
RDS(on) Driver on-resistance, FAULT
(1)
(2)
VI(EN) = 0 V, VO(FAULT) = 65 V
VI(ISENS) = 80 V, VO(FLTTIME) = 5 V
IO(FAULT) = 1 mA
50
All voltages are with respect to the −VIN terminal, unless otherwise stated.
Currents are positive into and negative out of the specified terminals.
TERMINAL FUNCTIONS
TERMINAL
I/O
DESCRIPTION
NAME
PW
DBT
DRAINSNS
13
41
I
Sense input for monitoring the load voltage status
EN
5
14
I
Enable input to turn on/off power to the load
FAULT
4
13
O
Open-drain, active-low indication of a load fault condition
FLTTIME
6
15
I/O
Connection for user-programming of the fault timeout period
GATE
10
28
O
Gate drive for external N−channel FET
INSA
2
1
I
Insertion detection input pin A
INSB
3
7
I
Insertion detection input pin B
IRAMP
7
16
I/O
ISENS
9
27
I
Current sense input
OVLO
14
43
I
Voltage sense input for supply overvoltage lockout (OVLO) protection
PG
12
40
O
Open-drain, active-low indication of load power-good condition
RTN
11
34
I
Positive supply input
UVLO
1
44
I
Voltage sense input for supply undervoltage lockout (UVLO) protection
−VIN
8
22
I
Negative supply input and reference pin
Programming input for setting the inrush current slew rate
5
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SLUS536C − AUGUST 2002 − REVISED AUGUST 2004
PIN ASSIGNMENTS
DRAINSNS: Sense input for monitoring the load voltage status. The DRAINSNS pin determines the load status by
sensing the voltage level on the external pass FET drain. DRAINSNS must be pulled low with repect to −VIN (less
than 1.35 V typically) to declare a power good condition. This corresponds to a low VDS across the FET, indicating
that the load voltage has successfully ramped up to the DC input level. DRAINSNS must be connected to the FET
drain through a small-signal blocking diode as shown in the typical application diagram. An internal pull-up maintains
a high logic level at the pin until overridden by a fully-enhanced external FET.
EN: Enable input to turn on/off power to the load. The EN pin is referenced to the −VIN potential of the circuit. When
this input is pulled high (above the nominal 1.4-V threshold), and all other input qualifications are met (supply above
device undervoltage lockout (UVLO), UVLO pin high and OVLO pin low, INSx pins pulled low) the device enables
the GATE output, and begins the ramp of current to the load. When this input is low, the linear current amplifier (LCA)
is disabled, and a large pull-down device is applied to the FET gate, disabling power to the load.
FAULT: Open-drain, active-low indication of a load fault condition. When the device EN is deasserted, or when
enabled and the load current is less than the programmed limit, this output is high impedance. If the device remains
in current regulation mode at the expiration of the fault timer, or if a fast-acting overload condition causes greater than
100-mV drop across the sense resistor, the fault is latched, the load is turned off, and the FAULT pin is pulled low
(to −VIN). The TPS2392 remains latched off for a fault, and can be reset by cycling either the EN pin or power to the
device. The TPS2393 retries the load at approximately a 1% duty cycle.
FLTTIME: Connection for user-programming of the fault timeout period. An external capacitor connected from
FLTTIME to −VIN establishes the timeout period to declare a fault condtion. This timeout protects against indefinite
current sourcing into a faulted load, and also provides a filter against nuisance trips from momentary current spikes
or surges. The TPS2392 and TPS2393 define a fault condition as voltage at the ISENS pin at or greater than the
40-mV fault threshold. When a fault condition exists, the timer is active. The devices manage fault timing by charging
the external capacitor to the 4-V fault threshold, then subsequently discharging it to reset the timer (TPS2392), or
discharging it at approximately 1% the charge rate to establish the duty cycle for retrying the load (TPS2393).
Whenever the internal fault latch is set (timer expired), the pass FET is rapidly turned off, and the FAULT output is
asserted.
GATE: Gate drive for external N−channel FET. When enabled, and the input supply is above the UVLO threshold,
the gate drive is enabled and the device begins charging an external capacitor connected to the IRAMP pin. This
pin voltage is used to develop the reference voltage at the non-inverting input of the internal LCA. The inverting input
is connected to the current sense node, ISENS. The LCA acts to slew the pass FET gate to force the ISENS voltage
to track the reference. The reference is internally clamped at 40 mV, so the maximum current that can be sourced
to the load is determined by the sense resistor value as IMAX ≤ 40 mV/RSENSE. Once the load voltage has ramped
up to the input dc potential, and current demand drops off, the LCA drives the GATE output to about 14 V to fully
enhance the pass FET, completing the low-impedance supply return path for the load.
INSA: Insertion detection input pin A. The INSA and INSB inputs work together to provide an insertion detection
function for TPS2392 and TPS2393 applications. In order to turn on the FET gate drive (the GATE output), both INSA
and INSB must be pulled below the detection threshold, approximatey 1.4 V. Implementations using this feature
provide a mechanism for pulling these pins directly to −VIN potential (device ground), eliminating any threshold
ambiguity. An on-chip pull-up is provided at each INSx pin; no additional pull-up is needed to hold the pins high during
the insertion process. The insertion inputs are debounced with a nominal 2.5-ms filter.
INSB: Insertion detection input pin B. See INSA description.
IRAMP: Programming input for setting the inrush current slew rate. An external capacitor connected between this
pin and −VIN establishes the load current slew rate whenever power to the load is enabled. The device charges the
external capacitor to establish the reference input to the LCA. The closed-loop control of the LCA and pass FET acts
to maintain the current sense voltage at ISENS at the reference potential. Since the sense voltage is developed as
the drop across a resistor, the load current slew rate is set by the voltage ramp rate at the IRAMP pin. When the output
is disabled for any reason (e.g., EN deassertion, voltage or current fault, etc.), the capacitor is discharged and held
low to initialize it for the next turn-on.
6
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SLUS536C − AUGUST 2002 − REVISED AUGUST 2004
PIN ASSIGNMENTS
ISENS: Current sense input. An external low-value resistor connected between this pin and −VIN is used to feed
back current magnitude information to the TPS2392 and TPS2393. There are two internal device thresholds
associated with the voltage at the ISENS pin. During ramp-up of the load’s input capacitance, or during other periods
of excessive demand, the HSPM acts to limit this voltage to 40 mV. Whenever the LCA is in current regulation mode,
the capacitor at FLTTIME is charged to activate the timer. If, when the LCA is driving to its supply rail, a fast-acting
fault such as a short-circuit, causes the ISENS voltage to exceed 100 mV (the overload threshold), the GATE pin
is pulled low rapidly, bypassing the fault timer.
OVLO: Voltage sense input for supply overvoltage lockout (OVLO) protection. Overvoltage protection can be
achieved by applying a divided down sample of the input supply voltage to this pin. In order to turn on gate drive to
the external FET, the OVLO pin must be below the 1.4-V typical threshold, while all other input qualifications are met.
If the OVLO pin is raised above this threshold, as with increasing supply voltage, the GATE output is pulled low,
interrupting the supply to the load. An internal 10-µA pull-up is switched to this pin when the threshold is exceeded,
providing a mechanism for setting the amount of OVLO hysteresis along with the trip threshold.
PG: Open-drain, active-low indication of load power good condition. The TPS2392 and TPS2393 devices define
power good as the voltage at the DRAINSNS pin below the power good threshold, and the voltage at the IRAMP pin
being above 5 V. This assures that full programmed sourcing current is available to the load prior to declaring power
good, even with very slow current ramp rates. The additional protection prevents potential discharging of the module
bulk capacitance during load turn-on.
RTN: Positive supply input for the TPS2392 and TPS2393. For negative voltage systems, the supply pin connects
directly to the return node of the input power bus. Internal regulators step down the input voltage to generate the
various supply levels used by the TPS2392 and TPS2393.
UVLO: Voltage sense input for supply uvervoltage lockout (UVLO) protection. Undervoltage protection can be
achieved by applying a divided down sample of the input supply voltage to this pin. In order to turn on the gate drive
to the external FET, the UVLO pin must be above the 1.4-V typical threshold, while all other input qualifications are
met. If the UVLO pin drops below this threshold, as with decreasing supply voltage, the GATE output is pulled low,
interrupting the supply to the load. An internal 10-µA pull−up is switched to this pin when the threshold is exceeded,
providing a mechanism for setting the amount of UVLO hysteresis along with the trip threshold.
For proper operation, a minimum 1500-pF capacitor, connected between the UVLO and −VIN pins, is required.
−VIN: Negative supply input and reference pin for the TPS2392 and TPS2393. This pin connects directly to the input
supply negative rail. The input and output pins and all internal circuitry are referenced to this pin, so it is essentially
the GND or VSS pin of the device.
7
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SLUS536C − AUGUST 2002 − REVISED AUGUST 2004
TYPICAL CHARACTERISTICS
EN (5 V/div)
EN (5 V/div)
VDRAIN
(20 V/div)
VDRAIN
(50 V/div)
CONTACT
BOUNCE
CONTACT
BOUNCE
CIRAMP = 3900 pF
CFLT = 0.1 µF
CLOAD = 50 µF
ILOAD
(500 mA/div)
CLOAD = 100 µF
t − Time − 2.5 ms / div
t − Time − 1 ms / div
Figure 1. Live Insertion Event − VIN = −48 V
Figure 2. Live Insertion Event − VIN = −70 V
TPS2392 ONLY
TPS2393 ONLY
VDRAIN (50 V/div)
VDRAIN (50 V/div)
FLTTIME (2 V/div)
FLTTIME (2 V/div)
ILOAD (1 A/div)
ILOAD (1 A/div)
FAULT (20 V/div)
FAULT (20 V/div)
CIRAMP = 3900 pF
CFLT = 0.047 µF
CIRAMP = 3900 pF
CFLT = 0.047 µF
t − Time − 1 ms / div
Figure 3. Turn−On Into Shorted Load
8
ILOAD
(500 mA/div)
t − Time − 1 ms / div
Figure 4. Turn−On Into Shorted Load (TPS2393)
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SLUS536C − AUGUST 2002 − REVISED AUGUST 2004
TYPICAL CHARACTERISTICS
CLOAD = 50 µF
VUVLO_L
VUVLO_H
RTN (5 V/div)
RTN (5 V/div)
GATE (5 V/div)
GATE (5 V/div)
CLOAD = 50µF
RLOAD = 1 kΩ
t − Time − 5 ms / div
t − Time − 5 ms / div
Figure 5. UVLO Protection, Supply Rising
Figure 6. UVLO Protection Supply Falling
IRAMP (2 V/div)
INSA (5 V/div)
CIRAMP=
3900 pF
Insertion
Delay
GATE (5 V/div)
CIRAMP = .022 µF
CIRAMP = .056 µF
CFLT = 0.33 µF
CLOAD = 600 µF
VDRAIN (20 V/div)
ILOAD
(500 mA/div)
t − Time − 1 ms / div
Figure 7. Inertion Detection Function
t − Time − 10 ms / div
Figure 8. Load Current Ramp Profiles
9
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SLUS536C − AUGUST 2002 − REVISED AUGUST 2004
TYPICAL CHARACTERISTICS
FAULT (50 V/div)
FAULT (50 V/div)
CIRAMP = 3900 pF
CFLT = 0.047 µF
CLOAD = 100 µF
RLOAD = 12.5 Ω
PG (50 V/div)
CIRAMP = 3900 pF
CFLT = 0.047 µF
CLOAD = 100 µF
PG (50 V/div)
FLTTIME (2 V/div)
FLTTIME (2 V/div)
VDRAIN (50 V/div)
VDRAIN
(50 V/div)
ILOAD
(1 A/div)
ILOAD (1 A/div)
t − Time − 50 ms / div
t − Time − 50 ms / div
Figure 9. Fault Retry Operation (TPS2393)
Figure 10. Fault Recovery (Large Scale View)
TPS2393 ONLY
CIRAMP = 3900 pF
CLOAD = 220 µF
FAULT (50 V/div)
CIRAMP = 3900 pF
CFLT = 0.047 µF
CLOAD = 100 µF
PG (50 V/div)
IRAMP (2 V/div)
FLTTIME (2 V/div)
VDRAIN (20 V/div)
VDRAIN (50 V/div)
PG (50 V/div)
ILOAD (1 A/div)
10
t − Time − 1 ms / div
t − Time − 1 ms / div
Figure 11. Fault Recovery − Expanded View
Figure 12. PG Output Timing, Voltage Qualified
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SLUS536C − AUGUST 2002 − REVISED AUGUST 2004
TYPICAL CHARACTERISTICS
SUPPLY CURRENT
vs
AMBIENT TEMPERATURE
1500
CIRAMP = 6800 pF
CLOAD = 50 µF
VTH_PG
VRTN = 80 V
ICC − Supply Current − µA
1200
IRAMP (2 V/div)
VDRAIN (20 V/div)
900
VRTN = 48 V
600
VRTN = 36 V
VRTN = 20 V
300
PG (50 V/div)
0
−40
t − Time − 1 ms / div
−15
10
35
60
TA − Ambient Temperature − °C
Figure 13. PG Output Timing, Current Qualified
IRAMP OUTPUT CURRENT
vs
AMBIENT TEMPERATURE, REDUCED RATE
GATE HIGH-LEVEL OUTPUT VOLTAGE
vs
AMBIENT TEMPERATURE
VI(ISENS) = 0 V
IO(GATE) = −10 µA
15
VOH − Output Voltage − V
Figure 14.
−500
VRTN = 80 V
VRTN = 48 V
ISRC1 − IRAMP Output Current − nA
16
14
13
VRTN = 36 V
12
VRTN = 20 V
11
10
−40
−15
10
35
60
TA − Ambient Temperature − °C
Figure 15.
85
85
VO(IRAMP) = 0.25 V
−520
VRTN = 48 V
VRTN = 20 V
−540
−560
VRTN = 80 V
−580
−600
−620
−40
−15
10
35
60
85
TA − Ambient Temperature − °C
Figure 16.
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SLUS536C − AUGUST 2002 − REVISED AUGUST 2004
TYPICAL CHARACTERISTICS
IRAMP OUTPUT CURRENT
vs
AMBIENT TEMPERATURE, NORMAL RATE
−47
−9.0
VI(ISENS) = 80 mV
VO(FLTTIME) =2V
Average for VO(IRAMP) = 1 V, 3 V
VRTN = 20 V to 80 V
−48
ICHG − Charging Current − µA
ISRC2 − IRAMP Output Current − µA
TIMER CHARGING CURRENT
vs
AMBIENT TEMPERATURE
−9.4
−9.8
−10.2
VRTN = 20 V
−49
−50
−51
VRTN = 36 V
−10.6
VRTN = 48 V
VRTN = 80 V
−52
−11.0
−40
−15
10
35
60
TA − Ambient Temperature − °C
−53
−40
85
−15
TIMER DISCHARGE CURRENT
vs
AMBIENT TEMPERATURE
360
320
280
240
−40
−15
10
35
60
TA − Ambient Temperature − °C
Figure 19.
12
60
85
FAULT LATCH THRESHOLD
vs
AMBIENT TEMPERATURE
4.25
TPS2393 ONLY
VI(ISENS) = 80 mV
VO(FLTTIME) = 2 V
VI(RTN) = 20 V to 80 V
VFLT − Fault Latch Threshold Voltage − V
IDSG − Charging Current − nA
400
35
Figure 18.
Figure 17.
440
10
TA − Ambient Temperature − °C
85
VI(RTN) = 48 V
4.15
4.05
3.95
3.85
3.75
−40
−15
10
35
60
TA − Ambient Temperature − °C
Figure 20.
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SLUS536C − AUGUST 2002 − REVISED AUGUST 2004
TYPICAL CHARACTERISTICS
UVLO PIN PULL-UP CURRENT
vs
AMBIENT TEMPERATURE
VOLTAGE COMPARATOR THRESHOLD
vs
AMBIENT TEMPERATURE
−9.3
ISRC_UV − Charging Current − µA
VTH − Voltage Comparator Threshold − V
1.44
VI(RTN) = 20 V to 48 V
Undervoltage
Comparator
1.42
1.40
Overvoltage
Comparator
VI(RTN) = 80 V
1.38
VI(UVLO) = 2.5 V
VI(RTN) = 20 V to 48 V
−9.5
−9.7
−9.9
−10.1
1.36
−40
−15
10
35
60
TA − Ambient Temperature − °C
85
−10.3
−40
−15
10
35
60
TA − Ambient Temperature − °C
85
Figure 22.
Figure 21.
INSA PIN INSERTION DELAY TIME
vs
AMBIENT TEMPERATURE
tD_INS − INSA Insertion Delay − ms
3.0
2.9
VRTN = 20 V
VRTN = 80 V
2.8
2.7
2.6
VRTN = 36 V
VRTN = 48 V
2.5
Input voltage falling
measured to GATE pull-up
2.4
−40
−15
10
35
60
TA − Ambient Temperature − °C
85
Figure 23.
13
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SLUS536C − AUGUST 2002 − REVISED AUGUST 2004
DETAILED DESCRIPTION
When a plug-in module or printed circuit card is inserted into a live chassis slot, discharged supply bulk capacitance
on the board can draw huge transient currents from the system supplies. Without some form of inrush limiting, these
currents can reach peak magnitudes ranging over 100 A, particularly in high-voltage systems. Such large transients
can damage connector pins, PCB etch, and plug-in and supply components. In addition, current spikes can cause
voltage droops on the power distribution bus, causing other boards in the system to reset.
The TPS2392 and TPS2393 are hot swap power managers that limit current peaks to preset levels, as well as control
the slew rate (di/dt) at which charging current ramps to the programmed limit. These devices use an external
N-channel pass FET and sense element to provide closed-loop control of current sourced to the load. Input
undervoltage lockout (UVLO) and overvoltage lockout (OVLO) functions control automatic turn-on when the input
supply voltage is within the specified operational window, otherwise inhibiting card operation by turning off the pass
FET. In addition, load power can be controlled with a system logic command via the EN input, allowing electrical
isolation of faulty cards from the power bus. Two active-low inputs can be connected to provide card insertion
detection. An internal overload comparator provides circuit breaker protection against short-circuits occurring during
steady-state (post-turn-on) operation of the card. Load power status is continuously monitored and reported via the
PG (powergood) and FAULT outputs.
The TPS2392 and TPS2393 operate directly from the input supply (nominal −48 Vdc rail). The −VIN pin connects
to the negative voltage rail, and the RTN pin connects to the supply return. Internal regulators convert input power
to the supply levels required by the device circuitry. An input UVLO circuit holds the GATE output low until the supply
voltage reaches a nominal 16-V level, regardless of the status of all other control inputs. A block of comparators
monitors input supply voltage and other output enable conditions. As shown in Figure 24, the status of these five
comparators is AND’d together in order to enable turning on power to the load. Two precision comparators monitor
the voltage levels at the UVLO and OVLO pins. Typically, these pins are driven with a divided-down sample of the
supply voltage to establish the UVLO and OVLO trip thresholds for the circuit. The UVLO input must be above the
internal 1.4-V reference, and the OVLO pin must remain below the reference voltage to enable the load. Both of these
inputs are provided with a small, 10-µA pull-up source, which is switched to the input pin whenever the associated
comparator is tripped. These current sources provide a mechanism for user-programming of the amount of hysteresis
for the UVLO and OVLO thresholds.
The same comparator circuit is also available at the EN pin, providing a third precision input. A switched pull-up is
also available at this pin for hysteresis programming. Alternatively, this input can be used as a logic enable command,
with a nominal 1.4-V logic threshold.
The INSA and INSB pins provide an optional insertion detection function to the hot swap circuit. Both these pins must
be pulled low, below 1.0 -V minimum, to enable a load start-up. Internal pull-ups at these inputs maintain a HI logic
level (about 6.5 V) at the device pins when floating. This eliminates the need for additional external components to
maintain the HI logic level during insertion and extraction events. An external mechanism for pulling these inputs low
completes the qualification for turning on power to the load.
Once the device is enabled (internal EN_A signal asserted), the GATE output pull-down is turned off, and the linear
control amplifier (LCA) is enabled. A current source in the ramp generator block begins charging an external capacitor
connected between the IRAMP and −VIN pins. The resultant voltage ramp at the IRAMP pin is scaled by a factor of
1/100, and applied to the non-inverting input of the LCA (the VLIM signal). Load current magnitude information at the
ISENS pin is applied to the inverting input. This sense voltage is developed by connecting the current sense resistor
between ISENS and −VIN. As the external FET begins to conduct, the LCA slews its gate to force the ISENS voltage
to track the internal reference (VLIM). Consequently, the load current slew rate tracks the linear voltage ramp at the
IRAMP pin, producing a linear di/dt of current to the load.
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SLUS536C − AUGUST 2002 − REVISED AUGUST 2004
DETAILED DESCRIPTION
RTN
11
10 µA
10 µA
1.35V
13
DRAINSNS
12
PG
+
+
INSA
2
+
10 µA
Q
R
Q
5V
+
INSB
S
FLT
3
10 µA
EN_A
H=CLOSED
UVLO
1
7
IRAMP
4
FAULT
6
FLTTIME
9
ISENS
10
GATE
RAMP
GENERATOR
+
VLIM
FLT
ON
10 µA
L=CLOSED
EN
OVLO
14
FAULT
TIMER
+
OC
10 µA
5
Q
R
R
Q
FT
OVERCURRENT
100 mV
OLC
+
+
VLIM
1.4V
−VIN
S
OVERLOAD
H=CLOSED
EN
OL
S
LCA
+
EN_A
8
UDG−02116
Figure 24. Block Diagram of PW Package
Under normal load and input supply conditions, this controlled current charges the module’s input bulk capacitance
up to the input dc voltage level. At this point, the load demand drops off, and the voltage at ISENS decreases. The
LCA now drives the GATE output to its supply rail. The 14-V typical output level ensures sufficient overdrive to fully
enhance the external FET, while not exceeding the typical 20-V VGS rating of common N-channel power MOSFETs.
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SLUS536C − AUGUST 2002 − REVISED AUGUST 2004
RTN
34
10 µA
10 µA
1.35V
41
DRAINSNS
40
PG
+
+
INSA
1
+
10 µA
Q
R
Q
5V
+
INSB
S
FLT
7
10 µA
EN_A
H=CLOSED
UVLO
44
16 IRAMP
RAMP
GENERATOR
+
VLIM
FLT
13
ON
10 µA
L=CLOSED
EN
OVLO
43
FAULT
TIMER
+
OC
OL
S
S
Q
R
R
Q
FAULT
FT
15 FLTTIME
10 µA
OVERLOAD
OVERCURRENT
H=CLOSED
EN
14
27 ISENS
+
VLIM
1.4V
−VIN
100 mV
OLC
+
LCA
+
28
GATE
EN_A
22
UDG−02116
Figure 25. Block Diagram of DBT Package
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SLUS536C − AUGUST 2002 − REVISED AUGUST 2004
DETAILED DESCRIPTION
Current fault response timing and retry duty cycle are accomplished by the fault timer block in conjunction with an
external capacitor connected between the FLTTIME and −VIN pins. Whenever the hot swap controller is in current
control mode, such as during inrush limiting at insertion, or in response to excessive demand during operation of the
plug-in, the LCA asserts the OVERCURRENT signal shown in Figure 24. This signal starts the charging of the
FLTTIME capacitor. If this capacitor charges to the pin’s 4-V trip threshold, the fault is latched. A latched fault disables
the LCA drive, and turns on a large pull-down device at the GATE output to rapidly turn off the external FET. The fault
condition is indicated by turning on the open-drain FAULT output driver. A latched fault also causes discharge of the
external capacitors at the IRAMP and FLTTIME pins, in order to reset the hot swap circuit for the next output enable
event, if and when conditions permit.
An internal overload comparator (OLC in Figure 24) also monitors the ISENS voltage against a nominal 100-mV
threshold. This comparator provides circuit breaker protection against sudden current fault conditions, such as a load
short-circuit. The OVERLOAD output of this comparator also drives the fault timer. The timer circuit applies a 4-µs
deglitch filter to help reduce nuisance trips. However, if the overload condition exceeds the filter length, the fault is
latched, the LCA disabled, and the FET gate rapidly pulled down, bypassing the programmed timeout period.
The PG pin is an open-drain, active-low indication of a load power good status. Load voltage sensing is provided at
the DRAINSNS pin. To assert PG, the device must not be in latched current fault status, the DRAINSNS pin must
be pulled below the 1.35-V nominal threshold, and the voltage at the IRAMP pin must be greater than approximately
5 V. This last criteria ensures that maximum allowed sourcing current is available to the load before declaring power
good. Once all the conditions are met, the PG status is latched on-chip. This prevents instances of momentary
current-limit operation (e.g., due to load surges or voltage spikes on the input supply) from propagating through to
the PG output. However, if input conditions are not met, or if a persistent load fault does result in fault timeout, the
PG latch will be cleared.
Additional details of the ramp generator operation are shown in Figure 25. To enable the generator, the large NMOS
device shown in this circuit is turned off. This allows a small current source to charge the external capacitor connected
at the IRAMP pin. The voltage ramp on the capacitor actually has two discrete, linear slopes. As shown in Figure 25,
current is supplied from either of two sources. An internal comparator monitors the IRAMP voltage level, and selects
the appropriate charging rate. Initially at turn-on, when the pin voltage is 0 V, the 600-nA source is selected, to provide
a slow turn-on (or reduced-rate) sourcing period. This slow turn-on ensures that the LCA is pulled out of saturation,
and is slewing to the voltage at its non-inverting input before normal rate load charging is allowed. This scheme helps
reduce or eliminate current steps at the external FET on-threshold. Once the voltage at the IRAMP pin reaches
approximately 0.5 V, the SLOW signal is deasserted, and the 10-µA source is selected for the remainder of the ramp
period.
The IRAMP pin voltage is divided down by a factor of 100, and applied to the non-inverting input of the LCA (see
Figure 24). Although the IRAMP capacitor is charged to about 6.5 V, the VLIM reference is clamped at 40 mV.
Therefore, current sourced to the load during turn-on is limited to a value given by IMAX ≤ 40 mV/RSENSE, where
RSENSE is the value of the external sense resistor. Therefore, both load current maximum slew rate and peak
magnitude are easily set with just two external components.
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SLUS536C − AUGUST 2002 − REVISED AUGUST 2004
DETAILED DESCRIPTION
10 µA
600 nA
SLOW
+
0.5 V
IRAMP
VLIM
99R
EN_A
R
40mV
UDG−20117
Figure 26. Ramp Generator Block Details
Note that any condition which causes turn-off of the external FET (EN_A signal goes low) also causes a rapid
discharge of the IRAMP capacitor. In this manner, the soft-start function is automatically reset by the TPS2392 and
TPS2393, and ready for the next load enable event.
Fault timer operation is further detailed in Figure 26. As described earlier, the LCA OVERCURRENT output drives
the OC input signal shown in Figure 26. Overcurrent fault timing is actually inhibited during the reduced rate (slow
turn-on) portion of the IRAMP voltage waveform. However, once the device transitions to the normal rate current ramp
(VO(IRAMP) ≥ 0.5 V), the FLTTIME capacitor is charge by the 50-µA current source, generating a second voltage ramp
at the FLTTIME pin. This voltage is monitored by the two comparators shown in the fault timer block. If this voltage
reaches the nominal 4-V comparator threshold, the fault is latched, the GATE pin pulled low rapidly, and the FAULT
output asserted. The filtered overload signal (OL) can also set the fault latch. Once a fault is latched, capacitor
charging ceases (ON signal deasserted) and the timing capacitor is discharged.
The TPS2392 latches off in response to faults. Once a fault timeout occurs, the RESET signal turns on a large
NMOS device to rapidly discharge the external capacitor, resetting the timer for any subsequent device reset.
The TPS2392 can be reset only by cycling power to the device, or by cycling the EN input.
In response to a latched fault condition, the TPS2393 enters a fault retry mode, wherein it periodically retries
the load to test for continued existence of the fault. In this mode, the FLTTIME capacitor is discharged slowly
by a about a 0.4-µA constant-current sink. When the voltage at the FLTTIME pin decays below 0.5 V, the ON
signal once again enables the LCA and ramp generator circuits, and a normal turn-on current ramp ensues.
Again, during the load charging, the OC signal causes charging of the FLTTIME capacitor until the next delay
period elapses. The sequential charging and discharging of the FLTTIME capacitor results in a typical 1% retry
duty cycle. If the current-limit fault subsides (GATE pin drives to high-level output), the timing cap is rapidly
discharged, duty-cycle operation stops, and the fault latch is reset. For an initial latched fault that was due to
an overload condition (i.e., overload comparator response), the latching action causes charging of the timer
capacitor, with GATE output already off, to initiate fault retry timing.
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SLUS536C − AUGUST 2002 − REVISED AUGUST 2004
DETAILED DESCRIPTION
4 µs
OL
50 µA
FAULT
S
4V
OC
S
Q
R
Q
+
FLTTIME
0.4 µA
RETRY
0.5V
+
RESET
ON
EN
FAULT
LOGIC
R
TPS2393 ONLY
See block diagram on page 15 and 16 for pinout.
UDG−20118
Figure 27. Fault Timer Block Operation
Note that because of the timing inhibit during the initial slow ramp period, the duty cycle in practice is slightly
greater than the nominal 1% value. However, sourced current during this period peaks at only about one-eighth
the maximum limit. The duty cycle of the normal ramp and constant-current periods will be about 1%.
The fault logic within the timer block automatically manages capacitor charge and discharge rates (RESET
signal), and the operational status of other device-internal circuits (ON signal). For the TPS2393, the FAULT
output remains asserted continuously during retry mode; it is only released if the fault condition clears.
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SLUS536C − AUGUST 2002 − REVISED AUGUST 2004
APPLICATION INFORMATION
setting the sense resistor value
Due to the current−limiting action of the internal LCA, the maximum allowable load current for an implementation
is easily programmed by selecting the appropriate sense resistor value. The LCA acts to limit the sense voltage
VI(ISENS) to its internal reference. Once the voltage at the IRAMP pin exceeds approximately 4 V, this limit is
the clamp voltage, VREF_K. Therefore, a maximum sense resistor value can be determined from equation (1).
R SENSE v 33 mV
IMAX
(1)
where:
D RSENSE is the resistor value
D IMAX is the desired current limit
When setting the sense resistor value, it is important to consider two factors, the minimum current that may
be imposed by the TPS2392 or TPS2393, and the maximum load under normal operation of the module. For
the first factor, the specification minimum clamp value is used, as seen in equation (1). This method accounts
for the tolerance in the sourced current limit below the typical level expected (40 mV/RSENSE). (The clamp
measurement includes LCA input offset voltage; therefore, this offset does not have to be factored into the
current limit again.) Second, if the load current varies over a range of values under normal operating conditions,
then the maximum load level must be allowed for by the value of RSENSE. One example of this is when the load
is a switching converter, or brick, which draws higher input current, for a given power output, when the
distribution bus is at the low end of its operating range, with decreasing draw at higher supply voltages. To avoid
current-limit operation under normal loading, some margin should be designed in between this maximum
anticipated load and the minimum current limit level, or IMAX > ILOAD(max), for equation (1).
For example, using a 20-mΩ sense resistor for a nominal 1-A load application provides a minimum of 650 mA
of overhead for load variance/margin. Typical bulk capacitor charging current during turn-on ia 2 A
(40 mV/20 mΩ).
setting the inrush slew rate
The TPS2392/93 devices enable user-programming of the maximum current slew rate during load start-up
events. A capacitor tied to the IRAMP pin (C1 in the typical application diagram) controls the di/dt rate. Once
the sense resistor value has been established, a value for ramp capacitor CIRAMP, in microfarads, can be
determined from equation (2).
C IRAMP +
11
100
R SENSE
ǒdtdiǓMAX
(2)
where:
D RSENSE is the sense resistor value in Ω
D (di/dt)MAX is the desired maximum slew rate in A/s
For example, if the desired slew rate for the typical application shown is 1500 mA/mS, the calculated value for
CIRAMP is about 3700 pF. Selecting the next larger standard value of 3900 pF (as shown in the diagram) provides
some margin for capacitor and sense resistor tolerances.
As described in the Detailed Description section of this datasheet, the TPS2392 and TPS2393 initiate ramp
capacitor charging, and consequently, load current di/dt at a reduced rate. This reduced rate applies until the
voltage on the IRAMP pin is about 0.5 V. The maximum di/dt rate, as set by equation (2), is effective once the
device has switched to the 10-µA charging source.
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SLUS536C − AUGUST 2002 − REVISED AUGUST 2004
APPLICATION INFORMATION
setting the fault timing capacitor
The fault timeout period is established by the value of the capacitor connected to the FLTTIME pin, CFLT. The
timeout period permits riding out spurious current glitches and surges that may occur during operation of the
system, and prevents indefinite sourcing into faulted loads swapped into a live system. However, to ensure
smooth voltage ramping under all conditions of load capacitance and input supply potential, the minimum
timeout should be set to accommodate these system variables. To do this, a rough estimate of the maximum
voltage ramp time for a completely discharged plug-in card provides a good basis for setting the minimum timer
delay.
Due to the three-phase nature of the load current at turn-on, the load voltage ramp has potentially three distinct
phases and is seen by comparing Figure 1 and Figure 2. This profile depends on the relative values of load
capacitance, input dc potential, maximum current limit and other factors. The first two phases are characterized
by the two different slopes of the current ramp; the third phase, if required to complete load charging, is the
constant-current charging at IMAX. Considering the two current ramp phases to be one period at an average
di/dt simplifies calculation of the required timing capacitor.
For the TPS2392 and TPS2393, the typical duration of the soft-start ramp period, tSS, is given by equation (3).
t SS + 1183
C IRAMP
(3)
where:
D tSS is the soft-start period in milliseconds, and
D CIRAMP is given in µF
During this current ramp period, the load voltage magnitude which is attained is estimated by equation (4).
V LSS +
iAVG
2
CL
C IRAMP
100
RSENSE
ǒt SSǓ
2
(4)
where:
D VLSS is the load voltage reached during soft-start
D iAVG is 3.38 µA for the TPS2392 and TPS2393
D CL is the amount of the load capacitance
D tSS is the soft−start period, in seconds
The quantity iAVG in equation (4) is a weighted average of the two charge currents applied to CIRAMP during
turn-on, considering the typical output values.
If the result of equation (4) is larger than the maximum input supply value, then the load can be expected to
charge completely during the inrush slewing portion of the insertion event. However, if this voltage is less than
the maximum supply input, VIN(max), the HSPM transitions to the constant-current charging of the load. The
remaining amount of time required at IMAX is determined from equation (5).
t CC +
CL
ǒVIN(max) * VLSSǓ
ǒ
Ǔ
V REF_K(min)
R SENSE
(5)
where:
D tCC is the constant-current voltage ramp time, in seconds
D VREF_K(min) is the minimum clamp voltage, 33 mV.
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SLUS536C − AUGUST 2002 − REVISED AUGUST 2004
APPLICATION INFORMATION
With this information, the minimum recommended value timing capacitor CFLT can be determined. The delay
time needed will be either tSS or the sum of tSS and tCC, according to the estimated time to charge the load.
Since fault timing is generated by the constant-current charging of CFLT, the capacitor value is determined by
equation (6) or (7).
C FLT(min) +
C FLT(min) +
55 t SS
3.75
55
(6)
ǒt SS ) t CCǓ
3.75
(7)
where:
D CFLT(min) is the recommended capacitor value, in microfarads
D tSS is the result of equation (3), in seconds
D tCC is the result of equation (5), in seconds
For the typical application example, with the 100-µF filter capacitor in front of the dc-to-dc converter, equations
(3) and (4) estimate the load voltage ramping to −46 V during the soft-start period. If the module should operate
down to −72-V input supply, approximately another 1.58 ms of constant-current charging may be required.
Therefore, equation (7) is used to determine CFLT(min), and the result is approximately 0.1 µF.
setting the undervoltage and overvoltage thresholds
The UVLO and OVLO pins can be used to set the undervoltage (VUV) and overvoltage (VOV) thresholds of the
hot swap circuit. When the input supply is below VUV or above VOV, the GATE pin is held low, disconnecting
power from the load, and deasserting the PG output. When input voltage is within the UV/OV window, the GATE
drive is enabled, assuming all other input conditions are valid for turn-on.
Threshold hysteresis is provided via two internal sources which are switched to either pin whenever the
corresponding input level exceeds the internal 1.4-V reference. The additional bias shifts the pin voltage in
proportion to the external resistance connected to it. This small voltage shift at the device pin is gained up by
the external divider to input supply levels.
GND
GND
R1
200 kΩ
1%
R1
R7
RTN
RTN
UVLO
R2
4.99 kΩ
1%
UVLO
TPS2392/93*
TPS2392/93*
OVLO
R2
OVLO
−VIN
R3
3.92 kΩ
1%
−VIN
R8
−48V
−48V
(a)
(b)
V UV_L + R1 ) R2 ) R3
R2 ) R3
V REF
V OV_L + R1 ) R2 ) R3
R3
V REF * I SRC_UV
R1
V UV_L + R1 ) R2
R2
VTH_UV
V OV_L + R7 ) R8
R8
VTH_OV
*Additional details omitted for clarity. See block diagram on page 15 and 16 for pinout.
Figure 28. Programming the Undervoltage and Overvoltage Thresholds
22
UDG−20119
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SLUS536C − AUGUST 2002 − REVISED AUGUST 2004
APPLICATION INFORMATION
The UV and OV thresholds can be individually programmed with a three-resistor divider connected to it as
shown in the typical application diagram, and again in Figure 27a. When the desired trip voltages and
undervoltage hysteresis have been established for the protected board, the resistor values needed can be
determined from the following equations. Generally, the process is simplest by first selecting the top leg of the
divider (R1 in the diagram) needed to obtain the threshold hysteresis. This value is calculated from equation (8).
R1 +
V HYS_UV
10 mA
(8)
where:
D VHYS_UV is the undervoltage hysteresis value
For example, assume the typical application design targets have been set to undervoltage turn-on at 33 V (input
supply rising), turn-off at 31 V (input voltage falling), and overvoltage shutdown at 72 V. Then equation (8) yields
R1 = 200 kΩ for the 2-V hysteresis. Once the value of R1 is selected, it is used to calculate resistors R2 and
R3.
R2 +
R3 +
ȱ
V UV_L
1*
ȧ
ǒV UV_L * 1.4Ǔ Ȳ ǒV OV_L ) 10*5
1.4
ȳ
ȧ
R1Ǔȴ
R1
1.4
R1
VUV_L
ǒV UV_L * 1.4Ǔ ǒVOV_L ) 10*5
Ǔ
R1
(9)
(10)
where:
D VUV_L is the UVLO threshold when the input supply is low; i.e., less than VUV
D VOV_L is the OVLO threshold when the input supply is low; .i.e., less than VOV
Again referring to the example schematic, equations (9) and (10) produce R2 = 4.909 kΩ (4.99 kΩ selected)
and R3 = 3.951 kΩ (3.92 kΩ selected), as shown. For the selected resistor values, the expected nominal supply
thresholds are as shown on the typical application diagram. The hysteresis on the overvoltage threshold, as
seen at the supply inputs, is given by the quantity (10 µA) * (R1 + R2). For the majority of applications, this value
will be very nearly the same as the UV hysteresis, since typically R1 >> R2.
If more independent control is needed for the OVLO hysteresis, there are several options. One option is to use
separate dividers for both the UVLO and OVLO pins, as shown in Figure 27b. In this case, once R1 and R7
have been selected for the required hysteresis per equation (8), the bottom resistors in the dividers (R2 and
R8 in Figure 27b) can be found from equation (11).
R XVLO +
VREF
ǒVXV_L * VREFǓ
RTOP
(11)
where:
D
D
D
D
RXVLO is R2 or R8
RTOP is R1 or R7 as appropriate for the threshold being set
VXV_L is the under (VUV_L ) or overvoltage (VOV_L ) threshold at the supply input
VREF is either VTH_UV or VTH_OV from the specification table, as required for the resistor being calculated
capacitor on UVLO pin
As shown in the typical application diagram, a minimum 1500 pF capacitor is required on the UVLO pin of the
TPS2392 or TPS2393. For some systems, it may be desirable to slow down the response of the controller to
undervoltage conditions. For example, if frequent voltage dips are anticipated due to other power events in the
system, it may be beneficial to delay somewhat the response of the detection circuit. For these situations, the
size of the capacitor can be increased accordingly, over the value shown.
23
www.ti.com
SLUS536C − AUGUST 2002 − REVISED AUGUST 2004
APPLICATION INFORMATION
using the PG output
The PG output is an indication of the load power status. PG is asserted after a load turn-on, once the load
voltage has ramped up to the input dc level, as indicated by a small VDS drop across the pass FET. The load
voltage is sensed by the DRAINSNS pin, which is connected to the pass FET drain through a small-signal
blocking diode. Also, the TPS2392 and TPS2393 first confirm that the full programmed sourcing current
(typically 40 mV/RSENSE) is available to the load electronics prior to declaring power good. The PG status is
latched once the power conditions are met, so that momentary current limiting operation due to input supply
transients is not reflected in this output status. This pin can be used to enable downstream converters, provide
a visual indication of load power good, or be level-translated or optocoupled to provide status reporting back
to the host controller.
When using PG to drive the enable input of a converter, care should be taken not to exceed the manufacturer’s
maximum voltage ratings for the pin. When asserted, the output driver pulls the PG pin to the −VIN pin potential.
Because this status in latched, subsequent current limit operation of the circuit could result in pulling the enable
input below the brick’s VIN− potential during the fault timeout period. If the brick does not provide an internal
clamp on this pin, a diode can be connected as shown in Figure 28 to externally limit the swing below VIN−.
In either case, a resistor (R7 in Figure 28) should be used to limit the current pulled from this pin, protecting
both the converter and the PG output. R7 should be large enough to limit the PG input current to less than
10 mA, while still allowing the brick enable to be pulled below its maximum VIL threshold.
DC/DC
CONVERTER
VIN+
GND
CIN
RTN
R7
43 kΩ
PG
TPS2392*
TPS2393*
EN
VOUT+
VDD
10 µA
D3
VIN−
VOUT−
D1
BAS19
DRNSNS
Q1
GATE
−VIN
ISENS
RSENSE
−48 V
*Additional details omitted for clarity.
See block diagram on page 15 and 16 for pinout.
Figure 29. TPS2392/TPS2393 Active-Low Converter Enable
24
UDG−20177
www.ti.com
SLUS536C − AUGUST 2002 − REVISED AUGUST 2004
APPLICATION INFORMATION
If the selected converter cannot tolerate any voltage excursions below VIN− potential, an alternative is to drive
the enable through an optocoupler. An implementation is shown in Figure 29.
DC/DC
CONVERTER
VIN+ VOUT+
GND
VDD
R7
CIN
EN
VIN−
RTN
PG
TPS2392*
TPS2393*
10 µA
VOUT−
D1
BAS19
DRNSNS
Q1
GATE
−VIN
ISENS
RSENSE
−48 V
*Additional details omitted for clarity.
See block diagram on page 15 and 16 for pinout.
UDG−20178
Figure 30. PG Driving An Optocoupler
25
PACKAGE OPTION ADDENDUM
www.ti.com
14-Oct-2022
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
Samples
(4/5)
(6)
TPS2393PW
ACTIVE
TSSOP
PW
14
90
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
TPS2393
Samples
TPS2393PWR
ACTIVE
TSSOP
PW
14
2000
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
TPS2393
Samples
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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