TPS5450
SLVS757E – MARCH 2007 – REVISED JULY 2022
TPS5450 5-A, Wide Input Range, Step-Down Converter
1 Features
3 Description
•
•
•
The TPS5450 is a high-output-current PWM converter
that integrates a low-resistance, high-side N-channel
MOSFET. Included on the substrate with the
listed features are a high-performance voltage
error amplifier that provides tight voltage regulation
accuracy under transient conditions; an undervoltagelockout circuit to prevent start-up until the input
voltage reaches 5.5 V; an internally set slow-start
circuit to limit inrush currents; and a voltage feedforward circuit to improve the transient response.
Using the ENA pin, shutdown supply current is
reduced to 18 μA typically. Other features include an
active-high enable, overcurrent limiting, overvoltage
protection and thermal shutdown. To reduce design
complexity and external component count, the
TPS5450 feedback loop is internally compensated.
•
•
•
•
•
•
•
2 Applications
•
•
•
•
High density point-of-load regulators
LCD displays, plasma displays
Battery chargers
12-V and 24-V distributed power systems
The TPS5450 device is available in a thermallyenhanced, 8-pin SOIC PowerPAD package. TI
provides evaluation modules and software tool to aid
in achieving high-performance power supply designs
to meet aggressive equipment development cycles.
Device Information(1)
PART NUMBER
TPS5450
(1)
PACKAGE
BODY SIZE (NOM)
HSOP (8)
4.89 mm × 3.90 mm
For all available packages, see the orderable addendum at
the end of the datasheet.
Efficiency vs Output Current
Simplified Schematic
100
VIN
PH
VIN
VOUT
95
90
NC
BOOT
NC
ENA VSENSE
GND
Efficiency - %
•
Wide input voltage range: 5.5 V to 36 V
Up to 5-A continuous (6-A peak) output current
High efficiency greater than 90% enabled by 110mΩ integrated MOSFET switch
Wide output voltage range: adjustable down to
1.22 V with 1.5% initial accuracy
Internal compensation minimizes external part
count
Fixed 500-kHz switching frequency for small filter
size
18-μA shutdown supply current
Improved line regulation and transient response by
input voltage feedforward
System protected by overcurrent limiting,
overvoltage protection, and thermal shutdown
–40°C to 125°C operating junction temperature
range
Available in small thermally enhanced 8-pin SOIC
PowerPAD™ package
85
80
75
70
VI = 12 V,
VO = 5 V,
fs = 500 kHz,
TA = 25°C
65
60
55
50
0
1
2
3
4
5
IO - Output Current - A
6
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
TPS5450
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Table of Contents
1 Features............................................................................1
2 Applications..................................................................... 1
3 Description.......................................................................1
4 Revision History.............................................................. 2
5 Pin Configuration and Functions...................................3
Pin Functions.................................................................... 3
6 Specifications.................................................................. 4
6.1 Absolute Maximum Ratings........................................ 4
6.2 ESD Ratings............................................................... 4
6.3 Recommended Operating Conditions.........................4
6.4 Thermal Information....................................................4
6.5 Electrical Characteristics.............................................6
6.6 Typical Characteristics................................................ 7
7 Detailed Description........................................................9
7.1 Overview..................................................................... 9
7.2 Functional Block Diagram........................................... 9
7.3 Feature Description...................................................10
7.4 Device Functional Modes..........................................11
8 Application and Implementation.................................. 12
8.1 Application Information............................................. 12
8.2 Typical Application.................................................... 12
9 Power Supply Recommendations................................19
10 Layout...........................................................................19
10.1 Layout Guidelines................................................... 19
10.2 Layout Example...................................................... 20
10.3 Thermal Calculations.............................................. 21
11 Device and Documentation Support..........................22
11.1 Device Support........................................................22
11.2 Trademarks............................................................. 22
11.3 Electrostatic Discharge Caution.............................. 22
11.4 Glossary.................................................................. 22
12 Mechanical, Packaging, and Orderable
Information.................................................................... 22
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision D (December 2014) to Revision E (July 2022)
Page
• Updated the numbering format for tables, figures, and cross-references throughout the document..................1
Changes from Revision C (October 2013) to Revision D (September 2014)
Page
• Added Pin Configuration and Functions section, ESD Ratings table, Feature Description section, Device
Functional Modes, Application and Implementation section, Power Supply Recommendations section, Layout
section, Device and Documentation Support section, and Mechanical, Packaging, and Orderable Information
section ............................................................................................................................................................... 1
2
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5 Pin Configuration and Functions
DDA PACKAGE
(TOP VIEW)
8
PH
7
VIN
3
6
GND
4
5
ENA
BOOT
1
NC
2
NC
VSENSE
PowerPAD
(Pin 9)
Pin Functions
PIN
I/O
DESCRIPTION
NAME
NO.
BOOT
1
O
Boost capacitor for the high-side FET gate driver. Connect 0.01-μF, low-ESR capacitor from BOOT pin to PH
pin.
NC
2, 3
–
Not connected internally.
VSENSE
4
I
Feedback voltage for the regulator. Connect to output voltage divider.
ENA
5
I
On and off control. Below 0.5 V, the device stops switching. Float the pin to enable.
GND
6
–
Ground. Connect to PowerPAD.
VIN
7
I
Input supply voltage. Bypass VIN pin to GND pin close to device package with a high-quality, low-ESR
ceramic capacitor.
PH
8
O
Source of the high-side power MOSFET. Connected to external inductor and diode.
PowerPAD
9
–
GND pin must be connected to the exposed pad for proper operation.
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6 Specifications
6.1 Absolute Maximum Ratings
over operating free-air temperature (unless otherwise noted)(1)
V
Voltage
MIN
MAX
UNIT
–0.3
40(2)
V
PH (steady-state)
–0.6
40(2)
PH (transient < 10 ns)
–1.2
ENA
–0.3
7
BOOT-PH
–0.3
10
–0.3
3
VIN
VSENSE
IO
Source current
PH
Ilkg
Leakage current
PH
10
μA
TJ
Operating virtual junction temperature
–40
150
°C
Tstg
Storage temperature
–65
150
°C
(1)
(2)
Internally Limited
Stresses beyond those listed under Section 6.1 may cause permanent damage to the device. These are stress ratings only and
functional operation of the device at these or any other conditions beyond those indicated under the Recommended Operating
Conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
Approaching the absolute maximum rating for the VIN pin may cause the voltage on the PH pin to exceed the absolute maximum
rating.
6.2 ESD Ratings
VALUE
V(ESD)
(1)
(2)
Electrostatic discharge
Human-body model (HBM), per ANSI/ESDA/JEDEC JS-001(1)
±2000
Charged-device model (CDM), per JEDEC specification JESD22C101(2)
±1500
UNIT
V
JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process.
6.3 Recommended Operating Conditions
MIN
MAX
VI
Input voltage range
5.5
36
UNIT
V
TJ
Operating junction temperature
–40
125
°C
6.4 Thermal Information
TPS5450
THERMAL METRIC(1) (2) (3)
DDA
UNIT
8 PINS
RθJA
Junction-to-ambient thermal resistance (custom board) (4)
30
RθJA
Junction-to-ambient thermal resistance (standard board)
42.3
ψJT
Junction-to-top characterization parameter
4.9
ψJB
Junction-to-board characterization parameter
20.7
RθJC(top)
Junction-to-case(top) thermal resistance
46.4
RθJC(bot)
Junction-to-case(bottom) thermal resistance
0.8
RθJB
Junction-to-board thermal resistance
20.8
(1)
(2)
(3)
4
°C/W
For more information about traditional and new thermal metrics, see the IC Package Thermal Metrics application report, SPRA953.
Maximum power dissipation may be limited by overcurrent protection
Power rating at a specific ambient temperature TA should be determined with a junction temperature of 125°C. This is the point where
distortion starts to substantially increase. Thermal management of the final PCB should strive to keep the junction temperature at or
below 125°C for best performance and long-term reliability. See Section 10.3 for more information.
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(4)
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Test boards conditions:
a.
b.
c.
d.
2 in x 1.85 in, 4 layers, thickness: 0.062 inch (1.57 mm).
2 oz. copper traces located on the top of the PCB
2 oz. copper ground planes on the 2 internal layers and bottom layer
4 thermal vias (10mil) located under the device package
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6.5 Electrical Characteristics
TJ = –40°C to 125°C, VIN = 5.5 V - 36 V (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
3
4.4
mA
18
50
μA
Start threshold voltage, UVLO
5.3
5.5
Hysteresis voltage, UVLO
330
SUPPLY VOLTAGE (VIN PIN)
IQ
Quiescent current
VSENSE = 2 V, Not switching,
PH pin open
Shutdown, ENA = 0 V
UNDERVOLTAGE LOCK OUT (UVLO)
V
mV
VOLTAGE REFERENCE
Voltage reference accuracy
TJ = 25°C
1.202
1.221
1.239
IO = 0 A – 5 A
1.196
1.221
1.245
V
OSCILLATOR
Internally set free-running frequency
400
Minimum controllable on time
Maximum duty cycle
87%
500
600
kHz
150
200
ns
1.3
V
89%
ENABLE (ENA PIN)
Start threshold voltage, ENA
Stop threshold voltage, ENA
0.5
Hysteresis voltage, ENA
V
450
Internal slow-start time (0~100%)
mV
6.6
8
10
ms
Current limit
6.0
7.5
9.0
A
Current limit hiccup time
13
16
20
ms
135
162
°C
14
°C
CURRENT LIMIT
THERMAL SHUTDOWN
Thermal shutdown trip point
Thermal shutdown hysteresis
OUTPUT MOSFET
rDS(on)
6
High-side power MOSFET switch
VIN = 5.5 V
150
110
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230
mΩ
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6.6 Typical Characteristics
530
3.5
VI = 12 V
I Q−Quiescent Current −mA
f − Oscillator Frequency − kHz
520
510
500
490
480
3.25
3
2.75
470
460
−50
−25
0
25
50
75
100
2.5
−50
125
−25
T − Junction Temperature − °C
Figure 6-1. Oscillator Frequency vs Junction
Temperature
T J = 125°C
15
T J = 27°C
T J = –40°C
10
5
0
5
10
15
20
25
30
35
100
125
1.220
1.215
1.210
-50
40
Figure 6-3. Shutdown Quiescent Current vs Input
Voltage
-25
0
25
50
75
100
TJ - Junction Temperature - °C
125
Figure 6-4. Voltage Reference vs Junction
Temperature
9
180
TSS − Internal Slow Start Time − ms
V I = 12 V
160
150
140
130
120
110
r
DS(on) −On Resistance −mΩ
75
1.225
V I −Input V oltage −V
170
50
1.230
ENA = 0 V
20
25
Figure 6-2. Non-Switching Quiescent Current vs
Junction Temperature
VREF - Voltage Reference - V
I SD −Shutdown Current
−µ A
25
0
T J −Junction T emperature − °C
100
8.5
8
7.5
90
80
−50
−25
0
25
50
75
100
T J −Junction Temperature − °C
125
Figure 6-5. On Resistance vs Junction
Temperature
7
−50
−25
0
25
50
75
100
TJ − Junction Temperature − °C
125
Figure 6-6. Internal Slow Start Time vs Junction
Temperature
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8
170
7.75
Minimum Duty Ratio - %
Minimum Controllable On Time − ns
180
160
150
140
7.25
130
120
−50
−25
0
25
50
75
100
TJ − Junction Temperature − °C
125
Figure 6-7. Minimum Controllable On Time vs
Junction Temperature
8
7.50
7
-50
-25
50
0
25
75
100
TJ - Junction Temperature - °C
125
Figure 6-8. Minimum Controllable Duty Ratio vs
Junction Temperature
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7 Detailed Description
7.1 Overview
The TPS5450 device is a 36-V, 5-A, step-down (buck) regulator with an integrated high-side n-channel MOSFET.
The device implements constant-frequency voltage-mode control with voltage feed forward for improved
line regulation and line transient response. Internal compensation reduces design complexity and external
component count.
The integrated 110-mΩ high-side MOSFET supports high-efficiency power-supply designs capable of delivering
5-A of continuous current to a load. The gate-drive bias voltage for the integrated high-side MOSFET is supplied
by a bootstrap capacitor connected from the BOOT to PH pins. The TPS5450 device reduces the external
component count by integrating the bootstrap recharge diode.
The TPS5450 device has a default input start-up voltage of 5.3 V typical. The ENA pin can be used to disable
the TPS5450 reducing the supply current to 18 µA. An internal pullup current source enables operation when
the EN pin is floating. The TPS5450 includes an internal slow-start circuit that slows the output rise time during
start-up to reduce in rush current and output voltage overshoot.
The minimum output voltage is the internal 1.221-V feedback reference. Output overvoltage transients are
minimized by an Overvoltage Protection (OVP) comparator. When the OVP comparator is activated, the highside MOSFET is turned off and remains off until the output voltage is less than 112.5% of the desired output
voltage.
Internal cycle-by-cycle overcurrent protection limits the peak current in the integrated high-side MOSFET. For
continuous overcurrent fault conditions the TPS5450 will enter hiccup mode overcurrent limiting. Thermal
protection protects the device from overheating.
7.2 Functional Block Diagram
VIN
VIN
1.221 V Bandgap
Reference
UVLO
VREF
SHDN
Slow Start
Boot
Regulator
BOOT
HICCUP
5 µA
ENA
ENABLE
SHDN
SHDN
VSENSE
Z1
Thermal
Protection
NC
SHDN
VIN
Ramp
Generator
NC
SHDN
VSENSE
OVP
Z2
Feed Forward
Gain = 25
PWM
Comparator
SHDN
GND
POWERPAD
Error
Amplifier
SHDN
HICCUP
Overcurrent
Protection
Oscillator
SHDN
Gate Drive
Control
112.5% VREF
Gate
Driver
SHDN
BOOT
PH
VOUT
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7.3 Feature Description
7.3.1 Oscillator Frequency
The internal free running oscillator sets the PWM switching frequency at 500 kHz. The 500-kHz switching
frequency allows less output inductance for the same output ripple requirement resulting in a smaller output
inductor.
7.3.2 Voltage Reference
The voltage reference system produces a precision reference signal by scaling the output of a temperature
stable band-gap circuit. The band-gap and scaling circuits are trimmed during production testing to an output of
1.221 V at room temperature.
7.3.3 Enable (ENA) and Internal Slow-Start
The ENA pin provides electrical on and off control of the regulator. Once the ENA pin voltage exceeds the
threshold voltage, the regulator starts operation and the internal slow-start begins to ramp. If the ENA pin voltage
is pulled below the threshold voltage, the regulator stops switching and the internal slow-start resets. Connecting
the pin-to-ground or to any voltage less than 0.5 V will disable the regulator and activate the shutdown mode.
The quiescent current of the TPS5450 in shutdown mode is typically 18 μA.
The ENA pin has an internal pullup current source, allowing the user to float the ENA pin. If an application
requires controlling the ENA pin, use open-drain or open-collector output logic to interface with the pin. To limit
the start-up inrush current, an internal slow-start circuit is used to ramp up the reference voltage from 0 V to its
final value, linearly. The internal slow-start time is 8 ms typically.
7.3.4 Undervoltage Lockout (UVLO)
The TPS5450 incorporates an UVLO circuit to keep the device disabled when VIN (the input voltage) is below
the UVLO start voltage threshold. During power-up, internal circuits are held inactive and the internal slow-start
is grounded until VIN exceeds the UVLO start threshold voltage. Once the UVLO start threshold voltage is
reached, the internal slow-start is released and device start-up begins. The device operates until VIN falls below
the UVLO stop threshold voltage. The typical hysteresis in the UVLO comparator is 330 mV.
7.3.5 Boost Capacitor (BOOT)
Connect a 0.01-μF, low-ESR ceramic capacitor between the BOOT pin and PH pin. This capacitor provides the
gate-drive voltage for the high-side MOSFET. X7R or X5R grade dielectrics are recommended due to their stable
values over temperature.
7.3.6 Output Feedback (VSENSE) and Internal Compensation
The output voltage of the regulator is set by feeding back the center point voltage of an external resistor divider
network to the VSENSE pin. In steady-state operation, the VSENSE pin voltage should be equal to the voltage
reference 1.221 V.
The TPS5450 implements internal compensation to simplify the regulator design. Since the TPS5450 uses
voltage mode control, a type 3 compensation network has been designed on chip to provide a high crossover
frequency and a high phase margin for good stability. See the Section 8.2.2.9.2 for more details.
7.3.7 Voltage Feed-Forward
The internal voltage feed-forward provides a constant DC power stage gain despite any variations with the input
voltage. This greatly simplifies the stability analysis and improves the transient response. Voltage feed-forward
varies the peak ramp voltage inversely with the input voltage so that the modulator and power stage gain are
constant at the feed-forward gain, that is.
Feed Forward Gain =
VIN
Ramp
pk – pk
(1)
The typical feed-forward gain of TPS5450 is 25.
10
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7.3.8 Pulse-Width-Modulation (PWM) Control
The regulator employs a fixed frequency pulse-width-modulator (PWM) control method. First, the feedback
voltage (VSENSE pin voltage) is compared to the constant voltage reference by the high-gain error amplifier and
compensation network to produce a error voltage. Then, the error voltage is compared to the ramp voltage by
the PWM comparator. In this way, the error voltage magnitude is converted to a pulse width which is the duty
cycle. Finally, the PWM output is fed into the gate-drive circuit to control the on-time of the high-side MOSFET.
7.3.9 Overcurrent Limiting
Overcurrent limiting is implemented by sensing the drain-to-source voltage across the high-side MOSFET. The
drain to source voltage is then compared to a voltage level representing the overcurrent threshold limit. If the
drain-to-source voltage exceeds the overcurrent threshold limit, the overcurrent indicator is set true. The system
will ignore the overcurrent indicator for the leading edge blanking time at the beginning of each cycle to avoid
any turn-on noise glitches.
Once overcurrent indicator is set true, overcurrent limiting is triggered. The high-side MOSFET is turned off for
the rest of the cycle after a propagation delay. The overcurrent limiting mode is called cycle-by-cycle current
limiting.
Sometimes under serious overload conditions such as short-circuit, the overcurrent runaway may still happen
when using cycle-by-cycle current limiting. A second mode of current limiting is used, that is, hiccup mode
overcurrent limiting. During hiccup mode overcurrent limiting, the voltage reference is grounded and the highside MOSFET is turned off for the hiccup time. Once the hiccup time duration is complete, the regulator restarts
under control of the slow-start circuit.
7.3.10 Overvoltage Protection
The TPS5450 has an overvoltage protection (OVP) circuit to minimize voltage overshoot when recovering from
output fault conditions. The OVP circuit includes an overvoltage comparator to compare the VSENSE pin voltage
and a threshold of 112.5% x VREF. Once the VSENSE pin voltage is higher than the threshold, the high-side
MOSFET will be forced off. When the VSENSE pin voltage drops lower than the threshold, the high-side
MOSFET will be enabled again.
7.3.11 Thermal Shutdown
The TPS5450 protects itself from overheating with an internal thermal shutdown circuit. If the junction
temperature exceeds the thermal shutdown trip point, the voltage reference is grounded and the high-side
MOSFET is turned off. The part is restarted under control of the slow-start circuit automatically when the junction
temperature drops 14°C below the thermal shutdown trip point.
7.4 Device Functional Modes
7.4.1 Operation near Minimum Input Voltage
The device is recommended to operate with input voltages above 5.5 V. The typical VIN UVLO threshold is
5.3 V and the device may operate at input voltages down to the UVLO voltage. At input voltages below the
actual UVLO voltage the device will not switch. If EN is floating or externally pulled up to greater up than 1.3 V,
when V(VIN) passes the UVLO threshold the device will become active. Switching is enabled and the slow-start
sequence is initiated. The TPS5450 device starts linearly ramping up the internal reference voltage from 0 V to
its final value over the internal slow-start time.
7.4.2 Operation With ENA Control
The enable start threshold voltage is 1.3 V maximum. With ENA held below the 0.5-V minimum stop threshold
voltage the device is disabled and switching is inhibited even if VIN is above its UVLO threshold. The IC
quiescent current is reduced in this state. If he EN voltage is increased above the threshold while VIN is above
its UVLO threshold, the device becomes active. Switching is enabled and the slow-start sequence is initiated.
The TPS5450 device starts linearly ramping up the internal reference voltage from 0 V to its final value over the
internal slow-start time.
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8 Application and Implementation
Note
Information in the following applications sections is not part of the TI component specification,
and TI does not warrant its accuracy or completeness. TI’s customers are responsible for
determining suitability of components for their purposes, as well as validating and testing their design
implementation to confirm system functionality.
8.1 Application Information
The TPS5450 device is a 36-V, 5-A, step-down regulator with an integrated high-side MOSFET. This device
is typically used to convert a higher DC voltage to a lower DC voltage with a maximum available output
current of 5 A. Example applications are: High Density Point-of-Load Regulators, LCD and Plasma Displays,
Battery Chargers, and 12-V and 24-V Distributed Power Systems. Use the following design procedure to select
component values for the TPS5450 device. This procedure illustrates the design of a high-frequency switching
regulator.
8.2 Typical Application
Figure 8-1 shows the schematic for a typical TPS5450 application. The TPS5450 can provide up to 5-A output
current at a nominal output voltage of 5 V. For proper thermal performance, the exposed PowerPAD underneath
the device must be soldered down to the printed-circuit board.
Figure 8-1. Application Circuit, 12 V to 5.0 V
8.2.1 Design Requirements
To begin the design process a few parameters must be decided upon. These requirements are typically
determined at the system levels. This example is designed to the following known parameters:
Table 8-1. Design Parameters
(1)
12
DESIGN PARAMETER(1)
EXAMPLE VALUE
Input voltage range
10 V to 31 V
Output voltage
5V
Input ripple voltage
400 mV
Output ripple voltage
30 mV
Output current rating
5A
Operating frequency
500 kHz
As an additional constraint, the design is set up to be small size and low component height.
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8.2.2 Detailed Design Procedure
The following design procedure can be used to select component values for the TPS5450. Alternately, use
the WEBENCH software to generate a complete design. The WEBENCH software uses an iterative design
procedure and accesses a comprehensive database of components when generating a design. This section
presents a simplified discussion of the design process.
8.2.2.1 Switching Frequency
The switching frequency for the TPS5450 is internally set to 500 kHz. It is not possible to adjust the switching
frequency.
8.2.2.2 Output Voltage Setpoint
The output voltage of the TPS5450 is set by a resistor divider (R1 and R2) from the output to the VSENSE pin.
Calculate the R2 resistor value for the output voltage of 5 V using Equation 2:
(2)
For any TPS5450 design, start with an R1 value of 10 kΩ. For an output voltage closest to but at least 5 V, R2 is
3.16 kΩ.
8.2.2.3 Input Capacitors
The TPS5450 requires an input decoupling capacitor and, depending on the application, a bulk input capacitor.
The minimum recommended decoupling capacitance is 4.7 μF. A high-quality ceramic type X5R or X7R is
required. For some applications, a smaller value decoupling capacitor may be used, so long as the input voltage
and current ripple ratings are not exceeded. The voltage rating must be greater than the maximum input voltage,
including ripple.
This input ripple voltage can be approximated by Equation 3:
ΔVIN =
I OUT(MAX) × 0.25
C BULK ×
ƒsw
= I OUT(MAX) × ESR MAX
(3)
where
•
•
•
•
IOUT(MAX) is the maximum load current
f SW is the switching frequency
CIN is the input capacitor value
ESRMAX is the maximum series resistance of the input capacitor
For this design, the input capacitance consists of two 4.7-μF capacitors, C1 and C4, in parallel. An additional
high frequency bypass capacitor, C5 is also used.
The maximum RMS ripple current also needs to be checked. For worst case conditions, this can be
approximated by Equation 4:
I
I
CIN
=
OUT(MAX)
2
(4)
In this case the input ripple voltage would be 281 mV and the RMS ripple current would be 2.5 A. The maximum
voltage across the input capacitors would be VIN max plus delta VIN/2. The chosen input decoupling capacitor
is rated for 50 V and the ripple current capacity is greater than 2.5 A each, providing ample margin. It is very
important that the maximum ratings for voltage and current are not exceeded under any circumstance.
Additionally some bulk capacitance may be needed, especially if the TPS5450 circuit is not located within about
2 inches from the input voltage source. The value for this capacitor is not critical but it also should be rated to
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handle the maximum input voltage including ripple voltage and should filter the output so that input ripple voltage
is acceptable.
8.2.2.4 Output Filter Components
Two components need to be selected for the output filter, L1 and C2. Since the TPS5450 is an internally
compensated device, a limited range of filter component types and values can be supported.
8.2.2.5 Inductor Selection
To calculate the minimum value of the output inductor, use Equation 5:
(
MIN
=
)
– V
OUT(MAX) × VIN(MAX)
OUT
V
×K
×I
×F
IN(MAX)
IND
OUT
SW(MIN)
V
L
(5)
KIND is a coefficient that represents the amount of inductor ripple current relative to the maximum output current.
Three things need to be considered when determining the amount of ripple current in the inductor: the peak
to peak ripple current affects the output ripple voltage amplitude, the ripple current affects the peak switch
current and the amount of ripple current determines at what point the circuit becomes discontinuous. For designs
using the TPS5450, KIND of 0.2 to 0.3 yields good results. Low-output ripple voltages can be obtained when
paired with the proper output capacitor, the peak switch current will be well below the current limit set point and
relatively low-load currents can be sourced before discontinuous operation.
For this design example use KIND = 0.2 and the minimum inductor value is calculated to be 10.4 μH. A higher
standard value is 15 μH, which is used in this design.
For the output filter inductor it is important that the RMS current and saturation current ratings not be exceeded.
The RMS inductor current can be found from Equation 6:
I
L(RMS)
+
Ǹ
ǒ
1
I2
)
OUT(MAX) 12
V
V
OUT
ǒVIN(MAX) * VOUTǓ
L
IN(MAX)
OUT
F
Ǔ
2
SW(MIN)
(6)
The peak inductor current can be determined with Equation 7:
V
I L(PK) + I
OUT(MAX)
)
1.6
OUT
ǒVIN(MAX) * VOUTǓ
V IN(MAX)
L
OUT
F
(7)
SW(MIN)
For this design, the RMS inductor current is 5.004 A, and the peak inductor current is 5.34 A. The chosen
inductor is a Sumida CDRH1127/LD-150 15μH. It has a minimum rated current of 5.65 A for both saturation and
RMS current. In general, inductor values for use with the TPS5450 are in the range of 10 μH to 100 μH.
8.2.2.6 Capacitor Selection
The important design factors for the output capacitor are DC voltage rating, ripple current rating, and equivalent
series resistance (ESR). The DC voltage and ripple current ratings cannot be exceeded. The ESR is important
because along with the inductor ripple current it determines the amount of output ripple voltage. The actual value
of the output capacitor is not critical, but some practical limits do exist. Consider the relationship between the
desired closed-loop crossover frequency of the design and LC corner frequency of the output filter. Due to the
design of the internal compensation, it is desirable to keep the closed-loop crossover frequency in the range 3
kHz to 30 kHz as this frequency range has adequate phase boost to allow for stable operation. For this design
example, it is assumed that the intended closed-loop crossover frequency will be between 2590 Hz and 24
kHz and also below the ESR zero of the output capacitor. Under these conditions the closed-loop crossover
frequency is related to the LC corner frequency by:
14
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f CO +
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f LC
2
85 VOUT
(8)
And the desired output capacitor value for the output filter to:
C OUT +
1
3357
L OUT
f CO
V OUT
(9)
For a desired crossover of 12 kHz and a 15-μH inductor, the calculated value for the output capacitor is 330
μF. The capacitor type should be chosen so that the ESR zero is above the loop crossover. The maximum ESR
should be:
ESR MAX +
2p
1
C OUT
f CO
(10)
The maximum ESR of the output capacitor also determines the amount of output ripple as specified in the initial
design parameters. The output ripple voltage is the inductor ripple current times the ESR of the output filter.
Check that the maximum specified ESR as listed in the capacitor data sheet results in an acceptable output
ripple voltage:
VPP (MAX) =
ESRMAX x VOUT x
( VIN(MAX)
- VOUT
)
NC x VIN(MAX) x LOUT x FSW
(11)
where
•
•
•
ΔVPP is the desired peak-to-peak output ripple.
NC is the number of parallel output capacitors.
FSW is the switching frequency.
For this design example, a single 330-μF output capacitor is chosen for C3. The calculated RMS ripple current
is 143 mA and the maximum ESR required is 40 mΩ. A capacitor that meets these requirements is a Sanyo
Poscap 10TPB330M, rated at 10 V with a maximum ESR of 35 mΩ and a ripple current rating of 3 A. An
additional small 0.1-μF ceramic bypass capacitor, C6 is also used in this design.
The minimum ESR of the output capacitor should also be considered. For good phase margin, the ESR zero
when the ESR is at a minimum should not be too far above the internal compensation poles at 24 kHz and 54
kHz.
The selected output capacitor must also be rated for a voltage greater than the desired output voltage plus one
half the ripple voltage. Any derating amount must also be included. The maximum RMS ripple current in the
output capacitor is given by Equation 12:
(
V
ICOUT(RMS) =
1
×
√12
V
OUT ×
IN(MAX)
( VIN(MAX) –
×L
V
)
)
OUT
×F
×N
OUT
SW
C
(12)
where
•
•
NC is the number of output capacitors in parallel.
FSW is the switching frequency.
Other capacitor types can be used with the TPS5450, depending on the needs of the application.
8.2.2.7 Boot Capacitor
The boot capacitor should be 0.01 μF.
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8.2.2.8 Catch Diode
The TPS5450 is designed to operate using an external catch diode between PH and GND. The selected diode
must meet the absolute maximum ratings for the application: Reverse voltage must be higher than the maximum
voltage at the PH pin, which is VINMAX + 0.5 V. Peak current must be greater than IOUTMAX plus on half the peak
to peak inductor current. Forward voltage drop should be small for higher efficiencies. It is important to note that
the catch diode conduction time is typically longer than the high-side FET on-time, so attention paid to diode
parameters can make a marked improvement in overall efficiency. Additionally, check that the device chosen is
capable of dissipating the power losses. For this design, a Diodes, Inc. B540A is chosen, with a reverse voltage
of 40 V, forward current of 5 A, and a forward voltage drop of 0.5 V.
8.2.2.9 Advanced Information
8.2.2.9.1 Output Voltage Limitations
Due to the internal design of the TPS5450, there are both upper and lower output voltage limits for any given
input voltage. The upper limit of the output voltage set point is constrained by the maximum duty cycle of 87%
and is given by:
V OUTMAX + 0.87
ǒǒVINMIN * I OMAX
Ǔ
Ǔ ǒ
0.230 ) VD * I OMAX
Ǔ
RL * VD
(13)
where
•
•
•
•
VINMIN = minimum input voltage
IOMAX = maximum load current
VD = catch diode forward voltage.
RL= output inductor series resistance.
This equation assumes maximum on resistance for the internal high-side FET.
The lower limit is constrained by the minimum controllable on time which may be as high as 200 ns. The
approximate minimum output voltage for a given input voltage and minimum load current is given by:
V OUTMIN + 0.12
ǒǒVINMAX * I OMIN
Ǔ
Ǔ ǒ
0.110 ) VD * I OMIN
Ǔ
RL * VD
(14)
where
•
•
•
•
VINMAX = maximum input voltage
IOMIN = minimum load current
VD = catch diode forward voltage.
RL= output inductor series resistance.
This equation assumes nominal on resistance for the high-side FET and accounts for worst case variation of
operating frequency set point. Any design operating near the operational limits of the device should be carefully
checked to assure proper functionality.
8.2.2.9.2 Internal Compensation Network
The design equations given in the example circuit can be used to generate circuits using the TPS5450. These
designs are based on certain assumptions and will tend to always select output capacitors within a limited
range of ESR values. If a different capacitor type is desired, it may be possible to fit one to the internal
compensation of the TPS5450. Equation 15 gives the nominal frequency response of the internal voltage-mode
type III compensation network:
H(s) +
16
ǒ1 ) 2p
Ǔ ǒ1 ) 2p
s
Fz1
Ǔ
s
Fz2
ǒ2p sFp0Ǔ ǒ1 ) 2p sFp1Ǔ ǒ1 ) 2p sFp2Ǔ ǒ1 ) 2p sFp3Ǔ
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where
•
•
•
Fp0 = 2165 Hz, Fz1 = 2170 Hz, Fz2 = 2590 Hz
Fp1 = 24 kHz, Fp2 = 54 kHz, Fp3 = 440 kHz
Fp3 represents the non-ideal parasitics effect.
Using this information along with the desired output voltage, feed-forward gain and output filter characteristics,
the closed-loop transfer function can be derived.
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8.2.3 Application Curves
The performance graphs (Figure 8-2 through Figure 8-8) are applicable to the circuit in Figure 8-1. TA = 25°C.
unless otherwise specified.
0.3
100
0.2
VI = 12 V
95
Output Regulation - %
Efficiency - %
VI = 15 V
90
VI = 24 V
85
VI = 28 V
80
0.1
0
-0.1
-0.2
-0.3
0
75
0
1
2
3
4
IO - Output Current - A
5
6
Figure 8-2. Efficiency vs. Output Current
0.5
1
1.5
2 2.5
3 3.5
IO - Output Current - A
4
4.5
5
Figure 8-3. Output Regulation % vs. Output
Current
0.3
VI = 200 mV/Div (AC Coupled)
Output Regulation - %
0.2
IO = 0 A
0.1
IO = 5 A
0
PH = 10 V/Div
-0.1
IO = 2.5 A
-0.2
-0.3
10
13
16
19
22
25
VI - Input Voltage - V
28
31
t - Time - 1 ms/Div
Figure 8-4. Output Regulation % vs. Input Voltage
Figure 8-5. Input Voltage Ripple and PH Node, IO =
5 A.
VOUT = 50 mV/div (AC Coupled, 20 MHz BWL)
VOUT = 50 mV/div (AC Coupled, 20 MHz BWL)
VPH = 10 V/div
18
IOUT = 1 A/div
t - Time = 1 ms/div
t - Time = 100 ms/div
Figure 8-6. Output Voltage Ripple and PH Node, IO
=5A
Figure 8-7. Transient Response, IO Step 1.25 to
3.75 A.
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TJ - Junction Temperature - °C
125
100
75
50
25
0
0.5
1
1.5
2
2.5
IC Power Dissipation - W
3
3.5
Figure 8-8. TPS5450 Power Dissipation vs Junction Temperature.
9 Power Supply Recommendations
The device is designed to operate from an input voltage supply range between 5.5 V and 36 V. This input supply
should be well regulated. If the input supply is located more than a few inches from the TPS5450 converter
additional bulk capacitance may be required in addition to the ceramic bypass capacitors. An electrolytic
capacitor with a value of 100 μF is a typical choice.
10 Layout
10.1 Layout Guidelines
Connect a low-ESR ceramic bypass capacitor to the VIN pin. Take care to minimize the loop area formed by the
bypass capacitor connections, the VIN pin, and the TPS5450 ground pin. The best way to do this is to extend the
top-side ground area from under the device adjacent to the VIN trace, and place the bypass capacitor as close
as possible to the VIN pin. The minimum recommended bypass capacitance is 4.7-μF ceramic with a X5R or
X7R dielectric.
There should be a ground area on the top layer directly underneath the IC, with an exposed area for connection
to the PowerPAD. Use vias to connect this ground area to any internal ground planes. Use additional vias at the
ground side of the input and output filter capacitors as well. The GND pin should be tied to the PCB ground by
connecting it to the ground area under the device as shown below.
The PH pin should be routed to the output inductor, catch diode and boot capacitor. Since the PH connection is
the switching node, the inductor should be located very close to the PH pin and the area of the PCB conductor
minimized to prevent excessive capacitive coupling. The catch diode should also be placed close to the device
to minimize the output current loop area. Connect the boot capacitor between the phase node and the BOOT
pin as shown. Keep the boot capacitor close to the IC and minimize the conductor trace lengths. The component
placements and connections shown work well, but other connection routings may also be effective.
Connect the output filter capacitor(s) as shown between the VOUT trace and GND. It is important to keep the
loop formed by the PH pin, Lout, Cout and GND as small as is practical.
Connect the VOUT trace to the VSENSE pin using the resistor divider network to set the output voltage. Do not
route this trace too close to the PH trace. Due to the size of the IC package and the device pin-out, the trace
may need to be routed under the output capacitor. Alternately, the routing may be done on an alternate layer if a
trace under the output capacitor is not desired.
If using the grounding scheme shown in Figure 10-1, use a via connection to a different layer to route to the ENA
pin.
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10.2 Layout Example
Feedback Trace
OUTPUT
INDUCTOR
BOOT
CAPACITOR
BOOT
RESISTOR
DIVIDER
EXPOSED
POWERPAD
AREA
PH
NC
VIN
NC
GND
VSENSE
ENA
INPUT
BYPASS
CAPACITOR
Vout
PH
Vin
OUTPUT
FILTER
CAPACITOR
CATCH
DIODE
TOPSIDE GROUND AREA
Route INPUT VOLTAGE
trace under the catch diode
and output capacitor
or on another layer
Signal VIA
Figure 10-1. Design Layout
0,45
1,27
2,15
3,10
2,49
5,75
Figure 10-2. TPS5450 Land Pattern
20
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10.3 Thermal Calculations
The following formulas show how to estimate the device power dissipation under continuous conduction mode
operations. They should not be used if the device is working at light loads in the discontinuous conduction mode.
Conduction Loss: Pcon = IOUT 2 x RDS(on) x VOUT/VIN
Switching Loss: Psw = VIN x IOUT x 0.01
Quiescent Current Loss: Pq = VIN x 0.01
Total Loss: Ptot = Pcon + Psw + Pq
Given TA → Estimated Junction Temperature: TJ = TA + Rth x Ptot
Given TJMAX = 125°C → Estimated Maximum Ambient Temperature: TAMAX = TJMAX – Rth x Ptot
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11 Device and Documentation Support
11.1 Device Support
11.1.1 Third-Party Products Disclaimer
TI'S PUBLICATION OF INFORMATION REGARDING THIRD-PARTY PRODUCTS OR SERVICES DOES NOT
CONSTITUTE AN ENDORSEMENT REGARDING THE SUITABILITY OF SUCH PRODUCTS OR SERVICES
OR A WARRANTY, REPRESENTATION OR ENDORSEMENT OF SUCH PRODUCTS OR SERVICES, EITHER
ALONE OR IN COMBINATION WITH ANY TI PRODUCT OR SERVICE.
11.2 Trademarks
PowerPAD™ is a trademark of Texas Instruments.
All trademarks are the property of their respective owners.
11.3 Electrostatic Discharge Caution
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled
with appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may
be more susceptible to damage because very small parametric changes could cause the device not to meet its published
specifications.
11.4 Glossary
TI Glossary
This glossary lists and explains terms, acronyms, and definitions.
12 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
22
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PACKAGE OPTION ADDENDUM
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11-Jul-2022
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
Samples
(4/5)
(6)
TPS5450DDA
ACTIVE SO PowerPAD
DDA
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 125
5450
Samples
TPS5450DDAG4
ACTIVE SO PowerPAD
DDA
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 125
5450
Samples
TPS5450DDAR
ACTIVE SO PowerPAD
DDA
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 125
5450
Samples
TPS5450DDARG4
ACTIVE SO PowerPAD
DDA
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 125
5450
Samples
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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