EVALUATION KIT AVAILABLE
MAX16831
High-Voltage, High-Power LED Driver with
Analog and PWM Dimming Control
General Description
The MAX16831 is a current-mode, high-brightness LED
(HBLED) driver designed to control two external
n-channel MOSFETs for the single-string LED current
regulation. The MAX16831 integrates all the building
blocks necessary to implement fixed-frequency HBLED
drivers with wide-range dimming control. The
MAX16831 is configurable to operate as a step-down
(buck), step-up (boost), or step-up/-down (buck-boost)
current regulator.
Current-mode control with leading-edge blanking simplifies control-loop design. Internal slope compensation
stabilizes the current loop when operating at duty cycles
above 50%. The MAX16831 operates over a wide input
voltage range and is capable of withstanding automotive load-dump events. Multiple MAX16831s can be
synchronized to each other or to an external clock. The
MAX16831 includes a floating dimming driver for
brightness control with an external n-channel MOSFET
in series with the LED string.
HBLED drivers using the MAX16831 achieve efficiencies of over 90% in automotive applications. The
MAX16831 also includes a 1.4A source and 2.5A sink
gate driver for driving switching MOSFETs in high-power
LED driver applications, such as front light assemblies.
The dimming control allows for wide PWM dimming at
frequencies up to 2kHz. Higher dimming ratios of up to
1000:1 are achievable at lower dimming frequencies.
The MAX16831 is available in a 32-pin thin QFN package
with exposed pad and operates over the -40°C to
+125°C automotive temperature range.
Features
o Wide Input Range: 6V to 76V With Cold-Start
Operation to 5.5V
o Integrated Differential LED Current-Sense
Amplifier
o Floating Dimming Driver Capable of Driving an
n-Channel MOSFET
o 5% LED Current Accuracy
o 200Hz On-Board Ramp Syncs to External PWM
Dimming Signal
o Programmable Switching Frequency (125kHz to
600kHz) and Synchronization
o Output Overvoltage Load Dump, LED Short,
Overtemperature Protection
o Low 107mV LED Current Sense for High
Efficiency
o Enable/Shutdown Input with Shutdown Current
Below 45µA
Ordering Information
TEMP RANGE
PIN-PACKAGE
MAX16831ATJ+
PART
-40°C to +125°C
32 TQFN-EP*
MAX16831ATJ/V+
-40°C to +125°C
32 TQFN-EP*
+Denotes a lead(Pb)-free/RoHS-compliant package.
*EP = Exposed pad.
/V denotes an automotive qualified part.
Typical Operating Circuits
Applications
Automotive Exterior Lighting:
High-Beam/Low-Beam/Signal Lights
Rear Combination Lights (RCL)
Daytime Running Lights (DRL)
Fog Light and Adaptive Front Light Assemblies
BUCK-BOOST CONFIGURATION
VIN
LO
VCC
RUV1
CS-
CS+
DGT
QS
RD
DRV
LEDs
SNS+
RSENSE
DIM
Emergency Lighting
DIM
SNSQGND
MAX16831
REG1
Navigation and Marine Indicators
CLMP
UVEN
CUVEN
Industrial and Architectural Lighting
Projectors with RGB LED Light Sources
RCS
CCLMP
RUV2
CREG1
HI
RT
CF
RTSYNC
ROV1
OV
COMP
CS
FB
AGND
SGND
REG2
R1
DRI
ROV2
CREG2
C2
R2
C1
Pin Configuration appears at end of data sheet.
Typical Operating Circuits continued at end of data sheet.
For pricing, delivery, and ordering information, please contact Maxim Direct
at 1-888-629-4642, or visit Maxim’s website at www.maximintegrated.com.
19-0809; Rev 2; 1/13
MAX16831
High-Voltage, High-Power LED Driver with
Analog and PWM Dimming Control
ABSOLUTE MAXIMUM RATINGS
VCC, HI, LO, CLMP to QGND .................................-0.3V to +80V
CS+, CS-, DGT, UVEN to QGND............................-0.3V to +80V
UVEN to QGND ..........................................-0.3V to (VCC + 0.3V)
DRV to SGND .........................................................-0.3V to +18V
DRI, REG2, DIM to AGND ......................................-0.3V to +18V
QGND, SGND to AGND ........................................-0.3V to +0.3V
SNS+ to SNS- ...........................................................-0.3V to +6V
CS, FB, COMP, SNS+, SNS-, OV, REF,
RTSYNC to AGND .................................................-0.3V to +6V
REG1, CLKOUT to AGND ........................................-0.3V to +6V
CS+ to CS- .............................................................-0.3V to +12V
HI to LO ..................................................................-0.3V to +36V
CS+, CS-, DGT, CLMP to LO .................................-0.3V to +12V
CS+, CS-, DGT, CLMP to LO ........................-0.3V to (HI + 0.3V)
HI to CLMP .............................................................-0.3V to +28V
Continuous Power Dissipation* (TA = +70°C)
32-Pin TQFN (derate 34.5mW/°C above +70°C) ........2758mW
Thermal Resistance*
θJA .................................................................................29°C/W
θJC ................................................................................1.7°C/W
Operating Temperature Range .........................-40°C to +125°C
Maximum Junction Temperature .....................................+150°C
Storage Temperature Range .............................-60°C to +150°C
Reflow Temperature.........................................................+240°C
Lead Temperature (soldering, 10s) .................................+300°C
*As per JEDEC 51 standard, multilayer board (PCB).
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional
operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device reliability.
ELECTRICAL CHARACTERISTICS
(VCC = VUVEN = 14V, CREG1 = 1µF, CREG2 = 1µF, CCLMP = 0.1µF, RT = 25kΩ, TA = TJ = -40°C to +125°C, unless otherwise noted.
Typical specifications are at TA = +25°C.)
PARAMETER
Input Voltage Range
Supply Current
Shutdown Current
SYMBOL
CONDITIONS
VCC
MIN
TYP
UNITS
76.0
V
IREG2 = 0A
2.7
4.5
mA
ISHDN
VUVEN ≤ 0.8V
25
45
µA
VCC_R
VCC rising
5.5
6.0
VCC_F
VCC falling
5.0
5.5
IQ
5.5
MAX
UVEN
VCC UVLO Threshold
VCC Threshold Hysteresis
UVEN Threshold
UVEN Input Current
VCC_HYS
0.4
V
V
VUVR
VUVEN rising
1.100
1.244
1.360
VUVF
VUVEN falling
1.000
1.145
1.260
IUVEN
VUVEN = 0V and VUVEN = 76V, VCC = 77V
-0.2
0 ≤ IREG1 ≤ 2mA, 7.5V ≤ VCC ≤ 76V
4.75
5.00
5.25
IREG1 = 2mA, VCC = 5.7V
4.00
4.50
5.25
0.5
1.0
V
25
Ω
+0.2
V
µA
REGULATORS
REG1 Regulator Output
VREG1
REG1 Dropout Voltage
IREG1 = 2mA (Note 1)
REG1 Load Regulation
∆V/∆I
REG2 Regulator Output
VREG2
REG2 Dropout Voltage
REG2 Load Regulation
VCC = 7.5V, 0 ≤ IREG1 ≤ 2mA
7.5V ≤ VCC ≤ 76V, IREG2 = 1mA
6.65
7.00
VCC = 5.7V, 0 ≤ IREG2 ≤ 20mA
4.5
5.0
IREG2 = 20mA (Note 1)
∆V/∆I
7.35
0.5
VCC = 7.5V, 0 ≤ IREG2 ≤ 20mA
V
V
V
25
Ω
3.0
V
HIGH-SIDE REGULATOR (CLMP) (All Voltages Referred to LO) (Note 2)
CLMP UVLO Threshold
VCLMPTH
CLMP UVLO Threshold
Hysteresis
VCLMPHYS
2
VCLMP rising
2.0
2.5
0.22
V
Maxim Integrated
MAX16831
High-Voltage, High-Power LED Driver with
Analog and PWM Dimming Control
ELECTRICAL CHARACTERISTICS (continued)
(VCC = VUVEN = 14V, CREG1 = 1µF, CREG2 = 1µF, CCLMP = 0.1µF, RT = 25kΩ, TA = TJ = -40°C to +125°C, unless otherwise noted.
Typical specifications are at TA = +25°C.)
PARAMETER
SYMBOL
CONDITIONS
8.7V ≤ (VHI - VLO) ≤ 36V, ICLMP = 1mA
CLMP Regulator Output Voltage
VCLMP
MIN
5.5
TYP
MAX
8.0
10.0
V
(VHI - VLO)
- 0.7
5.0V ≤ (VHI - VLO) ≤ 8.7V, ICLMP = 250µA
UNITS
CURRENT-SENSE AMPLIFIER (CSA)
Differential Input Voltage Range
VCS+ VCS-
0
Common-Mode Range
CS+ Input Bias Current
ICS+
VCS+ - VCS- = 0.3V
CS- Input Bias Current
ICS-
VCS+ - VCS- = 0.3V
Unity-Gain Bandwidth
0.3
V
0
VCC
V
-250
+250
µA
400
µA
From (CS+ - CS-) to CS
1.0
MHz
REF OUTPUT BUFFER
REF Output Voltage
VREF
-100µA ≤ IREF ≤ +100µA
2.85
3.00
VCLMP - VLO = 4V
5
20
VCLMP - VLO = 8V
30
67
VCLMP - VLO = 4V
10
22
VCLMP - VLO = 8V
40
76
DRI rising
4.0
3.15
V
DIM DRIVER
Source Current
Sink Current
mA
mA
GATE DRIVER
DRI UVLO Threshold
DRI UVLO Threshold Hysteresis
Driver Output Impedance
VUVLO_TH
VUVLO_HYST
4.2
4.4
0.3
V
V
ZOUT_L
VDRI = 7.0V, DRV sinking 250mA
2.8
4
ZOUT_H
VDRI = 7.0V, DRV sourcing 250mA
5.0
8
Ω
Peak Sink Current
ISK
VDRI = 7.0V
2.5
A
Peak Source Current
ISR
VDRI = 7.0V
1.4
A
VCOMP - (VSNS+ - VSNS-)
0.7
V
PWM, ILIM, AND HICCUP COMPARATOR
PWM Comparator Offset Voltage
Peak Current-Limit Comparator
Trip Threshold
160
Peak Current-Limit Comparator
Propagation Delay (Excluding
Blanking Time)
50mV overdrive
HICCUP Comparator Trip
Threshold
200
240
40
235
300
mV
ns
385
mV
SNS+ Input Bias Current
VSNS+ = 0V, VSNS- = 0V
-100
-65
µA
SNS- Input Bias Current
VSNS+ = 0V, VSNS- = 0V
-100
-65
µA
40
ns
Blanking Time
Maxim Integrated
tBLNK
3
MAX16831
High-Voltage, High-Power LED Driver with
Analog and PWM Dimming Control
ELECTRICAL CHARACTERISTICS (continued)
(VCC = VUVEN = 14V, CREG1 = 1µF, CREG2 = 1µF, CCLMP = 0.1µF, RT = 25kΩ, TA = TJ = -40°C to +125°C, unless otherwise noted.
Typical specifications are at TA = +25°C.)
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
ERROR AMPLIFIER
FB Input Bias Current
-100
+100
nA
EAMP Output Sink Current
VFB = 1.735V, VCOMP = 1V
3
7
mA
EAMP Output Source Current
VFB = 0.735V, VCOMP = 1V
2
7
mA
EAMP Input Common-Mode
Voltage
0
EAMP Output Clamp Voltage
Voltage Gain
Unity-Gain Bandwidth
1.1
AV
GBW
3.0
1.7
2.4
V
V
RCOMP = 100kΩ to AGND
80
dB
RCOMP = 100kΩ to AGND, CCOMP = 100pF
to AGND
0.5
MHz
OSCILLATOR, OSC SYNC, CLK, AND CLKOUT
RTSYNC Frequency Range
fSWMIN
125
fSWMAX
RTSYNC Oscillator Frequency
RTSYNC High-Level Voltage
VSIHL
RTSYNC Low-Level Voltage
VSILL
500
RT = 25kΩ
475
500
525
RT = 100kΩ
106
125
143
2.8
kHz
kHz
V
0.4
CLKOUT High Level
ISINK = 0.8mA
CLKOUT Low Level
ISOURCE = 1.6mA
0.4
V
fSW = 500kHz
500
pF
240
Hz
2000
Hz
3.2
V
300
mV
CLKOUT Maximum Load
Capacitance
CCLK_CAP
2.8
V
V
DIM SYNC, DIM RAMP, AND DIM PWM GEN
Internal Ramp Frequency
fRAMP
160
External Sync Frequency Range
fDIM
80
External Sync Low-Level Voltage
VLTH
0.4
External Sync High-Level Voltage
VHTH
DIM Comparator Offset
VDIMOS
170
200
V
200
DIGITAL SOFT-START
Soft-Start Duration
tSS
4.0
ms
OVERVOLTAGE COMPARATOR, LOAD OVERCURRENT COMPARATOR
OVP Overvoltage Comparator
Threshold
VOV
OVP Overvoltage Comparator
Hysteresis
VOV_HYST
VOV rising
1.20
1.235
1.27
V
63.5
mV
SLOPE COMPENSATION
Slope Compensation Peak
Voltage Per Cycle
Clock generated by RT
120
mV
Slope Compensation
External clock applied to RTSYNC
15
mV/µs
4
Maxim Integrated
MAX16831
High-Voltage, High-Power LED Driver with
Analog and PWM Dimming Control
ELECTRICAL CHARACTERISTICS (continued)
(VCC = VUVEN = 14V, CREG1 = 1µF, CREG2 = 1µF, CCLMP = 0.1µF, RT = 25kΩ, TA = TJ = -40°C to +125°C, unless otherwise noted.
Typical specifications are at TA = +25°C.)
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
THERMAL SHUTDOWN
Thermal Shutdown Temperature
TSHDN
Temperature rising
∆TSHDN
Hysteresis
+165
°C
20
°C
Note 1: Dropout voltage is defined as the input to output differential voltage at which the regulator output voltage drops 100mV below
the nominal output voltage.
Note 2: VCLMPTH determines the voltage required to operate the current-sense amplifier. The DIM driver requires 2.5V for (VCLMP - VLO)
to drive the external MOSFET. VHI is typically one diode drop above VCLMP. A large capacitor connected to VCLMP slows the
response of the LED current-sense circuitry, resulting in current overshoot. To ensure proper operation, connect a 0.1µF
capacitor from CLMP to LO.
Typical Operating Characteristics
(VCC = VUVEN = 14V, CREG1 = 1µF, CREG2 = 10µF, CCLMP = 0.1µF, RT = 25kΩ, RCS = 0.1Ω, TA = +25°C, unless otherwise noted.)
25
24
23
22
21
2.9
2.8
2.7
2.6
2.5
2.4
2.3
20
2.2
19
2.1
18
DGT AND DRV NOT
SWITCHING
2.0
-60 -40 -20 0
20 40 60 80 100 120 140
TEMPERATURE (°C)
Maxim Integrated
-60 -40 -20 0
20 40 60 80 100 120 140
TEMPERATURE (°C)
120
MAX16831 toc03
26
3.0
MAX16831 toc02
SHUTDOWN CURRENT (µA)
27
OPERATING CURRENT (mA)
MAX16831 toc01
28
VOLTAGE ACROSS LED CURRENT-SENSE
RESISTOR vs. SUPPLY VOLTAGE
OPERATING CURRENT
vs. TEMPERATURE
110
VOLTAGE ACROSS RCS (mV)
SHUTDOWN CURRENT
vs. TEMPERATURE
100
90
80
70
60
50
40
30
20
10
0
0
10
20
30
40
SUPPLY VOLTAGE (V)
5
MAX16831
High-Voltage, High-Power LED Driver with
Analog and PWM Dimming Control
Typical Operating Characteristics (continued)
(VCC = VUVEN = 14V, CREG1 = 1µF, CREG2 = 10µF, CCLMP = 0.1µF, RT = 25kΩ, RCS = 0.1Ω, TA = +25°C, unless otherwise noted.)
REG2 OUTPUT VOLTAGE
vs. TEMPERATURE
6.95
6.90
6.85
5
4
3
2
6.80
4.94
IREG1 = 2mA
4.90
20 40 60 80 100 120 140
0
5
10
15
20
25
30
35
40
-60 -40 -20 0
20 40 60 80 100 120 140
SUPPLY VOLTAGE (V)
TEMPERATURE (°C)
REG1 OUTPUT VOLTAGE
vs. SUPPLY VOLTAGE
CLMP REGULATOR VOLTAGE
vs. TEMPERATURE
CLMP REGULATOR VOLTAGE
vs. (VHI - VLO)
CLMP VOLTAGE (V)
8.1
4
3
2
8.0
7.9
7.8
7.7
1
15
20
25
30
35
6
5
4
3
CLMP REGULATOR VOLTAGE = VCLMP - VLO
1
-60 -40 -20 0
40
7
2
7.5
0
10
8
VHI - VLO = 11V
CLMP VOLTAGE = VCLMP - VLO
7.6
IREG1 = 2mA
MAX16831 toc09
8.2
9
CLMP REGULATOR VOLTAGE (V)
MAX16831 toc08
8.3
MAX16831 toc07
5
20 40 60 80 100 120 140
0
5
10
15
20
25
35
TEMPERATURE (°C)
VHI - VLO (V)
REF OUTPUT VOLTAGE
vs. TEMPERATURE
REF OUTPUT VOLTAGE
vs. LOAD CURRENT
PWM OSCILLATOR FREQUENCY
vs. TEMPERATURE
3.00
2.99
2.98
3.00
2.99
2.98
520
510
500
490
480
IREF = 100µA
RT = 25kΩ
2.96
2.96
-60 -40 -20 0
20 40 60 80 100 120 140
TEMPERATURE (°C)
40
MAX16831 toc12
3.01
2.97
2.97
530
PWM FREQUENCY (kHz)
3.01
3.02
MAX16831 toc11
MAX16831 toc10
3.02
6
30
SUPPLY VOLTAGE (V)
REF OUTPUT VOLTAGE (V)
5
4.96
TEMPERATURE (°C)
6
0
4.98
IREG2 = 20mA
0
-60 -40 -20 0
5.00
4.92
1
IREG2 = 20mA
REG1 OUTPUT VOLTAGE (V)
6
MAX16831 toc06
7
5.02
REG1 OUTPUT VOLTAGE (V)
7.00
MAX16831 toc05
7.05
8
REG2 OUTPUT VOLTAGE (V)
MAX16831 toc04
REG2 OUTPUT VOLTAGE (V)
7.10
REF OUTPUT VOLTAGE (V)
REG1 OUTPUT VOLTAGE
vs. TEMPERATURE
REG2 OUTPUT VOLTAGE
vs. SUPPLY VOLTAGE
470
-225 -175 -125 -75 -25 25
75 125 175 225
LOAD CURRENT (µA)
-60 -40 -20 0
20 40 60 80 100 120 140
TEMPERATURE (°C)
Maxim Integrated
MAX16831
High-Voltage, High-Power LED Driver with
Analog and PWM Dimming Control
Typical Operating Characteristics (continued)
(VCC = VUVEN = 14V, CREG1 = 1µF, CREG2 = 10µF, CCLMP = 0.1µF, RT = 25kΩ, RCS = 0.1Ω, TA = +25°C, unless otherwise noted.)
PWM OSCILLATOR FREQUENCY
vs. 1/RT CONDUCTANCE
MAX16831 toc14
450
10% DIMMING
1A/div
0A
400
50% DIMMING
1A/div
0A
350
300
90% DIMMING
1A/div
0A
250
200
90
150
100
0.005
MAX16831 toc15
500
100
LED CURRENT DUTY CYCLE (%)
MAX16831 toc13
550
PWM FREQUENCY (kHz)
LED CURRENT DUTY CYCLE
vs. DIM VOLTAGE
200Hz DIMMING OPERATION
80
70
60
50
40
30
20
10
0
0.015
0.025
0.035
0.045
0
2ms/div
1
DRIVER (DRV) RISE TIME
3
DRIVER (DRV) FALL TIME
MAX16831 toc16
MAX16831 toc17
DRV OUTPUT
RISING
2V/div
DRV OUTPUT
FALLING
2V/div
0V
0V
5nF CAPACITOR CONNECTED
FROM DRV TO AGND
5nF CAPACITOR CONNECTED
FROM DRV TO AGND
40ns/div
40ns/div
DGT RISE TIME
DGT FALL TIME
MAX16831 toc18
MAX16831 toc19
DGT OUTPUT
RISING
14V OFFSET
2V/div
DGT OUTPUT
FALLING
14V OFFSET
2V/div
14V
VHI - VLO = 11V
1nF CAPACITOR CONNECTED
FROM DGT TO AGND
40ns/div
Maxim Integrated
2
DIM VOLTAGE (V)
1/RT (kΩ-1)
14V
VHI - VLO = 11V
1nF CAPACITOR CONNECTED
FROM DGT TO AGND
40ns/div
7
MAX16831
High-Voltage, High-Power LED Driver with
Analog and PWM Dimming Control
Pin Description
8
PIN
NAME
1, 24
N.C.
FUNCTION
No Connection. Not internally connected.
2
UVEN
Undervoltage Lockout (UVLO) Threshold/Enable Input. UVEN is a dual-function adjustable UVLO
threshold input with an enable feature. Connect UVEN to VCC through a resistive voltage-divider to
program the UVLO threshold. Connect UVEN directly to VCC to use the 6.0V (max) default UVLO
threshold. Apply a voltage greater than 1.244V to UVEN to enable the device.
3
REG1
5V Regulator Output. REG1 is an internal low-dropout voltage regulator that generates a 5V (VCC >
6V) output voltage and supplies power to internal circuitry. Bypass REG1 to AGND through a 1µF
ceramic capacitor.
4
AGND
Analog Ground
5
REF
Accurate 3V Buffered Reference Output. Connect REF to DIM through a resistive voltage-divider to
apply a DC voltage for analog-controlled dimming functionality. Leave REF unconnected if unused.
6
DIM
Dimming Control Input. Connect DIM to an external PWM signal for PWM dimming. For analogcontrolled dimming, connect DIM to REF through a resistive voltage-divider. The dimming frequency
is 200Hz under these conditions. Connect DIM to AGND to turn off the LEDs.
7
RTSYNC
SYNC Input/Output. The PWM clock is generated by the RTSYNC oscillator. Connect an external
resistor to RTSYNC to select a clock switching frequency from 125kHz to 600kHz or connect RTSYNC
to an external clock to synchronize the MAX16831 with a master clock signal.
8
CLKOUT
Clock Output. CLKOUT buffers the oscillator/clock. Connect CLKOUT to the SYNC input of another
device to operate the MAX16831 in a multichannel configuration. CLKOUT is a logic output.
9, 10, 11
I.C.
12
COMP
13
CS
Current-Sense Amplifier Output. The current-sense amplifier (CSA) senses the differential voltage
across the load sense resistor, RCS, and generates a voltage, VCS, at CS proportional to the LED
current. Connect the proper compensation resistor from CS to FB.
14
FB
Error-Amplifier Inverting Input
15
OV
Overvoltage Protection Input. Connect OV to HI through a resistive voltage-divider to set the
overvoltage limit for the load. When the voltage at OV exceeds the 1.235V (typ) threshold, an
overvoltage fault is generated and the switching MOSFET turns off. The MOSFET is turned on again
when the voltage at OV drops below 1.17V (typ).
16, 17
SGND
18
DRV
Gate Driver Output. Connect DRV to the gate of an external n-channel MOSFET for switching.
19
DRI
Gate Driver Supply Input. Connect DRI to REG2 to power the primary switching MOSFET driver.
Bypass DRI to AGND through a 10µF ceramic capacitor.
20
SNS+
Positive Peak Current-Sense Input. Connect SNS+ to the positive side of the switch current-sense
resistor, RSENSE.
21
SNS-
Negative Peak Current-Sense Input. Connect SNS- to the negative side of the switch current-sense
resistor, RSENSE.
22
QGND
23
DGT
Internally Connected. Must be connected to AGND.
Error-Amplifier Output. Connect the compensation network from COMP to FB for stable closed-loop
control. Use low-leakage ceramic capacitors in the feedback network.
Switching Ground. SGND is the ground for non-analog and high-current gate driver circuitry.
Analog Ground. Ensure a low-impedance connection between QGND and AGND.
Dimming Gate Driver Output. Connect DGT to the gate of an external n-channel MOSFET for
dimming. DGT is powered by the internal regulator, CLMP, and is referenced to LO.
Maxim Integrated
MAX16831
High-Voltage, High-Power LED Driver with
Analog and PWM Dimming Control
Pin Description (continued)
PIN
NAME
FUNCTION
25
LO
Low-Voltage Input. LO is the return point for the LED current. When using the MAX16831 in a buckboost configuration, connect LO to VCC. When using the device in a boost configuration only,
connect LO to SGND. Connect LO to the junction of the inductor and LED current-sense resistor,
RCS, when using a buck configuration.
26
CS+
Noninverting Current-Sense Amplifier Input. Connect CS+ to the positive side of an external sense
resistor, RCS, connected in series with the load (LEDs).
27
CS-
Inverting Current-Sense Amplifier Input. Connect CS- to the negative side of an external sense
resistor, RCS, connected in series with the load (LEDs).
Internal CLMP Regulator Output. CLMP supplies an 8V (typ) output when VHI ≥ 9V. If VHI is lower than
9V, VCLMP is one diode drop below VHI. The CLMP regulator powers the current-sense amplifier and
provides the high reference for the dimming driver. VCLMP must be at least 2.5V higher than VLO to
enable the current-sense amplifier and dimming MOSFET driver. Bypass CLMP to LO with a 0.1µF
ceramic capacitor.
28
CLMP
29
HI
30
REG2
31
VCC
Supply Voltage Input
32
I.C.
Internally Connected. This pin is internally pulled to REG1 through a 10kΩ resistor. Leave this pin
unconnected or connect it to QGND using a resistor of any value. If it is directly connected to QGND,
400µA to 600µA of current will flow out of this pin from VCC. Any resistor between this pin and QGND
will reduce the current accordingly.
—
EP
Exposed Pad. Connect EP to AGND. EP also functions as a heatsink to maximize thermal dissipation.
Do not use as a ground connection.
High-Voltage Input. HI is referred to LO. HI supplies power to the current-sense amplifier and
dimming MOSFET gate driver through the CLMP regulator.
Internal Regulator Output. REG2 is an internal voltage regulator that generates a 7V output and
supplies power to internal circuitry. Connect REG2 to DRI to power the switching MOSFET driver
during normal operation. Bypass REG2 to AGND with a 10µF ceramic capacitor.
Detailed Description
The MAX16831 is a current-mode PWM LED driver
used for driving HBLEDs. By using two current regulation loops, 5% output current accuracy is achieved.
One current regulation loop controls the external
switching MOSFET peak current through a sense resistor, RSENSE, from SNS+ to SNS-, while the other current
regulation loop controls the average LED string current
through the sense resistor RCS in series with the LEDs.
The wide operating supply range of (6.0V/5.5V
ON/OFF) up to 76V makes the MAX16831 ideal in automotive applications.
The MAX16831 features a programmable undervoltage
lockout (UVEN) that ensures predictable operation during brownout conditions. The input UVEN circuit monitors
the supply voltage, VCC, and turns the driver off when
VCC drops below the UVLO threshold. Connect UVEN to
VCC to use the 5.7V (typ) default UVLO threshold. The
Maxim Integrated
MAX16831 includes a cycle-by-cycle current limit that
turns off the gate drive to the external switching MOSFET (Q S ) during an overcurrent condition. The
MAX16831 features a programmable oscillator that simplifies and optimizes the design of external magnetics.
The MAX16831 includes three internal voltage regulators, REG1, REG2, and CLMP, and a 3V buffered reference output, REF. Connect REG2 to the driver supply,
DRI, to power the switching MOSFET driver.
The MAX16831 is capable of synchronizing with an
external clock or operating in stand-alone mode. A single resistor, RT, can be used to adjust the switching frequency from 125kHz to 600kHz for stand-alone
operation. To synchronize the device with an external
clock, apply a clock signal directly to the RTSYNC
input. A buffered clock output, CLKOUT, is available to
configure the MAX16831 in multichannel applications.
9
MAX16831
High-Voltage, High-Power LED Driver with
Analog and PWM Dimming Control
The MAX16831 features a differential high-side level
shifter to drive an external n-channel MOSFET for dimming. Wide contrast “pulsed” dimming (1000:1) is possible by applying either a low-frequency PWM input
signal or a DC voltage to the dimming input (DIM).
Protection features include peak current limiting,
HICCUP mode current limiting, output overvoltage protection, short-circuit protection, and thermal shutdown.
The HICCUP current-limit circuitry reduces the power
delivered to the load during severe fault conditions.
Nonlatching overvoltage protection limits the voltage on
the external switching MOSFET (QS) under open-circuit
conditions in the LED string. During continuous operation at high input voltages, the power dissipation of the
MAX16831 could exceed the maximum rating and an
internal thermal shutdown circuitry safely turns off the
MAX16831 when the device junction temperature
exceeds +165°C. When the junction temperature drops
below the hysteresis temperature, the MAX16831 automatically re-initiates startup.
Undervoltage Lockout/Enable
The MAX16831 features a dual-purpose adjustable
UVLO input and enable function. Connect UVEN to VCC
through a resistive voltage-divider to set the undervoltage lockout (UVLO) threshold. The MAX16831 is
enabled when the UVEN exceeds the 1.244V (typ)
threshold. Drive UVEN to ground to disable the output.
Setting the UVLO Threshold
The MAX16831 features a programmable UVLO threshold. Connect UVEN directly to VCC to select the default
6.0V (max) UVLO threshold. Connect UVEN to V CC
through a resistive voltage-divider to select a UVLO
threshold (Figure 1). Calculate resistor values as follows:
⎛
⎞
VUVEN
R UV1 = R UV 2 × ⎜
⎟
V
V
⎝ UVLO
UVEN ⎠
where RUV1 + RUV2 ≤ 270kΩ, VUVEN is the 1.244V (typ)
threshold voltage, and V UVLO is the desired UVLO
threshold in volts at VCC (Figure 1).
The capacitor, CUVEN, is required to prevent chattering
at the UVLO threshold due to line impedance drops
during power-up and dimming. If the undervoltage setting is very close to the required minimum operating
voltage, then there can be jumps in the voltage at VCC
during dimming, which may cause the MAX16831 to
turn on and off when the dimming signal transitions
from low to high. The capacitor, C UVEN , should be
10
VIN
RUV2
VCC
UVEN
MAX16831
CUVEN
RUV1
QGND
Figure 1. Setting the UVLO Threshold
large enough to limit the ripple on UVEN to less than
the 100mV (min) UVEN hysteresis so that the device
does not turn off under these circumstances.
Soft-Start
The MAX16831 includes a factory-set 4ms (typ) softstart delay that allows the load current to ramp up in a
controlled manner, minimizing output overshoot. Softstart begins once the device is enabled and V CC
exceeds the UVLO threshold. Soft-start circuitry slowly
increases the internal soft-start voltage, VSS, resulting
in a controlled rise of the load current. Signals applied
to DIM are ignored until the soft-start duration is complete and a successive delay of 200µs has elapsed.
Internal Regulators
The MAX16831 includes a fixed 5V voltage regulator
REG1, a 7V voltage regulator REG2, and an internal 8V
regulator CLMP. REG1 and REG2 power up when VCC
exceeds the UVLO threshold. REG1 supplies power to
internal circuitry and remains on during PWM dimming.
It is capable of driving external loads up to 2mA.
REG2 is capable of delivering up to 20mA of current.
Connect REG2 to DRI to generate the supply voltage
for the primary switching MOSFET driver, DRV.
CLMP is powered by HI and supplies power to the current-sense amplifier (CSA). CSA is enabled when
V CLMP goes 2.5V above V LO and is disabled when
(VCLMP - VLO) falls below 2.28V. The CLMP regulator
also provides power to the dimming MOSFET control
circuitry. CLMP is the output of the CLMP regulator. Do
not use CLMP to power external circuitry. Bypass
CLMP to LO with a 0.1µF ceramic capacitor. A larger
capacitor will result in overshoots of the load current.
Maxim Integrated
MAX16831
High-Voltage, High-Power LED Driver with
Analog and PWM Dimming Control
Reference Voltage Output
The MAX16831 includes a 5% accurate 3V (typ)
buffered reference output, REF. REF is a push-pull output capable of sourcing/sinking 100µA of current and
can drive a maximum load capacitance of 100pF.
Connect REF to DIM through a resistive voltage-divider
to supply an analog signal for dimming. See the
Dimming Input (DIM) section.
REF
R3
MAX16831
DIM
AGND
R4
Dimming MOSFET Driver (DDR)
The MAX16831 requires an external n-channel
MOSFET for PWM dimming. Connect the MOSFET to
the output of the DDR dimming driver, DGT, for normal
operation. VDGT swings between VLO and VCLMP. The
DDR dimming driver is capable of sinking or sourcing
up to 20mA of current. The average current required to
drive the dimming MOSFET (IDRIVE_DIM) depends on
the MOSFET’s total gate charge (QG_DIM) and the dimming frequency of the converter, fDIM. Use the following equation to calculate the average gate drive current
for the n-channel dimming FET.
IDRIVE_DIM = QG_DIM x fDIM
n-Channel MOSFET Switch Driver (DRV)
The MAX16831 drives an external n-channel MOSFET.
Use an external supply or connect REG2 to DRI to
power the MOSFET driver. The driver output, VDRV,
swings between ground and VDRI. Ensure that VDRI
remains below the absolute maximum VGS rating of the
external MOSFET. DRV is capable of sinking 2.5A or
sourcing 1.4A of peak current, allowing the MAX16831
to switch MOSFETs in high-power applications. The
average current sourced to drive the external MOSFET
depends on the total gate charge (QG) and operating
frequency of the converter, fSW. The power dissipation
in the MAX16831 is a function of the average output
drive current (IDRIVE). Use the following equations to
calculate the power dissipation in the gate driver section of the MAX16831 due to IDRIVE:
IDRIVE = QG x fSW
PD = (IDRIVE + ICC) x VDRI
where VDRI is the supply voltage to the gate driver and
ICC is the operating supply current. IDRIVE should not
exceed 20mA.
Dimming Input (DIM)
The dimming input, DIM, functions with either analog or
PWM control signals. Once the internal pulse detector
detects three successive edges of a PWM signal with a
frequency between 80Hz and 2kHz, the MAX16831 synchronizes to the external signal and pulse-width-modulates the LED current at the external DIM input frequency
with the same duty cycle as the DIM input. If an analog
Maxim Integrated
Figure 2. Creating a DIM Input Signal from REF
control signal is applied to DIM, the MAX16831 compares the DC input to an internally generated 200Hz
ramp to pulse-width-modulate the LED current (fDIM =
200Hz). The output current duty cycle is linearly
adjustable from 0 to 100% (0.2V < VDIM < 2.8V).
Use the following formula to calculate the voltage, VDIM,
necessary for a given output-current duty cycle, D:
VDIM = (D x 2.6) + 0.2V
where VDIM is the voltage applied to DIM in volts.
Connect DIM to REF through a resistive voltage-divider
to apply a DC DIM control signal (Figure 2). Use the
required dimming input voltage, V DIM , calculated
above and select appropriate resistor values using the
following equation:
R4 = R3 x VDIM / (VREF - VDIM)
where V REF is the 3V reference output voltage and
30kΩ ≤ R3 + R4 ≤ 150kΩ.
For proper operation at startup or after toggling ENABLE,
the controller needs three clock edges or an analog voltage greater than 0.3V on the DIM input.
Oscillator, Clock, and Synchronization
The MAX16831 is capable of stand-alone operation or
synchronizing to an external clock, and driving external
devices in SYNC mode. For stand-alone operation, program the switching frequency by connecting a single
external resistor, RT, between RTSYNC and ground.
Select the switching frequency, fSW, from 125kHz to
600kHz and calculate RT using the following formula:
RT =
500kHz
× 25k Ω
fSW
where the switching frequency is in kHz and RT is in kΩ.
The MAX16831 is also capable of synchronizing to an
external clock signal ranging from 125kHz to 600kHz.
11
MAX16831
High-Voltage, High-Power LED Driver with
Analog and PWM Dimming Control
Connect the clock signal to the RTSYNC input. The
MAX16831 synchronizes to the external clock signal
after the detection of five successive clock edges at
RTSYNC.
A buffered clock output, CLKOUT, is capable of driving
the RTSYNC input of an external PWM controller for
multichannel applications. CLKOUT is capable of driving capacitive loads up to 500pF.
MASTER/PEER
SLAVE/PEER
MAX16831
MAX16831
RTSYNC
CLKOUT
RTSYNC
CLKOUT
RT
Multichannel Configuration
The MAX16831 is capable of multichannel operation.
Connect CLKOUT to the SYNC input of an external
device to use the MAX16831 as a master clock signal.
Connect an external clock signal to RTSYNC to configure the MAX16831 as a slave. To setup two or more
MAX16831 devices in a daisy-chain/peer-to-peer configuration, drive the RTSYNC input of one MAX16831
with the CLKOUT buffer of another (Figure 3).
ILIM and HICCUP Comparator
RSENSE sets the peak current through the inductor for
switching. The differential voltage across RSENSE is
compared to the 200mV voltage trip limit of the currentlimit comparator, ILIM. Set the current limit 20% higher
than the peak switch current at the rated output power
and minimum voltage. Use the following equation to
calculate RSENSE:
RSENSE = VSENSE / (1.2 x IPEAK)
where V SENSE is the 200mV differential voltage
between SNS+ and SNS- and IPEAK is the peak inductor current at full load and minimum input voltage.
When the voltage drop across RSENSE exceeds the
ILIM threshold, the MOSFET driver (DRV) terminates
the on-cycle and turns the switch off, reducing the current through the inductor. The FET is turned back on at
the beginning of the next switching cycle.
When the voltage across RSENSE exceeds the 300mV
(typ) HICCUP threshold, the HIC comparator terminates
the on-cycle of the device, turning the switching
MOSFET off. Following a startup delay of 4ms (typ), the
MAX16831 re-initiates soft-start. The device will continue to operate in HICCUP mode until the overcurrent
condition is removed.
A built-in 40ns leading-edge blanking circuit of the current-sense signal prevents these comparators from prematurely terminating the on-cycle of the external
switching MOSFET (QS). In some cases, this blanking
time may not be adequate and an additional RC filter
may be required to prevent spurious turn-off.
12
Figure 3. Master-Slave/Peer-Peer Clock Configuration
Load Current Sense
The load-sense resistor, R CS , monitors the current
through the LEDs. The internal floating current-sense
amplifier, CSA, measures the differential voltage across
RCS, and generates a voltage proportional to the LED
current through R CS at CS. This voltage on CS is
referred to AGND. The closed loop regulates the LED
current to a value, ILED, given by the following equation:
ILED = 0.107V / RCS
Slope Compensation
The MAX16831 uses an internal ramp generator for
slope compensation. The internal ramp signal is reset
to zero at the beginning of each cycle and has a peakto-peak voltage of 120mV per switching cycle. Use an
external resistor, RT, to set the switching frequency,
fSW, and calculate the slope of the compensating ramp,
mSLOPE, using the following equation:
mSLOPE = 120 x fSW [mV/s]
where fSW is the switching frequency in Hz. When the
MAX16831 is synchronized to an external clock, the
slope compensation ramp has a slope of 15mV/µs.
Internal Voltage-Error Amplifier (EAMP)
The MAX16831 includes a built-in voltage amplifier,
with tri-state output, which can be used to close the
feedback loop. The buffered output current-sense signal appears at CS, which is connected to the inverting
input, FB, of the error amplifier through resistor R1. The
noninverting input is connected to an internally trimmed
current reference.
The output of the error amplifier is controlled by the signal applied to DIM. When DIM is high, the output of the
amplifier is connected to COMP. The amplifier output is
open when DIM is low. This enables the integrating
Maxim Integrated
MAX16831
High-Voltage, High-Power LED Driver with
Analog and PWM Dimming Control
capacitor to hold the charge when the DIM signal has
turned off the gate drive. When DIM is high again, the
voltage on the compensation capacitors, C1 and C2,
will force the converter into steady-state instantaneously.
VLED+
MAX16831
ROV1
PWM Dimming
PWM dimming is achieved by driving DIM with either a
PWM signal or a DC signal. The PWM signal is internally connected to the error amplifier, the dimming
MOSFET gate driver, and the switching MOSFET gate
driver. When the DIM signal is high, the dimming
MOSFET and the switching MOSFET drivers are
enabled and the output of the voltage-error amplifier is
connected to the external compensation network. Also,
the buffered current-sense signal is connected to CS.
Preventing discharge of the compensation capacitor
when the DIM signal is low will allow the control loop to
return the LED current to its original value almost
instantaneously.
When the DIM signal goes low, the output of the error
amplifier is disconnected from the compensation network and the voltage of compensation capacitors, C1
and C2 is preserved. Choose low-leakage capacitors
for C1 and C2. The drivers for the external dimming
and switching MOSFETs are disabled, and the converter stops switching. The inductor energy is now transferred to the output capacitors.
When the DIM signal goes high and the gate drivers are
enabled, the additional voltage on the output capacitor
may cause a current spike on the LED string. A larger
output capacitor will result in a smaller current spike. The
MAX16831 thus achieves fast PWM dimming response.
Fault Protection
The MAX16831 features built-in overvoltage protection,
overcurrent protection, HICCUP mode current-limit protection, and thermal shutdown. Overvoltage protection
is achieved by connecting OV to HI through a resistive
voltage-divider. HICCUP mode limits the power dissipation in the external MOSFETs during severe fault
conditions. Internal thermal shutdown protection safely
turns off the converter when the junction temperature
exceeds +165°C.
Overvoltage Protection
The overvoltage protection (OVP) comparator compares the voltage at OV with a 1.235V (typ) internal reference. When the voltage at OV exceeds the internal
reference, the OVP comparator terminates PWM
switching and no further energy is transferred to the
Maxim Integrated
OV
AGND
ROV2
Figure 4. Setting the Overvoltage Threshold
load. The MAX16831 re-initiates soft-start once the
overvoltage condition is removed. Connect OV to HI
through a resistive voltage-divider to set the overvoltage threshold at the output.
Setting the Overvoltage Threshold
Connect OV to HI or to the high-side of the LEDs
through a resistive voltage-divider to set the overvoltage threshold at the output (Figure 4). The overvoltage
protection (OVP) comparator compares the voltage at
OV with a 1.235V (typ) internal reference. Use the following equation to calculate resistor values:
⎛ VOV _ LIM - VOV ⎞
R OV1 = R OV 2 × ⎜
⎟
VOV
⎝
⎠
where VOV is the 1.235V OV threshold. Choose ROV1
and ROV2 to be reasonably high-value resistors to prevent discharge of filter capacitors. This will prevent
unnecessary undervoltage and overvoltage conditions
during dimming.
Load-Dump Protection
The MAX16831 features load-dump protection up to
80V. LED drivers using the MAX16831 can sustain single fault load dump events. Repeated load dump events
within very short time intervals can cause damage to the
dimming MOSFET due to excess power dissipation.
Thermal Shutdown
The MAX16831 contains an internal temperature sensor
that turns off all outputs when the die temperature
exceeds +165°C. Outputs are enabled again when the
die temperature drops below +145°C.
13
MAX16831
High-Voltage, High-Power LED Driver with
Analog and PWM Dimming Control
Applications Information
Inductor Selection
The minimum required inductance is a function of operating frequency, input-to-output voltage differential, and
the peak-to-peak inductor current (∆I L ). Higher ∆I L
allows for a lower inductor value while a lower ∆I L
requires a higher inductor value. A lower inductor value
minimizes size and cost, improves large-signal transient response but reduces efficiency due to higher
peak currents and higher peak-to-peak output ripple
voltage for the same output capacitance. On the other
hand, higher inductance increases efficiency by reducing the ripple current, ∆IL. However, resistive losses
due to extra turns can exceed the benefit gained from
lower ripple current levels, especially when the inductance is increased without also allowing for larger
inductor dimensions. A good compromise is to choose
∆IL equal to 30% of the full load current. The inductor
saturating current is also important to avoid runaway
current during the output overload and continuous
short circuit. Select the ISAT to be higher than the maximum peak current limit.
Buck configuration: In a buck configuration, the average inductor current does not vary with the input. The
worst-case peak current occurs at a high input voltage.
In this case, the inductance L for continuous conduction mode is given by:
L=
VOUT × (VINMAX - VOUT )
VINMAX × fSW × ∆ IL
where VINMAX is the maximum input voltage, fSW is the
switching frequency, and VOUT is the output voltage.
Boost configuration: In the boost converter, the average
inductor current varies with line and the maximum average current occurs at low line. For the boost converter,
the average inductor current is equal to the input current. In this case, the inductance L is calculated as:
L=
VINMIN × (VOUT - VINMIN )
VOUT × fSW × ∆ IL
where VINMIN is the minimum input voltage, VOUT is the
output voltage, and fSW is the switching frequency.
Buck-boost configuration: In a buck-boost converter,
the average inductor current is equal to the sum of the
input current and the load current. In this case, the
inductance L is:
VOUT × VINMIN
L=
(VOUT + VINMIN ) × fSW × ∆ IL
14
where VINMIN is the minimum input voltage, VOUT is the
output voltage, and fSW is the switching frequency.
Output Capacitor
The function of the output capacitor is to reduce the
output ripple to acceptable levels. The ESR, ESL, and
the bulk capacitance of the output capacitor contribute
to the output ripple. In most of the applications, the output ESR and ESL effects can be dramatically reduced
by using low-ESR ceramic capacitors. To reduce the
ESL effects, connect multiple ceramic capacitors in
parallel to achieve the required bulk capacitance.
In a buck configuration, the output capacitance, CF, is
calculated using the following equation:
CF ≥
(VINMAX - VOUT ) × VOUT
∆ VR × 2 × L × VINMAX × fSW 2
where ∆VR is the maximum allowable output ripple.
In a boost configuration, the output capacitance, CF, is
calculated as:
CF ≥
(VOUT - VINMIN ) × 2 × IOUT
∆ VR × VOUT × fSW
where IOUT is the output current.
In a buck-boost configuration, the output capacitance,
CF, is calculated as:
CF ≥
2 × VOUT × IOUT
∆ VR × (VOUT + VINMIN ) × fSW
where VOUT is the voltage across the load and IOUT is
the output current. Connect the output capacitor(s)
from the output to ground in a buck-boost configuration
(not across the load as for other configurations).
Input Capacitor
A capacitor connected between the input line and
ground must be used when configuring the MAX16831
as a buck converter. Use a low-ESR input capacitor
that can handle the maximum input RMS ripple current.
Calculate the maximum allowable RMS ripple using the
following equation:
IIN(RMS) =
IOUT × VOUT × (VINMIN - VOUT )
VINMIN
In most of the cases, an additional electrolytic capacitor should be added to prevent input oscillations due to
line impedances.
Maxim Integrated
MAX16831
High-Voltage, High-Power LED Driver with
Analog and PWM Dimming Control
When using the MAX16831 in a boost or buck-boost
configuration, the input RMS current is low and the
input capacitance can be small.
Operating the MAX16831 Without the
Dimming Switch
The MAX16831 can also be used in the absence of the
dimming MOSFET. In this case, the PWM dimming performance is compromised but in applications that do
not require dimming, the MAX16831 can still be used.
A short circuit across the load will cause the MAX16831
to disable the gate drivers and they will remain off until
the input power is recycled.
Switching Power MOSFET Losses
When selecting MOSFETs for switching, consider the
total gate charge, power dissipation, the maximum
drain-to-source voltage, and package thermal impedance. The product of the MOSFET gate charge and
RDS(ON) is a figure of merit, with a lower number signifying better performance. Select MOSFETs optimized
for high-frequency switching applications.
MOSFET losses may be broken into three categories:
conduction loss, gate drive loss, and switching loss.
The following simplified power loss equation is true for
all the different configurations.
• Isolate the power components and high-current paths
from sensitive analog circuitry.
• Keep the high-current paths short, especially at the
ground terminals. This practice is essential for stable, jitter-free operation. Keep switching loops short.
• Connect AGND, SGND, and QGND to a ground
plane. Ensure a low-impedance connection between
all ground points.
• Keep the power traces and load connections short.
This practice is essential for high efficiency. Use
thick copper PCBs (2oz vs. 1oz) to enhance full-load
efficiency.
• Ensure that the feedback connection to FB is short
and direct.
• Route high-speed switching nodes away from the
sensitive analog areas.
• To prevent discharge of the compensation capacitors, C1 and C2, during the off-time of the dimming
cycle, ensure that the PCB area close to these components has extremely low leakage. Discharge of
these capacitors due to leakage may result in
degraded dimming performance.
Pin Configuration
PLOSS = PCONDUCTION + PGATEDRIVE + PSWITCH
I.C.
VCC
REG2
HI
CLMP
CS-
CS+
LO
TOP VIEW
32
31
30
29
28
27
26
25
+
24
N.C.
2
23
DGT
REG1
3
22
QGND
AGND
4
21
SNS-
REF
5
20
SNS+
DIM
6
19
DRI
RTSYNC
7
18
DRV
CLKOUT
8
17
SGND
MAX16831
9
10
11
12
13
14
15
16
CS
FB
OV
SGND
*EP
COMP
UVEN
I.C.
1
I.C.
N.C.
I.C.
Layout Recommendations
Typically, there are two sources of noise emission in a
switching power supply: high di/dt loops and high dv/dt
surfaces. For example, traces that carry the drain current often form high di/dt loops. Similarly, the heatsink
of the MOSFET connected to the device drain presents
a high dv/dt source; therefore, minimize the surface
area of the heatsink as much as possible. Keep all PCB
traces carrying switching currents as short as possible
to minimize current loops. Use ground planes for best
results.
Careful PCB layout is critical to achieve low switching
losses and clean, stable operation. Use a multilayer
board whenever possible for better noise performance
and power dissipation. Follow these guidelines for
good PCB layout:
• Use a large copper plane under the MAX16831
package. Ensure that all heat-dissipating components have adequate cooling. Connect the exposed
pad of the device to the ground plane.
TQFN
(5mm x 5mm)
*EP = EXPOSED PAD
Maxim Integrated
15
MAX16831
High-Voltage, High-Power LED Driver with
Analog and PWM Dimming Control
Functional Diagram
CLMP
VCC
CSA
7V
REG2
5V
REG1
+
UVLO
AND
EN
LO
CS- CS+
HI
CLMP
VLO
VCLMP
UVEN
REG2
THERMAL
SHUTDOWN
RLS
QGND
SLOPE
COMP
REG1
SLOPE
VCLMP
VCLMP
UGB
10uA
DDR
VBUF 3.0V
+
REF
+
CMP 1.3 x V
SS
-
DGT
TRI
VLO
RTSYNC
CS
OSC
OSC
OC
CLKOUT
POR
DRIVER
OV
OVP
OV
DRI
CONTROL
BLOCK
EN
VOV -
+
SGND
+
ILIM - 200mV
+
DIM
COMP
AGND
DRV
-
200mV
+
HIC
-
300mV
BLANKING
TIME
40ns
MAX16831
PWM
INTERNAL
TRIM
- 600mV +
SLOPE
SS
VOV
SNS-
+
-
200Hz
SNS+
X1
VSS
EAMP
COMP
FB
16
Maxim Integrated
MAX16831
High-Voltage, High-Power LED Driver with
Analog and PWM Dimming Control
Typical Operating Circuits (continued)
BOOST CONFIGURATION
VIN
RCS
CCLMP
RUV2
CF
VCC
RUV1
CS+
CS-
DGT
LO
CLMP
UVEN
QS
RD
DRV
CUVEN
LEDs
SNS+
RSENSE
REF
SNS-
R3
MAX16831
QGND
DIM
RTSYNC
R4
RT
ROV1
OV
REG1
CREG1
COMP
CS
AGND
FB
R1
SGND
HI
DRI
REG2
ROV2
CREG2
C2
R2
C1
Maxim Integrated
17
MAX16831
High-Voltage, High-Power LED Driver with
Analog and PWM Dimming Control
Typical Operating Circuits (continued)
BUCK CONFIGURATION
VIN
CCLMP
RUV2
VCC HI
RUV1
LO CLMP
RCS
CS- CS+ DGT
UVEN
QS
RD
DRV
CUVEN
LEDs
SNS+
RSENSE
SNS-
DIM
DIM
CF
QGND
MAX16831
REG1
CREG1
RT
RTSYNC
COMP
OV
CS
FB
AGND
R1
SGND
DRI
REG2
CREG2
C2
R2
C1
Package Information
Chip Information
PROCESS: BICMOS
18
For the latest package outline information and land patterns (footprints), go to www.maximintegrated.com/packages. Note that a
“+”, “#”, or “-” in the package code indicates RoHS status only.
Package drawings may show a different suffix character, but the
drawing pertains to the package regardless of RoHS status.
PACKAGE
TYPE
PACKAGE
CODE
OUTLINE NO.
LAND
PATTERN NO.
32 TQFN-EP
T3255M+4
21-0140
90-0014
Maxim Integrated
MAX16831
High-Voltage, High-Power LED Driver with
Analog and PWM Dimming Control
Revision History
REVISION
NUMBER
REVISION
DATE
0
4/07
Initial release
1
4/09
Updated Pin Description and Input Capacitor sections.
9, 14
2
1/13
Added automotive qualified part to Ordering Information
1
DESCRIPTION
PAGES
CHANGED
—
Maxim Integrated cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim Integrated product. No circuit patent
licenses are implied. Maxim Integrated reserves the right to change the circuitry and specifications without notice at any time. The parametric values (min and
max limits) shown in the Electrical Characteristics table are guaranteed. Other parametric values quoted in this data sheet are provided for guidance.
Maxim Integrated 160 Rio Robles, San Jose, CA 95134 USA 1-408-601-1000 ________________________________ 19
© 2013 Maxim Integrated Products, Inc.
Maxim Integrated and the Maxim Integrated logo are trademarks of Maxim Integrated Products, Inc.
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