IRU3004CW

Data Sheet No. PD94140 IRU3004 5-BIT PROGRAMMABLE SYNCHRONOUS BUCK CONTROLLER IC WITH DUAL LDO CONTROLLER DESCRIPTION FEATURES Meets latest VRM 8.4 specification for PentiumIII Provides single chip solution for Vcore, GTL+ and clock supply On-Board DAC programs the output voltage from 1.3V to 3.5V. The IRU3004 remains on for VID code of (11111) Dual linear regulator controller on-board for 1.5V GTL+ and 2.5V clock supplies Loss-less Short Circuit Protection Synchronous operation allows maximum efficiency Patented architecture allows fixed frequency operation as well as 100% duty cycle during dynamic load Minimum Part Count, No External Compensation Soft-Start Function High current totem pole driver for direct driving of the external power MOSFET Power Good Function The IRU3004 controller IC is specifically designed to meet Intel specifications for Pentium III microprocessor applications as well as the next generation P6 family processors. The IC provides a single chip controller IC for the Vcore, GTL+ and clock supplies required for the Pentium III applications. The IRU3004 features a patented topology, that in combination with a few external components as shown in the typical application circuit, will provide in excess of 20A of output current for an onboard DC-DC converter while automatically providing the right output voltage via the 5-bit internal DAC meeting the latest VRM specification. The IRU3004 also features loss-less current sensing by using the RDS(on) of the high side power MOSFET as the sensing resistor and a Power Good window comparator that switches its open collector output low when the output is outside of a ±10% window. Other features of the device are: under-voltage lockout for both 5V and 12V supplies, an external programmable soft-start function as well as programming the oscillator frequency by using an external capacitor. APPLICATIONS Pentium III & next generation processor DC to DC converter application Low Cost Pentium with AGP TYPICAL APPLICATION 5V R16 C5 C13 Note: Pentium III is trademark of Intel Corp. L2 Q1 L1 R1 C7 VOUT 3 R17 C16 Q2 C3 C10 R4 R2 R3 R12 R13 3.3V C4 12V C6 Q3 V OUT 1 C11 R18 V12 V5 CS+ HDrv CS- LDrv Gnd R7 VFB3 R11 Ct Lin1 C15 IRU3004 C1 SS C2 V FB1 D4 D3 D2 D1 D0 V FB2 PGd R8 Q4 V OUT 2 Lin2 C12 C9 3.3V R14 VID4 VID3 R5 VID2 C8 VID1 R9 C14 R15 Power Good VID0 Figure 1 - Typical application of the IRU3004. PACKAGE ORDER INFORMATION TA (8C) 0 To 70 0 To 70 Rev. 1.7 07/16/02 DEVICE IRU3004CW IRU3004CF PACKAGE 20-Pin Plastic SOIC (W) 20-Pin Plastic TSSOP (F) www.irf.com 1 IRU3004 ABSOLUTE MAXIMUM RATINGS V5 Supply Voltage .................................................... V12 Supply Voltage .................................................. Storage Temperature Range ...................................... Operating Junction Temperature Range ..................... 10V 20V -65°C To 150°C 0°C To 125°C PACKAGE INFORMATION 20-PIN WIDE BODY PLASTIC SOIC (W) 20-PIN PLASTIC TSSOP (F) TOP VIEW TOP VIEW Ct 1 Ct 1 20 Lin2 20 Lin2 Lin1 2 19 D 0 Lin1 2 19 D 0 V FB1 3 18 D 1 V FB1 3 18 D 1 V FB2 4 17 D 2 V FB2 4 17 D 2 V5 5 16 D 3 V5 5 16 D 3 PGd 6 15 D 4 PGd 6 15 D 4 CS- 7 14 V FB3 CS- 7 14 V FB3 CS+ 8 13 SS CS+ 8 13 SS HDrv 9 12 V12 HDrv 9 12 V12 Gnd 10 11 LDrv Gnd 10 uJA =858C/W 11 LDrv uJA =908C/W ELECTRICAL SPECIFICATIONS Unless otherwise specified, these specifications apply over V12=12V, V5=5V and TA=0 to 70°C. Typical values refer to TA=25°C. Low duty cycle pulse testing is used which keeps junction and case temperatures equal to the ambient temperature. PARAMETER VID Section DAC Output Voltage (Note 1) DAC Output Line Regulation DAC Output Temp Variation VID Input LO VID Input HI VID Input Internal Pull-Up Resistor to V5 Power Good Section Under-Voltage lower trip point Under-Voltage upper trip point UV Hysteresis Over-Voltage upper trip point Over-Voltage lower trip point OV Hysteresis Power Good Output LO Power Good Output HI Soft-Start Section Soft-Start Current 2 SYM TEST CONDITION MIN TYP MAX UNITS 0.98Vs Vs 1.02Vs 0.1 0.5 0.4 V % % V V KV 0.91Vs V V V V V V V V 2 27 VOUT Ramping Down VOUT Ramping Up VOUT Ramping Up VOUT Ramping Down 0.89Vs 0.015Vs 1.09Vs 0.015Vs RL=3mA RL=5K Pull-Up to 5V CS+=0V, CS-=5V www.irf.com 0.90Vs 0.92Vs 0.02Vs 1.10Vs 1.08Vs 0.02Vs 4.8 10 0.025Vs 1.11Vs 0.025Vs 0.4 mA Rev. 1.7 07/16/02 IRU3004 PARAMETER UVLO Section UVLO Threshold-12V UVLO Hysteresis-12V UVLO Threshold-5V UVLO Hysteresis-5V Error Comparator Section Input Bias Current Input Offset Voltage Delay to Output Current Limit Section CS Threshold Set Current CS Comp Offset Voltage Hiccup Duty Cycle Supply Current Operating Supply Current SYM TEST CONDITION Supply Ramping Up Supply Ramping Up MIN TYP MAX UNITS 9.2 0.3 4.1 0.2 10 0.4 4.3 0.3 10.8 0.5 4.5 0.4 V V V V 2 +2 100 mA mV ns 240 +5 2 mA mV % -2 VDIFF=10mV 160 -5 200 Css=0.1mF Output Drivers Section Rise Time Fall Time Dead Band Time Oscillator Section Osc Frequency Osc Valley Osc Peak LDO Controller Section VFB1 & VFB2 Input Bias Current Lin1 or Lin2 Drive Current CL=3000pF: V5 V12 20 14 CL=3000pF CL=3000pF CL=3000pF 100 70 70 200 100 130 300 ns ns ns Ct=150pF 160 220 260 0.2 KHz V V 1.545 2 V mA mA mA V5 1.455 1.500 50 Note 1: Vs refers to the set point voltage given in Table 1. D4 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 D3 1 1 1 1 1 1 1 1 0 0 0 0 0 0 0 0 D2 1 1 1 1 0 0 0 0 1 1 1 1 0 0 0 0 D1 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 D0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 Vs 1.30 1.35 1.40 1.45 1.50 1.55 1.60 1.65 1.70 1.75 1.80 1.85 1.90 1.95 2.00 2.05 D4 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 D3 1 1 1 1 1 1 1 1 0 0 0 0 0 0 0 0 D2 1 1 1 1 0 0 0 0 1 1 1 1 0 0 0 0 D1 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 D0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 Vs 2.0 2.1 2.2 2.3 2.4 2.5 2.6 2.7 2.8 2.9 3.0 3.1 3.2 3.3 3.4 3.5 Table 1 - Set point voltage vs. VID codes. Rev. 1.7 07/16/02 www.irf.com 3 IRU3004 PIN DESCRIPTIONS PIN# 1 PIN SYMBOL Ct 2 Lin1 3 4 5 6 VFB1 VFB2 V5 PGd 7 CS- 8 CS+ 9 10 HDrv Gnd 11 12 LDrv V12 13 SS 14 VFB3 15 D4 16 D3 17 D2 18 D1 19 D0 20 Lin2 4 PIN DESCRIPTION This pin programs the oscillator frequency in the range of 50KHz to 500KHz with an external capacitor connected from this pin to the ground. This pin controls the gate of an external transistor for either the GTL+ linear regulator or Clock supply. This pin provides the feedback for the linear regulator that its output drive is Lin1 pin. This pin provides the feedback for the linear regulator that its output drive is Lin2 pin. 5V supply voltage. This pin is an open collector output that switches LO when the output of the converter is not within ±10% (typical) of the nominal output voltage. When Power Good pin switches LO the sat voltage is less than 0.4V at 3mA. This pin is connected to the Source of the power MOSFET for the Core supply and it provides the negative sensing for the internal current sensing circuitry. This pin is connected to the Drain of the power MOSFET of the Core supply and it provides the positive sensing for the internal current sensing circuitry. An external resistor programs the CS threshold depending on the RDS of the power MOSFET. An external capacitor is placed in parallel with the programming resistor to provide high frequency noise filtering. Output driver for the high-side power MOSFET. This pin serves as the ground pin and must be connected directly to the ground plane. A high frequency capacitor (0.1 to 1mF) must be connected from V5 and V12 pins to this pin for noise free operation. Output driver for the synchronous power MOSFET. This pin is connected to the 12 V supply and serves as the power Vcc pin for the output drivers. A high frequency capacitor (0.1 to 1mF) must be connected directly from this pin to ground pin in order to supply the peak current to the power MOSFET duringthe transitions. This pin provides the soft-start for the switching regulator. An internal current source charges an external capacitor that is connected from this pin to the ground which ramps up the outputs of the switching regulator, preventing the outputs from overshooting as well as limiting the input current. The second function of the Soft-Start cap is to provide long off time (HICCUP) for the synchronous MOSFET during current limiting. This pin is connected directly to the output of the Core supply to provide feedback to the Error comparator. This pin selects a range of output voltages for the DAC. When in the LOW state the range is 1.3V to 2.05V. For VID codes of all "1" the IRU3004 keeps all the outputs on. MSB input to the DAC that programs the output voltage. This pin can be pulled-up externally by a 10K resistor to either 3.3V or 5V supply. Input to the DAC that programs the output voltage. This pin can be pulled up externally by a 10K resistor to either 3.3V or 5V supply. Input to the DAC that programs the output voltage. This pin can be pulled up externally by a 10KV resistor to either 3.3V or 5V supply. LSB input to the DAC that programs the output voltage. This pin can be pulled-up externally by a 10K resistor to either 3.3V or 5V supply. This pin controls the gate of an external transistor for either the GTL+ linear regulator or Clock supply. www.irf.com Rev. 1.7 07/16/02 IRU3004 BLOCK DIAGRAM Enable V12 12 V5 5 Vset Enable UVLO Vset 19 D1 18 D2 17 D3 16 D4 15 VFB2 4 Lin2 9 HDrv 11 LDrv 7 CS- 8 CS+ 1 Ct 13 SS 6 PGd 10 Gnd V12 Enable 5Bit DAC, Ctrl Logic VFB3 PWM Control + D0 14 V12 Slope Comp Soft Start & Fault Logic Osc Over Current 200uA Enable 20 1.1Vset 1.5V Lin1 2 0.9Vset VFB1 3 Figure 2 - Simplified block diagram of the IRU3004. Rev. 1.7 07/16/02 www.irf.com 5 IRU3004 TYPICAL APPLICATION Pentium III L2 L1 Q1 5V VO U T 3 R16 C5 R1 C7 C13 R17 C16 Q2 C3 C10 R4 R2 R3 R12 R13 3.3V C4 C6 Q3 VO U T 1 12V C11 R18 V12 V5 CS+ HDrv CS- LDrv Gnd R7 V FB3 R11 Ct Lin1 IRU3004 C1 C15 SS C2 D4 V FB1 D3 D2 D1 D0 V FB2 PGd Q4 VO U T 2 Lin2 C9 3.3V C12 VID4 VID3 R14 R5 R9 VID2 VID1 R8 C14 C8 R15 Power Good VID0 Figure 3 - Typical application of IRU3004 in an on-board DC-DC converter providing the Core, GTL+, and Clock supplies for the Pentium II microprocessor. 6 www.irf.com Rev. 1.7 07/16/02 IRU3004 IRU3004 APPLICATION PARTS LIST Ref Desig Description Qty Part # Manuf Q1 MOSFET 1 IRL3103S, TO-263 package IR Q2 MOSFET 1 IRL3103D1S, TO-263 package IR Q3 Bipolar Trans, GP 1 MPS2222A, SOT-23 package Q4 MOSFET 1 IRLR024, TO-252 package L1 Inductor 1 L=1mH, 5052 core with 4 turns of 1.0mm wire MicroMetal L2 Inductor 1 L=2.7mH, 5052B core with 7 turns of 1.2mm wire Micro Metal C1 Capacitor, Ceramic 1 150pF, 0603 C2, 6 Capacitor, Ceramic 2 1mF, 0603 C3 Capacitor, Electrolytic 2 10MV1200GX, 1200mF,10V C4 Capacitor, Ceramic 1 1mF, 0805 C5 Capacitor, Ceramic 1 220pF, 0603 C7, 14, 15 Capacitor, Ceramic 3 1000pF, 0603 C8 Capacitor, Ceramic 1 0.1mF, 0603 C9 Capacitor, Electrolytic 1 6MV1000GX, 1000mF, 6.3V Sanyo C10 Capacitor, Electrolytic 6 6MV1500GX, 1500mF, 6.3V Sanyo C11 Capacitor, Electrolytic 1 6MV150GX, 150mF, 6.3V Sanyo C12 Capacitor, Electrolytic 1 6MV1000GX, 1000mF, 6.3V Sanyo C13 Capacitor, Electrolytic 1 10MV470GX, 470mF, 10V Sanyo C16 Capacitor, Ceramic 1 4.7mF, 1206 R1 Resistor 1 3.3KV, 5%, 0603 R2, 3, 4 Resistor 3 4.7V, 5%, 1206 R5, 15 Resistor 2 10KV, 5%, 0603 R7, 12 Resistor 2 100V, 1%, 0603 R8 Resistor 1 150V, 1%, 0603 R9, 11, 14 Resistor 3 100V, 5%, 0603 R13 Resistor 1 22KV, 1%, 0603 R16 Resistor 1 220V, 1%, 0603 R17 Resistor 1 330V, 1%, 0603 R18 Resistor 1 10V, 5%, 0603 Motorola IR Sanyo Note 1: R16, R17, C16, R12, and R13 set the Vcore 2% higher for level shift to reduce CPU transient voltage. Note 2: R14 and R15 set the 1.5V approximately 1% higher to account for the trace resistance drop. Rev. 1.7 07/16/02 www.irf.com 7 IRU3004 TYPICAL APPLICATION Pentium with AGP L2 L1 Q1 5V VOUT3 R16 C5 C13 R1 C7 R17 C16 Q2 C3 C10 R4 R2 R3 R12 R13 3.3V C4 Q3 C6 12V C9 R18 V12 V5 CS+ HDrv CS- LDrv Gnd C11 R7 VFB3 R11 Ct Lin1 C15 IRU3004 C1 R8 VFB1 SS C2 D4 D3 D2 D1 D0 VFB2 PGd Q4 Lin2 3.3V 3.3V VID4 VID3 C12 R14 R5 VID2 VID1 R9 C14 C8 Power Good R15 VID0 Figure 4 - Typical application of IRU3004 in a Pentium with AGP where the power dissipation of the 3.3V linear regulator is equally distributed between Q3 and Q4 pass transistors. This equal distribution is possible by accurately regulating the first regulator using the IRU3004 linear controller and its internal 1% reference voltage while the second controller regulates the output of the first regulator from 4.17V to 3.3V, thereby distributing the power dissipation equally. 8 www.irf.com Rev. 1.7 07/16/02 IRU3004 IRU3004 APPLICATION PARTS LIST Ref Desig Description Q1 MOSFET Qty 1 Part # IRL3103s, TO-263 package Manuf IR Q2 MOSFET 1 IRL3103D1S, TO-263 package IR Q3, 4 MOSFET 2 IRL3303S, TO-263 package IR L1 Inductor 1 L=1mH, 5052 core with 4 turns of Micro Metal 1.0mm wire L2 Inductor 1 L=2.7mH, 5052B core with 7 turns of Micro Metal 1.2mm wire C1 Capacitor, Ceramic 1 150pF, 0603 C2, 6 Capacitor, Ceramic 2 1mF, 0603 C3 Capacitor, Electrolytic 2 10MV1200GX, 1200mF, 10V C4 Capacitor, Ceramic 1 1mF, 0805 C5 Sanyo Capacitor, Ceramic 1 220pF, 0603 C7, 14, 15 Capacitor, Ceramic 3 1000pF, 0603 C8 Capacitor, Ceramic 1 0.1mF, 0603 C9 Capacitor, Electrolytic 1 6MV1000GX, 1000mF, 6.3V Sanyo C10 Capacitor, Electrolytic 6 6MV1500GX, 1500mF, 6.3V Sanyo C11 Capacitor, Electrolytic 1 6MV150GX, 150mF, 6.3V Sanyo C12 Capacitor, Electrolytic 1 6MV1000GX, 1000mF, 6.3V Sanyo C13 Capacitor, Electrolytic 1 10MV470GX, 470mF, 10V Sanyo C16 Capacitor, Ceramic 1 4.7mF, 1206 R1 Resistor 1 3.3KV, 5%, 0603 R2, 3, 4 Resistor 3 4.7V, 5%, 1206 R5, 15 Resistor 2 10KV, 5%, 0603 R7 Resistor 1 267V, 1%, 0603 R8 Resistor 2 150V, 1%, 0603 R9, 11, 14 Resistor 3 100V, 5%, 0603 R12 Resistor 1 100V, 1%, 0603 R13 Resistor 1 22KV, 1%, 0603 R16 Resistor 1 220V, 1%, 0603 R17 Resistor 1 330V, 1%, 0603 R18 Resistor 1 10V, 5%, 0603 Note 1: R16, R17, C16, R12, and R13 set the Vcore 2% higher for level shift to reduce CPU transient voltage. Rev. 1.7 07/16/02 www.irf.com 9 IRU3004 APPLICATION INFORMATION An example of how to calculate the components for the application circuit is given below. Assuming, two sets of output conditions that this regulator must meet: a) Vo=2.8V, Io=14.2A, DVo=185mV, DIo=14.2A b) Vo=2V, Io=14.2A, DVo=140mV, DIo=14.2A The regulator design will be done such that it meets the worst case requirement of each condition. Output Capacitor Selection The first step is to select the output capacitor. This is done primarily by selecting the maximum ESR value that meets the transient voltage budget of the total DVo specification. Assuming that the regulators DC initial accuracy plus the output ripple is 2% of the output voltage, then the maximum ESR of the output capacitor is calculated as: ESR [ 100 = 7mV 14.2 The Sanyo MVGX series is a good choice to achieve both the price and performance goals. The 6MV1500GX, 1500mF, 6.3V has an ESR of less than 36mV typical. Selecting 6 of these capacitors in parallel has an ESR of ≈ 6mV which achieves our low ESR goal. output capacitor ESR at the cost of load regulation. One can show that the new ESR requirement eases up by half the total trace resistance. For example, if the ESR requirement of the output capacitors without voltage level shifting must be 7mV, then after level shifting the new ESR will only need to be 9.5mV if the trace resistance is 5mV (7 + 5/2=9.5). However, one must be careful that the combined “voltage level shifting” and the transient response is still within the maximum tolerance of the Intel specification. To insure this, the maximum trace resistance must be less than: Rs [ 2 3 (Vspec - 0.02 3 Vo - DVo) DI Where: Rs = Total maximum trace resistance allowed Vspec = Intel total voltage specification Vo = Output voltage DVo = Output ripple voltage DI = load current step For example, assuming: Vspec = ±140mV = ±0.1V for 2V output Vo = 2V DVo = assume 10mV = 0.01V DI = 14.2A Then the Rs is calculated to be: Other type of Electrolytic capacitors from other manufacturers to consider are the Panasonic FA series or the Nichicon PL series. Reducing the Output Capacitors Using Voltage Level Shifting Technique The trace resistance or an external resistor from the output of the switching regulator to the Slot 1 can be used to the circuit advantage and possibly reduce the number of output capacitors, by level shifting the DC regulation point when transition from light load to full load and vice versa. To accomplish this, the output of the regulator is typically set about half the DC drop that results from light load to full load. For example, if the total resistance from the output capacitors to the Slot 1 and back to the Gnd pin of the device is 5mV and if the total DI, the change from light load to full load is 14A, then the output voltage measured at the top of the resistor divider which is also connected to the output capacitors in this case, must be set at half of the 70mV or 35mV higher than the DAC voltage setting. This intentional voltage level shifting during the load transient eases the requirement for the 10 Rs [ 2 3 (0.140 - 0.02 3 2 - 0.01) = 12.6mV 14.2 However, if a resistor of this value is used, the maximum power dissipated in the trace (or if an external resistor is being used) must also be considered. For example if Rs=12.6mV, the power dissipated is: Io23Rs = 14.22312.6 = 2.54W This is a lot of power to be dissipated in a system. So, if the Rs=5mV, then the power dissipated is about 1W which is much more acceptable. If level shifting is not implemented, then the maximum output capacitor ESR was shown previously to be 7mV which translated to ≈ 6 of the 1500mF, 6MV1500GX type Sanyo capacitors. With Rs=5mV, the maximum ESR becomes 9.5mV which is equivalent to ≈ 4 caps. Another important consideration is that if a trace is being used to implement the resistor, the power dissipated by the trace increases the case temperature of the output capacitors which could seriously effect the life time of the output capacitors. www.irf.com Rev. 1.7 07/16/02 IRU3004 Output Inductor Selection The output inductance must be selected such that under low line and the maximum output voltage condition, the inductor current slope times the output capacitor ESR is ramping up faster than the capacitor voltage is drooping during a load current step. However, if the inductor is too small, the output ripple current and ripple voltage become too large. One solution to bring the ripple current down is to increase the switching frequency, however that will be at the cost of reduced efficiency and higher system cost. The following set of formulas are derived to achieve the optimum performance without many design iterations. The maximum output inductance is calculated using the following equation: (V L = ESR3C3 - Vo(MAX) 2 3 ∆I IN(MIN) ) Where: VIN(MIN) = Minimum input voltage Vo = 2.8V , DI = 14.2A L = 0.006390003 In our example for Vo=2.8V and 14.2A load, assuming IRL3103 MOSFET for both switches with maximum onresistance of 19mV, we have: 1 = 5ms 200000 Vsw = Vsync = 14.2 3 0.019 = 0.27V 2.8 + 0.27 D≈ = 0.61 5 - 0.27 + 0.27 TON = 0.61 3 5 = 3.1ms T= TOFF = 5 - 3.1 = 1.9ms 1.9 DIr = (2.8 + 0.27) 3 = 1.94A 3 DVo = 1.94 3 0.006 = 0.011V = 11mV Power Component Selection Assuming IRL3103 MOSFETs as power components, we will calculate the maximum power dissipation as follows: For high-side switch the maximum power dissipation happens at maximum Vo and maximum duty cycle. - 2.8 = 3.7mH ( 4.75 2314.2 ) Assuming that the programmed switching frequency is set at 200KHz, an inductor is designed using the Micrometals’ powder iron core material. The summary of the design is outlined below: The selected core material is Powder Iron, the selected core is T50-52D from Micro Metal wound with 8 turns of #16 AWG wire, resulting in 3mH inductance with ≈ 3mV of DC resistance. (2.8 + 0.27) DMAX ≈ (4.75 - 0.27 + 0.27) = 0.65 PDH = DMAX 3 Io2 3 RDS(MAX) PDH = 0.65 3 14.22 3 0.029 = 3.8W RDS(MAX) = Maximum RDS(ON) of the MOSFET (1258C) For synchronous MOSFET, maximum power dissipation happens at minimum Vo and minimum duty cycle. DMIN ≈ (2 + 0.27) = 0.43 (5.25 - 0.27 + 0.27) PDS = (1 - DMIN) 3 Io2 3 RDS(MAX) Assuming L=3mH and Fsw=200KHz (switching frequency), the inductor ripple current and the output ripple voltage is calculated using the following set of equations: T ≡ Switching Period D ≡ Duty Cycle Vsw ≡ High side Mosfet ON Voltage RDS ≡ Mosfet On Resistance Vsync ≡ Synchronous MOSFET ON Voltage DIr ≡ Inductor Ripple Current DVo ≡ Output Ripple Voltage 1 Fsw Vsw = Vsync = Io3RDS T= D≈ Vo + Vsync VIN - Vsw + Vsync Rev. 1.7 07/16/02 TON = D3T TOFF = T - TON DIr = (Vo + Vsync)3 TOFF L PDS = (1 - 0.43) 3 14.22 3 0.029 = 3.33W Heat Sink Selection Selection of the heat sink is based on the maximum allowable junction temperature of the MOSFETS. Since we previously selected the maximum RDS(on) at 1258C, then we must keep the junction below this temperature. Selecting TO-220 package gives uJC=1.88C/W (from the venders’ data sheet) and assuming that the selected heat sink is black anodized, the heat-sink-to-case thermal resistance is uCS=0.058C/W, the maximum heat sink temperature is then calculated as: Ts = TJ - PD 3 (uJC + uCS) Ts = 125 - 3.82 3 (1.8 + 0.05) = 1188C DVo = DIr3ESR www.irf.com 11 IRU3004 With the maximum heat sink temperature calculated in the previous step, the heat-sink-to-air thermal resistance (uSA) is calculated as follows: Assuming TA = 358C: Switcher Timing Capacitor Selection The switching frequency can be programmed using an external timing capacitor. The value of Ct can be approximated using the equation below: Fsw y DT = Ts - TA = 118 - 35 = 838C Temperature Rise Above Ambient 3.5 3 10-5 Ct Where: Ct = Timing Capacitor Fsw = Switching Frequency DT 83 uSA = = = 228C/W PD 3.82 Next, a heat sink with lower uSA than the one calculated in the previous step must be selected. One way is to simply look at the graphs of the “Heat Sink Temp Rise Above the Ambient” vs. the “Power Dissipation” given in the heat sink manufacturers’ catalog and select a heat sink that results in lower temperature rise than the one calculated in previous step. The following heat sinks from AAVID and Thermalloy meet this criteria. Company Part # Thermalloy............................6078B AAVID..................................577002 If Fsw = 200KHz: Ct y 3.5 3 10-5 = 175pF 200 3 103 LDO Power MOSFET Selection The first step in selecting the power MOSFET for the linear regulators is to select its maximum RDS(ON) based on the input to output Dropout voltage and the maximum load current. For Vo = 1.5V, VIN = 3.3V and IL = 2A: RDS(max) = Following the same procedure for the Schottky diode results in a heat sink with uSA=258C/W. Although it is possible to select a slightly smaller heat sink, for simplicity, the same heat sink as the one for the high side MOSFET is also selected for the synchronous MOSFET. Switcher Current Limit Protection The PWM controller uses the MOSFET RDS(ON) as the sensing resistor to sense the MOSFET current and compares to a programmed voltage which is set externally via a resistor (Rcs) placed between the drain of the MOSFET and the “CS+” terminal of the IC as shown in the application circuit. For example, if the desired current limit point is set to be 22A and from our previous selection, the maximum MOSFET RDS(ON)=19mV, then the current sense resistor, Rcs is calculated as: Where: IB = 200mA is the internal current setting of the device (V IN - Vo) (3.3 - 1.5) = = 0.9V IL 2 Note that since the MOSFETs RDS(ON) increases with temperature, this number must be divided by ≈ 1.5, in order to find the RDS(on) max at room temperature. The Motorola MTP3055VL has a maximum of 0.18V RDS(ON) at room temperature, which meets our requirement. To select the heat sink for the LDO MOSFET the first step is to calculate the maximum power dissipation of the device and then follow the same procedure as for the switcher. PD = (V IN - Vo) 3 IL Where: PD = Power Dissipation of the Linear Regulator IL = Linear Regulator Load Current For the 1.5V and 2A load: PD = (3.3 - 1.5) 3 2 = 3.6W Assuming TJ(max) = 1258C then: Vcs = ICL 3 RDS = 22 3 0.019 = 0.418V Vcs 0.418V Rcs = = = 2.1KV IB 200mA 12 Ts = TJ - PD 3 (uJC + uCS) Ts = 125 - 3.6 3 (1.8 + 0.05) = 118°C www.irf.com Rev. 1.7 07/16/02 IRU3004 With the maximum heat sink temperature calculated in the previous step, the heat-sink-to-air thermal resistance (uSA) is calculated as follows: Disabling the LDO Regulators The LDO controllers can easily be disabled by connecting the feedback pins (V FB1 and VFB2) to a voltage higher than 1.5V such as 5V for all devices. Assuming TA = 358C: DT = Ts - TA = 118 - 35 = 838C Temperature Rise Above Ambient θSA = DT 83 = = 238C/W PD 3.6 The same heat sink as the one selected for the switcher MOSFETs is also suitable for the 1.5V regulator. It is also possible to use TO-263 package or even the MTD3055VL in D-Pak if the load current is less than 1.5A. For the 2.5V regulator, since the dropout voltage is only 0.8V and the load current is less than 0.5A, for most applications, the same MOSFET without heat sink or for low cost applications, one can use PN2222A in TO-92 or SOT-23 package. LDO Regulator Component Selection Since the internal voltage reference for the linear regulators is set at 1.5V for all devices, there is no need to divide the output voltage for the 1.5V, GTL+ regulator. For the 2.5V Clock supply, the resistor dividers are selected per following: ( Vo = 1+ Rt RB )3V V (V) ((Vo - 1.0043V )) 2.8 R13 = 1003 ( = 11.76KV (2.835 - 1.00432.800)) = DAC DAC Select 11.8KV, 1% Note: The value of the top resistor must not exceed 100V. The bottom resistor can then be adjusted to raise the output voltage. Assuming Rt = 100V, for Vo = 2.5V: Rt Vo VREF For example, if DAC voltage setting is for 2.8V and the desired output under light load is 2.835V, then R13 is calculated using the following formula: R13 = 1003 REF Where: Rt = Top resistor divider RB = Bottom resistor divider Vref = 1.5V typical RB = Switcher Output Voltage Adjust As was discussed earlier, the trace resistance from the output of the switching regulator to the Slot 1 can be used to the circuit advantage and possibly reduce the number of output capacitors, by level shifting the DC regulation point when transitioning from light load to full load and vice versa. To account for the DC drop, the output of the regulator is typically set about half the DC drop that results from light load to full load. For example, if the total resistance from the output capacitors to the Slot 1 and back to the Gnd pin of the part is 5mV and if the total DI, the change from light load to full load is 14A, then the output voltage measured at the top of the resistor divider which is also connected to the output capacitors in this case, must be set at half of the 70mV or 35mV higher than the DAC voltage setting. To do this, the top resistor of the resistor divider (R12 in the application circuit) is set at 100V, and the R13 is calculated. 100 = 150V 2.5 - 1 1.5 ( )-1 ( ) For 1.5V output, Rt can be shorted and RB left open. However, it is recommended to leave the resistor dividers as shown in the typical application circuit so that the output voltage can be adjusted higher to account for the trace resistance in the final board layout. It is also recommended that an external filter be added on the linear regulators to reduce the amount of the high frequency ripple at the output of the regulators. This can simply be done by the resistor capacitor combination as shown in the application circuit. Soft-Start Capacitor Selection The soft-start capacitor must be selected such that during the start up, when the output capacitors are charging up, the peak inductor current does not reach the current limit threshold. A minimum of 1mF capacitor insures this for most applications. An internal 10mA current source charges the soft-start capacitor which slowly ramps up the inverting input of the PWM comparator VFB3. This insures the output voltage to ramp at the same rate as the soft-start cap thereby limiting the input current. For example, with 1mF and the 10mA internal current source the ramp up rate is (DV/Dt)=(I/C)=1V/100ms. Assuming that the output capacitance is 9000mF, the maximum start up current will be: I = 9000mF 3 (1V / 100ms) = 0.09A Rev. 1.7 07/16/02 www.irf.com 13 IRU3004 Input Filter It is recommended to place an inductor between the system 5V supply and the input capacitors of the switching regulator to isolate the 5V supply from the switching noise that occurs during the turn on and off of the switching components. Typically an inductor in the range of 1 to 3mH will be sufficient in this type of application. Switcher External Shutdown The best way to shutdown the switcher is to pull down on the soft-start pin using an external small signal transistor such as 2N3904 or 2N7002 small signal MOSFET. This allows slow ramp up of the output, the same as the power up. Layout Considerations Switching regulators require careful attention to the layout of the components, specifically power components since they switch large currents. These switching components can create large amount of voltage spikes and high frequency harmonics if some of the critical components are far away from each other and are connected with inductive traces. The following is a guideline of how to place the critical components and the connections between them in order to minimize the above issues. Start the layout by first placing the power components: 1) Place the input capacitors C3 and the high side MOSFET, Q1 as close to each other as possible. 2) Place the synchronous MOSFET, Q2 and the Q1 as close to each other as possible with the intention that the source of Q1 and drain of the Q2 has the shortest length. 3) Place the snubber R4 & C7 between Q1 & Q2. 4) Place the output inductor, L2 and the output capacitors, C10 between the MOSFET and the load with output capacitors distributed along the slot 1 and close to it. 5) Place the bypass capacitors, C4 and C6 right next to 12V and 5V pins. C4 next to the 12V, pin 12 and C6 next to the 5V, pin 5. 6) Place the controller IC such that the PWM output drives, pins 9 and 11 are relatively short distance from gates of Q1 and Q2. 7) Place resistor dividers, R7 & R8 close to pin 3, R12 & R13 (see note) close to pin 14 and R14 and R15 (see note) close to pin 20. 14 Note: Although, the PWM controller does not require R12-15 resistors, and the feedback pins 3 and 14 can be directly connected to their respective outputs, they can be used to set the outputs slightly higher to account for any output drop at the load due to the trace resistance. 8) Place R11, C15, Q3 and C11 close to each other and do the same with R9, C14, Q4 and C12. Note: It is better to place the linear regulator components close to the IC and then run a trace from the output of each regulator to its respective load such as 2.5V to the clock and 1.5V for GTL + termination. However, if this is not possible then the trace from the linear drive output pins, pins 2 and 20 must be routed away from any high frequency data signals. It is critical, to place high frequency ceramic capacitors close to the clock chip and termination resistors to provide local bypassing. 9) Place timing capacitor C1 close to pin 1 and soft start capacitor C2 close to pin 13. Component connections: Note: It is extremely important that no data bus should be passing through the switching regulator section specifically close to the fast transition nodes such as PWM drives or the inductor voltage. Using the 4 layer board, dedicate on layer to ground, another layer as the power layer for the 5V, 3.3V, Vcore, 1.5V and if it is possible for the 2.5V. Connect all grounds to the ground plane using direct vias to the ground plane. Use large low inductance/low impedance plane to connect the following connections either using component side or the solder side: a) b) c) d) e) f) g) h) C3 to Q1 Drain Q1 Source to Q2 Drain Q2 drain to L2 L2 to the output capacitors, C10 C10 to the slot 1 Input filter L1 to the C3 C9 to Q4 drain C12 to the Q4 source Connect the rest of the components using the shortest connection possible. IR WORLD HEADQUARTERS : 233 Kansas St., El Segundo,California 90245, USA Tel: (310) 252-7105 TAC Fax: (310) 252-7903 Visit us at www.irf.com for sales contact information Data and specifications subject to change without notice. 02/01 www.irf.com Rev. 1.7 07/16/02 IRU3004 (F) TSSOP Package 20-Pin A L Q R1 B 1.0 DIA C R E N M P O PIN NUMBER 1 F D DETAIL A DETAIL A G J H K SYMBOL DESIG A B C D E F G H J K L M N O P Q R R1 MIN 4.30 0.19 6.40 --0.85 0.05 08 0.50 0.09 0.09 20-PIN NOM MAX 0.65 BSC 4.40 6.40 BSC --1.00 1.00 6.50 --0.90 --128 REF 128 REF --1.00 REF 0.60 0.20 ----- 4.50 0.30 6.60 1.10 0.95 0.15 88 0.75 ----- NOTE: ALL MEASUREMENTS ARE IN MILLIMETERS. Rev. 1.7 07/16/02 www.irf.com 15 IRU3004 (W) SOIC Package 20-Pin Surface Mount, Wide Body H A B C R E DETAIL-A PIN NO. 1 L D 0.516 0.020 x 458 DETAIL-A K F G SYMBOL A B C D E F G I J K L R T I T J 20-PIN MIN MAX 12.598 12.979 1.018 1.524 0.66 REF 0.33 0.508 7.40 7.60 2.032 2.64 0.10 0.30 0.229 0.32 10.008 10.654 08 88 0.406 1.270 0.63 0.89 2.337 2.642 NOTE: ALL MEASUREMENTS ARE IN MILLIMETERS. 16 www.irf.com Rev. 1.7 07/16/02 IRU3004 PACKAGE SHIPMENT METHOD PKG DESIG PACKAGE DESCRIPTION PIN COUNT PARTS PER TUBE PARTS PER REEL T&R Orientation F TSSOP Plastic 20 74 2500 Fig A W SOIC, Wide Body 20 38 1000 Fig B 1 1 1 1 Feed Direction Figure A 1 1 Feed Direction Figure B IR WORLD HEADQUARTERS: 233 Kansas St., El Segundo, California 90245, USA Tel: (310) 252-7105 TAC Fax: (310) 252-7903 Visit us at www.irf.com for sales contact information Data and specifications subject to change without notice. 02/01 Rev. 1.7 07/16/02 www.irf.com 17