LT1513-2CT7#PBF

LT1513/LT1513-2 SEPIC Constant- or Programmable-Current/ Constant-Voltage Battery Charger U DESCRIPTION FEATURES ■ ■ ■ ■ ■ ■ ■ ■ Charger Input Voltage May Be Higher, Equal to or Lower Than Battery Voltage Charges Any Number of Cells Up to 20V 1% Voltage Accuracy for Rechargeable Lithium Batteries 100mV Current Sense Voltage for High Efficiency (LT1513) 0mV Current Sense Voltage for Easy Current Programming (LT1513-2) Battery Can Be Directly Grounded 500kHz Switching Frequency Minimizes Inductor Size Charging Current Easily Programmable or Shut Down U APPLICATIONS ■ ■ ■ ■ ■ The LT ®1513 is a 500kHz current mode switching regulator specially configured to create a constant- or programmable-current/constant-voltage battery charger. In addition to the usual voltage feedback node, it has a current sense feedback circuit for accurately controlling output current of a flyback or SEPIC (Single-Ended Primary Inductance Converter) topology charger. These topologies allow the current sense circuit to be ground referred and completely separated from the battery itself, simplifying battery switching and system grounding problems. In addition, these topologies allow charging even when the input voltage is lower than the battery voltage. The LT1513 can also drive a CCFL Royer converter with high efficiency in floating or grounded mode. Maximum switch current on the LT1513 is 3A. This allows battery charging currents up to 2A for a single lithium-ion cell. Accuracy of 1% in constant-voltage mode is perfect for lithium battery applications. Charging current can be easily programmed for all battery types. Charging of NiCd, NiMH, Lead-Acid or Lithium Rechargeable Cells Precision Current Limited Power Supply Constant-Voltage/Constant-Current Supply Transducer Excitation Universal Input CCFL Driver , LTC and LT are registered trademarks of Linear Technology Corporation. U TYPICAL APPLICATION • + Maximum Charging Current L1A* VIN CHARGE VSW 6 S/S GND 4 TAB 1 R5 270Ω C5 0.1µF 3 2.2 D1 2 1.8 R1 • R2 R4 39Ω C4 0.22µF SINGLE Li-Ion CELL (4.1V) 2.0 1.25A L1B* VFB IFB VC † 5 LT1513 SYNC AND/OR SHUTDOWN SHUTDOWN 2.4 C2** 4.7µF 7 C3 22µF 25V + C1 22µF 25V ×2 R3 0.08Ω DOUBLE Li-Ion CELL (8.2V) 1.6 1.4 12V 1.2 16V 1.0 20V 0.8 BATTERY VOLTAGE 0.6 LT1513 • TA01 * L1A, L1B ARE TWO 10µH WINDINGS ON A COMMON CORE: COILTRONICS CTX10-4 ** CERAMIC MARCON THCR40EIE475Z OR TOKIN 1E475ZY5U-C304 † MBRD340 OR MBRS340T3. MBRD340 HAS 5µA TYPICAL LEAKAGE, MBRS340T3 50µA TYPICAL Figure 1. SEPIC Charger with 1.25A Output Current CURRENT (A) WALL ADAPTER INPUT 0.4 0 5 10 20 15 INPUT VOLTAGE (V) 25 30 INDUCTOR = 10µH ACTUAL PROGRAMMED CHARGING CURRENT WILL BE INDEPENDENT OF INPUT VOLTAGE IF IT DOES NOT EXCEED VALUES SHOWN LT1513 • TA02 sn1513 1513fas 1 LT1513/LT1513-2 W W W AXI U U ABSOLUTE RATI GS Supply Voltage ....................................................... 30V Switch Voltage ........................................................ 40V S/S Pin Voltage ....................................................... 30V FB Pin Voltage (Transient, 10ms) ......................... ±10V VFB Pin Current .................................................... 10mA IFB Pin Voltage (Transient, 10ms) ......................... ±10V Operating Junction Temperature Range LT1513C ............................................... 0°C to 125°C LT1513I ............................................ – 40°C to 125°C Short Circuit ......................................... 0°C to 150°C Storage Temperature Range ................ – 65°C to 150°C Lead Temperature (Soldering, 10 sec)................. 300°C U W U PACKAGE/ORDER I FOR ATIO FRONT VIEW 7 6 5 4 3 2 1 TAB IS GND VIN S/S VSW GND IFB FB VC R PACKAGE 7-LEAD PLASTIC DD ORDER PART NUMBER ORDER PART NUMBER FRONT VIEW LT1513CR LT1513-2CR LT1513IR LT1513-2IR LT1513-2CT7 LT1513-2IT7 VIN S/S VSW GND IFB FB VC 7 6 5 4 3 2 1 T7 PACKAGE 7-LEAD TO-220 TJMAX = 125°C, θJA = 30°C/ W TJMAX = 125°C, θJA = 50°C/ W, θJC = 4°C/W WITH PACKAGE SOLDERED TO 0.5INCH2 COPPER AREA OVER BACKSIDE GROUND PLANE OR INTERNAL POWER PLANE, θJA CAN VARY FROM 20°C/W TO > 40°C/W DEPENDING ON MOUNTING TECHNIQUE Consult factory for Military grade parts. ELECTRICAL CHARACTERISTICS VIN = 5V, VC = 0.6V, VFB = VREF, IFB = 0V, VSW and S/S pins open, unless otherwise noted. SYMBOL PARAMETER CONDITIONS VREF FB Reference Voltage Measured at FB Pin VC = 0.8V FB Input Current ● MIN TYP MAX UNITS 1.233 1.228 1.245 1.245 1.257 1.262 V V 300 550 600 nA nA VFB = VREF ● VIREF IFBVOS gm FB Reference Voltage Line Regulation 2.7V ≤ VIN ≤ 25V, VC = 0.8V IFB Reference Voltage (LT1513) Measured at IFB Pin VFB = 0V, VC = 0.8V IFB Input Current 0.01 0.03 %/V ● – 107 –110 – 100 – 100 – 93 – 90 mV mV VIFB = VIREF (Note 2) ● 10 25 35 µA IFB Reference Voltage Line Regulation 2.7V ≤ VIN ≤ 25V, VC = 0.8V ● 0.01 0.05 %/V IFB Voltage Offset (LT1513-2) (Note 3) IVFB = 60µA (Note 4) ● – 7.5 2.5 12.5 mV IFB Input Current VIFB = VIREF ● – 200 – 10 0 nA VFB Source Current VIREF = – 10mV, VFB = 1.2V ● – 700 – 300 – 100 µA Error Amplifier Transconductance ∆IC = ±25µA 1500 ● 1100 700 1900 2300 µmho µmho 120 200 350 µA 1400 2400 µA ● Error Amplifier Source Current VFB = VREF – 150mV, VC = 1.5V ● Error Amplifier Sink Current VFB = VREF + 150mV, VC = 1.5V ● sn1513 1513fas 2 LT1513/LT1513-2 ELECTRICAL CHARACTERISTICS VIN = 5V, VC = 0.6V, VFB = VREF, IFB = 0V, VSW and S/S pins open, unless otherwise noted. SYMBOL PARAMETER MIN TYP MAX UNITS High Clamp, VFB = 1V Low Clamp, VFB = 1.5V 1.70 0.25 1.95 0.40 2.30 0.52 V V VC Pin Threshold Duty Cycle = 0% 0.8 1 1.25 V Switching Frequency 2.7V ≤ VIN ≤ 25V 0°C ≤ TJ ≤ 125°C TJ < 0°C 450 430 400 500 500 550 580 580 kHz kHz kHz 260 ns Error Amplifier Clamp Voltage AV f CONDITIONS Error Amplifier Voltage Gain 500 Maximum Switch Duty Cycle ● 85 Switch Current Limit Blanking Time Output Switch Breakdown Voltage 0°C ≤ TJ ≤ 125°C TJ < 0°C VSAT Output Switch ON Resistance ISW = 2A ● ILIM Switch Current Limit Duty Cycle = 50% Duty Cycle = 80% (Note 1) ● ● 40 35 3.0 2.6 ∆IIN/∆ISW Supply Current Increase During Switch ON Time Control Voltage to Switch Current Transconductance % 47 V V 0.25 0.45 Ω 3.8 3.4 5.4 5.0 A A 15 25 mA/A 4 Minimum Input Voltage IQ 95 130 BV V/ V A/V ● 2.4 2.7 V Supply Current 2.7V ≤ VIN ≤ 25V ● 4 5.5 mA Shutdown Supply Current 2.7V ≤ VIN ≤ 25V, VS/S ≤ 0.6V, TJ ≥ 0°C TJ < 0°C ● 12 30 50 µA µA Shutdown Threshold 2.7V ≤ VIN ≤ 25V ● 0.6 1.3 2 V ● 5 12 25 µs ● – 10 15 µA ● 600 800 kHz Shutdown Delay S/S Pin Input Current Synchronization Frequency Range The ● denotes specifications which apply over the full operating temperature range. Note 1: For duty cycles (DC) between 50% and 85%, minimum guaranteed switch current is given by ILIM = 1.33 (2.75 – DC). 0V ≤ VS/S ≤ 5V Note 2: The IFB pin is servoed to its regulating state with VC = 0.8V. Note 3: Consult factory for grade selected parts. Note 4: The IFB pin is sevoed to regulate FB to 1.245V sn1513 1513fas 3 LT1513/LT1513-2 U W TYPICAL PERFORMANCE CHARACTERISTICS Switch Saturation Voltage vs Switch Current Switch Current Limit vs Duty Cycle 6 150°C 100°C 0.9 25°C SWITCH CURRENT LIMIT (A) 0.8 0.7 0.6 0.5 –55°C 0.4 0.3 0.2 3.0 5 2.8 25°C AND 125°C 4 INPUT VOLTAGE (V) 1.0 SWITCH SATURATION VOLTAGE (V) Minimum Input Voltage vs Temperature –55°C 3 2 2.6 2.4 2.2 2.0 1 0.1 0 0 0 0 0.4 0.8 1.2 1.6 2.0 2.4 2.8 3.2 3.6 4.0 SWITCH CURRENT (A) LT1513 • G03 12 0 BATTERY VOLTAGE (V) –10 –20 –30 VIN = 12V MAXIMUM AVAILABLE CHARGING CURRENT WITH 12V INPUT 10 8 (A) (A) 8.4V BATTERY ICHRG = 0.5A (B) 6 (B) 8.4V BATTERY ICHRG = 1A 4 (C) 4.2V BATTERY ICHRG = 1.5A (C) –40 2 –50 –50 –25 0 0 25 50 75 100 125 150 TEMPERATURE (°C) 0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 CHARGING CURRENT (A) 1513 G07 LT1513 • G06 Feedback Input Current vs Temperature Minimum Peak-to-Peak Synchronization Voltage vs Temperature 800 fSYNC = 700kHz FEEDBACK INPUT CURRENT (nA) MINIMUM SYNCHRONIZATION VOLTAGE (VP-P) 25 50 75 100 125 150 TEMPERATURE (°C) Output Charging Characteristics Showing Constant-Current and Constant-Voltage Operation Negative Feedback Input Current vs Temperature 3.0 0 LT1513 • G02 LT1513 • G01 NEGATIVE FEEDBACK INPUT CURRENT (µA) 1.8 –50 –25 10 20 30 40 50 60 70 80 90 100 DUTY CYCLE (%) 2.5 2.0 1.5 1.0 0.5 0 –50 –25 700 VFB = VREF 600 500 400 300 200 100 0 25 50 75 100 125 150 TEMPERATURE (°C) LT1513 • G04 0 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) LT1513 • G05 sn1513 1513fas 4 LT1513/LT1513-2 U U U PIN FUNCTIONS VC (Pin 1): The compensation pin is primarily used for frequency compensation, but it can also be used for soft starting and current limiting. It is the output of the error amplifier and the input of the current comparator. Peak switch current increases from 0A to 3.6A as the VC voltage varies from 1V to 1.9V. Current out of the VC pin is about 200µA when the pin is externally clamped below the internal 1.9V clamp level. Loop frequency compensation is performed with a capacitor or series RC network from the VC pin directly to the ground pin (avoid ground loops). FB (Pin 2): The feedback pin is used for positive output voltage sensing. The R1/R2 voltage divider connected to FB defines Li-Ion float voltage at full charge, or acts as a voltage limiter for NiCd or NiMH applications. FB is the inverting input to the voltage error amplifier. Input bias current is typically 300nA, so divider current is normally set to 100µA to swamp out any output voltage errors due to bias current. The noninverting input of this amplifier is tied internally to a 1.245V reference. The grounded end of the output voltage divider should be connected directly to the LT1513 ground pin (avoid ground loops). IFB (Pin 3): The current feedback pin is used to sense charging current. It is the input to a current sense amplifier that controls charging current when the battery voltage is below a programmed limit. During constant-current operation, the LT1513 IFB pin regulates at – 100mV. Input resistance of this pin is 5kΩ, so filter resistance (R4, Figure 1) should be less than 50Ω. The 39Ω, 0.22µF filter shown in Figure 1 is used to convert the pulsating current in the sense resistor to a smooth DC current feedback signal. The LT1513-2 IFB pin regulates at 0mV to provide programmable current limit. The current through R5, Figure 5, is balanced by the current through R4, programming the maximum voltage across R3. GND (Pin 4): The ground pin is common to both control circuitry and switch current. VC, FB and S/S signals must be Kelvin and connected as close as possible to this pin. The TAB of the R package should also be connected to the power ground. VSW (Pin 5): The switch pin is the collector of the power switch, carrying up to 3A of current with fast rise and fall times. Keep the traces on this pin as short as possible to minimize radiation and voltage spikes. In particular, the path in Figure 1 which includes SW to C2, D1, C1 and around to the LT1513 ground pin should be as short as possible to minimize voltage spikes at switch turn-off. S/S (Pin 6): This pin can be used for shutdown and/or synchronization. It is logic level compatible, but can be tied to VIN if desired. It defaults to a high ON state when floated. A logic low state will shut down the charger to a micropower state. Driving the S/S pin with a continuous logic signal of 600kHz to 800kHz will synchronize switching frequency to the external signal. Shutdown is avoided in this mode with an internal timer. VIN (Pin 7): The input supply pin should be bypassed with a low ESR capacitor located right next to the IC chip. The grounded end of the capacitor must be connected directly to the ground plane to which the TAB is connected. TAB: The TAB on the surface mount R package is electrically connected to the ground pin, but a low inductance connection must be made to both the TAB and the pin for proper circuit operation. See suggested PC layout in Figure 4. sn1513 1513fas 5 LT1513/LT1513-2 W BLOCK DIAGRAM VIN SHUTDOWN DELAY AND RESET S/S SYNC SW LOW DROPOUT 2.3V REG 500kHz OSC ANTISAT LOGIC DRIVER SWITCH + 4k IFB IFBA – COMP 50k* – – + VFB + EA IA AV ≈ 6 VC 1.245V REF *REMOVE ON LT1513-2 0.04Ω – LT1513 • BD Figure 2 U OPERATION The LT1513 is a current mode switcher. This means that switch duty cycle is directly controlled by switch current rather than by output voltage or current. Referring to the Block Diagram, the switch is turned “on” at the start of each oscillator cycle. It is turned “off” when switch current reaches a predetermined level. Control of output voltage and current is obtained by using the output of a dual feedback voltage sensing error amplifier to set switch current trip level. This technique has the advantage of simplified loop frequency compensation. A low dropout internal regulator provides a 2.3V supply for all internal circuitry on the LT1513. This low dropout design allows input voltage to vary from 2.7V to 25V. A 500kHz oscillator is the basic clock for all internal timing. It turns “on” the output switch via the logic and driver circuitry. Special adaptive antisat circuitry detects onset of saturation in the power switch and adjusts driver current instantaneously to limit switch saturation. This minimizes driver dissipation and provides very rapid turn-off of the switch. A unique error amplifier design has two inverting inputs which allow for sensing both output voltage and current. A 1.245V bandgap reference biases the noninverting input. The first inverting input of the error amplifier is brought out for positive output voltage sensing. The second inverting input is driven by a “current” amplifier which is sensing output current via an external current sense resistor. The current amplifier is set to a fixed gain of – 12.5 which provides a – 100mV current limit sense voltage. The LT1513-2 option removes the feedback resistors around the IFB amplifier and connects its output to the FB signal. This provides a ground referenced current sense voltage suitable for external current programming and makes amplifier input and output available for external loop compensation. The error signal developed at the amplifier output is brought out externally and is used for frequency compensation. During normal regulator operation this pin sits at a voltage between 1V (low output current) and 1.9V (high output current). Switch duty cycle goes to zero if the VC pin is pulled below the VC pin threshold, placing the LT1513 in an idle mode. sn1513 1513fas 6 LT1513/LT1513-2 U W U U APPLICATIONS INFORMATION The LT1513 is an IC battery charger chip specifically optimized to use the SEPIC converter topology. A complete charger schematic is shown in Figure 1. The SEPIC topology has unique advantages for battery charging. It will operate with input voltages above, equal to or below the battery voltage, has no path for battery discharge when turned off, and eliminates the snubber losses of flyback designs. It also has a current sense point that is ground referred and need not be connected directly to the battery. The two inductors shown are actually just two identical windings on one inductor core, although two separate inductors can be used. A current sense voltage is generated with respect to ground across R3 in Figure 1. The average current through R3 is always identical to the current delivered to the battery. The LT1513 current limit loop will servo the voltage across R3 to – 100mV when the battery voltage is below the voltage limit set by the output divider R1/R2. Constant-current charging is therefore set at 100mV/R3. R4 and C4 filter the current signal to deliver a smooth feedback voltage to the IFB pin. R1 and R2 form a divider for battery voltage sensing and set the battery float voltage. The suggested value for R2 is 12.4k. R1 is calculated from: R2(VBAT – 1.245) 1.245 + R2(0.3µA) VBAT = battery float voltage 0.3µA = typical FB pin bias current R1 = A value of 12.4k for R2 sets divider current at 100µA. This is a constant drain on the battery when power to the charger is off. If this drain is too high, R2 can be increased to 41.2k, reducing divider current to 30µA. This introduces an additional uncorrectable error to the constant voltage float mode of about ±0.5% as calculated by: VBAT Error = ± 0.15µA(R1)(R2) 1.245(R1+ R2) ±0.15µA = expected variation in FB bias current around the nominal 0.3µA typical value. With R2 = 41.2k and R1 = 228k, (VBAT = 8.2V), the error due to variations in bias current would be ±0.42%. A second option is to disconnect the divider when charger power is off. This can be done with a small NFET as shown in Figure 3. D2, C6 and R6 form a peak detector to drive the gate of the FET to about the same as the battery voltage. If power is turned off, the gate will drop to 0V and the only drain on the battery will be the reverse leakage of the catch diode D1. See Diode Selection for a discussion of diode leakage. L1A ADAPTER INPUT VIN C2 D1 R1 D2 VSW L1B + LT1513 C1 VFB GND R3 C6 470pF R6 470k SCHEMATIC SIMPLIFIED FOR CLARITY D2 = 1N914, 1N4148 OR EQUIVALENT R2 1513 F03 Figure 3. Eliminating Divider Current Maximum Input Voltage Maximum input voltage for the LT1513 is partly determined by battery voltage. A SEPIC converter has a maximum switch voltage equal to input voltage plus output voltage. The LT1513 has a maximum input voltage of 30V and a maximum switch voltage of 40V, so this limits maximum input voltage to 30V, or 40V – VBAT, whichever is less. Shutdown and Synchronization The dual function S/S pin provides easy shutdown and synchronization. It is logic level compatible and can be pulled high or left floating for normal operation. A logic low on the S/S pin activates shutdown, reducing input supply current to 12µA. To synchronize switching, drive the S/S pin between 600kHz and 800kHz. Inductor Selection L1A and L1B are normally just two identical windings on one core, although two separate inductors can be used. A typical value is 10µH, which gives about 0.5A peak-to-peak inductor current. Lower values will give higher ripple current, which reduces maximum charging current. 5µH can be used if charging currents are at least 20% lower than the values sn1513 1513fas 7 LT1513/LT1513-2 U W U U APPLICATIONS INFORMATION shown in the maximum charging current graph. Higher inductance values give slightly higher maximum charging current, but are larger and more expensive. A low loss toroid core such as Kool Mµ®, Molypermalloy or Metglas® is recommended. Series resistance should be less than 0.04Ω for each winding. “Open core” inductors, such as rods or barrels are not recommended because they generate large magnetic fields which may interfere with other electronics close to the charger. Input Capacitor The SEPIC topology has relatively low input ripple current compared to other topologies and higher harmonics are especially low. RMS ripple current in the input capacitor is less than 0.25A with L = 10µH and less than 0.5A with L = 5µH. A low ESR 22µF, 25V solid tantalum capacitor (AVX type TPS or Sprague type 593D) is adequate for most applications with the following caveat. Solid tantalum capacitors can be destroyed with a very high turn-on surge current such as would be generated if a low impedance input source were “hot switched” to the charger input. If this condition can occur, the input capacitor should have the highest possible voltage rating, at least twice the surge input voltage if possible. Consult with the capacitor manufacturer before a final choice is made. A 4.7µF ceramic capacitor such as the one used for the coupling capacitor can also be used. These capacitors do not have a turn-on surge limitation. The input capacitor must be connected directly to the VIN pin and the ground plane close to the LT1513. in Figure 1. These are AVX type TPS or Sprague type 593D surface mount solid tantalum units intended for switching applications. Do not substitute other types without ensuring that they have adequate ripple current ratings. See Input Capacitor section for details of surge limitation on solid tantalum capacitors if the battery may be “hot switched” to the output of the charger. Coupling Capacitor C2 in Figure 1 is the coupling capacitor that allows a SEPIC converter topology to work with input voltages either higher or lower than the battery voltage. DC bias on the capacitor is equal to input voltage. RMS ripple current in the coupling capacitor has a maximum value of about 1A at full charging current. A conservative formula to calculate this is: I (V + V )(1.1) ICOUP(RMS) = CHRG IN BAT 2(VIN ) (1.1 is a fudge factor to account for inductor ripple current and other losses) With ICHRG = 1.2A, VIN = 15V and VBAT = 8.2V, ICOUP = 1.02A. The recommended capacitor is a 4.7µF ceramic type from Marcon or Tokin. These capacitors have extremely low ESR and high ripple current ratings in a small package. Solid tantalum units can be substituted if their ripple current rating is adequate, but typical values will increase to 22µF or more to meet the ripple current requirements. Output Capacitor Diode Selection It is assumed as a worst case that all the switching output ripple current from the battery charger could flow in the output capacitor. This is a desirable situation if it is necessary to have very low switching ripple current in the battery itself. Ferrite beads or line chokes are often inserted in series with the battery leads to eliminate high frequency currents that could create EMI problems. This forces all the ripple current into the output capacitor. Total RMS current into the capacitor has a maximum value of about 1A, and this is handled with the two paralleled 22µF, 25V capacitors shown The switching diode should be a Schottky type to minimize both forward and reverse recovery losses. Average diode current is the same as output charging current, so this will be under 2A. A 3A diode is recommended for most applications, although smaller devices could be used at reduced charging current. Maximum diode reverse voltage will be equal to input voltage plus battery voltage. Diode reverse leakage current will be of some concern during charger shutdown. This leakage current is a direct drain on the battery when the charger is not powered. High Kool Mµ is a registered trademark of Magnetics, Inc. Metglas is a registered trademark of AlliedSignal Inc. sn1513 1513fas 8 LT1513/LT1513-2 U W U U APPLICATIONS INFORMATION vary depending on the mounting technique (copper area, airflow, etc.). current Schottky diodes have relatively high leakage currents (5µA to 500µA) even at room temperature. The latest very-low-forward devices have especially high leakage currents. It has been noted that surface mount versions of some Schottky diodes have as much as ten times the leakage of their through-hole counterparts. This may be because a low forward voltage process is used to reduce power dissipation in the surface mount package. In any case, check leakage specifications carefully before making a final choice for the switching diode. Be aware that diode manufacturers want to specify a maximum leakage current that is ten times higher than the typical leakage. It is very difficult to get them to specify a low leakage current in high volume production. This is an on going problem for all battery charger circuits and most customers have to settle for a diode whose typical leakage is adequate, but theoretically has a worst-case condition of higher than desired battery drain. Average supply current (including driver current) is: IIN = 4mA + (VBAT )(ICHRG )(0.024) VIN Switch power dissipation is given by: PSW = (ICHRG )2 (RSW )(VBAT + VIN )(VBAT) (VIN )2 RSW = Output switch ON resistance Total power dissipation of the die is equal to supply current times supply voltage, plus switch power: PD(TOTAL) = (IIN)(VIN) + PSW For VIN = 10V, VBAT = 8.2V, ICHRG = 1.2A, RSW = 0.3Ω, Thermal Considerations IIN = 4mA + 24mA = 28mA Care should be taken to ensure that worst-case conditions do not cause excessive die temperatures. Typical thermal resistance is 30°C/W for the R package but this number will PSW = 0.64W PD = (10)(0.028) + 0.64 = 0.92W GROUND PLANE VBAT LT1513 TAB AND GROUND PIN SOLDERED TO GROUND PLANE + + C1 C1 D1 C2 C5 L1A + R5 R3 2 WINDING INDUCTOR L1B C3 C1,C3,C5 AND R3 TIED DIRECTLY TO GROUND PLANE LT1513 • F04 VIN Figure 4. LT1513 Suggested Partial Layout for Critical Thermal and Electrical Paths sn1513 1513fas 9 LT1513/LT1513-2 U U W U APPLICATIONS INFORMATION Programmed Charging Current exceed 60µA to maintain a sharp constant voltage to constant current crossover characteristic. ICHARGE can also be controlled by a PWM input. Assuming the signal is a CMOS rail-to-rail output with a source impedance of less than a few hundred ohms, effective ISET is VCC multiplied by the PWM ratio. ICHARGE has good linearity over the entire 0% to 100% range. LT1513-2 charging current can be programmed with a DC voltage source or equivalent PWM signal, as shown in Figure 5. In constant-current mode, IFB acts as a virtual ground. The ISET voltage across R5 is balanced by the voltage across R4 in the ratio R4/R5. Charging current is given by: Voltage Mode Loop Stability ( V )( R4 / R5 )– I FBVOS ICHARGE = ISET R3 The LT1513 operates in constant-voltage mode during the final phase of charging lithium-ion and lead-acid batteries. This feedback loop is stabilized with a series resistor and capacitor on the VC pin of the chip. Figure 6 shows the simplified model for the voltage loop. The error amplifier is modeled as a transconductance stage with gm = 1500µmho IFB input current is small and can normally be ignored, but IFB offset voltage must be considered if operating over a wide range of program currents. The voltage across R3 at maximum charge current can be increased to reduce offset errors at lower charge currents. In Figure 5, ISET from 0V to 5V corresponds to an ICHARGE of 0A to 1A +37/– 62mA. C4 and R4 smooth the switch current waveform. During constant-current operation, the voltage feedback network loads the FB pin, which is held at VREF by the IFB amplifier. It is recommended that this load does not LT1513-2 IFB R5 249k R4 10k L1B ISET C4 0.1µF R3 0.2Ω 1513 F05 Figure 5 MODULATOR SECTION V1 CP** 3pF IP I 4(VIN) gm = P = V1 VIN + VBAT R1* 71.5k VIN = DC INPUT VOLTAGE VBAT = DC BATTERY VOLTAGE RP** 1M + EA gm 1500µmho + RG 330k C1 – VC R5 330Ω C5 0.1µF C1 FB 1.245V R2 12.5k + RCAP ≈0.15Ω EACH C1 22µF EACH RBAT 0.1Ω BATTERY 1513 F06 * FOR 8.4V BATTERY. ADJUST VALUE OF R1 FOR ACTUAL BATTERY VOLTAGE ** RP AND CP MODEL PHASE DELAY IN THE MODULATOR THIS IS A SIMPLIFIED AC MODEL FOR THE LT1513 IN CONSTANTVOLTAGE MODE. RESISTOR AND CAPACITOR NUMBERS CORRESPOND TO THOSE USED IN FIGURE 1. RP AND CP MODEL THE PHASE DELAY IN THE MODULATOR. C3 IS 3pF FOR A 10µH INDUCTOR. IT SHOULD BE SCALED PROPORTIONALLY FOR OTHER INDUCTOR VALUES (6pF FOR 20µH). THE MODULATOR IS A TRANSCONDUCTANCE WHOSE GAIN IS A FUNCTION OF INPUT AND BATTERY VOLTAGE AS SHOWN. AS SHOWN, THIS LOOP HAS A UNITY-GAIN FREQUENCY OF ABOUT 250Hz. UNITY-GAIN WILL MOVE OUT TO SEVERAL KILOHERTZ IF BATTERY RESISTANCE INCREASES TO SEVERAL OHMS. R5 IS NOT USED IN ALL APPLICATIONS, BUT IT GIVES BETTER PHASE MARGIN IN CONSTANT-VOLTAGE MODE WITH HIGH BATTERY RESISTANCE. Figure 6. Constant-Voltage Small-Signal Model sn1513 1513fas 10 LT1513/LT1513-2 U U W U APPLICATIONS INFORMATION problem, and indeed small signal loop stability can be excellent even in the presence of subharmonic switching. The primary issue with subharmonics is the presence of EMI at frequencies below 500kHz. (from the Electrical Characteristics). Amplifier output resistance is modeled with a 330k resistor. The power stage (modulator section) of the LT1513 is modeled as a transconductance whose value is 4(VIN)/(VIN + VBAT). This is a very simplified model of the actual power stage, but it is sufficient when the unity-gain frequency of the loop is low compared to the switching frequency. The output filter capacitor model includes its ESR (RCAP). A series resistance (RBAT) is also assigned to the battery model. Constant-Current Mode Loop Stability The LT1513 is normally very stable when operating in constant-current mode (see Figure 7), but there are certain conditions which may create instabilities. The combination of higher value current sense resistors (low programmed charging current), higher input voltages, and the addition of a loop compensation resistor (R5) on the VC pin may create an unstable current mode loop. (A resistor is sometimes added in series with C5 to improve loop phase margin when the loop is operating in voltage mode.) Instability results because loop gain is too high in the 50kHz to 150kHz region where excess phase occurs in the current sensing amplifier and the modulator. The IFBA amplifier (gain of –12.5) has a pole at approximately 150kHz. The modulator section consisting of the current comparator, the power switch and the magnetics, has a pole at approximately 50kHz when the coupled inductor value is 10µH. Higher inductance will reduce the pole frequency proportionally. The design procedure presented here is to roll off the loop to unity-gain at a frequency of 25kHz or lower to avoid these excess phase regions. Analysis of this loop normally shows an extremely stable system for all conditions, even with 0Ω for R5. The one condition which can cause reduced phase margin is with a very large battery resistance (>5Ω), or with the battery replaced with a resistive load. The addition of R5 gives good phase margin even under these unusual conditions. R5 should not be increased above 330Ω without checking for two possible problems. The first is instability in the constant current region (see Constant-Current Mode Loop Stability), and the second is subharmonic switching where switch duty cycle varies from cycle to cycle. This duty cycle instability is caused by excess switching frequency ripple voltage on the VC pin. Normally this ripple is very low because of the filtering effect of C5, but large values of R5 can allow high ripple on the VC pin. Normal loop analysis does not show this V1 RP** 1M CP 3pF FB IP MODULATOR SECTION IP = 4(V1)(VIN) VIN + VBAT IFB – IFBA + VC EA RG 330k + R5 330Ω C5 0.1µF – RA 100k R4 24Ω gm 1500µmho CA 10pF 1.245V THIS IS A SIMPLIFIED AC MODEL FOR THE LT1513 IN CONSTANT-CURRENT MODE. RESISTOR AND CAPACITOR NUMBERS CORRESPOND TO THOSE USED IN FIGURE 1. RP AND CP MODEL THE PHASE DELAY IN THE PowerPath. C3 IS 3pF FOR A 10µH INDUCTOR. IT SHOULD BE SCALED PROPORTIONALLY FOR OTHER INDUCTOR VALUES (6pF FOR 20µH). THE PowerPath IS A TRANSCONDUCTANCE WHOSE GAIN IS A FUNCTION OF INPUT AND BATTERY VOLTAGE AS SHOWN. VOLTAGE GAIN = 12 C4 0.22µF R3 0.1Ω THE CURRENT AMPLIFIER HAS A FIXED VOLTAGE GAIN OF 12. ITS PHASE DELAY IS MODELED WITH RA AND CA. THE ERROR AMPLIFIER HAS A TRANSCONDUCTANCE OF 1500µmho AND AN INTERNAL OUTPUT SHUNT RESISTANCE OF 330k. AS SHOWN, THIS LOOP HAS A UNITY-GAIN FREQUENCY OF ABOUT 27kHz. R5 IS NOT USED IN ALL APPLICATIONS, BUT IT GIVES BETTER PHASE MARGIN IN CONSTANT VOLTAGE MODE. 1513 F07 Figure 7. Constant-Current Small-Signal Model sn1513 1513fas 11 LT1513/LT1513-2 U W U U APPLICATIONS INFORMATION The suggested way to control unity loop frequency is to increase the filter time constant on the IFB pin (R4/C4 in Figures 1 and 7). The filter resistor cannot be arbitrarily increased because high values will affect charging current accuracy. Charging current will increase by 1% for each 40Ω increase in R4. There is no inherent limitation on the value of C4, but if this capacitor is ceramic, it should be an X7R type to maintain its value over temperature. X7R dielectric requires a larger footprint. The formula for calculating the minimum value for the filter capacitor C4 is: C4 = (R3)(4)(VIN)(12)(1500µ)(R5) 2π(f)(R4)(VIN + VBAT ) VIN = Highest input voltage 1500µ = Transconductance of error amplifier (EA)f = Desired unity-gain frequency VBAT = Battery voltage For example, assume VIN(MAX) = 15V, R3 = 0.4Ω (charging current set to 0.25A), R4 = 24Ω, R5 = 330Ω and VBAT = 8V, C4 = 0.4(4)(15)(12)(0.0015)(330) = 1µF 6.3(25000)(39)(15 + 8) The value for C4 could be reduced to a more manageable size by increasing R4 to 75Ω and reducing R5 to 300Ω, yielding 0.47µF for C4. The 2% increase in charging current can be ignored or factored into the value for R3. More Help Linear Technology Field Application Engineers have a CAD spreadsheet program for detailed calculations of circuit operating conditions. In addition, our Applications Department is always ready to lend a helping hand. The LT1371 data sheet may also be helpful. The LT1513 is identical except for the current amplifier circuitry. sn1513 1513fas 12 LT1513/LT1513-2 U TYPICAL APPLICATIONS Lithium-Ion Battery Charger with Switchable Charge Current (I )(R4) – IFBVOS ICHARGE = R5 R3 Many battery chemistries require several constant-current settings during the charging cycle. The circuit shown in Figure 8 uses the LT1513-2 to provide switchable 1.35A and 0.13A constant-current modes. The circuit is based on a standard SEPIC battery charger circuit set to a single lithium-ion cell charge voltage of 4.1V. The LT1513-2 has IFB referenced to ground allowing a simple resistor network to set the charging current values. In constant-current mode, the IFB error amplifier drives the FB pin, increasing charging current, until IR4 is balanced by IR5. There are several ways to control IR5 including DAC, PWM or resistor network as shown here. If the lithium cell requires precharging, Q1 is turned on, setting a constant current of 0.13A. When charge voltage is reached, Q1 is turned off, programming the full charge current of 1.35A. As the cell voltage approaches 4.1V, the voltage sensing network (R1, R2) starts driving the VFB pin, changing the LT1513-2 to constant-voltage mode. As charging current falls, the output remains in constant-voltage mode for the remainder of the charging cycle. When charging is complete, the LT1513-2 can be shut down with the S/S pin. C2 4.7µF L1A CTX10-4 VIN + CHARGE SHUTDOWN PRECHARGE • C3 25µF 25V 7 6 1 3.3V R6 10k R7 910Ω Q1 D1 MBRS330T3 R3 330Ω C5 0.1µF VIN S/S VSW LT1513-2 VC VFB IFB GND 5 L1B 2 R1 78.7k 0.5% • + 3 R4 4.7k 4 R5 36k Li-Ion RECHARGABLE CELL C4 0.22µF R3 0.25Ω C1 22µF 25V ×2 R2 34k 0.5% GND CHARGE 1513 F08 Figure 8. Lithium-Ion Battery Charger sn1513 1513fas 13 LT1513/LT1513-2 U TYPICAL APPLICATIONS This Cold Cathode Fluorescent Lamp driver uses a Royer class self-oscillating sine wave converter to driver a high voltage lamp with an AC waveform. CCFL Royer converters have significantly degraded efficiency if they must operate at low input voltages, and this circuit was designed to handle input voltages as low as 2.7V. Therefore, the LT1513 is connected to generate a negative current through L2 that allows the Royer to operate as if it were connected to a constant higher voltage input. 2.7V C1 47µF TO 20V ELECT L1 20µH + 7 6 2 5 VIN S/S VSW R2 0.25Ω C6 0.1µF 4, TAB D2 15V C9 1µF R7 2k 2 3• R5 330k C2 4.7µF CERAMIC L2 20µH R4 20k C5 4.7nF D1 3A 1 T1 C3 0.082µF WIMA GND 1 C10 10nF IFB R3 10k 3 LT1513-2 VFB VC The Royer output winding and the bulb are allowed to float in this circuit. This can yield significantly higher efficiency in situations where the stray bulb capacitance to surrounding enclosure is high. To regulate bulb current in Figure 9, Royer input current is sensed with R2 and filtered with R3 and C6. This negative feedback signal is applied to the IFB pin of the LT1513. For more information on this circuit contact the LTC Applications Department and see Design Note 133. Considerable written application literature on Royer CCFL circuits is also available from other LTC Application and Design Notes. Q2 •6 10 C4 27pF 3kV • 4 R1 470Ω 5 Q1 CCFL LAMP CURRENT 5.6mA NEGATIVE VOLTAGE IS GENERATED HERE PWM DIMMING (≈1kHz) 1513 F09 C2: TOKIN MULTILAYER CERAMIC C3: MUST BE A LOW LOSS CAPACITOR, WIMA MKP-20 OR EQUIVALENT L1, L2: COILTRONICS CTX20-4 (MUST BE SEPARATE INDUCTORS) Q1, Q2: ZETEX ZTX849 OR FZT849 T1: COILTRONICS CTX110605 (67:1) Figure 9. CCFL Power Supply for Floaing Lamp Configuration Operates on 2.7V sn1513 1513fas 14 LT1513/LT1513-2 U PACKAGE DESCRIPTION Dimensions in inches (millimeters) unless otherwise noted. R Package 7-Lead Plastic DD Pak (LTC DWG # 05-08-1462) 0.060 (1.524) TYP 0.060 (1.524) 0.256 (6.502) 0.390 – 0.415 (9.906 – 10.541) 0.165 – 0.180 (4.191 – 4.572) 0.045 – 0.055 (1.143 – 1.397) 15° TYP 0.060 (1.524) 0.183 (4.648) 0.059 (1.499) TYP 0.330 – 0.370 (8.382 – 9.398) ( +0.203 0.102 –0.102 BOTTOM VIEW OF DD PAK HATCHED AREA IS SOLDER PLATED COPPER HEAT SINK ) 0.095 – 0.115 (2.413 – 2.921) 0.075 (1.905) 0.300 (7.620) +0.008 0.004 –0.004 ( +0.012 0.143 –0.020 +0.305 3.632 –0.508 0.040 – 0.060 (1.016 – 1.524) 0.026 – 0.036 (0.660 – 0.914) ) 0.013 – 0.023 (0.330 – 0.584) 0.050 ± 0.012 (1.270 ± 0.305) R (DD7) 0396 T7 Package 7-Lead Plastic TO-220 (Standard) (LTC DWG # 05-08-1422) 0.390 – 0.415 (9.906 – 10.541) 0.165 – 0.180 (4.191 – 4.572) 0.147 – 0.155 (3.734 – 3.937) DIA 0.045 – 0.055 (1.143 – 1.397) 0.230 – 0.270 (5.842 – 6.858) 0.460 – 0.500 (11.684 – 12.700) 0.570 – 0.620 (14.478 – 15.748) 0.330 – 0.370 (8.382 – 9.398) 0.620 (15.75) TYP 0.700 – 0.728 (17.780 – 18.491) 0.152 – 0.202 0.260 – 0.320 (3.860 – 5.130) (6.604 – 8.128) 0.040 – 0.060 (1.016 – 1.524) 0.095 – 0.115 (2.413 – 2.921) 0.013 – 0.023 (0.330 – 0.584) 0.026 – 0.036 (0.660 – 0.914) 0.135 – 0.165 (3.429 – 4.191) 0.155 – 0.195 (3.937 – 4.953) T7 (TO-220) (FORMED) 1197 sn1513 1513fas Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 15 LT1513/LT1513-2 RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LT1239 Backup Battery Management System Charges Backup Battery and Regulates Backup Battery Output when Main Battery Removed LTC®1325 Microprocessor Controlled Battery Management System Can Charge, Discharge and Gas Gauge NiCd, NiMH and Pb-Acid Batteries with Software Charging Profiles LT1510 1.5A Constant-Current/Constant-Voltage Battery Charger Step-Down Charger for Li-Ion, NiCd and NiMH LT1511 3.0A Constant-Current/Constant-Voltage Battery Charger with Input Current Limiting Step-Down Charger that Allows Charging During Computer Operation and Prevents Wall-Adapter Overload LT1512 SEPIC Constant-Current/Constant-Voltage Battery Charger Step-Up/Step-Down Charger for Up to 1A Current sn1513 1513fas 16 Linear Technology Corporation 1630 McCarthy Blvd., Milpitas, CA 95035-7417 ● (408) 432-1900 FAX: (408) 434-0507● TELEX: 499-3977 ● www.linear-tech.com LT/TP 0198 REV A 4K • PRINTED IN THE USA  LINEAR TECHNOLOGY CORPORATION 1996