LT1934/LT1934-1
Micropower Step-Down
Switching Regulators
in ThinSOT and DFN
FEATURES
DESCRIPTION
n
The LT ®1934 is a micropower step-down DC/DC converter
with internal 400mA power switch, packaged in a low
profile (1mm) ThinSOT. With its wide input range of 3.2V
to 34V, the LT1934 can regulate a wide variety of power
sources, from 4-cell alkaline batteries and 5V logic rails
to unregulated wall transformers and lead-acid batteries.
Quiescent current is just 12μA and a zero current shutdown mode disconnects the load from the input source,
simplifying power management in battery-powered systems. Burst Mode® operation and the low drop internal
power switch result in high efficiency over a broad range
of load current.
n
n
n
n
n
n
n
n
Wide Input Voltage Range: 3.2V to 34V
Micropower Operation: IQ = 12μA
5V at 250mA from 6.5V to 34V Input (LT1934)
5V at 60mA from 6.5V to 34V Input (LT1934-1)
3.3V at 250mA from 4.5V to 34V Input (LT1934)
3.3V at 60mA from 4.5V to 34V Input (LT1934-1)
Low Shutdown Current: 1V
VFB = 0V
Maximum Duty Cycle
VFB = 1V
Switch VCESAT
ISW = 300mA (LT1934, S6 Package)
ISW = 300mA (LT1934, DCB Package)
ISW = 75mA (LT1934-1, S6 Package)
ISW = 75mA (LT1934-1, DCB Package)
l
l
2
2
mV
±15
±60
0.007
–40°C ≤ TA ≤ 85°C
–40°C ≤ TA ≤ 125°C
l
l
1.4
1.8
12
85
83
88
88
350
90
nA
nA
%/V
2.3
μs
μs
%
%
200
225
65
70
300
400
120
490
160
mA
mA
120
mV
mV
mV
mV
Switch Current Limit
LT1934
LT1934-1
BOOST Pin Current
ISW = 300mA (LT1934)
ISW = 75mA (LT1934-1)
8.5
6.0
12
10
mA
mA
Minimum Boost Voltage (Note 3)
ISW = 300mA (LT1934)
ISW = 75mA (LT1934-1)
1.8
1.7
2.5
2.5
V
V
VSHDN = 2.3V
VSHDN = 34V
0.5
1.5
Switch Leakage Current
SHDN Pin Current
SHDN Input Voltage High
SHDN Input Voltage Low
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LT1934E and LT1934E-1 are guaranteed to meet performance
specifications from 0°C to 85°C. Specifications over the –40°C to 85°C
2
μA
5
μA
μA
2.3
V
0.25
V
operating temperature range are assured by design, characterization and
correlation with statistical process controls. The LT1934I and LT1934I-1
specifications are guaranteed over the –40°C to 125°C temperature range.
Note 3: This is the minimum voltage across the boost capacitor needed to
guarantee full saturation of the internal power switch.
1934fe
3
LT1934/LT1934-1
TYPICAL PERFORMANCE CHARACTERISTICS
LT1934 Efficiency, VOUT = 5V
LT1934 Efficiency, VOUT = 3.3V
LT1934
VOUT = 3.3V
L = 47μH
90
TA = 25°C
VIN = 12V
80
EFFICIENCY (%)
EFFICIENCY (%)
LT1934
VOUT = 5V
L = 47μH
90
TA = 25°C
VIN = 24V
70
60
90
80
VIN = 24V
VIN = 12V
70
50
100
1934 G03
Off Time vs Temperature
3.0
VIN = 12V
VIN = 24V
70
60
10
300
200
100
2.0
1.5
1.0
LT1934-1
100
0.5
0
–50
50
1
2.5
400
OFF TIME (μs)
SWITCH CURRENT LIMIT (mA)
LT1934
80
–25
LOAD CURRENT (mA)
50
25
0
75
TEMPERATURE (°C)
1934 G04
100
0
–50 –25
125
50
25
75
0
TEMPERATURE (°C)
100
SHDN Bias Current
vs SHDN Voltage
VFB vs Temperature
1.27
TA = 25oC
125
1934 G06
1934 G05
Frequency Foldback
16
10
LOAD CURRENT (mA)
Current Limit vs Temperature
LT1934-1
VOUT = 3.3V
L = 100μH
TA = 25°C
1
0.1
500
0.1
70
1934 G02
LT1934-1 Efficiency, VOUT = 3.3V
90
VIN = 24V
1
10
100
LOAD CURRENT (mA)
1934 G01
100
VIN = 12V
80
60
50
0.1
1
10
100
LOAD CURRENT (mA)
LT1934-1
VOUT = 5V
L = 150μH
TA = 25°C
VIN = 5V
60
50
0.1
EFFICIENCY (%)
LT1934-1 Efficiency, VOUT = 5V
100
100
EFFICIENCY (%)
100
2.0
TA = 25°C
14
10
8
6
4
SHDN PIN CURRENT (μA)
FEEDBACK VOLTAGE (V)
SWITCH OFF TIME (μs)
1.26
12
1.25
1.24
1.5
1.0
0.5
1.23
2
0
0
0.2
0.4
0.8
1.0
0.6
FEEDBACK PIN VOLTAGE (V)
1.2
1934 G07
1.22
–50
0
–25
50
25
0
75
TEMPERATURE (°C)
100
125
1934 G08
0
2
4
6
8
SHDN PIN VOLTAGE (V)
10
12
1934 G09
1934fe
4
LT1934/LT1934-1
TYPICAL PERFORMANCE CHARACTERISTICS
Undervoltage Lockout
vs Temperature
20
4.0
15
3.5
UVLO (V)
QUIESCENT CURRENT (μA)
Quiescent Current
vs Temperature
10
5
3.0
2.5
0
–50
–25
75
0
25
50
TEMPERATURE (°C)
100
2.0
–50
125
–25
75
0
25
50
TEMPERATURE (°C)
100
1934 G10
1934 G11
Minimum Input Voltage
VOUT = 5V
Minimum Input Voltage
VOUT = 3.3V
6.0
8
LT1934
VOUT = 5V
TA = 25°C
7 BOOST DIODE TIED TO OUTPUT
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
LT1934
VOUT = 3.3V
5.5 TA = 25°C
BOOST DIODE TIED TO OUTPUT
5.0
125
VIN TO START
4.5
4.0
VIN TO RUN
VIN TO START
6
VIN TO RUN
5
3.5
3.0
0.1
4
0.1
1
10
100
LOAD CURRENT (mA)
1934 G12
PIN FUNCTIONS
1
10
100
LOAD CURRENT (mA)
1934 G13
(TSOT-23/DFN)
BOOST (Pin 1/Pin 1): The BOOST pin is used to provide a
drive voltage, higher than the input voltage, to the internal
bipolar NPN power switch.
in shutdown mode. Tie to ground to shut down the LT1934.
Apply 2.3V or more for normal operation. If the shutdown
feature is not used, tie this pin to the VIN pin.
GND (Pin 2/Pin 5): Tie the GND pin to a local ground plane
below the LT1934 and the circuit components. Return the
feedback divider to this pin.
VIN (Pin 5/Pin 3): The VIN pin supplies current to the
LT1934’s internal regulator and to the internal power
switch. This pin must be locally bypassed.
FB (Pin 3/Pin 6): The LT1934 regulates its feedback pin
to 1.25V. Connect the feedback resistor divider tap to this
pin. Set the output voltage according to VOUT = 1.25V
(1 + R1/R2) or R1 = R2 (VOUT/1.25 – 1).
SW (Pin 6/Pin 2): The SW pin is the output of the internal
power switch. Connect this pin to the inductor, catch diode
and boost capacitor.
SHDN (Pin 4/Pin 4): The SHDN pin is used to put the LT1934
Exposed Pad (Pin 7, DFN Package): This pin must be
soldered to ground plane.
1934fe
5
LT1934/LT1934-1
BLOCK DIAGRAM
VIN
VIN
+
C2
+
–
D2
BOOST
ON TIME
R Qa
12μs DELAY
S
OFF TIME
SHDN
L1
SW
1.8μs DELAY
ON OFF
C3
Q
VOUT
D1
C1
+
VREF 1.25V
ENABLE
–
GND
FEEDBACK
COMPARATOR
FB
R2
R1
1934 BD
1934fe
6
LT1934/LT1934-1
OPERATION
(Refer to Block Diagram)
The LT1934 uses Burst Mode control, combining both low
quiescent current operation and high switching frequency,
which result in high efficiency across a wide range of load
currents and a small total circuit size.
A comparator monitors the voltage at the FB pin of the
LT1934. If this voltage is higher than the internal 1.25V
reference, the comparator disables the oscillator and power
switch. In this state, only the comparator, reference and
undervoltage lockout circuits are active, and the current into
the VIN pin is just 12μA. As the load current discharges the
output capacitor, the voltage at the FB pin falls below 1.25V
and the comparator enables the oscillator. The LT1934
begins to switch, delivering current to the output capacitor. The output voltage rises, and when it overcomes the
feedback comparator’s hysteresis, the oscillator is disabled
and the LT1934 returns to its micropower state.
The oscillator consists of two one-shots and a flip-flop.
A rising edge from the off-time one-shot sets the flip-flop,
which turns on the internal NPN power switch. The switch
remains on until either the on-time one-shot trips or the
current limit is reached. A sense resistor and amplifier
monitor the current through the switch and resets the
flip-flop when this current reaches 400mA (120mA for
the LT1934-1). After the 1.8μs delay of the off-time oneshot, the cycle repeats. Generally, the LT1934 will reach
current limit on every cycle—the off time is fixed and
the on time is regulated so that the LT1934 operates at
the correct duty cycle. The 1.8μs off time is lengthened
when the FB pin voltage falls below 0.8V; this foldback
behavior helps control the output current during start-up
and overload. Figure 1 shows several waveforms of an
LT1934 producing 3.3V from a 10V input. When the switch
is on, the SW pin voltage is at 10V. When the switch is
off, the inductor current pulls the SW pin down until it is
clamped near ground by the external catch diode.
The switch driver operates from either the input or from
the BOOST pin. An external capacitor and diode are used
to generate a voltage at the BOOST pin that is higher than
the input supply. This allows the driver to fully saturate
the bipolar switch for efficient operation.
If the SHDN pin is grounded, all internal circuits are turned
off and VIN current reduces to the device leakage current,
typically a few nA.
VOUT
50mV/DIV
VSW
10V/DIV
ISW
0.5A/DIV
ILI
0.5A/DIV
1934 F01a
5μs/DIV
Figure 1. Operating Waveforms of the LT1934 Converting
10V to 3.3V at 180mA (Front Page Schematic)
1934fe
7
LT1934/LT1934-1
APPLICATIONS INFORMATION
Which One to Use: LT1934 or LT1934-1?
The only difference between the LT1934 and LT1934-1
is the peak current through the internal switch and the
inductor. If your maximum load current is less than 60mA,
use the LT1934-1. If your maximum load is higher, use
the LT1934; it can supply up to ~300mA.
While the LT1934-1 can’t deliver as much output current,
it has other advantages. The lower peak switch current
allows the use of smaller components (input capacitor,
inductor and output capacitor). The ripple current at the
input of the LT1934-1 circuit will be smaller and may be
an important consideration if the input supply is current
limited or has high impedance. The LT1934-1’s current
draw during faults (output overload or short) and startup is lower.
The maximum load current that the LT1934 or LT1934-1
can deliver depends on the value of the inductor used.
Table 1 lists inductor value, minimum output capacitor
and maximum load for 3.3V and 5V circuits. Increasing
the value of the capacitor will lower the output voltage
ripple. Component selection is covered in more detail in
the following sections.
Minimum Input Voltage
The minimum input voltage required to generate a particular output voltage is determined by either the LT1934’s
undervoltage lockout of ~3V or by its maximum duty cycle.
The duty cycle is the fraction of time that the internal
switch is on and is determined by the input and output
voltages:
DC = (VOUT + VD)/(VIN – VSW + VD)
where VD is the forward voltage drop of the catch diode
(~0.4V) and VSW is the voltage drop of the internal switch
(~0.3V at maximum load for the LT1934, ~0.1V for the
LT1934-1). This leads to a minimum input voltage of:
VIN(MIN) = (VOUT + VD)/DCMAX – VD + VSW
with DCMAX = 0.85.
Inductor Selection
A good first choice for the inductor value is:
L = 2.5 • (VOUT + VD) • 1.8μs/ILIM
where ILIM is the switch current limit (400mA for the
LT1934 and 120mA for the LT1934-1). This choice provides
a worst-case maximum load current of 250mA (60mA for
the LT1934-1). The inductor’s RMS current rating must
be greater than the load current and its saturation current
should be greater than ILIM . To keep efficiency high, the
series resistance (DCR) should be less than 0.3Ω (1Ω
for the LT1934-1). Table 2 lists several vendors and types
that are suitable.
PART
VOUT
L
MINIMUM
COUT
MAXIMUM
LOAD
LT1934
3.3V
100μH
47μH
33μH
100μH
47μH
33μH
300mA
250mA
200mA
5V
150μH
68μH
47μH
47μH
33μH
22μH
300mA
250mA
200mA
This simple rule may not provide the optimum value for
your application. If the load current is less, then you can
relax the value of the inductor and operate with higher
ripple current. This allows you to use a physically smaller
inductor, or one with a lower DCR resulting in higher
efficiency. The following provides more details to guide
inductor selection. First, the value must be chosen so that
the LT1934 can supply the maximum load current drawn
from the output. Second, the inductor must be rated appropriately so that the LT1934 will function reliably and
the inductor itself will not be overly stressed.
3.3V
150μH
100μH
68μH
15μH
10μH
10μH
60mA
45mA
20mA
Detailed Inductor Selection and
Maximum Load Current
5V
220μH
150μH
100μH
10μH
4.7μH
4.7μH
60mA
45mA
20mA
Table 1
LT1934-1
The square wave that the LT1934 produces at its switch
pin results in a triangle wave of current in the inductor. The
LT1934 limits the peak inductor current to ILIM . Because
1934fe
8
LT1934/LT1934-1
APPLICATIONS INFORMATION
Table 2. Inductor Vendors
VENDOR
PHONE
URL
PART SERIES
COMMENTS
Murata
(404) 426-1300
www.murata.com
LQH3C
Small, Low Cost, 2mm Height
Sumida
(847) 956-0666
www.sumida.com
CR43
CDRH4D28
CDRH5D28
Coilcraft
(847) 639-6400
www.coilcraft.com
DO1607C
DO1608C
DT1608C
Würth
Electronics
(866) 362-6673
www.we-online.com
WE-PD1, 2, 3, 4
the average inductor current equals the load current, the
maximum load current is:
IOUT(MAX) = IPK – ΔIL /2
where IPK is the peak inductor current and ΔIL is the
peak-to-peak ripple current in the inductor. The ripple
current is determined by the off time, tOFF = 1.8μs, and
the inductor value:
ΔIL = (VOUT + VD) • tOFF /L
IPK is nominally equal to ILIM . However, there is a slight
delay in the control circuitry that results in a higher peak
current and a more accurate value is:
IPK = ILIM + 150ns • (VIN – VOUT)/L
These expressions are combined to give the maximum
load current that the LT1934 will deliver:
IOUT(MAX) = 350mA + 150ns • (VIN – VOUT)/L – 1.8μs
• (VOUT + VD)/2L (LT1934)
IOUT(MAX) = 90mA + 150ns • (VIN – VOUT)/L – 1.8μs
• (VOUT + VD)/2L (LT1934-1)
The minimum current limit is used here to be conservative.
The third term is generally larger than the second term,
so that increasing the inductor value results in a higher
output current. This equation can be used to evaluate
a chosen inductor or it can be used to choose L for a
given maximum load current. The simple, single equation
rule given above for choosing L was found by setting
ΔIL = ILIM /2.5. This results in IOUT(MAX) ~0.8ILIM (ignoring
the delay term). Note that this analysis assumes that the
inductor current is continuous, which is true if the ripple
current is less than the peak current or ΔIL < IPK.
The inductor must carry the peak current without saturating excessively. When an inductor carries too much
current, its core material can no longer generate additional magnetic flux (it saturates) and the inductance
drops, sometimes very rapidly with increasing current.
This condition allows the inductor current to increase
at a very high rate, leading to high ripple current and
decreased overload protection.
Inductor vendors provide current ratings for power inductors. These are based on either the saturation current or
on the RMS current that the inductor can carry without
dissipating too much power. In some cases it is not clear
which of these two determine the current rating. Some data
sheets are more thorough and show two current ratings,
one for saturation and one for dissipation. For LT1934 applications, the RMS current rating should be higher than
the load current, while the saturation current should be
higher than the peak inductor current calculated above.
Input Capacitor
Step-down regulators draw current from the input supply in pulses with very fast rise and fall times. The input
capacitor is required to reduce the resulting voltage ripple
at the LT1934 and to force this switching current into
a tight local loop, minimizing EMI. The input capacitor
must have low impedance at the switching frequency to
do this effectively. A 2.2μF ceramic capacitor (1μF for the
LT1934-1) satisfies these requirements.
If the input source impedance is high, a larger value capacitor may be required to keep input ripple low. In this
case, an electrolytic of 10μF or more in parallel with a 1μF
ceramic is a good combination. Be aware that the input
1934fe
9
LT1934/LT1934-1
APPLICATIONS INFORMATION
capacitor is subject to large surge currents if the LT1934
circuit is connected to a low impedance supply, and that
some electrolytic capacitors (in particular tantalum) must
be specified for such use.
Output Capacitor and Output Ripple
The output capacitor filters the inductor’s ripple current and
stores energy to satisfy the load current when the LT1934
is quiescent. In order to keep output voltage ripple low, the
impedance of the capacitor must be low at the LT1934’s
switching frequency. The capacitor’s equivalent series
resistance (ESR) determines this impedance. Choose one
with low ESR intended for use in switching regulators. The
contribution to ripple voltage due to the ESR is approximately ILIM • ESR. ESR should be less than ~150mΩ for
the LT1934 and less than ~500mΩ for the LT1934-1.
The value of the output capacitor must be large enough
to accept the energy stored in the inductor without a large
change in output voltage. Setting this voltage step equal to
1% of the output voltage, the output capacitor must be:
COUT > 50 • L • (ILIM /VOUT)2
For example, an LT1934 producing 3.3V with L = 47μH
requires 33μF. This value can be relaxed if small circuit
size is more important than low output ripple.
Sanyo’s POSCAP series in B-case and C-case sizes
provides very good performance in a small package for
the LT1934. Similar performance in traditional tantalum
capacitors requires a larger package (C- or D-case).
The LT1934-1, with its lower switch current, can use a
B-case tantalum capacitor.
With a high quality capacitor filtering the ripple current
from the inductor, the output voltage ripple is determined
by the hysteresis and delay in the LT1934’s feedback
comparator. This ripple can be reduced further by adding
a small (typically 10pF) phase lead capacitor between the
output and the feedback pin.
Ceramic Capacitors
Ceramic capacitors are small, robust and have very low
ESR. However, ceramic capacitors can cause problems
when used with the LT1934.
Not all ceramic capacitors are suitable. X5R and X7R
types are stable over temperature and applied voltage
and give dependable service. Other types (Y5V and Z5U)
have very large temperature and voltage coefficients of
capacitance. In the application circuit they may have only
a small fraction of their nominal capacitance and voltage
ripple may be much larger than expected.
Ceramic capacitors are piezoelectric. The LT1934’s switching frequency depends on the load current, and at light
loads the LT1934 can excite the ceramic capacitor at audio
frequencies, generating audible noise. If this is unacceptable, use a high performance electrolytic capacitor at the
output. The input capacitor can be a parallel combination
of a 2.2μF ceramic capacitor and a low cost electrolytic
capacitor. The level of noise produced by the LT1934-1
Table 3. Capacitor Vendors
VENDOR
PHONE
URL
PART SERIES
COMMENTS
Panasonic
(714) 373-7366
www.panasonic.com
Ceramic,
Polymer,
Tantalum
EEF Series
Kemet
Sanyo
Murata
(864) 963-6300
(408) 749-9714
(404) 436-1300
AVX
Taiyo Yuden
(864) 963-6300
www.kemet.com
www.sanyovideo.com
Ceramic,
Tantalum
Ceramic,
Polymer,
Tantalum
www.murata.com
Ceramic,
www.avxcorp.com
Ceramic,
Tantalum
T494, T495
POSCAP
TPS Series
www.taiyo-yuden.com Ceramic
1934fe
10
LT1934/LT1934-1
APPLICATIONS INFORMATION
when used with ceramic capacitors will be lower and may
be acceptable.
A final precaution regarding ceramic capacitors concerns
the maximum input voltage rating of the LT1934. A ceramic
input capacitor combined with trace or cable inductance
forms a high quality (under damped) tank circuit. If the
LT1934 circuit is plugged into a live supply, the input voltage can ring to twice its nominal value, possibly exceeding
the LT1934’s rating. This situation is easily avoided; see
the Hot Plugging Safely section.
D2
BOOST
C3
LT1934
VIN
VIN
VOUT
SW
GND
VBOOST – VSW VOUT
MAX VBOOST VIN + VOUT
(2a)
D2
Catch Diode
C3
BOOST
LT1934
A 0.5A Schottky diode is recommended for the catch
diode, D1. The diode must have a reverse voltage rating
equal to or greater than the maximum input voltage. The
ON Semiconductor MBR0540 is a good choice; it is rated
for 0.5A forward current and a maximum reverse voltage
of 40V.
Schottky diodes with lower reverse voltage ratings usually have a lower forward drop and may result in higher
efficiency with moderate to high load currents. However,
these diodes also have higher leakage currents. This leakage
current mimics a load current at the output and can raise
the quiescent current of the LT1934 circuit, especially at
elevated temperatures.
BOOST Pin Considerations
Capacitor C3 and diode D2 are used to generate a boost
voltage that is higher than the input voltage. In most cases
a 0.1μF capacitor and fast switching diode (such as the
1N4148 or 1N914) will work well. Figure 2 shows two
ways to arrange the boost circuit. The BOOST pin must
be more than 2.5V above the SW pin for best efficiency.
For outputs of 3.3V and above, the standard circuit (Figure 2a) is best. For outputs between 2.8V and 3V, use a
0.22μF capacitor and a small Schottky diode (such as the
BAT-54). For lower output voltages the boost diode can be
tied to the input (Figure 2b). The circuit in Figure 2a is more
efficient because the BOOST pin current comes from a lower
voltage source. You must also be sure that the maximum
voltage rating of the BOOST pin is not exceeded.
The minimum operating voltage of an LT1934 application is limited by the undervoltage lockout (~3V) and by
VIN
VIN
SW
VOUT
GND
1934 F02
VBOOST – VSW VIN
MAX VBOOST 2VIN
(2b)
Figure 2. Two Circuits for Generating the Boost Voltage
the maximum duty cycle as outlined above. For proper
start-up, the minimum input voltage is also limited by the
boost circuit. If the input voltage is ramped slowly, or the
LT1934 is turned on with its SHDN pin when the output
is already in regulation, then the boost capacitor may not
be fully charged. Because the boost capacitor is charged
with the energy stored in the inductor, the circuit will rely
on some minimum load current to get the boost circuit
running properly. This minimum load will depend on input
and output voltages, and on the arrangement of the boost
circuit. The minimum load generally goes to zero once the
circuit has started. Figure 3 shows a plot of minimum load
to start and to run as a function of input voltage. In many
cases the discharged output capacitor will present a load
to the switcher which will allow it to start. The plots show
the worst-case situation where VIN is ramping very slowly.
Use a Schottky diode (such as the BAT-54) for the lowest
start-up voltage.
At light loads, the inductor current becomes discontinuous and the effective duty cycle can be very high. This
reduces the minimum input voltage to approximately
300mV above VOUT. At higher load currents, the inductor
current is continuous and the duty cycle is limited by the
1934fe
11
LT1934/LT1934-1
APPLICATIONS INFORMATION
Minimum Input Voltage VOUT = 3.3V
6.0
INPUT VOLTAGE (V)
LT1934
VOUT = 3.3V
5.5 TA = 25°C
BOOST DIODE TIED TO OUTPUT
5.0
VIN TO START
4.5
4.0
VIN TO RUN
3.5
3.0
0.1
to VIN), then the LT1934’s internal circuitry will pull its
quiescent current through its SW pin. This is fine if your
system can tolerate a few mA in this state. If you ground
the SHDN pin, the SW pin current will drop to essentially
zero. However, if the VIN pin is grounded while the output
is held high, then parasitic diodes inside the LT1934 can
pull large currents from the output through the SW pin
and the VIN pin. Figure 4 shows a circuit that will run only
when the input voltage is present and that protects against
a shorted or reversed input.
D4
10
100
1
LOAD CURRENT (mA)
5
VIN
VIN
100k 4
Minimum Input Voltage VOUT = 5V
INPUT VOLTAGE (V)
GND
2
SW
6
VOUT
FB
3
BACKUP
VIN TO START
D4: MBR0530
6
1934 F07
Figure 4. Diode D4 Prevents a Shorted Input from Discharging
a Backup Battery Tied to the Output; It Also Protects the Circuit
from a Reversed Input. The LT1934 Runs Only When the Input
is Present
VIN TO RUN
5
4
0.1
SHDN
1M
LT1934
VOUT = 5V
TA = 25°C
7 BOOST DIODE TIED TO OUTPUT
1
LT1934
1934 G12
8
BOOST
PCB Layout
1
10
100
LOAD CURRENT (mA)
1934 G13
Figure 3. The Minimum Input Voltage Depends
on Output Voltage, Load Current and Boost Circuit
maximum duty cycle of the LT1934, requiring a higher
input voltage to maintain regulation.
Shorted Input Protection
If the inductor is chosen so that it won’t saturate excessively, an LT1934 buck regulator will tolerate a shorted
output. There is another situation to consider in systems
where the output will be held high when the input to the
LT1934 is absent. This may occur in battery charging applications or in battery backup systems where a battery
or some other supply is diode OR-ed with the LT1934’s
output. If the VIN pin is allowed to float and the SHDN pin
is held high (either by a logic signal or because it is tied
For proper operation and minimum EMI, care must be
taken during printed circuit board layout. Figure 5 shows
the high current paths in the buck regulator circuit. Note
that large, switched currents flow in the power switch,
the catch diode (D1) and the input capacitor (C2). The
loop formed by these components should be as small as
possible. Furthermore, the system ground should be tied
to the regulator ground in only one place; this prevents
the switched current from injecting noise into the system
ground. These components, along with the inductor and
output capacitor, should be placed on the same side of
the circuit board, and their connections should be made
on that layer. Place a local, unbroken ground plane below
these components, and tie this ground plane to system
ground at one location, ideally at the ground terminal of
the output capacitor C1. Additionally, the SW and BOOST
nodes should be kept as small as possible. Finally, keep
the FB node as small as possible so that the ground pin
1934fe
12
LT1934/LT1934-1
APPLICATIONS INFORMATION
VIN
VIN
SW
GND
SW
GND
(5a)
(5b)
IC1
VSW
VIN
C2
L1
SW
D1
GND
C1
1934 F05
(5c)
Figure 5. Subtracting the Current When the Switch is On (a) from the Current When the Switch is Off (b) Reveals the Path of the High
Frequency Switching Current (c). Keep This Loop Small. The Voltage on the SW and BOOST Nodes Will Also be Switched; Keep These
Nodes as Small as Possible. Finally, Make Sure the Circuit is Shielded with a Local Ground Plane
SHUTDOWN
VIN
VOUT
SYSTEM
GROUND
VIAS TO LOCAL GROUND PLANE
OUTLINE OF LOCAL GROUND PLANE
1934 F06
Figure 6. A Good PCB Layout Ensures Proper, Low EMI Operation
and ground traces will shield it from the SW and BOOST
nodes. Figure 6 shows component placement with trace,
ground plane and via locations. Include two vias near
the GND pin of the LT1934 to help remove heat from the
LT1934 to the ground plane.
Hot Plugging Safely
The small size, robustness and low impedance of ceramic
capacitors make them an attractive option for the input
bypass capacitor of LT1934 and LT1934-1 circuits. However, these capacitors can cause problems if the LT1934
is plugged into a live supply (see Linear Technology
Application Note 88 for a complete discussion). The low
loss ceramic capacitor combined with stray inductance in
series with the power source forms an under damped tank
circuit, and the voltage at the VIN pin of the LT1934 can
ring to twice the nominal input voltage, possibly exceeding
the LT1934’s rating and damaging the part. If the input
supply is poorly controlled or the user will be plugging
the LT1934 into an energized supply, the input network
should be designed to prevent this overshoot.
1934fe
13
LT1934/LT1934-1
APPLICATIONS INFORMATION
Figure 7 shows the waveforms that result when an LT1934
circuit is connected to a 24V supply through six feet of
24-gauge twisted pair. The first plot is the response with
a 2.2μF ceramic capacitor at the input. The input voltage
rings as high as 35V and the input current peaks at 20A.
One method of damping the tank circuit is to add another
capacitor with a series resistor to the circuit. In Figure 7b
an aluminum electrolytic capacitor has been added. This
capacitor’s high equivalent series resistance damps the
circuit and eliminates the voltage overshoot. The extra
capacitor improves low frequency ripple filtering and can
slightly improve the efficiency of the circuit, though it is
likely to be the largest component in the circuit. An alternative solution is shown in Figure 7c. A 1Ω resistor is added
CLOSING SWITCH
SIMULATES HOT PLUG
IIN
VIN
LT1934
+
VIN
10V/DIV
2.2μF
LOW
IMPEDANCE
ENERGIZED
24V SUPPLY
IIN
10A/DIV
STRAY
INDUCTANCE
DUE TO 6 FEET
(2 METERS) OF
TWISTED PAIR
10μs/DIV
(7a)
LT1934
10μF
35V
AI.EI.
+
2.2μF
(7b)
1Ω
LT1934
0.1μF
2.2μF
(7c)
LT1934-1
1μF
(7d)
4.7Ω
LT1934-1
0.1μF
1μF
(7e)
1934 F07
Figure 7. A Well Chosen Input Network Prevents Input Voltage Overshoot and
Ensures Reliable Operation When the LT1934 is Connected to a Live Supply
1934fe
14
LT1934/LT1934-1
APPLICATIONS INFORMATION
in series with the input to eliminate the voltage overshoot
(it also reduces the peak input current). A 0.1μF capacitor
improves high frequency filtering. This solution is smaller
and less expensive than the electrolytic capacitor. For high
input voltages its impact on efficiency is minor, reducing
efficiency less than one half percent for a 5V output at full
load operating from 24V.
Voltage overshoot gets worse with reduced input capacitance. Figure 7d shows the hot plug response with a 1μF
ceramic input capacitor, with the input ringing above 40V.
The LT1934-1 can tolerate a larger input resistance, such
as shown in Figure 7e where a 4.7Ω resistor damps the
voltage transient and greatly reduces the input current
glitch on the 24V supply.
High Temperature Considerations
The die temperature of the LT1934 must be lower than the
maximum rating of 125°C. This is generally not a concern
unless the ambient temperature is above 85°C. For higher
temperatures, care should be taken in the layout of the
circuit to ensure good heat sinking of the LT1934. The
maximum load current should be derated as the ambient
temperature approaches 125°C.
The die temperature is calculated by multiplying the LT1934
power dissipation by the thermal resistance from junction
to ambient. Power dissipation within the LT1934 can be
estimated by calculating the total power loss from an
efficiency measurement and subtracting the catch diode
loss. The resulting temperature rise at full load is nearly
independent of input voltage. Thermal resistance depends
on the layout of the circuit board, but a value of 150°C/W
is typical for the TSOT-23 and 75°C/W for the DFN.
The temperature rise for an LT1934 (TSOT-23) producing
5V at 250mA is approximately 25°C, allowing it to deliver
full load to 100°C ambient. Above this temperature the
load current should be reduced. For 3.3V at 250mA the
temperature rise is 15°C. The DFN temperature rise will
be roughly one-half of these values.
Finally, be aware that at high ambient temperatures the
external Schottky diode, D1, is likely to have significant
leakage current, increasing the quiescent current of the
LT1934 converter.
Outputs Greater Than 6V
For outputs greater than 6V, tie a diode (such as a 1N4148)
from the SW pin to VIN to prevent the SW pin from ringing
above VIN during discontinuous mode operation. The 12V
output circuit in Typical Applications shows the location of
this diode. Also note that for outputs above 6V, the input
voltage range will be limited by the maximum rating of
the BOOST pin. The 12V circuit shows how to overcome
this limitation using an additional Zener diode.
1934fe
15
LT1934/LT1934-1
TYPICAL APPLICATIONS
3.3V Step-Down Converter
D2
0.1μF
BOOST
VIN
4.5V TO 34V
D1
LT1934-1
SHDN
ON OFF
VOUT
3.3V
45mA
SW
VIN
C2
1μF
L1
100μH
10pF
1M
+
FB
C1
22μF
604k
GND
C1: TAIYO YUDEN JMK316BJ226ML
C2: TAIYO YUDEN GMK316BJ105ML
D1: ZETEX ZHCS400 OR ON SEMI MBR0540
D2: CENTRAL CMDSH-3
L1: COILCRAFT DO1608C-104 OR
WURTH ELECTRONICS WE-PD4 TYPE S
1934 TA04
5V Step-Down Converter
D2
0.1μF
BOOST
VIN
6.5V TO 34V
ON OFF
VOUT
5V
45mA
SW
VIN
C2
1μF
L1
150μH
D1
LT1934-1
SHDN
10pF
1M
+
FB
GND
C1: TAIYO YUDEN JMK316BJ226ML
C2: TAIYO YUDEN GMK316BJ105ML
D1: ZETEX ZHCS400 OR ON SEMI MBR0540
D2: CENTRAL CMPD914
L1: COILCRAFT DO1608C-154 OR
WURTH ELECTRONICS WE-PD4 TYPE S
C1
22μF
332k
1934 TA05
1934fe
16
LT1934/LT1934-1
TYPICAL APPLICATIONS
1.8V Step-Down Converter
D2
0.1μF
BOOST
VIN
3.6V TO 16V
VIN
C2
2.2μF
VOUT
1.8V
250mA
SW
D1
LT1934
147k
SHDN
ON OFF
L1
33μH
+
FB
C1
100μF
332k
GND
C1: SANYO 2R5TPB100M
C2: TAIYO YUDEN EMK316BJ225ML
D1: ZETEX ZHCS400 OR ON SEMI MBR0540
D2: CENTRAL CMPD914
L1: SUMIDA CR43-330
1934 TA06
Loop Powered 3.3V Supply with Additional Isolated Output
D3
L1B
50μH •
SW
VIN
D4
10V
390k
GND
•
D1
LT1934-1
SHDN
1μF
D2
C1 L1A
50μH
BOOST
VIN
14V TO 32V
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