LTC3606BIDD#PBF

LTC3606B 800mA Synchronous Step-Down DC/DC with Average Input Current Limit DESCRIPTION FEATURES n n n n n n n n n n n n n n Programmable Average Input Current Limit: ±5% Accuracy Step-Down Output: Up to 96% Efficiency Low Noise Pulse-Skipping Operation at Light Loads Input Voltage Range: 2.5V to 5.5V Output Voltage Range: 0.6V to 5V 2.25MHz Constant-Frequency Operation Power Good Output Voltage Monitor Low Dropout Operation: 100% Duty Cycle Internal Soft-Start Current Mode Operation for Excellent Line and Load Transient Response ±2% Output Voltage Accuracy Short-Circuit Protected Shutdown Current ≤ 1μA Available in Small Thermally Enhanced 8-Lead 3mm × 3mm DFN Package APPLICATIONS n n n n The LTC®3606B is an 800mA monolithic synchronous buck regulator using a constant frequency current mode architecture. The input supply voltage range is 2.5V to 5.5V, making it ideal for Li-Ion and USB powered applications. 100% duty cycle capability provides low dropout operation, extending the run time in battery-operated systems. Low output voltages are supported with the 0.6V feedback reference voltage. The LTC3606B can supply 800mA output current. The LTC3606B’s programmable average input current limit is ideal for USB applications and for point-of-load power supplies because the LTC3606B’s limited input current will still allow its output to deliver high peak load currents without collapsing the input supply. The operating frequency is internally set at 2.25MHz allowing the use of small surface mount inductors. Internal soft-start reduces in-rush current during start-up. The LTC3606B is available in an 8-Lead 3mm × 3mm DFN package. L, LT, LTC, LTM, Linear Technology, the Linear logo and Burst Mode are registered trademarks and Hot Swap is a trademark of Linear Technology Corporation. All other trademarks are the property of their respective owners. Protected by U.S.Patents, including 5481178, 6127815, 6304066, 6498466, 6580258, 6611131. High Peak Load Current Applications USB Powered Devices Supercapacitor Charging Radio Transmitters and Other Handheld Devices TYPICAL APPLICATION Monolithic Buck Regulator with Input Current Limit 1.5μH VIN 3.4V TO 5.5V VIN CIN 10μF RUN + PGOOD VFB RLIM PGOOD GND 1000pF VOUT 3.4V AT 800mA SW LTC3606B 499k 116k 1210k GSM Pulse Load 2.2mF s2 SuperCap VIN AC-COUPLED 1V/DIV IOUT 500mA/DIV 255k 3606B TA01 ILIM = 475mA VOUT 200mV/DIV IIN 500mA/DIV 1ms/DIV 3606B TA01b VIN = 5V, 500mA COMPLIANT ILOAD = 0A to 2.2A 3606bfb 1 LTC3606B ABSOLUTE MAXIMUM RATINGS PIN CONFIGURATION (Note 1) TOP VIEW Input Supply Voltage (VIN) ........................... –0.3V to 6V VFB ................................................... –0.3V to VIN + 0.3V RUN, RLIM ....................................... –0.3V to VIN + 0.3V SW ................................................... –0.3V to VIN + 0.3V PGOOD............................................. –0.3V to VIN + 0.3V P-Channel SW Source Current (DC) (Note 2) ..............1A N-Channel SW Source Current (DC) (Note 2)..............1A Peak SW Source and Sink Current (Note 2) ............. 2.7A Operating Junction Temperature Range (Notes 3, 6, 8) ........................................ –40°C to 125°C Storage Temperature Range .................. –65°C to 125°C Reflow Peak Body Temperature ............................ 260°C GND 1 RLIM 2 GND 3 SW 4 8 VFB 9 GND 7 RUN 6 PGOOD 5 VIN DD PACKAGE 8-LEAD (3mm s 3mm) PLASTIC DFN TJMAX = 125°C, θJA = 40°C/W EXPOSED PAD (PIN 9) IS GND, MUST BE SOLDERED TO PCB ORDER INFORMATION LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE LTC3606BEDD#PBF LTC3606BEDD#TRPBF LFMB 8-Lead (3mm × 3mm) Plastic DFN –40°C to 85°C LTC3606BIDD#PBF LTC3606BIDD#TRPBF LFMB 8-Lead (3mm × 3mm) Plastic DFN –40°C to 125°C Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. Consult LTC Marketing for information on non-standard lead based finish parts. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ 3606bfb 2 LTC3606B ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating junction temperature range, otherwise specifications are at TA = 25°C, VIN = 5V, unless otherwise noted. SYMBOL PARAMETER VIN VIN Operating Voltage Range CONDITIONS MIN VUV VIN Undervoltage Lockout IFB Feedback Pin Input Current VFBREG Feedback Voltage LTC3606BE, –40°C < TJ < 85°C (Note 7) LTC3606BI, –40°C < TJ < 125°C (Note 7) VLINEREG VFB Line Regulation VLOADREG l TYP MAX UNITS 5.5 V 2.5 V ±30 nA 0.600 0.600 0.612 0.618 V V VIN = 2.5V to 5.5V (Note 7) 0.01 0.25 %/V VFB Load Regulation ILOAD = 0mA to 800mA (Note 7) 0.5 Supply Current Active Mode (Note 4) Shutdown VFB = 0.95 × VFBREG VRUN = 0V, VIN = 5.5V 420 650 1 μA μA fOSC Oscillator Frequency VFB = VFBREG 1.8 2.25 2.7 MHz ILIM(PEAK) Peak Switch Current Limit VIN = 5V, VFB < VFBREG , Duty Cycle 100pF at the RLIM pin. Each application input current limit will call for different CLIM value to optimize its response time. Using a large CLIM capacitor requires longer time for the RLIM pin voltage to charge. For example, consider the application 500mA input current limit, 5V input and 1A, 2.5V output with a 50% duty cycle. When an instantaneous 1A output pulse is applied, the current out of the RLIM pin becomes 1A/55k = 18.2μA during the 50% on-time or 9.1μA full duty cycle. With a CLIM capacitor of 1μF, RLIM of 116k, and using I = CdV/dt, it will take 110ms for CLIM to charge from 0V to 1V. This is the time after which the LTC3606B will start input current limiting. Any current within this time must be considered in each application to determine if it is tolerable. 3606bfb 8 LTC3606B OPERATION Figure 1a shows VIN (IIN) current below input current limit with a CLIM capacitor of 0.1μF. When the load pulse is applied, under the specified condition, ILIM current is 1.1A/55k • 0.66 = 13.2μA, where 0.66 is the duty cycle. It will take a little more than 7.5ms to charge the CLIM capacitor from 0V to 1V, after which the LTC3606B begins to limit input current. The IIN current is not limited during this 7.5ms time and is more than 725mA. This current transient may cause the input supply to temporarily droop if the supply current compliance is exceeded, but recovers after the input current limit engages. The output will continue to deliver the required current load while the output voltage droops to allow the input voltage to remain regulated during input current limit. and the output must deliver the required current load. This may cause the input voltage to droop if the current compliance is exceeded. Depending on how long this time is, the VIN supply decoupling capacitor can provide some of this current before VIN droops too much. In applications with a bigger VIN supply decoupling capacitor and where VIN supply is allow to droop closer to dropout, the CLIM capacitor can be increased slightly. This will delay the start of input current limit and artificially regulated VOUT before input current limit is engaged. In this case, within the 577μs load pulse, the VOUT voltage will stay artificially regulated for 92μs out of the total 577μs before the input current limit activates. This approach may be used if a faster recovery on the output is desired. For applications with short load pulse duration, a smaller CLIM capacitor may be the better choice as in the example shown in Figure 1b. In this example, a 577μs, 0A to 2A output pulse is applied once every 4.7ms. A CLIM capacitor of 2.2nF requires 92μs for VRLIM to charge from 0V to 1V. During this 92μs, the input current limit is not yet engaged Selecting a very small CLIM will speed up response time but it can put the device within threshold of interfering with normal operation and input current limit in every few switching cycles. This may be undesirable in terms of noise. Use 2πRC >> 100/clock frequency (2.25MHz) as a starting point, R being RLIM, C being CLIM. VOUT 2V/DIV VOUT 200mV/DIV IIN 500mA/DIV VIN AC-COUPLED 1V/DIV VRLIM 1V/DIV IOUT 500mA/DIV IIN 500mA/DIV IL 1A/DIV 50ms/DIV 3606B F01a VIN = 5V, 500mA COMPLIANT RLIM = 116k, CLIM = 0.1μF ILOAD = 0A to 1.1A, COUT = 2.2mF, VOUT = 3.3V ILIM = 475mA Figure 1a. Input Current Limit Within 100ms Load Pulses 1ms/DIV 3606B F01b VIN = 5V, 500mA COMPLIANT RLIM = 116k, CLIM = 2200pF ILOAD = 0A to 2A, COUT = 2.2mF, VOUT = 3.3V ILIM = 475mA Figure 1b. Input Current Limit Within 577μs, 2A Repeating Load Pulses 3606bfb 9 LTC3606B APPLICATIONS INFORMATION A general LTC3606B application circuit is shown in Figure 2. External component selection is driven by the load requirement, and begins with the selection of the inductor L. Once the inductor is chosen, CIN and COUT can be selected. Inductor Selection Although the inductor does not influence the operating frequency, the inductor value has a direct effect on ripple current. The inductor ripple current IL decreases with higher inductance and increases with higher VIN or VOUT :  V  V
IL = OUT • 1 OUT  (1) fO • L  VIN  Accepting larger values of IL allows the use of low inductances, but results in higher output voltage ripple, greater core losses, and lower output current capability. A reasonable starting point for setting ripple current is 40% of the maximum output load current. So, for a 800mA regulator, IL = 320mA (40% of 800mA). The inductor value will also have an effect on Burst Mode operation. The transition to low current operation begins when the peak inductor current falls below a level set by the internal burst clamp. Lower inductor values result in higher ripple current which causes the transition to occur at lower load currents. This causes a dip in efficiency in the upper range of low current operation. Furthermore, lower inductance values will cause the bursts to occur with increased frequency. L1 VIN 2.5V TO 5.5V VIN RPGD CIN VOUT SW LTC3606B CF RUN COUT PGOOD VFB RLIM PGOOD GND RLIM R2 R1 CLIM 3606B F02 Figure 2. LTC3606B General Schematic Inductor Core Selection Different core materials and shapes will change the size/ current and price/current relationship of an inductor. Toroid or shielded pot cores in ferrite or permalloy materials are small and do not radiate much energy, but generally cost more than powdered iron core inductors with similar electrical characteristics. The choice of which style inductor to use often depends more on the price versus size requirements, and any radiated field/EMI requirements, than on what the LTC3606B requires to operate. Table 1 shows some typical surface mount inductors that work well in LTC3606B applications. Table 1. Representative Surface Mount Inductors MANUFACTURER PART NUMBER MAX DC VALUE CURRENT DCR HEIGHT Coilcraft LPS4012-152ML LPS4012-222ML LPS4012-332ML LPS4012-472ML LPS4018-222ML LPS4018-332ML LPS4018-472ML 1.5μH 2.2μH 3.3μH 4.7μH 2.2μH 3.3μH 4.7μH 2200mA 1750mA 1450mA 1450mA 2300mA 2000mA 1800mA 0.070Ω 0.100Ω 0.100Ω 0.170Ω 0.070Ω 0.080Ω 0.125Ω 1.2mm 1.2mm 1.2mm 1.2mm 1.8mm 1.8mm 1.8mm FDK FDKMIPF2520D FDKMIPF2520D FDKMIPF2520D 4.7μH 3.3μH 2.2μH 1100mA 1200mA 1300mA 0.11Ω 0.1Ω 0.08Ω 1mm 1mm 1mm LQH32CN4R7M23 4.7μH 450mA 0.2Ω 2mm ELT5KT4R7M 4.7μH 950mA 0.2Ω 1.2mm CDRH2D18/LD CDH38D11SNP3R3M CDH38D11SNP2R2M 4.7μH 3.3μH 630mA 1560mA 0.086Ω 0.115Ω 2mm 1.2mm 2.2μH 1900mA 0.082Ω 1.2mm 2.2μH 2.2μH 3.3μH 2.2μH 4.7μH 510mA 530mA 410mA 1100mA 750mA 0.13Ω 0.33Ω 0.27Ω 0.1Ω 0.19Ω 1.6mm 1.25mm 1.6mm 1mm 1mm 4.7μH 700mA 0.28Ω 1mm 3.3μH 870mA 0.17Ω 1mm 2.2μH 1000mA 0.12Ω 1mm 2.2μH 1500mA 0.076Ω 1.2mm 3.3μH 1700mA 0.095Ω 1.2mm 2.2μH 2300mA 0.059Ω 1.4mm Murata Panasonic Sumida Taiyo Yuden CB2016T2R2M CB2012T2R2M CB2016T3R3M NR30102R2M NR30104R7M TDK VLF3010AT4R7MR70 VLF3010AT3R3MR87 VLF3010AT2R2M1R0 VLF4012AT-2R2 M1R5 VLF5012ST-3R3 M1R7 VLF5014ST-2R2 M2R3 3606bfb 10 LTC3606B APPLICATIONS INFORMATION Input Capacitor (CIN) Selection In continuous mode, the input current of the converter is a square wave with a duty cycle of approximately VOUT / VIN . To prevent large voltage transients, a low equivalent series resistance (ESR) input capacitor sized for the maximum RMS current must be used. The maximum RMS capacitor current is given by: IRMS
IMAX VOUT (VIN  VOUT ) VIN Where the maximum average output current IMAX equals the peak current minus half the peak-to-peak ripple current, IMAX = ILIM – IL /2. This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT/2. This simple worst-case is commonly used to design because even significant deviations do not offer much relief. Note that capacitor manufacturer’s ripple current ratings are often based on only 2000 hours lifetime. This makes it advisable to further derate the capacitor, or choose a capacitor rated at a higher temperature than required. Several capacitors may also be paralleled to meet the size or height requirements of the design. An additional 0.1μF to 1μF ceramic capacitor is also recommended on VIN for high frequency decoupling when not using an all-ceramic capacitor solution. Output Capacitor (COUT) Selection The selection of COUT is driven by the required effective series resistance (ESR). Typically, once the ESR requirement for COUT has been met, the RMS current rating generally far exceeds the IRIPPLE(P-P) requirement. The output ripple VOUT is determined by:  1  VOUT
IL ESR+  8fOCOUT   where fO = operating frequency, COUT = output capacitance and IL = ripple current in the inductor. For a fixed output voltage, the output ripple is highest at maximum input voltage since IL increases with input voltage. If tantalum capacitors are used, it is critical that the capacitors are surge tested for use in switching power supplies. An excellent choice is the AVX TPS series of surface mount tantalum. These are specially constructed and tested for low ESR so they give the lowest ESR for a given volume. Other capacitor types include Sanyo POSCAP, Kemet T510 and T495 series, and Sprague 593D and 595D series. Consult the manufacturer for other specific recommendations. Using Ceramic Input and Output Capacitors Higher values, lower cost ceramic capacitors are now becoming available in smaller case sizes. Their high ripple current, high voltage rating and low ESR make them ideal for switching regulator applications. Because the LTC3606B control loop does not depend on the output capacitor’s ESR for stable operation, ceramic capacitors can be used freely to achieve very low output ripple and small circuit size. However, care must be taken when ceramic capacitors are used at the input. When a ceramic capacitor is used at the input and the power is supplied by a wall adapter through long wires, a load step at the output can induce ringing at the input, VIN. At best, this ringing can couple to the output and be mistaken as loop instability. At worst, a sudden inrush of current through the long wires can potentially cause a voltage spike at VIN, large enough to damage the part. For more information, see Application Note 88. When choosing the input and output ceramic capacitors, choose the X5R or X7R dielectric formulations. These dielectrics have the best temperature and voltage characteristics of all the ceramics for a given value and size. 3606bfb 11 LTC3606B APPLICATIONS INFORMATION Setting the Output Voltage The LTC3606B regulates the VFB pin to 0.6V during regulation. Thus, the output voltage is set by a resistive divider, Figure 2, according to the following formula: VOUT = 0.6V 1+ R2 R1 (2) The output voltage settling behavior is related to the stability of the closed-loop system and will demonstrate the actual overall supply performance. For a detailed explanation of optimizing the compensation components, including a review of control loop theory, refer to Application Note 76. To improve the frequency response of the main control loop, a feedback capacitor (CF) may also be used. Great care should be taken to route the VFB line away from noise sources, such as the inductor or the SW line. In some applications, a more severe transient can be caused by switching in loads with large (>1μF) input capacitors. The discharged input capacitors are effectively put in parallel with COUT, causing a rapid drop in VOUT. No regulator can deliver enough current to prevent this problem if the switch connecting the load has low resistance and is driven quickly. The solution is to limit the turn-on speed of the load switch driver. A Hot Swap™ controller is designed specifically for this purpose and usually incorporates current limiting, short-circuit protection, and soft-starting. Checking Transient Response Efficiency Considerations The regulator loop response can be checked by looking at the load transient response. Switching regulators take several cycles to respond to a step in load current. When a load step occurs, VOUT immediately shifts by an amount equal to ILOAD • ESR, where ESR is the effective series resistance of COUT. ILOAD also begins to charge or discharge COUT generating a feedback error signal used by the regulator to return VOUT to its steady-state value. During this recovery time, VOUT can be monitored for overshoot or ringing that would indicate a stability problem. The percent efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Percent efficiency can be expressed as: The initial output voltage step may not be within the bandwidth of the feedback loop, so the standard second order overshoot/DC ratio cannot be used to determine the phase margin. In addition, feedback capacitors (CF) can be added to improve the high frequency response, as shown in Figure 2. Capacitor CF provides phase lead by creating a high frequency zero with R2 which improves the phase margin. Although all dissipative elements in the circuit produce losses, four sources usually account for the losses in LTC3606B circuits: 1) VIN quiescent current, 2) switching losses, 3) I2R losses, 4) other system losses. Keeping the current small (< 10μA) in these resistors maximizes efficiency, but making it too small may allow stray capacitance to cause noise problems or reduce the phase margin of the error amp loop. % Efficiency = 100% – (L1 + L2 + L3 + ...) where L1, L2, etc., are the individual losses as a percentage of input power. 1. The VIN current is the DC supply current given in the Electrical Characteristics which excludes MOSFET driver and control currents. VIN current results in a small (