LT3688IFEPBF

FEATURES n n n n n n n n LT3688 Dual 800mA Step-Down Switching Regulator with Power-On Reset and Watchdog Timer DESCRIPTION The LT®3688 is an adjustable frequency (350kHz to 2.2MHz) dual monolithic step-down switching regulator with two power-on reset timers and a watchdog timer. The regulator operates off inputs up to 36V. Low ripple Burst Mode® operation maintains high efficiency at low output current while keeping output ripple below 25mV in a typical application, with input quiescent current of just 115μA. Shutdown circuitry reduces input supply current to less than 1μA while EN/UVLO is pulled low. The reset and watchdog timeout periods are both adjustable using external capacitors. Tight accuracy specifications and glitch immunity ensure reliable reset operation without false triggering. The open collector RST pins will pull down if the monitored output voltage drops 10% below the programmed value. The LT3688 watchdog timer monitors for watchdog falling edges grouped too close together or too far apart. The LT3688 is available in 24-Pin TSSOP and 4mm × 4mm QFN packages, each with an exposed pad for low thermal resistance. Wide Input Range: Operation from 3.8V to 36V Low Ripple ( 125°C). As the switching frequency is increased above fSW(MAX), the part is required to switch for shorter periods to maintain the same duty cycle. Delays associated with turning off the power switch dictate the minimum on-time of the part. When the required on-time decreases below the minimum on-time of 140ns, the switch pulse width remains fixed at 140ns (instead of becoming narrower) to accommodate the same duty cycle requirement. The inductor current ramps up to a value exceeding the load current and the output ripple increases. The part then remains off until the output voltage dips below the programmed value before it begins switching again. Maximum Operating Voltage Range The maximum input voltage for LT3688 applications depends on switching frequency, the absolute maximum ratings of the VIN and BST pins, and by the minimum duty cycle (DCMIN). The LT3688 can operate from input voltages up to 36V. DCMIN = tON(MIN) • fSW where tON(MIN) is equal to 140ns and fSW is the switching frequency. Running at a lower switching frequency allows a lower minimum duty cycle. The maximum input voltage before pulse-skipping occurs depends on the output voltage and the minimum duty cycle: V + VF VIN(PS) = OUT – VF + VSW DCMIN Example: f = 2.1MHz, VOUT = 3.3V DCMIN = 140ns • 2.1MHz = 0.294 3.3V + 0.5V VIN(PS) = – 0.5V + 0.3V = 12.7V 0.294 3688f Operating Frequency Tradeoffs Selection of the operating frequency is a tradeoff between efficiency, component size and maximum input voltage. The advantage of high frequency operation is that smaller inductor and capacitor values may be used. The disadvantages are lower efficiency, and narrower input voltage range at constant-frequency. The highest constantswitching frequency (fSW(MAX)) for a given application can be calculated as follows: VOUT + VF fSW(MAX) = tON(MIN) ( VIN + VF – VSW ) 13 LT3688 APPLICATIONS INFORMATION The LT3688 will regulate the output voltage at input voltages greater than VIN(PS). For example, an application with an output voltage of 3.3V and switching frequency of 2.1MHz has a VIN(PS) of 12.7V, as shown in Figure 1. Figure 2 shows operation at 27V. Output ripple and peak inductor current have significantly increased. A saturating inductor may further reduce performance. In pulse skip mode, the LT3688 skips switching pulses to maintain output regulation. The LT3688 will also skip pulses at very low load currents. VIN(PS) vs load current is plotted in the Typical Performance section. VOUT 50mV/DIV (AC) IL 500mA/DIV Unlike many fixed frequency regulators, the LT3688 can extend its duty cycle by remaining on for multiple cycles. The LT3688 will not switch off at the end of each clock cycle if there is sufficient voltage across the boost capacitor (C3 in the Block Diagram). Eventually, the voltage on the boost capacitor falls and requires refreshing. Circuitry detects this condition and forces the switch to turn off, allowing the inductor current to charge up the boost capacitor. This places a limitation on the maximum duty cycle as follows: DCMAX = 90% This leads to a minimum input voltage of: V + VF VIN(MIN) = OUT – VF + VSW DCMAX where VF is the forward voltage drop of the catch diode (~0.4V) and VSW is the voltage drop of the internal switch (~0.3V at maximum load). Example: ISW=0.8A and VOUT = 3.3V 3.3V + 0.4V – 0.4 + 0.3V = 4V 90% For best performance in dropout, use a 1μF or larger boost capacitor. VIN(MIN) = Inductor Selection and Maximum Output Current 2μs/DIV 3688 F01 Figure 1. Operation Below Pulse-Skipping Voltage. VOUT = 3.3V and fSW = 2.1MHz VOUT 50mV/DIV (AC) IL 500mA/DIV 2μs/DIV 3688 F02 A good first choice for the inductor value is L = ( VOUT + VF ) • 1.8MHz fSW Figure 2. Operation Above VIN(ps). VIN = 27V, VOUT = 3.3V and fSW = 2.1MHz. Output Ripple and Peak Inductor Current Increase Minimum Operating Voltage Range The minimum input voltage is determined either by the LT3688’s minimum operating voltage of ~3.6V or by its maximum duty cycle. The duty cycle is the fraction of time that the internal switch is on and is determined by the input and output voltages: DC = VOUT + VF VIN – VSW + VF where VF is the voltage drop of the catch diode (~0.4V), fSW is in MHz, and L is in μH. The inductor’s RMS current rating must be greater than the maximum load current and its saturation current should be at least 30% higher. For robust operation in fault conditions (start-up or shortcircuit) and high input voltage (>30V), use an 8.2μH or greater inductor with a saturation rating of 2.2A, or higher. For highest efficiency, the series resistance (DCR) should be less than 0.1Ω. Table 2 lists several vendors and types that are suitable. 3688f 14 LT3688 APPLICATIONS INFORMATION Table 2. Inductor Vendors VENDOR Murata TDK Toko PART SERIES LQH55D SLF7045 SLF10145 DC62CB D63CB D75C D75F CR54 CDRH74 CDRH6D38 CR75 TYPE Open Shielded Shielded Shielded Shielded Shielded Open Open Shielded Shielded Open URL www.murata.com www.component.tdk.com www.toko.com When the switch is off, the potential across the inductor is the output voltage plus the catch diode drop. This gives the peak-to-peak ripple current in the inductor ΔIL = (1– DC) ( VOUT + VF ) L•f Sumida www.sumida.com where f is the switching frequency of the LT3688 and L is the value of the inductor. The peak inductor and switch current is ΔI ISW(PK) = IL(PK) = IOUT + L 2 To maintain output regulation, this peak current must be less than the LT3688’s switch current limit ILIM . ILIM is at least 1.25A for at low duty cycles and decreases linearly to 0.9A at DC = 0.9. The maximum output current is a function of the chosen inductor value: IOUT(MAX) = ILIM – ΔIL 2 ΔIL 2 The optimum inductor for a given application may differ from the one indicated by this simple design guide. A larger value inductor provides a higher maximum load current, and reduces the output voltage ripple. If your load is lower than the maximum load current, then you can relax the value of the inductor and operate with higher ripple current. This allows you to use a physically smaller inductor, or one with a lower DCR resulting in higher efficiency. Be aware that if the inductance differs from the simple rule above, then the maximum load current will depend on input voltage. In addition, low inductance may result in discontinuous mode operation, which further reduces maximum load current. Discontinuous operation occurs when IOUT is less than ΔIL / 2. For details of maximum output current and discontinuous mode operation, see Linear Technology’s Application Note AN44. Finally, for duty cycles greater than 50% (VOUT/VIN > 0.5), a minimum inductance is required to avoid sub-harmonic oscillations: 1.2MHz LMIN = ( VOUT + VF ) • fSW where VF is the voltage drop of the catch diode (~0.4V), fSW is in MHz, and LMIN is in μH. The current in the inductor is a triangle wave with an average value equal to the load current. The peak switch current is equal to the output current plus half the peak-to-peak inductor ripple current. The LT3688 limits its switch current in order to protect itself and the system from overload faults. Therefore, the maximum output current that the LT3688 will deliver depends on the switch current limit, the inductor value, and the input and output voltages. = 1.25A • (1– 0.3DC) – Choosing an inductor value so that the ripple current is small will allow a maximum output current near the switch current limit. One approach to choosing the inductor is to start with the simple rule given above, look at the available inductors, and choose one to meet cost or space goals. Then use these equations to check that the LT3688 will be able to deliver the required output current. Note again that these equations assume that the inductor current is continuous. Input Capacitor Bypass the input of the LT3688 circuit with a ceramic capacitor of an X7R or X5R type. Y5V types have poor performance over temperature and applied voltage, and should not be used. A 2.2μF to 4.7μF ceramic capacitor is adequate to bypass the LT3688 and will easily handle the ripple current. Note that larger input capacitance is required when a lower switching frequency is used. If the input power source has high impedance, or there is significant inductance due to long wires or cables, 3688f 15 LT3688 APPLICATIONS INFORMATION additional bulk capacitance may be necessary. This can be provided with a lower performance electrolytic capacitor. Step-down regulators draw current from the input supply in pulses with very fast rise and fall times. The input capacitor is required to reduce the resulting voltage ripple at the LT3688 input and to force this very high frequency switching current into a tight local loop, minimizing EMI. A 2.2μF capacitor is capable of this task, but only if it is placed close to the LT3688 and the catch diode (see the PCB Layout section). A second precaution regarding the ceramic input capacitor concerns the maximum input voltage rating of the LT3688. A ceramic input capacitor combined with trace or cable inductance forms a high quality (under damped) tank circuit. If the LT3688 circuit is plugged into a live supply, the input voltage can ring to twice its nominal value, possibly exceeding the LT3688’s voltage rating. See Linear Technology’s Application Note 88 for details. Output Capacitor and Output Ripple The output capacitor has two essential functions. Along with the inductor, it filters the square wave generated by the LT3688 to produce the DC output. In this role it determines the output ripple, and low impedance at the switching frequency is important. The second function is to store energy in order to satisfy transient loads and stabilize the LT3688’s control loop. Ceramic capacitors have very low equivalent series resistance (ESR) and provide the best ripple performance. A good starting value is: COUT = 50 VOUT • fSW Taiyo Yuden High performance electrolytic capacitors can be used for the output capacitor. Low ESR is important, so choose one that is intended for use in switching regulators. The ESR should be specified by the supplier and should be 0.1Ω or less. Such a capacitor will be larger than a ceramic capacitor and will have a larger capacitance because the capacitor must be large to achieve low ESR. Table 3 lists several capacitor vendors. Table 3. Capacitor Vendors VENDOR Panasonic PART SERIES Ceramic Polymer Tantalum Ceramic Tantalum Ceramic Polymer Tantalum Ceramic Ceramic Tantalum Ceramic COMMENTS EEEF Series Kemet Sanyo T494, T495 POSCAP Murata AVX TPS Series Catch Diode The catch diode conducts current only during switch-off time. Average forward current in normal operation can be calculated from: ID(AVG) = IOUT ( VIN – VOUT ) VIN where fSW is in MHz and COUT is the recommended output capacitance in μF. Use X5R or X7R types, which will provide low output ripple and good transient response. Transient performance can be improved with a high value capacitor, but a phase lead capacitor across the feedback resistor R1 may be required to get the full benefit (see the Compensation section). where IOUT is the output load current. The only reason to consider a diode with a larger current rating than necessary for nominal operation is for the worst-case condition of shorted output. The diode current will then increase to the typical peak switch current limit. Peak reverse voltage is equal to the regulator input voltage. Use a Schottky diode with a reverse voltage rating greater than the input voltage. Table 4 lists several Schottky diodes and their manufacturers. 3688f 16 LT3688 APPLICATIONS INFORMATION Table 4. Capacitor Vendors Part Number On Semiconductor MBR0520L MBR0540 MBRM120E MBRM140 Diodes Inc. B0530W B120 B130 B140HB DFLS140 30 20 30 40 40 0.5 1 1 1 1.1 510 500 500 20 40 20 40 0.5 0.5 1 1 620 530 550 VR (V) IAVE (A) VF at 1A (mV) Ceramic Capacitors Ceramic capacitors are small, robust and have very low ESR. However, ceramic capacitors can cause problems when used with the LT3688 due to their piezoelectric nature. When in Burst Mode operation, the LT3688’s switching frequency depends on the load current, and at very light loads the LT3688 can excite the ceramic capacitor at audio frequencies, generating audible noise. Since the LT3688 operates at a lower current limit during Burst Mode operation, the noise is typically very quiet. If this is unacceptable, use a high performance tantalum or electrolytic capacitor at the output. Frequency Compensation The LT3688 uses current mode control to regulate the output, which simplifies loop compensation. In particular, the LT3688 does not require the ESR of the output capacitor for stability, allowing the use of ceramic capacitors to achieve low output ripple and small circuit size. Figure 3 shows an equivalent circuit for the LT3688 control loop. The error amp is a transconductance amplifier with finite output impedance. The power section, consisting of the modulator, power switch and inductor, is modeled as a transconductance amplifier generating an output current proportional to the voltage at the VC node. Note that the output capacitor, C1, integrates this current, and that the capacitor on the VC node (CC) integrates the error amplifier output current, resulting in two poles in the loop. RC provides a zero. With the recommended output capacitor, the loop crossover occurs above the RCCC zero. This simple model works well as long as the value of the inductor is not too high and the loop crossover frequency is much lower than the switching frequency. With a larger ceramic capacitor (very low ESR), crossover may be lower and a phase lead capacitor (CPL) across the feedback divider may improve the phase margin and transient response. At minimum, use a 10pF phase lead capacitor to reduce noise injection to the FB pin. If the output capacitor is different than the recommended capacitor, stability should be checked across all operating conditions, including load current, input voltage and temperature. The LT1375 data sheet contains a more thorough discussion of loop compensation and describes how to test the stability using a transient load. Figure 4 shows the transient response when the load current is stepped from 300mA to 600mA and back to 300mA. LT3688 0.7V CURRENT MODE POWER STAGE OUT FB R1 CPL VC RC 80k CC 100pF GND 3M ERROR AMPLIFIER Figure 3. Model for the Loop Response VOUT 100mV/DIV ILOAD 200mA/DIV Figure 4. Transient Load Response of the LT3688 Front Page Application as the Load Current is Stepped from 300mA to 600mA + gm = 300μA/V – – + gm = 1.6A/V 800mV ESR + C1 R2 C1 TANTALUM OR CERAMIC ELECTROLYTIC 3688 F03 50μs/DIV 3688 F04 3688f 17 LT3688 APPLICATIONS INFORMATION Low Ripple Burst Mode Operation To enhance efficiency at light loads, the LT3688 operates in low ripple Burst Mode operation that keeps the output capacitor charged to the proper voltage while minimizing the input quiescent current. During Burst Mode operation, the LT3688 delivers single cycle bursts of current to the output capacitor followed by sleep periods where the output power is delivered to the load by the output capacitor. Because the LT3688 delivers power to the output with single, low current pulses, the output ripple is kept below 25mV for a typical application. In addition, VIN and BIAS quiescent currents are reduced to typically 65μA and 155μA, respectively, during the sleep time. As the load current decreases towards a no-load condition, the percentage of time that the LT3688 operates in sleep mode increases and the average input current is greatly reduced, resulting in high efficiency even at very low loads (see Figure 5). At higher output loads the LT3688 will be running at the frequency programmed by the RT resistor, and will be operating in standard PWM mode. The transition between PWM and low ripple Burst Mode operation is seamless, and will not disturb the output voltage. The front page application circuit will switch at full frequency at output loads higher than about 60mA. three ways to arrange the boost circuit. The BST pin must be more than 2.3V above the SW pin for best efficiency. For outputs of 3V and above, the standard circuit (Figure 6a) is best. For outputs between 2.8V and 3V, use a 1μF boost capacitor. A 2.5V output presents a special case because it is marginally adequate to support the boosted drive stage while using the internal boost diode. For reliable BST pin operation with 2.5V outputs, use a good external Schottky diode (such as the ON semi MBR0540), and a 1μF boost capacitor (see Figure 6b). For lower output voltages, the boost diode can be tied to the input (Figure 6c), or to another supply greater than 2.8V. The circuit in Figure 6a is more efficient because the BST pin current and BIAS pin quiescent current comes from a lower voltVOUT BIAS BST VIN VIN LT3688 SW 4.7μF GND C3 (6a) For VOUT > 2.8V VOUT BIAS BST D2 IL 0.2A/DIV VSW 5V/DIV VOUT 10mV/DIV VIN VIN LT3688 SW C3 4.7μF GND (6b) For 2.5V < VOUT < 2.8V 5μs/DIV 3688 F05 VOUT Figure 5. Burst Mode Operation VIN BIAS BST VIN LT3688 SW BST and BIAS Pin Considerations Capacitor C3 and the internal boost Schottky diodes (see the Block Diagram) are used to generate boost voltages that are higher than the input voltage. In most cases, a 0.22μF capacitor will work well. For the best performance in dropout, use a 1μF or larger capacitor. Figure 6 shows 4.7μF GND C3 3688 F06 (6c) For VOUT < 2.5V; VIN(MAX) = 30V Figure 6. Three Circuits for Generating the Boost Voltage 3688f 18 LT3688 APPLICATIONS INFORMATION age source. However, the full benefit of the BIAS pin is not realized unless it is at least 3V. Ensure that the maximum voltage ratings of the BST and BIAS pins are not exceeded. The minimum operating voltage of an LT3688 application is limited by the minimum input voltage (3.6V) and by the maximum duty cycle, as outlined in a previous section. For proper start-up, the minimum input voltage is also limited by the boost circuit. If the input voltage is ramped slowly, or the LT3688 is turned on with its EN/UVLO pin when the output is already in regulation, then the boost capacitor may not be fully charged. Because the boost capacitor is charged with the energy stored in the inductor, the circuit will rely on some minimum load current to get the boost circuit running properly. This minimum load will depend on input and output voltages, and on the arrangement of the boost circuit. The minimum load generally goes to zero once the circuit has started. Figure 7 shows a plot of minimum load to start and to run as a function of input voltage. In many cases, the discharged output capacitor will present a load to the switcher, which will allow it to start. The plots show the worst-case situation where VIN is ramping very slowly. For lower start-up voltage, the boost diode can be tied to VIN ; however, this restricts the input range to one-half of the absolute maximum rating of the BST pin. At light loads, the inductor current becomes discontinuous and the effective duty cycle can be very high. This reduces the minimum input voltage to approximately 300mV above VOUT. At higher load currents, the inductor current is continuous and the duty cycle is limited by the maximum duty cycle of the LT3688, requiring a higher input voltage to maintain regulation. There is one particular issue to note if sequencing is used. If the BIAS pin is tied to VOUT2, it will be low during the startup of VOUT1. This will prevent the boost circuit from working on VOUT1 until it has risen to 90% of its programmed value, increasing the required startup voltage. Using circuit in Figure 6b for VOUT1 will reduce the startup voltage to its normal value. An alternative is to tie BIAS to VOUT1, if it is greater than 2.8V. Soft-Start and Individual Channel Shutdown The RUN/SS (Run/Soft-Start) pins are used to place the individual switching regulators in shutdown mode. They also provide a soft-start function. To shut down either INPUT VOLTAGE (V) 8 7.5 TO START 7 6.5 6 5.5 5 4.5 4 1 10 LOAD (mA) 3688 F07a VOUT = 5V TO RUN 100 1000 7.0 6.5 6.0 INPUT VOLTAGE (V) 5.5 5.0 4.5 4.0 VOUT = 3.3V TO START TO RUN 3.5 3.0 1 10 LOAD (mA) 3688 F07b 100 1000 Figure 7. The Minimum Input Voltage Depends on Output Voltage, Load Current and Boost Circuit regulator, pull the RUN/SS pin to ground with an opendrain or collector. Note that if CONFIG is tied high or low (not open), shutting down Channel 1 will also shut down Channel 2 because of the sequencing function (See the Configuration and Sequencing section for more details). 2.5μA current sources pull up on each pin. If the RUN/SS pin reaches ~0.2V, the channel will begin to switch If a capacitor is tied from the RUN/SS pin to ground, then the internal pull-up current will generate a voltage ramp on this pin. This voltage clamps the VC pin, limiting the peak switch current and therefore input current during start up. A good value for the soft-start capacitor is COUT/10,000, where COUT is the value of the output capacitor. The RUN/SS pins can be left floating if the Soft-Start feature is not used. They can also be tied together with a single capacitor providing soft-start. The internal current sources 3688f 19 LT3688 APPLICATIONS INFORMATION will charge these pins to ~2V. The RUN/SS pins provide a soft-start function that limits peak input current to the circuit during start-up. This helps to avoid drawing more current than the input source can supply or glitching the input supply when the LT3688 is enabled. The RUN/SS pins do not provide an accurate delay to start or an accurately controlled ramp at the output voltage, both of which depend on the output capacitance and the load current. Synchronization Synchronizing the LT3688 oscillator to an external frequency can be done by connecting a square wave (with positive and negative pulse width > 150ns) to the SYNC pin. The square wave amplitude should have valleys that are below 0.4V and peaks that are above 1.3V (up to 6V). The LT3688 may be synchronized over a 350kHz to 2.5MHz range. The RT resistor should be chosen to set the LT3688 switching frequency 20% below the lowest synchronization input. For example, if the synchronization signal will be 350kHz and higher, RT should be chosen for 280kHz. To assure reliable and safe operation, the LT3688 will only synchronize when the output voltage is above 90% of its regulated voltage. It is therefore necessary to choose a large enough inductor value to supply the required output current at the frequency set by the RT resistor (see the Inductor Selection section). It is also important to note that the slope compensation is set by the RT value. When the sync frequency is much higher than the one set by RT, the slope compensation will be significantly reduced, which may require a larger inductor value to prevent subharmonic oscillation. Shutdown and Undervoltage Lockout Figure 8 shows how to add undervoltage lockout (UVLO) to the LT3688. Typically, UVLO is used in situations where the input supply is current limited, or has a relatively high source resistance. A switching regulator draws constant power from the source, so source current increases as source voltage drops. This looks like a negative resistance load to the source and can cause the source to current limit or latch low under low source voltage conditions. UVLO prevents the regulator from operating at source voltages where the problems might occur. An internal comparator will force the part into shutdown below the minimum VIN of 3.5V. This feature can be used to prevent excessive discharge of battery-operated systems. If an adjustable UVLO threshold is required, the EN/UVLO pin can be used. The threshold voltage of the EN/UVLO pin comparator is 1.25V. Current hysteresis is added above the EN threshold. This can be used to set voltage hysteresis of the UVLO using the following: R3 = R4 = VH – VL 3.7μA R3 • 1.25V VH – 1.25V – R3 • 0.3μA Example: switching should not start until the input is above 4.40V, and is to stop if the input falls below 4V. VH = 4.40V, VL = 4V R3 = R4 = 4.40V – 4V = 107k 3.7μA 107k • 1.25V = 43.2k 4.40V – 1.25V – 107k • 0.3μA LT3688 1.25V R3 EN/UVLO VIN VC C1 R4 0.3μA 3.7μA Figure 8. Undervoltage Lockout Keep the connection from the resistor to the EN/UVLO pin short and make sure the interplane or surface capacitance to switching nodes is minimized. If high resistor values are used, the EN/UVLO pin should be bypassed with a 1nF capacitor to prevent coupling problems from the switch node. 3688f 20 + RUN/SS 3688 F08 – LT3688 APPLICATIONS INFORMATION Output Voltage Monitoring The LT3688 provides power supply monitoring for microprocessor-based systems. The features include power-on reset (POR) and watchdog timing. A precise internal voltage reference and glitch immune precision POR comparator circuits monitor the LT3688 output voltages. Each channel’s output voltage must be above 90% of the programmed value for RST not to be asserted (refer to the Timing Diagram). The LT3688 will assert RST during power-up, power-down and brownout conditions. Once the output voltage rises above the RST threshold, the adjustable reset timer is started and RST is released after the reset timeout period. On power-down, once the output voltage drops below RST threshold, RST is held at a logic low. The reset timer is adjustable using external capacitors. This capability helps hold the microprocessor in a stable shutdown condition. The RST pin has weak pull-up to the BIAS pin. The above discussion is concerned only with the DC value of the monitored supply. Real supplies also have relatively high-frequency variation, from sources such as load transients, noise, and pickup. These variations should not be considered by the monitor in determining whether a supply voltage is valid or not. The variations may cause spurious outputs at RST, particularly if the supply voltage is near its trip threshold. Two techniques are used to combat spurious reset without sacrificing threshold accuracy. First, the timeout period helps prevent high-frequency variation whose frequency is above 1/ tRST from appearing at the RST output. When the voltage at FB goes below the threshold, the RST pin asserts low. When the supply recovers past the threshold, the reset timer starts (assuming it is not disabled), and RST does not go high until it finishes. If the supply becomes invalid any time during the timeout period, the timer resets and starts fresh when the supply next becomes valid. While the reset timeout is useful for preventing toggling of the reset output in most cases, it is not effective at preventing nuisance resets due to short glitches (due to load transients or other effects) on a valid supply. To reduce sensitivity to these short glitches, the comparator has additional anti-glitch circuitry. Any transient at the input of the comparator needs to be of sufficient magnitude and duration (tUV) before it can change the monitor state. The combination of the reset timeout and anti-glitch circuitry prevents spurious changes in output state without sacrificing threshold accuracy. Watchdog Timer The LT3688 includes an adjustable watchdog timer that monitors a μP’s activity. If a code execution error occurs in a μP, the watchdog will detect this error and will set the WDO low. This signal can be used to interrupt a routine or to reset a μP . The watchdog circuitry is triggered by negative edges on the WDI pin. The window mode restricts the WDI pin’s negative going pulses to appear inside a programmed time window (see the Timing Diagram) to prevent WDO from going low. If more than two pulses are registered in the window’s fast period, the WDO is forced to go low. The WDO also goes low if no negative edge is supplied to the WDI pin in the window’s slow timer period. During a code execution error, the microprocessor will output WDI pulses that would be either too fast or too slow. This condition will assert WDO and force the microprocessor to reset the program. In window mode, the WDI signal frequency is bounded by an upper and lower limit for normal operation. The WDI input frequency period should be higher than the window mode’s fast period and lower than the window mode’s slow period to keep WDO high under normal conditions. The window mode’s fast and slow times have a fixed ratio of 16 between them. These times can be increased or decreased by adjusting an external capacitor on the CWDT pin. When WDO is asserted, a timer is enabled for a time equivalent to 1/8th of the watchdog window upper boundary. Any WDI pulses that appear while the reset timer is running are ignored. When the timer expires, the WDO is allowed to go high again. Therefore, if no input is applied to the WDI pin, then the watchdog circuitry produces a train of pulses on the WDO pin. The high time of this pulse train is equal to the watchdog window upper boundary, and low time is equal to the 1/8th of the watchdog window upper boundary. 3688f 21 LT3688 APPLICATIONS INFORMATION If WDO is low and RST goes low, then WDO will go high. The WDE pin allows the user to turn on and off the watchdog function. Leaving this pin open is okay and will automatically enable the watchdog. It has an internal weak pull-down to ground. The WDI pin has an internal weak pull-up that keeps the WDI pin high. If watchdog is disabled, leaving this pin open is acceptable. Configuration and Sequencing Use the CONFIG pin to adjust the sequencing and the behavior of the power-on reset and watchdog timers. The table below shows all of the configuration options. Table 5. Configuration Options CONFIG CONDITION Channel 1 starts before Channel 2 Channel 1 and Channel 2 start simultaneously Watchdog operates only if Reset 1 Expires Watchdog operates only if Reset 1 and Reset 2 Expire RST1 and RST2 high only if Timer 1 Expires RST1 and RST2 use independent timers × × × × × VOUT1 (10V/DIV) VOUT2 (10V/DIV) RST1 (5V/DIV) RST2 (5V/DIV) WDO (5V/DIV) WDI (10V/DIV) VOUT1 (10V/DIV) VOUT2 (10V/DIV) RST1 (5V/DIV) RST2 (5V/DIV) WDO (5V/DIV) WDI (10V/DIV) 10ms/DIV 3688 F09a Figure 9a. CONFIG = HIGH VOUT1 (10V/DIV) VOUT2 (10V/DIV) RST1 (5V/DIV) RST2 (5V/DIV) WDO (5V/DIV) HIGH LOW OPEN × × × × WDI (10V/DIV) 10ms/DIV 3688 F09b Figure 9b. CONFIG = LOW With the CONFIG pin tied high, VOUT1 will rise first, as shown in Figure 9a. After VOUT1 reaches VUV, VOUT2 will start increasing. In addition, the reset timer for Channel 1 starts. Once VOUT2 reaches VUV, the reset timer for Channel 2 starts. Once the reset timers for both Channel 1 and Channel 2 have expired, the Watchdog will start operation. With the CONFIG pin tied low, VOUT1 will rise first. After VOUT1 reaches VUV, VOUT2 will start increasing. The reset timer will only start if both VOUT1 and VOUT2 are above VUV, as shown in Figure 9b. Once the reset timer programmed by CPOR1 expires, both RST1 and RST2 can pull high, and the Watchdog will start operation. In this mode, tie CPOR2 to GND. With the CONFIG pin open, VOUT1 and VOUT2 can rise simultaneously, as shown in figure 9c. After VOUT1 reaches VUV the reset timer for Channel 1 starts. Once VOUT2 reaches VUV, the reset timer for Channel 2 starts. Once the reset timer for Channel 1 has expired, the Watchdog will start operation. 10ms/DIV 3688 F09c Figure 9c. CONFIG = OPEN Figure 9. Startup Waveforms with the Three Configuration Settings Selecting the Reset Timing Capacitors The reset timeout period is adjustable in order to accommodate a variety of microprocessor applications. The reset timeout period, tRST, is adjusted by connecting a capacitor, CPOR, between the CPOR pin and ground. The value of this capacitor is determined by: 3688f 22 LT3688 APPLICATIONS INFORMATION ⎛ pF ⎞ CPOR = t RST • 200 ⎜ ⎟ ⎝ ms ⎠ This equation is accurate for reset timeout periods of 1ms, or greater. To program faster timeout periods, see the Reset Timeout Period vs Capacitance graph in the Typical Characteristics section. Leaving the CPOR pin unconnected will generate a minimum reset timeout of approximately 65μs. Maximum reset timeout is limited by the largest available low leakage capacitor. The accuracy of the timeout period will be affected by capacitor leakage (the nominal charging current is 2.5μA), capacitor tolerance and temperature coefficient. A low leakage, low tempco, capacitor is recommended. Selecting the Watchdog Timing Capacitor The watchdog timeout period is adjustable and can be optimized for software execution. The watchdog window upper boundary, tWDU is adjusted by connecting a capacitor, CWDT, between the CWDT pin and ground. Given a specified watchdog timeout period, the capacitor is determined by: ⎛ pF ⎞ CWDT = t WDU • 50 ⎜ ⎟ ⎝ ms ⎠ The window lower boundary (tWDL) and the watchdog timeout (tWDTO) have a fixed relationship to tWDU for a given capacitor. The window lower boundary is related to tWDU by the following: 1 t WDL = • t WDU 16 The watchdog timeout is related to tWDU by the following: 1 tWDTO = • tWDU 8 Leaving the CWDT pin unconnected will generate a minimum watchdog window upper boundary of approximately 200μs. Maximum window upper boundary is limited by the largest available low leakage capacitor. The timing accuracy of the reset and watchdog signals depends on the initial accuracy and stability of the programing capacitors. Use capacitors with specified accuracy, leakage and voltage and temperature coefficients. For surface mount ceramic capacitors C0G and NP0 types are superior to alternatives such as X5R and X7R. Shorted and Reversed Input Protection If an inductor is chosen to prevent excessive saturation, the LT3688 will tolerate a shorted output. When operating in short-circuit condition, the LT3688 will reduce its frequency until the valley current is at a typical value of 1.2A (see Figure 12). There is another situation to consider in systems where the output will be held high when the input to the LT3688 is absent. This may occur in battery charging applications or in battery backup systems where a battery or some other supply is diode ORed with the LT3688’s output. If the VIN pin is allowed to float and the EN/UVLO pin is held high (either by a logic signal or because it is tied to VIN), then the LT3688’s internal circuitry will pull its quiescent current through its SW pin. This is fine if the system can tolerate a few mA in this state. If the EN/UVLO pin is grounded, the SW pin current will drop to essentially zero. However, if the VIN pin is grounded while the output is held high, then parasitic diodes inside the LT3688 can pull large currents from the output through the SW pin and the VIN pin. Figure 13 shows a circuit that will run only when the input voltage is present and that protects against a shorted or reversed input. VSW 10V/DIV IL 500mA/DIV 5μs/DIV 3688 F12 Figure 12. The LT3688 Reduces Its Frequency to Below 70kHz to Protect Against Shorted Output with 36V Input PCB Layout For proper operation and minimum EMI, care must be taken during printed circuit board layout. Figure 14 shows the recommended component placement with trace, ground plane and via locations. Note that large, switched currents flow in the LT3688’s VIN, DA and SW pins, the catch diode (D1) and the input capacitor (C1). The loop formed by 3688f 23 LT3688 APPLICATIONS INFORMATION D4 VIN VIN BIAS BOOST LTC3688 SW DA EN/UVLO GND FB VOUT keep thermal resistance low, extend the ground plane as much as possible, and add thermal vias under and near the LT3688 to additional ground planes within the circuit board and on the bottom side. High Temperature Considerations The PCB must provide heat sinking to keep the LT3688 cool. The exposed pad on the bottom of the package must be soldered to a ground plane. This ground should be tied to large copper layers below with thermal vias; these layers will spread the heat dissipated by the LT3688. Placing additional vias can reduce thermal resistance further. With these steps, the thermal resistance from die (or junction) to ambient can be reduced to θJA = 40°C/W or less. With 100 LFPM airflow, this resistance can fall by another 25%. Further increases in airflow will lead to lower thermal resistance. Because of the large output current capability of the LT3688, it is possible to dissipate enough heat to raise the junction temperature beyond the absolute maximum of 125°C (150°C for H Grade). When operating at high ambient temperatures, the maximum load current should be derated as the ambient temperature approaches 125°C (150°C for H Grade). Power dissipation within the LT3688 can be estimated by calculating the total power loss from an efficiency measurement and subtracting the catch diode loss. The die temperature is calculated by multiplying the LT3688 power dissipation by the thermal resistance from junction-to-ambient. Thermal resistance depends on the layout of the circuit board, but values from 30°C/W to 60°C/W are typical. Die temperature rise was measured on a 4-layer, 5cm • 7.5cm circuit board in still air at a load current of 0.8A (fSW = 800kHz). For a 12V input to 3.3V output the die temperature elevation above ambient was 14°C; for 12VIN to 5VOUT the rise was 15°C and for 12VIN to 5VOUT and 3.3VOUT the rise was 30°C. Other Linear Technology Publications Application Notes 19, 35 and 44 contain more detailed descriptions and design information for buck regulators and other switching regulators. The LT1376 data sheet has a more extensive discussion of output ripple, loop compensation and stability testing. Design Note 318 shows how to generate a bipolar output supply using a buck regulator. 3688f + 3688 F13 Figure 13. Diode D4 Prevents a Shorted Input from Discharging a Backup Battery Tied to the Output; It Also Protects the Circuit from a Reversed Input. The LT3688 Runs Only When the Input Is Present 3688 F14 Figure 14. Top Layer PCB Layout in the LT3688 Demonstration Board these components should be as small as possible. These components, along with the inductor and output capacitor, should be placed on the same side of the circuit board. Place a local, unbroken ground plane below these components. The SW and BST nodes should be as small as possible. Finally, keep the FB node small so that the ground traces will shield them from the SW and BST nodes. The exposed pad on the bottom of the package must be soldered to ground so that the pad acts as a heat sink. To 24 LT3688 TYPICAL APPLICATIONS VIN 6V TO 36V C1 4.7μF EN/UVLO BST1 SW1 C6 10pF C4 22μF R1 523k C8 1nF D1 DA1 FB1 CONFIG RUN/SS1 WDE I/O I/O RESET μP WDI WDO RST1 RST2 RUN/SS2 CWDT CPOR1 CPOR2 RT SYNC LT3688 DA2 FB2 VIN BIAS BST2 SW2 L2 12μH R3 316k R4 100k C7 10pF VOUT1 5V 800mA ON OFF L1 18μH C2 0.22μF C3 0.22μF D2 VOUT2 3.3V 800mA R2 100k C9 1nF C5 22μF C10 4.7nF R5 110k C11 4.7nF C12 1nF 3688 TA02 GND C1-C5: X5R OR X7R D1, D2: DIODES INC. B140 fSW = 500kHz 5V and 3.3V Regulator with Power-On Reset and Watchdog Timers VIN 8V TO 36V C1 4.7μF R6 475k R7 100k L1 10μH VIN EN/UVLO C2 0.22μF D1 DA1 FB1 BST1 SW1 CONFIG BIAS BST2 SW2 C3 0.22μF D2 DA2 FB2 RUN/SS2 CWDT CPOR1 CPOR2 RT SYNC L2 12μH R3 316k R4 100k C7 10pF VOUT 1.8V 800mA C6 10pF C4 47μF R1 187k R2 150k C8 1nF LT3688 VOUT2 3.3V 800mA C9 1nF C5 22μF RUN/SS1 WDE I/O I/O RESET μP WDI WDO RST1 RST2 C10 4.7nF R5 110k fSW = 500kHz C11 4.7nF C12 1nF 3688 TA02 GND C1-C5: X5R OR X7R D1, D2: DIODES INC. B140 3.3V and 1.8V Regulator with Power-On Reset and Watchdog Timers and Input Under Voltage Lockout 3688f 25 LT3688 TYPICAL APPLICATIONS VIN 6V TO 36V C1 4.7μF EN/UVLO BST1 SW1 C6 10μF C4 10μF R1 523k R2 100k C8 1nF D1 DA1 FB1 CONFIG RUN/SS1 WDE I/O I/O RESET μP WDI WDO RST1 RST2 RUN/SS2 CWDT CPOR1 CPOR2 RT SYNC LT3688 DA2 FB2 VIN BIAS BST2 SW2 L2 8.2μH R3 316k R4 100k C7 10pF ON OFF VOUT1 5V 800mA L1 8.2μH C2 0.1μF C3 0.1μF D2 VOUT2 3.3V 800mA C9 1nF C5 22μF C10 4.7nF R5 20k C11 4.7nF C12 1nF 3688 TA04 GND C1-C5: X5R OR X7R D1, D2: DIODES INC. B140 fSW = 2MHz: 8V < VIN < 16V, TJ < 85°C 2MHz Switching Frequency, 5V and 3.3V Regulator with Power-On Reset and Watchdog Timers VIN 4V TO 36V C1 4.7μF EN/UVLO BST1 SW1 VIN BIAS BST2 SW2 LT3688 DA1 FB1 CONFIG RUN/SS1 RUN/SS2 CWDT CPOR1 CPOR2 RT SYNC WDE WDI WDO RST1 RST2 DA2 FB2 L2 15μH R3 316k R4 100k C7 10pF VOUT1 1.2V 800mA ON OFF L1 8.2μH C6 10pF C4 100μF D1 R1 90.9k R2 182k C8 1nF C2 0.22μF C3 0.22μF D2 VOUT2 3.3V 800mA C9 1nF C5 22μF WATCHDOG DEFEAT μP I/O C10 4.7nF R5 143k C11 4.7nF C12 1nF 3688 TA05 RESET GND C1-C5: X5R OR X7R D1, D2: DIODES INC. B140 fSW = 400kHz: VIN < 25V 3.3V and 1.2V Regulator with Power-On Reset Timer and Defeatable Watchdog, Timing Error Resets Microprocessor 3688f 26 LT3688 PACKAGE DESCRIPTION FE Package 24-Lead Plastic TSSOP (4.4mm) (Reference LTC DWG # 05-08-1771 Rev Ø) Variation AA 3.25 (.128) 7.70 – 7.90* (.303 – .311) 3.25 (.128) 24 23 22 21 20 19 18 17 16 15 14 13 6.60 0.10 4.50 0.10 SEE NOTE 4 2.74 (.108) 0.45 0.05 1.05 0.10 0.65 BSC RECOMMENDED SOLDER PAD LAYOUT 6.40 2.74 (.252) (.108) BSC 1 2 3 4 5 6 7 8 9 10 11 12 1.20 (.047) MAX 0 –8 4.30 – 4.50* (.169 – .177) 0.25 REF 0.09 – 0.20 (.0035 – .0079) 0.50 – 0.75 (.020 – .030) 0.65 (.0256) BSC NOTE: 1. CONTROLLING DIMENSION: MILLIMETERS MILLIMETERS 2. DIMENSIONS ARE IN (INCHES) 3. DRAWING NOT TO SCALE 0.195 – 0.30 (.0077 – .0118) TYP 0.05 – 0.15 (.002 – .006) FE24 (AA) TSSOP 0208 REV Ø 4. RECOMMENDED MINIMUM PCB METAL SIZE FOR EXPOSED PAD ATTACHMENT *DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.150mm (.006") PER SIDE UF Package 24-Lead Plastic QFN (4mm × 4mm) (Reference LTC DWG # 05-08-1697) BOTTOM VIEW—EXPOSED PAD PIN 1 NOTCH R = 0.20 TYP OR 0.35 × 45° CHAMFER 23 24 0.40 ± 0.10 1 2 4.50 ± 0.05 2.45 ± 0.05 3.10 ± 0.05 (4 SIDES) 2.45 ± 0.10 (4-SIDES) 4.00 ± 0.10 (4 SIDES) 0.70 ±0.05 PIN 1 TOP MARK (NOTE 6) 0.75 ± 0.05 R = 0.115 TYP PACKAGE OUTLINE 0.25 ±0.05 0.50 BSC RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS 0.200 REF 0.00 – 0.05 NOTE: 1. DRAWING PROPOSED TO BE MADE A JEDEC PACKAGE OUTLINE MO-220 VARIATION (WGGD-X)—TO BE APPROVED 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE, IF PRESENT 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE (UF24) QFN 0105 0.25 ± 0.05 0.50 BSC 3688f Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 27 LT3688 TYPICAL APPLICATION VIN 6V TO 36V C1 4.7μF EN/UVLO BST1 SW1 C6 10pF C4 22μF R1 523k C8 1nF D1 DA1 FB1 CONFIG RUN/SS1 WDE I/O I/O RESET C1-C5: X5R OR X7R D1, D2: DFLS140 μP WDI WDO RST1 RST2 RUN/SS2 CWDT CPOR1 CPOR2 RT SYNC LT3688 DA2 FB2 VIN BIAS BST2 SW2 L2 12μH R3 340k R4 100k C7 10pF VOUT1 5V 800mA ON OFF L1 18μH C2 0.22μF C3 0.22μF D2 VOUT2 3.3V 800mA R2 100k C9 1nF C5 22μF C10 4.7nF R5 49.9k C11 4.7nF C12 1nF 3688 TA06 GND fSW = 1MHz: 7V < VIN < 21V 1MHz 5V and 3.3V Regulator with Power-On Reset and Watchdog Timers RELATED PARTS PART NUMBER LT3640 LT3689/LT3689–5 DESCRIPTION 35V, 55V MAX, Dual (1.3A, 1.1A), 2.5MHz High Efficiency StepDown DC/DC Converter with POR Reset and Watchdog Timer 36V, 60V Transient Protection, 800mA, 2.2MHz High Efficiency MicroPower Step-Down DC/DC Converter with POR Reset and Watchdog Timer 37V, 55Vmax, 1.2A, 2.5MHz High Efficiency Step-Down DC/DC Converter 36V, 60Vmax, 1A, 2.2MHz High Efficiency Micropower StepDown DC/DC Converter 38V, 1.2A (IOUT), 2MHz, High Efficiency Step-Down DC/DC Converter with only 2.8uA of Quiescent Current 55V, 1.2A (IOUT), 2MHz, High Efficiency Step-Down DC/DC Converter with only 2.8uA of Quiescent Current 40V, 350mA (IOUT), 2MHz, High Efficiency Step-Down DC/DC Converter with only 2.5uA of Quiescent Current 60V, 350mA (IOUT), 2MHz, High Efficiency Step-Down DC/DC Converter with only 2.5uA of Quiescent Current COMMENTS VIN Min = 4V, VIN Max = 35V, Transient to 55V, VOUT(MIN) = 0.6V, IQ =