LTC3831

LTC3831 High Power Synchronous Switching Regulator Controller for DDR Memory Termination FEATURES s s s s s s s s DESCRIPTIO s s s s s High Power Switching Regulator Controller for DDR Memory Termination VOUT Tracks 1/2 of VIN or External VREF No Current Sense Resistor Required Low Input Supply Voltage Range: 3V to 8V Maximum Duty Cycle > 91% Over Temperature Drives All N-Channel External MOSFETs High Efficiency: Over 95% Possible Programmable Fixed Frequency Operation: 100kHz to 500kHz External Clock Synchronization Operation Programmable Soft-Start Low Shutdown Current: 91%. It includes a fixed frequency PWM oscillator for low output ripple operation. The 200kHz free-running clock frequency can be externally adjusted or synchronized with an external signal from 100kHz to above 500kHz. In shutdown mode, the LTC3831 supply current drops to 91%. Similarly, the MAX 3831f W UU LTC3831 APPLICATIO S I FOR ATIO comparator forces the output to 0% duty cycle if the feedback signal is greater than 3% above VREF. To prevent these two comparators from triggering due to noise, the MIN and MAX comparators’ response times are deliberately delayed by two to three microseconds. These two comparators help prevent extreme output perturbations with fast output load current transients, while allowing the main feedback loop to be optimally compensated for stability. Thermal Shutdown The LTC3831 has a thermal protection circuit that disables both gate drivers if activated. If the chip junction temperature reaches 150°C, both TG and BG are pulled low. TG and BG remain low until the junction temperature drops below 125°C, after which, the chip resumes normal operation. Soft-Start and Current Limit The LTC3831 includes a soft-start circuit that is used for start-up and current limit operation. The SS pin requires an external capacitor, CSS, to GND with the value determined by the required soft-start time. An internal 12µA current source is included to charge CSS. During powerup, the COMP pin is clamped to a diode drop (B-E junction of QSS in the Block Diagram) above the voltage at the SS pin. This prevents the error amplifier from forcing the loop to maximum duty cycle. The LTC3831 operates at low duty cycle as the SS pin rises above 0.6V (VCOMP ≈ 1.2V). As SS continues to rise, QSS turns off and the error amplifier takes over to regulate the output. The MIN comparator is disabled during soft-start to prevent it from overriding the soft-start function. The LTC3831 includes yet another feedback loop to control operation in current limit. Just before every falling edge of TG, the current comparator, CC, samples and holds the voltage drop measured across the external upper MOSFET, Q1, at the IFB pin. CC compares the voltage at IFB to the voltage at the IMAX pin. As the peak current rises, the measured voltage across Q1 increases due to the drop across the RDS(ON) of Q1. When the voltage at IFB drops below IMAX, indicating that Q1’s drain current has exceeded the maximum level, CC starts to pull current out of CSS, cutting the duty cycle and controlling the output U current level. The CC comparator pulls current out of the SS pin in proportion to the voltage difference between IFB and IMAX. Under minor overload conditions, the SS pin falls gradually, creating a time delay before current limit takes effect. Very short, mild overloads may not affect the output voltage at all. More significant overload conditions allow the SS pin to reach a steady state, and the output remains at a reduced voltage until the overload is removed. Serious overloads generate a large overdrive at CC, allowing it to pull SS down quickly and preventing damage to the output components. By using the RDS(ON) of Q1 to measure the output current, the current limiting circuit eliminates an expensive discrete sense resistor that would otherwise be required. This helps minimize the number of components in the high current path. The current limit threshold can be set by connecting an external resistor RIMAX from the IMAX pin to the main VIN supply at the drain of Q1. The value of RIMAX is determined by: RIMAX = (ILMAX)(RDS(ON)Q1)/IIMAX where: ILMAX = ILOAD + (IRIPPLE/2) ILOAD= Maximum load current IRIPPLE = Inductor ripple current = W UU ( VIN – VOUT )( VOUT ) (fOSC )(LO )(VIN) fOSC = LTC3831 oscillator frequency = 200kHz LO = Inductor value RDS(ON)Q1 = On-resistance of Q1 at ILMAX IIMAX = Internal 12µA sink current at IMAX The RDS(ON) of Q1 usually increases with temperature. To keep the current limit threshold constant, the internal 12µA sink current at IMAX is designed with a positive temperature coefficient to provide first order correction for the temperature coefficient of RDS(ON)Q1. In order for the current limit circuit to operate properly and to obtain a reasonably accurate current limit threshold, the 3831f 9 LTC3831 APPLICATIO S I FOR ATIO IIMAX and IFB pins must be Kelvin sensed at Q1’s drain and source pins. In addition, connect a 0.1µF decoupling capacitor across RIMAX to filter switching noise. Otherwise, noise spikes or ringing at Q1’s source can cause the actual current limit to be greater than the desired current limit set point. Due to switching noise and variation of RDS(ON), the actual current limit trip point is not highly accurate. The current limiting circuitry is primarily meant to prevent damage to the power supply circuitry during fault conditions. The exact current level where the limiting circuit begins to take effect will vary from unit to unit as the RDS(ON) of Q1 varies. Typically, RDS(ON) varies as much as ±40% and with ± 25% variation on the LTC3831’s IMAX current, this can give a ±65% variation on the current limit threshold. The RDS(ON) is high if the VGS applied to the MOSFET is low. This occurs during power up, when PVCC1 is ramping up. To prevent the high RDS(ON) from activating the current limit, the LTC3831 disables the current limit circuit if PVCC1 is less than 2.5V above VCC. To ensure proper operation of the current limit circuit, PVCC1 must be at least 2.5V above VCC when TG is high. PVCC1 can go low when TG is low, allowing the use of an external charge pump to power PVCC1. VIN LTC3831 RIMAX 0.1µF + CC 12µA 12 IMAX IFB TG 1k BG Q2 Q1 LO – 13 Figure 3. Current Limit Setting Oscillator Frequency The LTC3831 includes an onboard current controlled oscillator that typically free-runs at 200kHz. The oscillator frequency can be adjusted by forcing current into or out of the FREQSET pin. With the pin floating, the oscillator runs at about 200kHz. Every additional 1µA of current into/out of the FREQSET pin decreases/increases the frequency by 10 U 10kHz. The pin is internally servoed to 1.265V, connecting a 50k resistor from FREQSET to ground forces 25µA out of the pin, causing the internal oscillator to run at approximately 450kHz. Forcing an external 10µA current into FREQSET cuts the internal frequency to 100kHz. An internal clamp prevents the oscillator from running slower than about 50kHz. Tying FREQSET to VCC forces the chip to run at this minimum speed. Shutdown The LTC3831 includes a low power shutdown mode, controlled by the logic at the SHDN pin. A high at SHDN allows the part to operate normally. A low level at SHDN for more than 100µs forces the LTC3831 into shutdown mode. In this mode, all internal switching stops, the COMP and SS pins pull to ground and Q1 and Q2 turn off. The LTC3831 supply current drops to 100µs or else the LTC3831 enters into the shutdown mode. Figure 4 describes the operation of the external synchronization function. A negative transition at the SHDN pin forces the internal ramp signal low to restart a new PWM cycle. Notice that the ramp amplitude is lowered as the external clock frequency goes higher. The effect of this 3831f W UU + CIN + VOUT COUT 3831 F03 LTC3831 APPLICATIO S I FOR ATIO SHDN TRADITIONAL SYNC METHOD WITH EARLY RAMP TERMINATION 200kHz FREE RUNNING RAMP SIGNAL RAMP SIGNAL WITH EXT SYNC RAMP AMPLITUDE ADJUSTED LTC3831 KEEPS RAMP AMPLITUDE CONSTANT UNDER SYNC Figure 4. External Synchronization Operation decrease in ramp amplitude increases the open-loop gain of the controller feedback loop. As a result, the loop crossover frequency increases and it may cause the feedback loop to be unstable if the phase margin is insufficient. To overcome this problem, the LTC3831 monitors the peak voltage of the ramp signal and adjust the oscillator charging current to maintain a constant ramp peak. Input Supply Considerations/Charge Pump The LTC3831 requires four supply voltages to operate: VIN for the main power input, PVCC1 and PVCC2 for MOSFET gate drive and a clean, low ripple VCC for the LTC3831 internal circuitry (Figure 5). In many applications, VCC can be powered from VIN through an RC filter. This supply can be as low as 3V. The low quiescent current (typically 800µA) allows the use of relatively large filter resistors and correspondingly small filter capacitors. 100Ω and 4.7µF usually provide adequate filtering for VCC. For best performance, connect the 4.7µF bypass capacitor as close to the LTC3831 VCC pin as possible. U Gate drive for the top N-channel MOSFET Q1 is supplied from PVCC1. This supply must be above VIN (the main power supply input) by at least one power MOSFET VGS(ON) for efficient operation. An internal level shifter allows PVCC1 to operate at voltages above VCC and VIN, up to 14V maximum. This higher voltage can be supplied with a separate supply, or it can be generated using a charge pump. Gate drive for the bottom MOSFET Q2 is provided through PVCC2. This supply only need to be above the power MOSFET VGS(ON) for efficient operation. PVCC2 can also be driven from the same supply/charge pump for the PVCC1, or it can be connected to a lower supply to improve efficiency. Figure 6 shows a doubling charge pump circuit that can be used to provide 2VIN gate drive for Q1. The charge pump consists of a Schottky diode from VIN to PVCC1 and a 0.1µF capacitor from PVCC1 to the switching node at the drain of VCC PVCC2 PVCC1 VIN 3831 F04 W UU TG Q1 LO VOUT INTERNAL CIRCUITRY BG Q2 + COUT 3831 F05 LTC3831 Figure 5. Supplies Input VIN OPTIONAL USE FOR VIN ≥ 7V DZ 12V 1N5242 PVCC2 PVCC1 TG MBR0530T1 0.1µF Q1 LO VOUT BG + Q2 COUT 3831 F06a LTC3831 Figure 6. Doubling Charge Pump 3831f 11 LTC3831 APPLICATIO S I FOR ATIO Q2. This circuit provides 2VIN – VF to PVCC1 while Q1 is ON and VIN – VF while Q1 is OFF where VF is the forward voltage of the Schottky diode. Ringing at the drain of Q2 can cause transients above 2VIN at PVCC1; if VIN is higher than 7V, a 12V zener diode should be included from PVCC1 to PGND to prevent transients from damaging the circuitry at PVCC1 or the gate of Q1. For applications with a lower VIN supply, a tripling charge pump circuit shown in Figure 7 can be used to provide 2VIN and 3VIN gate drive for the external top and bottom MOSFETs respectively. This circuit provides 3VIN – 3VF to PVCC1 while Q1 is ON and 2VIN – 2VF to PVCC2 where VF is the forward voltage of the Schottky diode. The circuit requires the use of Schottky diodes to minimize forward drop across the diodes at start-up. The tripling charge pump circuit can rectify any ringing at the drain of Q2 and provide more than 3VIN at PVCC1; a 12V zener diode should be included from PVCC1 to PGND to prevent transients from damaging the circuitry at PVCC1 or the gate of Q1. The charge pump capacitors for PVCC1 refresh when the BG pin goes high and the switch node is pulled low by Q2. The BG on time becomes narrow when the LTC3831 operates at maximum duty cycle (95% typical) which can occur if the input supply rises more slowly than the softstart capacitor or the input voltage droops during load transients. If the BG on time gets so narrow that the switch node fails to pull completely to ground, the charge pump voltage may collapse or fail to start causing excessive dissipation in external MOSFET Q1. This is most likely with low VCC voltages and high switching frequencies, coupled with large external MOSFETs that slow the BG and switch node slew rates. The LTC3831 overcomes this problem by sensing the PVCC1 voltage when TG is high. If PVCC1 is less than 2.5V above VCC, the maximum TG duty cycle is reduced to 70% by clamping the COMP pin at 1.8V (QC in the Block Diagram). This increases the BG on time and allows the charge pump capacitors to be refreshed. 12 U DZ 12V 1N5242 1N5817 1N5817 0.1µF 0.1µF Q1 LO VOUT BG VIN 1N5817 10µF PVCC2 PVCC1 TG W UU + Q2 COUT 3831 F07 LTC3831 Figure 7. Tripling Charge Pump For applications using an external supply to power PVCC1, this supply must also be higher than VCC by at least 2.5V to ensure normal operation. Connecting the Ratiometric Reference Input The LTC3831 derives its ratiometric reference, V REF, using an internal resistor divider. The top and bottom of the resistor divider is connected to the R+ and R – pins respectively. This permits the output voltage to track at a ratio of the differential voltage at R+ and R –. The LTC3831 can operate with a minimum VFB of 1.1V and maximum VFB of (VCC – 1.75V). With R– connected to GND, this gives a VR+ input range of 2.2V to (2 • VCC – 3.5V). If VR+ is higher than the permitted input voltage, increase the VCC voltage to raise the input range. In a typical DDR memory termination application as shown in Figure 1, R+ is connected to VDDQ, the supply voltage of the interface, and R – to GND. The output voltage VTT is connected to the FB pin, so VTT = 0.5 • VDDQ. If a ratio greater than 0.5 is desired, it can be achieved using an external resistor divider connected to VTT and FB pin. Figure 8 shows an application that generates a VTT of 0.6 • VDDQ. 3831f LTC3831 APPLICATIO S I FOR ATIO 5V MBR0530T1 1µF 0.1µF PVCC2 VCC SS PVCC1 TG IMAX + 4.7µF 0.01µF 130k SHDN LTC3831 IFB FREQSET SHDN COMP BG PGND GND R+ R– FB CIN: SANYO POSCAP 6TPB330M COUT: SANYO POSCAP 4TPB470M Q1, Q2: SILICONIX Si4410DY Q2 MBRS340T3 C1 33pF RC 15k CC 1500pF Figure 8. Typical Application with VTT = 0.6 • VDDQ Power MOSFETs Two N-channel power MOSFETs are required for most LTC3831 circuits. These should be selected based primarily on threshold voltage and on-resistance considerations. Thermal dissipation is often a secondary concern in high efficiency designs. The required MOSFET threshold should be determined based on the available power supply voltages and/or the complexity of the gate drive charge pump scheme. In 3.3V input designs where an auxiliary 12V supply is available to power PVCC1 and PVCC2, standard MOSFETs with RDS(ON) specified at VGS = 5V or 6V can be used with good results. The current drawn from this supply varies with the MOSFETs used and the LTC3831’s operating frequency, but is generally less than 50mA. LTC3831 applications that use 5V or lower VIN voltage and doubling/tripling charge pumps to generate PVCC1 and PVCC2, do not provide enough gate drive voltage to fully enhance standard power MOSFETs. Under this condition, the effective MOSFET RDS(ON) may be quite high, raising the dissipation in the FETs and reducing efficiency. Logiclevel FETs are the recommended choice for 5V or lower voltage systems. Logic-level FETs can be fully enhanced with a doubler/tripling charge pump and will operate at maximum efficiency. U VDDQ 2.5V W UU + 10k 0.1µF CIN 330µF ×2 Q1 0.1µF MBRS340T3 LO 1.2µH 1k VTT 1.5V ± 6A COUT 470µF ×3 2k 1% 10k 1% 3831 F08 + After the MOSFET threshold voltage is selected, choose the RDS(ON) based on the input voltage, the output voltage, allowable power dissipation and maximum output current. In a typical LTC3831 circuit operating in continuous mode, the average inductor current is equal to the output load current. This current flows through either Q1 or Q2 with the power dissipation split up according to the duty cycle: VOUT VIN V V –V DC(Q2) = 1 – OUT = IN OUT VIN VIN DC(Q1) = The RDS(ON) required for a given conduction loss can now be calculated by rearranging the relation P = I2R. RDS(ON)Q1 = RDS(ON)Q2 = PMAX(Q1) DC(Q1) • (ILOAD )2 PMAX(Q2 ) DC(Q2) • (ILOAD)2 = = VIN • PMAX(Q1) VOUT • (ILOAD )2 VIN • PMAX(Q2 ) ( VIN – VOUT ) • (ILOAD)2 PMAX should be calculated based primarily on required efficiency or allowable thermal dissipation. A typical high 3831f 13 LTC3831 APPLICATIO S I FOR ATIO efficiency circuit designed for 2.5V input and 1.25V at 5A output might allow no more than 3% efficiency loss at full load for each MOSFET. Assuming roughly 90% efficiency at this current level, this gives a PMAX value of: (1.25V)(5A/0.9)(0.03) = 0.21W per FET and a required RDS(ON) of: (2.5V) • (0.21W) = 0.017Ω (1.25V)(5A)2 (2.5V) • (0.21W) = 0.017Ω RDS(ON)Q2 = (2.5V – 1.25V)(5A)2 RDS(ON)Q1 = Note that while the required RDS(ON) values suggest large MOSFETs, the power dissipation numbers are only 0.21W per device or less; large TO-220 packages and heat sinks are not necessarily required in high efficiency applications. Siliconix Si4410DY or International Rectifier IRF7413 (both in SO-8) or Siliconix SUD50N03-10 (TO-252) or ON Semiconductor MTD20N03HDL (DPAK) are small footprint surface mount devices with RDS(ON) values below 0.03Ω at 5V of VGS that work well in LTC3831 circuits. Using a higher PMAX value in the RDS(ON) calculations generally Table 1. Recommended MOSFETs for LTC3831 Applications RDS(ON) AT 25°C (mΩ) 19 20 35 8 10 9 19 28 37 PARTS Siliconix SUD50N03-10 TO-252 Siliconix Si4410DY SO-8 ON Semiconductor MTD20N03HDL D PAK Fairchild FDS6670A S0-8 Fairchild FDS6680 SO-8 ON Semiconductor MTB75N03HDL DD PAK IR IRL3103S DD PAK IR IRLZ44 TO-220 Fuji 2SK1388 TO-220 Note: Please refer to the manufacturer’s data sheet for testing conditions and detailed information. 3831f 14 U decreases the MOSFET cost and the circuit efficiency and increases the MOSFET heat sink requirements. Table 1 highlights a variety of power MOSFETs that are for use in LTC3831 applications. Inductor Selection The inductor is often the largest component in an LTC3831 design and must be chosen carefully. Choose the inductor value and type based on output slew rate requirements. The maximum rate of rise of inductor current is set by the inductor’s value, the input-to-output voltage differential and the LTC3831’s maximum duty cycle. In a typical 2.5V input 1.25V output application, the maximum rise time will be: W UU DCMAX • ( VIN – VOUT ) 1.138 A = LO LO µs where LO is the inductor value in µH. With proper frequency compensation, the combination of the inductor and output capacitor values determine the transient recovery time. In general, a smaller value inductor improves transient response at the expense of ripple and inductor core saturation rating. A 2µH inductor has a 0.57A/µs rise RATED CURRENT (A) 15 at 25°C 10 at 100°C 10 at 25°C 8 at 70°C 20 at 25°C 16 at 100°C 13 at 25°C 11.5 at 25°C 75 at 25°C 59 at 100°C 64 at 25°C 45 at 100°C 50 at 25°C 36 at 100°C 35 at 25°C TYPICAL INPUT CAPACITANCE CISS (pF) 3200 2700 880 3200 2070 4025 1600 3300 1750 θJC (°C/W) 1.8 TJMAX (°C) 175 150 1.67 25 25 1 1.4 1 2.08 150 150 150 150 175 175 150 LTC3831 APPLICATIO S I FOR ATIO time in this application, resulting in a 8.8µs delay in responding to a 5A load current step. During this 8.8µs, the difference between the inductor current and the output current is made up by the output capacitor. This action causes a temporary voltage droop at the output. To minimize this effect, the inductor value should usually be in the 1µH to 5µH range for most LTC3831 circuits. To optimize performance, different combinations of input and output voltages and expected loads may require different inductor values. Once the required value is known, the inductor core type can be chosen based on peak current and efficiency requirements. Peak current in the inductor will be equal to the maximum output load current plus half of the peak-topeak inductor ripple current. Ripple current is set by the inductor value, the input and output voltage and the operating frequency. The ripple current is approximately equal to: IRIPPLE = ( VIN − VOUT ) • ( VOUT ) fOSC • LO • VIN fOSC = LTC3831 oscillator frequency = 200kHz LO = Inductor value Solving this equation with our typical 2.5V to 1.25V application with 2µH inductor, we get: (2.5V – 1.25V) • 1.25V = 1.56 AP-P 200kHz • 2µH • 2.5V Peak inductor current at 5A load: 5A + (1.56A/2) = 5.78A The ripple current should generally be between 10% and 40% of the output current. The inductor must be able to withstand this peak current without saturating, and the copper resistance in the winding should be kept as low as possible to minimize resistive power loss. Note that in circuits not employing the current limit function, the current in the inductor may rise above this maximum under short circuit or fault conditions; the inductor should be sized accordingly to withstand this additional current. Inductors with gradual saturation characteristics are often the best choice. U Input and Output Capacitors A typical LTC3831 design places significant demands on both the input and the output capacitors. During normal steady load operation, a buck converter like the LTC3831 draws square waves of current from the input supply at the switching frequency. The peak current value is equal to the output load current plus 1/2 the peak-to-peak ripple current. Most of this current is supplied by the input bypass capacitor. The resulting RMS current flow in the input capacitor heats it and causes premature capacitor failure in extreme cases. Maximum RMS current occurs with 50% PWM duty cycle, giving an RMS current value equal to IOUT/2. A low ESR input capacitor with an adequate ripple current rating must be used to ensure reliable operation. Note that capacitor manufacturers’ ripple current ratings are often based on only 2000 hours (3 months) lifetime at rated temperature. Further derating of the input capacitor ripple current beyond the manufacturer’s specification is recommended to extend the useful life of the circuit. Lower operating temperature has the largest effect on capacitor longevity. The output capacitor in a buck converter under steadystate conditions sees much less ripple current than the input capacitor. Peak-to-peak current is equal to inductor ripple current, usually 10% to 40% of the total load current. Output capacitor duty places a premium not on power dissipation but on ESR. During an output load transient, the output capacitor must supply all of the additional load current demanded by the load until the LTC3831 adjusts the inductor current to the new value. ESR in the output capacitor results in a step in the output voltage equal to the ESR value multiplied by the change in load current. A 5A load step with a 0.05Ω ESR output capacitor results in a 250mV output voltage shift; this is 20% of the output voltage for a 1.25V supply! Because of the strong relationship between output capacitor ESR and output load transient response, choose the output capacitor for ESR, not for capacitance value. A capacitor with suitable ESR will usually have a larger capacitance value than is needed to control steady-state output ripple. Electrolytic capacitors, such as the Sanyo MV-WX series, rated for use in switching power supplies with specified ripple current ratings and ESR, can be used effectively in 3831f W UU 15 LTC3831 APPLICATIO S I FOR ATIO LTC3831 applications. OS-CON electrolytic capacitors from Sanyo and other manufacturers give excellent performance and have a very high performance/size ratio for electrolytic capacitors. Surface mount applications can use either electrolytic or dry tantalum capacitors. Tantalum capacitors must be surge tested and specified for use in switching power supplies. Low cost, generic tantalums are known to have very short lives followed by explosive deaths in switching power supply applications. Other capacitor series that can be used include Sanyo POSCAPs and the Panasonic SP line. A common way to lower ESR and raise ripple current capability is to parallel several capacitors. A typical LTC3831 application might exhibit 5A input ripple current. Sanyo OS-CON capacitors, part number 10SA220M (220µF/10V), feature 2.3A allowable ripple current at 85°C; three in parallel at the input (to withstand the input ripple current) meet the above requirements. Similarly, Sanyo POSCAP 4TPB470M (470µF/4V) capacitors have a maximum rated ESR of 0.04Ω, three in parallel lower the net output capacitor ESR to 0.013Ω. Feedback Loop Compensation The LTC3831 voltage feedback loop is compensated at the COMP pin, which is the output node of the error amplifier. The feedback loop is generally compensated with an RC + C network from COMP to GND as shown in Figure 9a. Loop stability is affected by the values of the inductor, the output capacitor, the output capacitor ESR, the error amplifier transconductance and the error amplifier compensation network. The inductor and the output capacitor create a double pole at the frequency: fLC = 1/ 2π (LO )(COUT ) COMP 10 RC CC C1 ERR Figure 9a. Compensation Pin Hook-Up fZ LOOP GAIN fSW = LTC3831 SWITCHING FREQUENCY fCO = CLOSED-LOOP CROSSOVER FREQUENCY [ ] 20dB/DECADE The ESR of the output capacitor and the output capacitor value form a zero at the frequency: fESR = 1/ [2π (ESR)(COUT )] The compensation network used with the error amplifier must provide enough phase margin at the 0dB crossover fLC fESR fCO Figure 9b. Bode Plot of the LTC3831 Overall Transfer Function 16 – + U frequency for the overall open-loop transfer function. The zero and pole from the compensation network are: fZ = 1/[2π(RC)(CC)] and fP = 1/[2π(RC)(C1)] respectively. Figure 9b shows the Bode plot of the overall transfer function. Although a mathematical approach to frequency compensation can be used, the added complication of input and/or output filters, unknown capacitor ESR, and gross operating point changes with input voltage, load current variations, all suggest a more practical empirical method. This can be done by injecting a transient current at the load and using an RC network box to iterate toward the final values, or by obtaining the optimum loop response using a network analyzer to find the actual loop poles and zeros. LTC3831 VFB W UU 6 VTT VREF 3831 F09a fP FREQUENCY 3830 F10b 3831f LTC3831 APPLICATIO S I FOR ATIO Table 2 shows the suggested compensation component value for 2.5V to 1.25V applications based on the 470µF Sanyo POSCAP 4TPB470M output capacitors. Table 3 shows the suggested compensation component values for 2.5V to 1.25V applications based on 1500µF Sanyo MV-WX output capacitors. Table 2. Recommended Compensation Network for 2.5V to 1.25V Applications Using Multiple Paralleled 470µF Sanyo POSCAP 4TPB470M Output Capacitors L1 (µH) 1.2 1.2 1.2 2.4 2.4 2.4 4.7 4.7 4.7 COUT (µF) 1410 2820 4700 1410 2820 4700 1410 2820 4700 RC (kΩ) 6.8 15 22 15 36 47 33 68 120 CC (nF) 3.3 3.3 1.5 10 3.3 4.7 10 22 10 C1 (pF) 33 33 33 33 10 10 10 10 10 PVCC 100Ω 4.7µF + 1µF GND NC VCC PVCC2 PVCC1 LTC3831 FREQSET SHDN COMP TG IMAX IFB R+ BG FB R– PGND C1 RC CC CSS GND SS GND PGND PGND 3830 F11 Figure 10. Typical Schematic Showing Layout Considerations U LAYOUT CONSIDERATIONS When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC3831. These items are also illustrated graphically in the layout diagram of Figure 10. The thicker lines show the high current paths. Note that at 5A current levels or above, Table 3. Recommended Compensation Network for 2.5V to 1.25V Applications Using Multiple Paralleled 1500µF Sanyo MV-WX Output Capacitors L1 (µH) 1.2 1.2 1.2 2.4 2.4 2.4 4.7 4.7 4.7 COUT (µF) 4500 6000 9000 4500 6000 9000 4500 6000 9000 RC (kΩ) 20 27 43 51 62 82 82 100 150 CC (nF) 1.5 1 0.47 1 1 0.47 3.3 1 1 C1 (pF) 120 82 56 56 33 27 33 15 15 VIN 1µF 10k 0.1µF Q1 1k MBRS340T3 Q2 OPTIONAL MBRS340T3 LO VOUT W UU + CIN + COUT 3831f 17 LTC3831 APPLICATIO S I FOR ATIO current density in the PC board itself is a serious concern. Traces carrying high current should be as wide as possible. For example, a PCB fabricated with 2oz copper requires a minimum trace width of 0.15" to carry 5A. 1. In general, layout should begin with the location of the power devices. Be sure to orient the power circuitry so that a clean power flow path is achieved. Conductor widths should be maximized and lengths minimized. After you are satisfied with the power path, the control circuitry should be laid out. It is much easier to find routes for the relatively small traces in the control circuits than it is to find circuitous routes for high current paths. 2. The GND and PGND pins should be shorted directly at the LTC3831. This helps to minimize internal ground disturbances in the LTC3831 and prevents differences in ground potential from disrupting internal circuit operation. This connection should then tie into the ground plane at a single point, preferably at a fairly quiet point in the circuit such as close to the output capacitors. This is not always practical, however, due to physical constraints. Another reasonably good point to make this connection is between the output capacitors and the source connection of the bottom MOSFET Q2. Do not tie this single point ground in the trace run between the Q2 source and the input capacitor ground, as this area of the ground plane will be very noisy. 18 U 3. The small-signal resistors and capacitors for frequency compensation and soft-start should be located very close to their respective pins and the ground ends connected to the signal ground pin through a separate trace. Do not connect these parts to the ground plane! 4. The VCC, PVCC1 and PVCC2 decoupling capacitors should be as close to the LTC3831 as possible. The 4.7µF and 1µF bypass capacitors shown at VCC, PVCC1 and PVCC2 will help provide optimum regulation performance. 5. The (+) plate of CIN should be connected as close as possible to the drain of the upper MOSFET, Q1. An additional 1µF ceramic capacitor between VIN and power ground is recommended. 6. The VFB pin is very sensitive to pickup from the switching node. Care should be taken to isolate VFB from possible capacitive coupling to the inductor switching signal. 7. In a typical SSTL application, if the R+ pin is to be connected to VDDQ, which is also the main supply voltage for the switching regulator, do not connect R+ along the high current flow path; it should be connected to the SSTL interface supply output. R– should be connected to the interface supply GND. 8. Kelvin sense IMAX and IFB at Q1’s drain and source pins. 3831f W UU LTC3831 PACKAGE DESCRIPTIO .254 MIN .0165 ± .0015 RECOMMENDED SOLDER PAD LAYOUT 1 .015 ± .004 × 45° (0.38 ± 0.10) .007 – .0098 (0.178 – 0.249) .016 – .050 (0.406 – 1.270) NOTE: 1. CONTROLLING DIMENSION: INCHES INCHES 2. DIMENSIONS ARE IN (MILLIMETERS) 3. DRAWING NOT TO SCALE *DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE **DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE 0° – 8° TYP .053 – .068 (1.351 – 1.727) 23 4 56 7 8 .004 – .0098 (0.102 – 0.249) Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. U GN Package 16-Lead Plastic SSOP (Narrow .150 Inch) (Reference LTC DWG # 05-08-1641) .045 ± .005 .189 – .196* (4.801 – 4.978) 16 15 14 13 12 11 10 9 .009 (0.229) REF .150 – .165 .229 – .244 (5.817 – 6.198) .150 – .157** (3.810 – 3.988) .0250 TYP .008 – .012 (0.203 – 0.305) .0250 (0.635) BSC GN16 (SSOP) 0502 3831f 19 LTC3831 RELATED PARTS PART NUMBER LTC1530 LTC1702 LTC1703 LTC1705 DESCRIPTION High Power Synchronous Switching Regulator Controller Dual High Efficiency 2-Phase Synchronous Step-Down Controller Dual 550kHz Synchronous 2-Phase Switching Regulator Controller with Mobile VID Dual 550kHz Synchronous 2-Phase Switching Regulator Controller with 5-Bit VID Plus LDO Synchronous Step-Down Controller with 5-Bit Mobile VID Control 5-Bit Desktop VID Programmable Synchronous Switching Regulator Wide Operating Range/Step-Down Controller, No RSENSE Dual Synchronous Switching Regulator with 5-Bit Desktop VID 2-Phase, Synchronous High Efficiency Converter with Mobile VID 3A, Monolithic Synchronous Regulator for DDR/QDR Memory Termination Low Input Voltage, High Power, No RSENSE, Step-Down Synchronous Controller Wide Operating Range, No RSENSE, Step-Down Controller Wide VIN Step-Down Controller for DDR Memory Termination COMMENTS SO-8 with Current Limit. No RSENSETM required Constant Frequency, Standby 5V and 3.3V LDOs, 3.5V ≤ VIN ≤ 36V 550kHz, 25MHz GBW Voltage Mode, VIN ≤ 7V, No RSENSE LTC1702 with Mobile VID for Portable Systems Provides Core, I/O and CLK Supplies for Portable Systems Current Mode, VIN to 36V, IOUT Up to 42A, Various VID Tables Fault Protection, Power Good, 3.5V to 36V Input, Current Mode 1.3V to 3.5V Programmable Output Using Internal 5-Bit DAC VIN Up to 36V, Current Mode, Power Good 1.3V to 3.5V Programmable Core Output Plus I/O Output Current Mode Ensures Accurate Current Sensing VIN Up to 36V, IOUT Up to 40A Low RDS(ON) Internal Switch: 85mΩ, ±3A Output Current (Sink and Source), VOUT = VREF/2 Minimum VIN: 1.5V, Uses Standard Logic-Level N-Channel MOSFETs VIN Up to 36V, Current Mode, Power Good, Stable with Ceramic COUT Current Mode Operation, VOUT = 1/2 VIN, VOUT (VTT) Tracks VIN (VDDQ), No RSENSE, Symmetrical Sink and Source Output Current Limit 1.5V ≤ VIN, Generates 5V Gate Drive for Standard N-Ch MOSFETs, 2A ≤ IOUT ≤ 25A VOUT as low as 0.6V LTC1628 Family Dual High Efficiency 2-Phase Synchronous Step-Down Controllers LTC1709 Family 2-Phase, 5-Bit Desktop VID Synchronous Step-Down Controllers LTC1736 LTC1753 LTC1778 LTC1873 LTC1929 LTC3413 LTC3713 LTC3778 LTC3717 LTC3718 LTC3832 Bus termination Supply for Low Votlage VIN High Power Synchronous Switching Regulator Controller No RSENSE is a trademark of Linear Technology Corporation. 3831f 20 Linear Technology Corporation 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 q FAX: (408) 434-0507 q LT/TP 0503 1K • PRINTED IN USA www.linear.com © LINEAR TECHNOLOGY CORPORATION 2001