MC34118DW

MOTOROLA Order this document by MC34118/D SEMICONDUCTOR TECHNICAL DATA ARCHIVED - 6/2010 MC34118 Voice Switched Speakerphone Circuit Silicon Monolithic Integrated Circuit The MC34118 Voice Switched Speakerphone Circuit incorporates the necessary amplifiers, attenuators, level detectors, and control algorithm to form the heart of a high quality hands--free speakerphone system. Included are a microphone amplifier with adjustable gain and MUTE control, Transmit and Receive attenuators which operate in a complementary manner, level detectors at both input and output of both attenuators, and background noise monitors for both the transmit and receive channels. A Dial Tone Detector prevents the dial tone from being attenuated by the Receive background noise monitor circuit. Also included are two line driver amplifiers which can be used to form a hybrid network in conjunction with an external coupling transformer. A high--pass filter can be used to filter out 60 Hz noise in the receive channel, or for other filtering functions. A Chip Disable pin permits powering down the entire circuit to conserve power on long loops where loop current is at a minimum. The MC34118 may be operated from a power supply, or it can be powered from the telephone line, requiring typically 5.0 mA. The MC34118 can be interfaced directly to Tip and Ring (through a coupling transformer) for stand--alone operation, or it can be used in conjunction with a handset speech network and/or other features of a featurephone. • • • • • • • • • • • 1 1 PIN CONNECTIONS (Top View) TIP BACKGROUND NOISE MONITOR VOLUME CONTROL LEVEL DETECTORS ATTENUATOR CONTROL ZB Rx ATTENUATOR 28 GND 2 27 CPR CD 3 26 RLI1 VCC 4 25 RLO1 HTO+ 5 24 TLO1 HTO-- 6 23 TLI1 HTI 7 22 RXO TXO 8 21 RXI TXI 9 20 RLI2 MCO 10 19 RLO2 MCI 11 18 TLO2 MUT 12 17 TLI2 VLC 13 16 CPT CT 14 15 VB Device MC34118DW/EG/R2 --1.0 LEVEL DETECTORS MC34118P Operating Temperature Range TA = --20° to +60°C Package 28 SOICW 28 PDIP RING BACKGROUND NOISE MONITOR DIAL TONE DETECTOR SPEAKER 1 FI ORDERING INFORMATION Tx ATTENUATOR AGC FO (Pin assignments same for both packages) SIMPLIFIED BLOCK DIAGRAM MIKE EG (Pb--free) SUFFIX DW SUFFIX PLASTIC PACKAGE CASE 751F 28--SOICW 28 Improved Attenuator Gain Range: 52 dB Between Transmit and Receive Low Voltage Operation for Line--Powered Applications (3.0 -- 6.5 V) 4--Point Signal Sensing for Improved Sensitivity Background Noise Monitors for Both Transmit and Receive Paths Microphone Amplifier Gain Set by External Resistors — Mute Function Included Chip Disable for Active/Standby Operation On Board Filter Pinned--Out for User Defined Function Dial Tone Detector to Inhibit Receive Idle Mode During Dial Tone Presence Standard 28--Pin Plastic DIP Package and SOIC Package Available Compatible with MC34119 Speaker Amplifier Motorola now offers Pb--free packages with suffix code EG MUTE P SUFFIX PLASTIC PACKAGE CASE 710 28--PDIP 28 BIAS FILTER VCC CHIP DISABLE POWER AMP (EXTERNAL) REV 1 10/03 © Motorola, Inc. 2003 MOTOROLA MC34118 1 ABSOLUTE MAXIMUM RATINGS Parameter Value Units -- 1.0, + 7.0 Vdc Voltage at CD (Pin 3), MUT (Pin 12) -- 1.0, VCC + 1.0 Vdc Voltage at VLC (Pin 13) -- 1.0, VCC + 0.5 Vdc Voltage at TXI (Pin 9), RXI (Pin 21), FI (Pin 2) -- 0.5, VCC + 0.5 Vdc -- 65 to + 150 °C Supply Voltage (Pin 4) Storage Temperature Range Devices should not be operated at or outside these values. The “Recommended Operating Limits” provide for actual device operation. RECOMMENDED OPERATING LIMITS Parameter Min Typ Max Units 3.5 — 6.5 Vdc CD Input (Pin 3), MUT Input (Pin 12) 0 — VCC Vdc IVB Current (Pin 15) — — 500 µA 0.3 x VB — VB Vdc Attenuator Input Signal Voltage (Pins 9, 21) 0 — 350 mVrms Microphone Amplifier, Hybrid Amplifier Gain 0 — 40 dB Load Current @ RXO, TXO (Pins 8, 22) @ MCO (Pin 10) @ HTO--, HTO+ (Pins 6, 5) 0 0 0 — — — ± 2.0 ± 1.0 ± 5.0 mA Ambient Operating Temperature Range -- 20 — + 60 °C Supply Voltage (Pin 4) (See Text) VLC (Pin 13) ELECTRICAL CHARACTERISTICS (TA = + 25°C, VCC = 5.0 V, CD ≤ 0.8 V, unless noted) Symbol Min Typ Max Units ICC — — 5.5 600 8.0 800 mA µA RCD VCDH VCDL 50 2.0 0 90 — — — VCC 0.8 kΩ Vdc Vdc VB — 1.8 1.3 2.1 — 2.4 Vdc VB Output Resistance (IVB = 1.0 mA) ROVB — 400 — Ω VB Power Supply Rejection Ratio (CVB = 220 µF, f = 1.0 kHz) PSRR — 54 — dB GRX GRX ∆GRX1 ∆GRX2 GRXI ∆GRX3 + 4.0 + 4.0 -- 0.5 — -- 22 49 + 6.0 + 6.0 0 -- 25 -- 20 52 + 8.0 + 8.0 + 0.5 -- 15 -- 17 54 VCR 27 35 — dB Parameter POWER SUPPLY VCC Supply Current (VCC = 6.5 V, CD = 0.8 V) (VCC = 6.5 V, CD = 2.0 V) CD Input Resistance (VCC = VCD = 6.5 V) CD Input Voltage — High — Low VB Output Voltage (VCC = 3.5 V) (VCC = 5.0 V) ATTENUATORS (TA = + 25°C) Receive Attenuator Gain (f = 1.0 kHz, VLC = VB) Rx Mode, RXI = 150 mVrms (VCC = 5.0 V) Rx Mode, RXI = 150 mVrms (VCC = 3.5 V) Gain Change -- VCC = 3.5 V versus VCC = 5.0 V AGC Gain Change -- VCC = 2.8 V versus VCC = 5.0 V* Idle Mode, RXI = 150 mVrms Range (Rx to Tx Mode) Volume Control Range (Rx Mode, 0.3 VB < VLC < VB) RXO DC Voltage (Rx Mode) dB VRXO — VB — Vdc ∆RXO DC Voltage (Rx to Tx Mode) ∆VRXO — ± 10 ± 150 mV RXO High Voltage (Iout = -- 1.0 mA RXI = VB + 1.5 V) VRXOH 3.7 — — Vdc RXO Low Voltage (Iout = + 1.0 mA, RXI = VB -- 1.0, Output measured with respect to VB)* VRXOL — -- 1.5 -- 1.0 Vdc RRXI 7.0 10 14 kΩ RXI Input Resistance (RXI < 350 mVrms) (continued) MC34118 2 MOTOROLA ELECTRICAL CHARACTERISTICS (TA = + 25°C, VCC = 5.0 V, CD ≤ 0.8 V, unless noted) Symbol Min Typ Max Transmit Attenuator Gain (f = 1.0 kHz) Tx Mode, TXI = 150 mVrms Idle Mode, TXI = 150 mVrms Range (Tx to Rx Mode) GTX GTXI ∆GTXI + 4.0 -- 22 49 + 6.0 -- 20 52 + 8.0 -- 17 54 TXO DC Voltage (Tx Mode) VTXO — VB — Parameter Units ATTENUATORS (continued) (TA = + 25°C) dB Vdc ∆TXO DC Voltage (Tx to Rx Mode) ∆VTXO — ± 30 ± 150 mV TXO High Voltage (Iout = -- 1.0 mA TXI = VB + 1.5 V) VTXOH 3.7 — — Vdc TXO Low Voltage (Iout = + 1.0 mA, TXI = VB -- 1.0 V, Output measured with respect to VB)* VTXOL — -- 1.5 -- 1.0 Vdc TXI Input Resistance (TXI < 350 mVrms) RTXI 7.0 10 14 kΩ Gain Tracking (GRX + GTX, @ Tx, Idle, Rx)* GTR — ± 0.1 — dB — — — + 240 0 -- 240 — — — * See text for explanation. ATTENUATOR CONTROL (TA = + 25°C) CT Voltage (Pin 14 -- VB) Rx Mode (VLC = VB) Idle Mode Tx Mode VCT CT Source Current (switching to Rx mode) ICTR -- 85 -- 60 -- 40 µA CT Sink Current (switching to Tx mode) ICTT + 40 + 60 + 85 µA CT Slow Idle Current ICTS — 0 — µA CT Fast Idle Internal Resistance RFI 1.5 2.0 3.6 kΩ VLC Input Current IVLC — -- 60 — nA Dial Tone Detector Threshold VDT 10 15 20 mV 0 + 50 mVdc mV MICROPHONE AMPLIFIER (TA = + 25°C, VMUT ≤ 0.8 V, AVCL = 31 dB unless otherwise noted) Output Offset (VMCO -- VB, Feedback R = 180 kΩ) MCOVOS -- 50 Open Loop Gain (f < 100 Hz) AVOLM 70 80 — dB Gain Bandwidth GBWM — 1.0 — MHz Output High Voltage (Iout = -- 1.0 mA, VCC = 5.0 V) VMCOH 3.7 — — Vdc Output Low Voltage (Iout = + 1.0 mA) VMCOL — — 200 mVdc IBM — -- 40 — nA GMT -- 55 — — -- 68 — — dB Input Bias Current (@ MCI) Muting (∆Gain) (f = 1.0 kHz, VMUT = 2.0 V) (300 Hz < f < 10 kHz) MUT Input Resistance (VCC = VMUT = 6.5 V) RMUT 50 90 — kΩ MUT Input — High VMUTH 2.0 — VCC Vdc MUT Input — Low VMUTL 0 — 0.8 Vdc Distortion (300 Hz < f < 10 kHz) THDM — 0.15 — % HYBRID AMPLIFIERS (TA = + 25°C) HTO -- Offset (VHTO-- -- VB, Feedback R = 51 kΩ) HVOS -- 20 0 + 20 mVdc HBVOS -- 30 0 + 30 mVdc Open Loop Gain (HTI to HTO--, f < 100 Hz) AVOLH 60 80 — dB Gain Bandwidth GBWH — 1.0 — MHz Closed Loop Gain (HTO-- to HTO+) AVCLH -- 0.35 0 + 0.35 dB IBH — -- 30 — nA HTO-- High Voltage (Iout = -- 5.0 mA) VHT--H 3.7 — — Vdc HTO-- Low Voltage (Iout = + 5.0 mA) VHT--L — — 250 mVdc HTO+ High Voltage (Iout = -- 5.0 mA) VHT+H 3.7 — — Vdc HTO+ Low Voltage (Iout = + 5.0 mA) VHT+L — — 450 mVdc Distortion (300 Hz < f < 10 kHz, See Figure 1) THDH — 0.3 — % HTO-- to HTO+ Offset (Feedback R = 51 kΩ) Input Bias Current (@ HTI) MOTOROLA MC34118 3 Parameter Symbol Min Typ Max Units LEVEL DETECTORS AND BACKGROUND NOISE MONITORS (TA = + 25°C) Transmit--Receive Switching Threshold (Ratio of Current at RLI1 + RLI2 to 20 µA at TLI1 + TLI2 to switch from Tx to Rx) ITH 0.8 1.0 1.2 Source Current at RLO1, RLO2, TLO1, TLO2 ILSO — -- 2.0 — mA Sink Current at RLO1, RLO2, TLO1, TLO2 ILSK — 4.0 — µA CPR, CPT Output Resistance (Iout = 1.5 mA) RCP — 35 — Ω CPR, CPT Leakage Current ICPLK — -- 0.2 — µA FILTER (TA = + 25°C) FOVOS -- 200 -- 90 0 mV FO Sink Current Voltage Offset at FO (VFO -- VB, 220 kΩ from VB to FI) IFO 150 260 400 µA FI Bias Current IFI — -- 50 — nA Rx Mode (From FI to RXO, FO connected to RXI) THDR — 0.5 3.0 % Tx Mode (From MCI to HTO--/HTO+, includes Tx attenuator) THDT — 0.8 3.0 % SYSTEM DISTORTION (TA = + 25°C, f = 1.0 kHz) 1. All currents into a device pin are positive, those out of a pin are negative. Algebraic convention rather than magnitude is used to define limits. 51 k 0.1 10 k -+ HTI VB R R VB -+ 1200 Ω Vin HTO-X1 INSTR. AMPLIFIER TO DIST. ANALYZER HTO+ Figure 1. Hybrid Amplifier Distortion Test TEMPERATURE CHARACTERISTICS Typical Value @ 25°C Typical Change -- 20 to + 60°C VCC Supply Current (CD = 0.8 V) 5.0 mA -- 0.3%/°C VCC Supply Current (CD = 2.0 V) 400 µA -- 0.4%/°C VB Output Voltage (VCC = 5.0 V) 2.1 V + 0.8%/°C Attenuator Gain (Max Gain) + 6.0 dB 0.0008 dB/°C Attenuator Gain (Max Attenuation) -- 46 dB 0.004 dB/°C Parameter Attenuator Input Resistance (@ TXI, RXI) 10 kΩ + 0.6%/°C Dial Tone Detector Threshold 15 mV + 20 µV/°C CT Source, Sink Current ± 60 µA -- 0.15 %/°C 0 mV ± 4.0 µV/°C 1.0 ± 0.02%/°C 4.0 µA -- 10 nA/°C 0 dB 0.001%/°C Microphone, Hybrid Amplifier Offset Transmit--Receive Switching Threshold Sink Current at RLO1, RLO2, TLO1, TLO2 Closed Loop Gain (HTO-- to HTO+) MC34118 4 MOTOROLA PIN DESCRIPTION Pin Name 1 FO Filter output. Output impedance is less than 50 ohms. 2 FI Filter input. Input impedance is greater than 1.0 Mohm. 3 CD Chip Disable. A logic low (< 0.8 V) sets normal operation. A logic high (> 2.0 V) disables the IC to conserve power. Input impedance is nominally 90 kΩ. 4 5 VCC HTO+ Description Pin Name Description 13 VLC Volume control input. When VLC = VB, the receive attenuator is at maximum gain when in the receive mode. When VLC = 0.3 VB, the receive gain is down 35 dB. Does not affect the transmit mode. 14 CT An RC at this pin sets the response time for the circuit to switch modes. 15 VB A supply voltage of + 2.8 to + 6.5 volts is required, at ≈5.0 mA. As VCC falls from 3.5 to 2.8 volts, an AGC circuit reduces the receive attenuator gain by ≈25 dB (when in the receive mode). An output voltage ≈VCC/2. This voltage is a system ac ground, and biases the volume control. A filter cap is required. 16 CPT An RC at this pin sets the time constant for the transmit background monitor. Output of the second hybrid amplifier. The gain is internally set at -- 1.0 to provide a differential output, in conjunction with HTO--, to the hybrid transformer. 17 TLI2 Input to the transmit level detector on the mike/speaker side. 18 TLO2 Output of the first hybrid amplifier. The gain of the amp is set by external resistors. Output of the transmit level detector on the mike/speaker side, and input to the transmit background monitor. 19 RLO2 Output of the receive level detector on the mike/speaker side. 6 HTO-- 7 HTI Input and summing node for the first hybrid amplifier. DC level is ≈VB. 20 RLI2 8 TXO Output of the transmit attenuator. DC level is approximately VB. Input to the receive level detector on the mike/speaker side. 21 RXI 9 TXI Input to the transmit attenuator. Max. signal level is 350 mVrms. Input impedance is ≈10 kΩ. Input to the receive attenuator and dial tone detector. Max input level is 350 mV RMS. Input impedance is ≈10 kΩ. 22 RXO Output of the receive attenuator. DC level is approximately VB. 10 MCO Output of the microphone amplifier. The gain of the amplifier is set by external resistors. 23 TLI1 11 MCI Input and summing node of the microphone amplifier. DC level is ≈VB. Input to the transmit level detector on the line side. 24 TLO1 12 MUT Mute input. A logic low (< 0.8 V) sets normal operation. A logic high (> 2.0 V) mutes the microphone amplifier without affecting the rest of the circuit. Input impedance is nominally 90 kΩ. Output of the transmit level detector on the line side. 25 RLO1 Output of the receive level detector on the line side, and input to the receive background monitor. 26 RLI1 Input to the receive level detector on the line side. 27 CPR An RC at this pin sets the time constant for the receive background monitor. 28 GND Ground pin for the entire IC. NOTE: Pin numbers are identical for the DIP package and the SOIC package. MOTOROLA MC34118 5 Figure 2. MC34118 Block Diagram MC34118 6 MOTOROLA VCC VCC MIKE R R VB BIAS 400 15 20 LEVEL DETECTORS BACKGROUND NOISE MONITOR 10 + TLI2 MCO -- RMF 11 V B MCI SPEAKER AMP MC34119 GND 28 CD 3 4 VCC 18 19 RLO2 TLO2 16 CPT MUTE 12 RMI VB RLI2 17 VB CT 14 TXI 9 VLC 13 RXO 22 ATTENUATOR CONTROL AGC VCC VOLUME VB DIAL TONE DETECTOR -+ VB Rx ATTENUATOR 15 mV Tx ATTENUATOR 23 8 21 RXI 26 VB -+ VB 1 FO R -+ R ZBAL +1 FILTER 6 HTO-- BACKGROUND NOISE MONITOR HTI 7 RLI1 LEVEL DETECTORS TLI1 RHI TXO RHF FI 2 VB TLO1 RLO1 25 24 27 CPR HTO+ 5 VCC RING TIP FUNCTIONAL DESCRIPTION INTRODUCTION The fundamental difference between the operation of a speakerphone and a handset is that of half--duplex versus full--duplex. The handset is full duplex since conversation can occur in both directions (transmit and receive) simultaneously. A speakerphone has higher gain levels in both paths, and attempting to converse full duplex results in oscillatory problems due to the loop that exists within the system. The loop is formed by the receive and transmit paths, the hybrid, and the acoustic coupling (speaker to microphone). The only practical and economical solution used to date is to design the speakerphone to function in a half duplex mode — i.e., only one person speaks at a time, while the other listens. To achieve this requires a circuit which can detect who is talking, switch on the appropriate path (transmit or receive), and switch off (attenuate) the other path. In this way, the loop gain is maintained less than unity. When the talkers exchange function, the circuit must quickly detect this, and switch the circuit appropriately. By providing speech level detectors, the circuit operates in a “hands--free” mode, eliminating the need for a “push--to--talk” switch. The handset, by the way, has the same loop as the speakerphone. But since the gains are considerably lower, and since the acoustic coupling from the earpiece to the mouth piece is almost non--existent (the receiver is normally held against a person’s ear), oscillations don’t occur. The MC34118 provides the necessary level detectors, attenuators, and switching control for a properly operating speakerphone. The detection sensitivity and timing are externally controllable. Additionally, the MC34118 provides background noise monitors which make the circuit insensitive to room and line noise, hybrid amplifiers for interfacing to Tip and Ring, the microphone amplifier, and other associated functions. Please refer to the Block Diagram (Figure 2) when reading the following sections. ATTENUATORS The transmit and receive attenuators are complementary in function, i.e., when one is at maximum gain (+ 6.0 dB), the other is at maximum attenuation (-- 46 dB), and vice versa. They are never both fully on or both fully off. The sum of their gains remains constant (within a nominal error band of ± 0.1 dB) at a typical value of -- 40 dB (see Figure 10). Their purpose is to control the transmit and receive paths to provide the half--duplex operation required in a speakerphone. The attenuators are non--inverting, and have a -- 3.0 dB (from max gain) frequency of ≈100 kHz. The input impedance of each attenuator (TXI and RXI) is nominally 10 kΩ (see Figure 3), and the input signal should be limited to 350 mVrms (990 mVp--p) to prevent distortion. That maximum recommended input signal is independent of the volume control setting. The diode clamp on the inputs limits the input swing, and therefore the maximum negative output swing. This is the reason for VRXOL and VTXOL specification MOTOROLA being defined as they are in the Electrical Characteristics. The output impedance is < 10 Ω until the output current limit (typically 2.5 mA) is reached. VB 10 k TXI (RXI) 4.0 k 96 k TO ATTENUATOR INPUT Figure 3. Attenuator Input Stage The attenuators are controlled by the single output of the Control Block, which is measurable at the CT pin (Pin 14). When the CT pin is at + 240 millivolts with respect to VB, the circuit is in the receive mode (receive attenuator is at + 6.0 dB). When the CT pin is at -- 240 millivolts with respect to VB, the circuit is in the transmit mode (transmit attenuator is at + 6.0 dB). The circuit is in an idle mode when the CT voltage is equal to VB, causing the attenuators’ gains to be halfway between their fully on and fully off positions (-- 20 dB each). Monitoring the CT voltage (with respect to VB) is the most direct method of monitoring the circuit’s mode. The inputs to the Control Block are seven: 2 from the comparators operated by the level detectors, 2 from the background noise monitors, the volume control, the dial-tone detector, and the AGC circuit. These seven inputs are described below. LEVEL DETECTORS There are four level detectors — two on the receive side and two on the transmit side. Refer to Figure 4 — the terms in parentheses form one system, and the other terms form the second system. Each level detector is a high gain amplifier with back--to--back diodes in the feedback path, resulting in non--linear gain, which permits operation over a wide dynamic range of speech levels. Refer to the graphs of Figures 11, 12 and 13 for their dc and ac transfer characteristics. The sensitivity of each level detector is determined by the external resistor and capacitor at each input (TLI1, TLI2, RLI1, and RLI2). Each output charges an external capacitor through a diode and limiting resistor, thus providing a dc representation of the input ac signal level. The outputs have a quick rise time (determined by the capacitor and an internal 350 Ω resistor), and a slow decay time set by an internal current source and the capacitor. The capacitors on the four outputs should have the same value (± 10%) to prevent timing problems. Referring to Figure 2, on the receive side, one level detector (RLI1) is at the receive input receiving the same signal as at Tip and Ring, and the other (RLI2) is at the output of the MC34118 7 LEVEL DETECTOR RLI1 (TLI2) 5.1 k SIGNAL INPUT SIGNAL INPUT VB -+ 350 Ω RLO1 (TLO2) 0.1 µF 2.0 µF -+ + -4.0 µA CPR (CPT) BACKGROUND NOISE MONITOR 36 mV -+ 100 k VCC 47 µF 56 k 33 k VB 0.1 µF LEVEL DETECTOR C4 (C3) 5.1 k TLI1 (RLI2) VB -+ -+ 350 Ω TLO1 (RLO2) 2.0 µF 4.0 µA C2 (C1) TO ATTENUATOR CONTROL BLOCK COMPARATOR NOTE: External component values are applicaton dependent. Figure 4. Level Detectors speaker amplifier. On the transmit side, one level detector (TLI2) is at the output of the microphone amplifier, while the other (TLI1) is at the hybrid output. Outputs RLO1 and TLO1 feed a comparator, the output of which goes to the Attenuator Control Block. Likewise, outputs RLO2 and TLO2 feed a second comparator which also goes to the Attenuator Control Block. The truth table for the effects of the level detectors on the Control Block is given in the section describing the Control Block. BACKGROUND NOISE MONITORS The purpose of the background noise monitors is to distinguish speech (which consists of bursts) from background noise (a relatively constant signal level). There are two background noise monitors — one for the receive path and one for the transmit path. Referring to Figure 4, the receive background noise monitor is operated on by the RLI1--RLO1 level detector, while the transmit background noise monitor is operated on by the TLI2--TLO2 level detector. They monitor the background noise by storing a dc voltage representative of the respective noise levels in capacitors at CPR and CPT. The voltages at these pins have slow rise times (determined by the external RC), but fast decay times. If the signal at RLI1 (or TLI2) changes slowly, the voltage at CPR (or CPT) will remain more positive than the voltage at the non--inverting input of the monitor’s output comparator. When speech is present, the voltage on the noninverting input of the comparator will rise quicker than the voltage at the inverting input (due to the burst characteristic of speech), causing its output to change. This output is sensed by the Attenuator Control Block. The 36 mV offset at the comparator’s input keeps the comparator from changing state unless the speech level exceeds the background noise by ≈4.0 dB. The time constant MC34118 8 of the external RC (≈4.7 seconds) determines the response time to background noise variations. VOLUME CONTROL The volume control input at VLC (Pin 13) is sensed as a voltage with respect to VB. The volume control affects the attenuators only in the receive mode. It has no effect in the idle or transmit modes. When in the receive mode, the gain of the receive attenuator will be + 6.0 dB, and the gain of the transmit attenuator will be -- 46 dB only when VLC is equal to VB. As VLC is reduced below VB, the gain of the receive attenuator is reduced (see Figure 14), and the gain of the transmit attenuator is increased such that their sum remains constant. Changing the voltage at VLC changes the voltage at CT (see the Attenuator Control Block section), which in turn controls the attenuators. The volume control setting does not affect the maximum attenuator input signal at which noticeable distortion occurs. The bias current at VLC is typically 60 nA out of the pin, and does not vary significantly with the VLC voltage or with VCC. DIAL TONE DETECTOR The dial tone detector is a comparator with one side connected to the receive input (RXI) and the other input connected to VB with a 15 mV offset (see Figure 5). If the circuit is in the receive mode, and the incoming signal is greater than 15 mV (10 mVrms), the comparator’s output will change, disabling the receive idle mode. The receive attenuator will then be at a setting determined solely by the volume control. The purpose of this circuit is to prevent the dial tone MOTOROLA (which would be considered as continuous noise) from fading away as the circuit would have the tendency to switch to the idle mode. By disabling the receive idle mode, the dial tone remains at the normally expected full level. TO Rx ATTENUATOR RXI TO ATTENUATOR CONTROL BLOCK -+ 15 mV C4 VB Figure 5. Dial Tone Detector AGC The AGC circuit affects the circuit only in the receive mode, and only when the supply voltage (VCC) is less than 3.5 volts. As VCC falls below 3.5 volts, the gain of the receive attenuator is reduced according to the graph of Figure 15. The transmit path attenuation changes such that the sum of the transmit and receive gains remains constant. The purpose of this feature is to reduce the power (and current) used by the speaker when a line--powered speakerphone is connected to a long line, where the available power is limited. By reducing the speaker power, the voltage sag at VCC is controlled, preventing possible erratic operation. ATTENUATOR CONTROL BLOCK The Attenuator Control Block has the seven inputs described above: — The output of the comparator operated by RLO2 and TLO2 (microphone/speaker side) — designated C1. — The output of the comparator operated by RLO1 and TLO1 (Tip/Ring side) — designated C2. — The output of the transmit background noise monitor — designated C3. — The output of the receive background noise monitor — designated C4. — The volume control. — The dial tone detector. — The AGC circuit. The single output of the Control Block controls the two attenuators. The effect of C1 -- C4 is as follows: Inputs C1 C2 C3 C4 Output Mode Tx Tx Rx Rx Tx Tx Rx Rx Tx Rx Tx Rx Tx Rx Tx Rx 1 y y X 0 0 0 X X y y 1 X 0 0 0 Transmit Fast Idle Fast Idle Receive Slow Idle Slow Idle Slow Idle Slow Idle X = Don’t Care; y = C3 and C4 are not both 0. MOTOROLA A definition of the above terms: 1) “Transmit” means the transmit attenuator is fully on (+ 6.0 dB), and the receive attenuator is at max. attenuation (-- 46 dB). 2) “Receive” means both attenuators are controlled by the volume control. At max. volume, the receive attenuator is fully on (+ 6.0 dB), and the transmit attenuator is at max. attenuation (-- 46 dB). 3) “Fast Idle” means both transmit and receive speech are present in approximately equal levels. The attenuators are quickly switched (30 ms) to idle until one speech level dominates the other. 4) “Slow Idle” means speech has ceased in both transmit and receive paths. The attenuators are then slowly switched (1 second) to the idle mode. 5) Switching to the full transmit or receive modes from any other mode is at the fast rate (≈30 ms). A summary of the truth table is as follows: 1) The circuit will switch to transmit if: a) both transmit level detectors sense higher signal levels relative to the respective receive level detectors (TLI1 versus RLI1, TLI2 versus RLI2), and b) the transmit background noise monitor indicates the presence of speech. 2) The circuit will switch to receive if: a) both receive level detectors sense higher signal levels relative to the respective transmit level detectors, and b) the receive background noise monitor indicates the presence of speech. 3) The circuit will switch to the fast idle mode if the level detectors disagree on the relative strengths of the signal levels, and at least one of the background noise monitors indicates speech. For example, referring to the Block Diagram (Figure 2), if there is sufficient signal at the microphone amp output (TLI2) and there is sufficient signal at the receive input (RLI1) to override the signal at the hybrid output (TLI1), and either or both background monitors indicate speech, then the circuit will be in the fast idle mode. Two conditions which can cause the fast idle mode to occur are a) when both talkers are attempting to gain control of the system by talking at the same time, and b) when one talker is in a very noisy environment, forcing the other talker to continually override that noise level. In general, the fast idle mode will occur infrequently. 4) The circuit will switch to the slow idle mode when a) both talkers are quiet (no speech present), or b) when one talker’s speech level is continuously overriden by noise at the other speaker’s location. The time required to switch the circuit between transmit, receive, fast idle and slow idle is determined in part by the components at the CT pin (Pin 14). (See the section on Switching Times for a more complete explanation of the switching time components.) A schematic of the CT circuitry is shown in Figure 6, and operates as follows: — RT is typically 120 kΩ, and CT is typically 5.0 µF. — To switch to the receive mode, I 1 is turned on (I 2 is off), charging the external capacitor to + 240 mV above V B. (An internal clamp prevents further charging of the capacitor.) — To switch to the transmit mode, I2 is turned on (I1 is off) bringing down the voltage on the capacitor to -- 240 mV with respect to VB. MC34118 9 VB RT -+ 2.0 k CT CT TO ATTENUATORS CONTROL CIRCUIT I1 I2 4 C1 -- C4 VOL. CONTROL DIAL TONE DET. AGC 60 µA EACH Figure 6. CT Attenuator Control Block Circuit MICROPHONE AMPLIFIER The microphone amplifier (Pins 10, 11) has the noninverting input internally connected to VB, while the inverting input and the output are pinned out. Unlike most op--amps, the amplifier has an all--NPN output stage, which maximizes phase margin and gain--bandwidth. This feature ensures stability at gains less than unity, as well as with a wide range of reactive loads. The open loop gain is typically 80 dB (f < 100 Hz), and the gain--bandwidth is typically 1.0 MHz (See Figure 16). The maximum p--p output swing is typically 1.0 volt less than VCC with an output impedance of < 10 Ω until current limiting is reached (typically 1.5 mA). Input bias current at MCI is typically 40 nA out of the pin. RMF VCC FROM MIKE VB RMI MCI + -- MCO VCC MUTE GAIN = – 75 k 75 k shorting the output to the inverting input (see Figure 7). The mute input has a threshold of ≈1.5 volts, and the voltage at this pin must be kept within the range of ground and VCC (see Figure 17). If the mute function is not used, the pin should be grounded. HYBRID AMPLIFIERS The two hybrid amplifiers (at HTO+, HTO--, and HTI), in conjunction with an external transformer, provide the two--to-four wire converter for interfacing to the telephone line. The gain of the first amplifier (HTI to HTO--) is set by external resistors (gain = -- RHF/RHI in Figure 2), and its output drives the second amplifier, the gain of which is internally set at -- 1.0. Unlike most op--amps, the amplifiers have an all--NPN output stage, which maximizes phase margin and gain-bandwidth. This feature ensures stability at gains less than unity, as well as with a wide range of reactive loads. The open loop gain of the first amplifier is typically 80 dB, and the gain bandwidth of each amplifier is ≈1.0 MHz (see Figure 16). The maximum p--p output swing of each amplifier is typically 1.2 volts less than VCC with an output impedance of < 10 Ω until current limiting is reached (typically 8.0 mA). The output current capability is guaranteed to be a minimum of 5.0 mA. The bias current at HTI is typically 30 nA out of the pin. The connections to the coupling transformer are shown in the Block Diagram (Figure 2). The block labeled ZBal is the balancing network necessary to match the line impedance. FILTER The operation of the filter circuit is determined by the external components. The circuit within the MC34118, from pins FI to FO is a buffer with a high input impedance (> 1.0 MΩ), and a low output impedance (< 50 Ω). The configuration of the external components determines whether the circuit is a high--pass filter (as shown in Figure 2), a low--pass filter, or a band--pass filter. As a high pass filter, with the components shown in Figure 8, the filter will keep out 60 Hz (and 120 Hz) hum which can be picked up by the external telephone lines. As a low pass filter (Figure 9), it can be used to roll off the high end frequencies in the receive circuit, which aids in protecting against acoustic feedback problems. 0 -- 3.0 50 305 Hz VB R1 56 k R MF R MI -- 30 f Figure 7. Microphone Amplifier and Mute = 1 N 2π  FOR C1 = C2 The muting function (Pin 12), when activated, will reduce the gain of the amplifier to ≈-- 39 dB (with RMI = 5.1 kΩ) by MC34118 10 C1 fN 1 C 2R1R2 VCC 100 k R2 220 k — To switch to idle quickly (fast idle), the current sources are turned off, and the internal 2.0 kΩ resistor is switched in, discharging the capacitor to VB with a time constant = 2.0 k x CT. — To switch to idle slowly (slow idle), the current sources are turned off, the switch at the 2.0 kΩ resistor is open, and the capacitor discharges to VB through the external resistor RT with a time constant = RT x CT. C2 FI FO 4700 pF 4700 pF 260 µA MC34118 Figure 8. High Pass Filter MOTOROLA With an appropriate choice of an input coupling capacitor to the low pass filter, a band pass filter is formed. 4.0 k 0 20 k -- 3.0 C1 -- 30 f VB = 1 2π 0.01 R2 FI 13 k R1 0.001 13 k fN N 220 k  1 C1C2R 2 C2 +1 FO MC34118 Vin FOR R1 = R2 Figure 9. Low Pass Filter POWER SUPPLY, VB, AND CHIP DISABLE The power supply voltage at VCC (Pin 4) is to be between 3.5 and 6.5 volts for normal operation, with reduced operation possible down to 2.8 volts (see Figure 15 and the AGC section). The power supply current is shown in Figure 18 for both the power--up and power--down mode. The output voltage at VB (Pin 15 is ≈(VCC -- 0.7)/2, and provides the ac ground for the system. The output impedance at VB is ≈400 Ω (see Figure 19), and in conjunction with the external capacitor at VB, forms a low pass filter for power supply rejection. Figure 20 indicates the amount of rejection with different capacitors. The choice of capacitor is application dependent based on whether the circuit is powered by the telephone line or a power supply. Since VB biases the microphone and hybrid amplifiers, the amount of supply rejection at their outputs is directly related to the rejection at VB, as well as their respective gains. Figure 21 depicts this graphically. The Chip Disable (Pin 3) permits powering down the IC to conserve power and/or for muting purposes. With CD ≤ 0.8 volts, normal operation is in effect. With CD ≥ 2.0 volts and ≤ VCC, the IC is powered down. In the powered down mode, the microphone and the hybrid amplifiers are disabled, and their outputs go to a high impedance state. Additionally, the bias is removed from the level detectors. The bias is not removed from the filter (Pins 1, 2), the attenuators (Pins 8, 9, 21, 22), or from Pins 13, 14, and 15 (the attenuators are disabled, however, and will not pass a signal). The input impedance at CD is typically 90 kΩ, has a threshold of ≈1.5 volts, and the voltage at this pin must be kept within the range of ground and VCC (see Figure 17). If CD is not used, the pin should be grounded. 140 + 10 TRANSMIT ATTENUATOR 120 GAIN (dB) -- 10 OUTPUT CHANGE (mV) 0 RECEIVE ATTENUATOR -- 20 -- 30 -- 40 -- 160 -- 80 0 + 80 + 160 VCT -- VB (mV) Figure 10. Attenuator Gain versus VCT (Pin 14) MOTOROLA 80 60 VB Iin (RLI1, RLI2 TLI1, TLI2) 40 20 VCC ≥ 3.5 V -- 50 -- 240 100 + 240 0 0 -- 20 -- 40 350 Ω -+ 2.0 µF Vout (RLO1, RLO2 TLO1, TLO2) -- 60 -- 80 4.0 µA -- 100 DC INPUT CURRENT (µA) Figure 11. Level Detector DC Transfer Characteristics MC34118 11 + 5.0 160 OUTPUT CHANGE FROM 1.0 kHz (mV) R = 5.1 k, C = 0.1 µF R = 10 k, C = 0.1 µF OUTPUT CHANGE (mV) 120 R = 10 k, C = 0.047 µF 80 C VB R 40 Vin @ 1.0 kHz 0 0 20 -+ (RLI1, RLI2 TLI1, TLI2) 350 Ω 2.0 µF Vout (RLO1, RLO2 TLO1, TLO2) 40 60 4.0 µA Vin = 1.0 mVrms 10 mVrms 40 mVrms 0 (RLI1, RLI2 TLI1, TLI2) -- 10 0.047 -- 15 80 VB 4.0 µA 2.0 µF Vout (RLO1, RLO2 TLO1, TLO2) 300 1.0K 3.0K 10K f, FREQUENCY (Hz) Figure 13. Level Detector AC Transfer Characteristics versus Frequency + 10 0 0 GAIN CHANGE (dB) -- 6.0 -- 10 GAIN (dB) 10 k 350 Ω -+ Vin @ 1.0 kHz -- 20 Figure 12. Level Detector AC Transfer Characteristics -- 20 -- 30 CIRCUIT IN RECEIVE MODE -- 12 -- 18 CIRCUIT IN RECEIVE MODE 0.5 VB ≤ VLC ≤ VB -- 24 -- 40 MINIMUM RECOMMENDED LEVEL 0.2 0.4 0.6 VLC/VB 0.8 -- 30 1.0 36 MICROPHONE AMP PHASE 80 72 HYBRID AMP PHASE 60 GAIN 108 144 180 40 20 2.9 3.0 3.1 3.2 VCC (VOLTS) 3.3 3.4 3.5 100 80 INPUT CURRENT ( µA) 0 EXCESS PHASE (DEGREES) + 100 2.8 Figure 15. Receive Attenuation Gain versus VCC Figure 14. Receive Attenuator versus Volume Control A VOL (dB) Vin = 10 mVrms -- 5.0 INPUT SIGNAL (mVrms) -- 50 Vin ≥ 10 mV VALID FOR 0 ≤ CD, MUT ≤ VCC 60 40 20 0 100 1.0K 10K 100K FREQUENCY (Hz) Figure 16. Microphone Amplifier and 1st Hybrid Amplifier Open Loop Gain and Phase MC34118 12 1.0M 0 0 1.0 2.0 3.0 4.0 5.0 6.5 VCC + 1.0 V INPUT VOLTAGE (VOLTS) Figure 17. Input Characteristics @ CD, MUT MOTOROLA 3.0 6.0 5.0 2.5 CD ≤ 0.8 V V B (VOLTS) ICC (mA) 4.0 3.0 2.0 2.0 1.5 VCC = 3.5 V 1.0 1.0 0 VCC = 6.0 V 0.5 2.0 ≤ CD ≤ VCC 0 1.0 2.0 3.0 4.0 5.0 0 6.0 0.5 0 VCC (VOLTS) 1.0 1.5 2.0 IB LOAD CURRENT (mA) Figure 18. Supply Current versus Supply Voltage Figure 19. VB Output Characteristics 60 60 40 220 µF 20 0 300 100 µF CVB = 50 µF 1.0K 2.0K HTO--, CVB = 220 µF 50 PSRR (dB) PSRR (dB) 1000 µF 500 µF 80 40 30 MCO, CVB = 1000 µF MCO, CVB = 220 µF 20 3.0K HTO--, CVB = 1000 µF MIC AMP GAIN = 31 dB HYBRID AMP GAIN = 14 dB SEE TEXT  10 300 1.0K FREQUENCY (Hz) 3.0K FREQUENCY (Hz) Figure 20. VB Power Supply Rejection versus Frequency and VB Capacitor Figure 21. Power Supply Rejection of the Microphone and Hybrid Amplifiers P--P OUTPUT SWING (VOLTS) 6.0 MCO 5.0 4.0 TXO, RXO HTO--, + 3.0 2.0 TXO 1.0 0 3.0 FO RXO 3.5 TYPICAL @ 25°C 4.0 4.5 5.0 5.5 6.0 6.5 VCC (VOLTS) Figure 22. Typical Output Swing versus VCC MOTOROLA MC34118 13 DESIGN GUIDELINES SWITCHING TIME The switching time of the MC34118 circuit is dominated by the components at CT (Pin 14, refer to Figure 6), and secondarily by the capacitors at the level detector outputs (RLO1, RLO2, TLO1, TLO2). The time to switch to receive or to transmit from idle is determined by the capacitor at CT, together with the internal current sources (refer to Figure 6). The switching time is: ∆T = ∆V x C I T For the typical case where ∆V = 240 mV, I = 60 µA, and CT is 5.0 µF, ∆T = 20 ms. If the circuit switches directly from receive to transmit (or vice--versa), the total switching time would be 40 ms. The switching time from either receive or transmit to idle depends on which type of idle mode is in effect. If the circuit is going to “fast idle,” the time constant is determined by the CT capacitor, and the internal 2.0 kΩ resistor (Figure 6). With CT = 5.0 µF, the time constant is ≈10 ms, giving a switching time to idle of ≈30 ms (for 95% change). Fast idle is an infrequent occurrence, however, occurring when both speakers are talking and competing for control of the circuit. The switching time from idle back to either transmit or receive is described above. If the circuit is switching to “slow idle,” the time constant is determined by the CT capacitor and RT, the external resistor (see Figure 6). With CT = 5.0 µF, and RT = 120 kΩ, the time constant is ≈ 600 ms, giving a switching time of ≈1.8 seconds (for a 95% change). The switching period to slow idle begins when both speakers have stopped talking. The switching time back to the original mode will depend on how soon that speaker begins speaking again. The sooner the speaking starts during the 1.8 second period, the quicker the switching time since a smaller voltage excursion is required. That switching time is determined by the internal current sources as described above. The above switching times occur, however, after the level detectors have detected the appropriate signal levels, since their outputs operate the Attenuator Control Block. Referring to Figure 4, the rise time of the level detectors’ outputs to new speech is quick by comparison (≈1.0 ms), determined by the internal 350 Ω resistor and the external capacitor (typically 2.0 µF). The output’s decay time is determined by the external capacitor, and an internal 4.0 µA current source giving a decay rate of ≈60 ms for a 120 mV excursion at RLO or TLO. However, the overall response time of the circuit is not a constant since it depends on the relative strength of the signals at the different level detectors, as well as the timing of the signals with respect to each other. The capacitors at the four outputs (RLO1, RLO2, TLO1, TLO2) must be equal value (± 10%) to prevent problems in timing and level response. MC34118 14 The rise time of the level detector’s outputs is not significant since it is so short. The decay time, however, provides a significant part of the “hold time” necessary to hold the circuit during the normal pauses in speech. The components at the inputs of the level detectors (RLI1, RLI2, TLI1, TLI2) do not affect the switching time, but rather affect the relative signal levels required to switch the circuit, as well as the frequency response of the detectors. DESIGN EQUATIONS Referring to Figure 24 (the coupling capacitors have been omitted for simplicity), and the circuit of Figure 23, the following definitions will be used (all measurements are at 1.0 kHz): — GMA is the gain of the microphone amplifier measured from the microphone output to TXI (typically 35 V/V, or 31 dB); — GTX is the gain of the transmit attenuator, measured from TXI to TXO; — GHA is the gain of hybrid amplifiers, measured from TXO to the HTO--/HTO+ differential output (typically 10.2 V/V, or 20.1 dB); — GHT is the gain from HTO--/HTO+ to Tip/Ring for transmit signals, and includes the balance network (measured at 0.4 V/V, or -- 8.0 dB); — GST is the sidetone gain, measured from HTO--/HTO+ to the filter input (measured at 0.18 V/V, or --15 dB); — GHR is the gain from Tip/Ring to the filter input for receive signals (measured at 0.833 V/V or -- 1.6 dB); — GFO is the gain of the filter stage, measured from the input of the filter to RXI, typically 0 dB at 1.0 kHz; — GRX is the gain of the receive attenuator measured from RXI to RXO; — GSA is the gain of the speaker amplifier, measured from RXO to the differential output of the MC34119 (typically 22 V/V or 26.8 dB); — GAC is the acoustic coupling, measured from the speaker differential voltage to the microphone output voltage. I) Transmit Gain The transmit gain, from the microphone output (VM) to Tip and Ring, is determined by the output characteristics of the microphone, and the desired transmit level. For example, a typical electret microphone will produce ≈0.35 mVrms under normal speech conditions. To achieve 100 mVrms at Tip/ Ring, an overall gain of 285 V/V is necessary. The gain of the transmit attenuator is fixed at 2.0 (+ 6.0 dB), and the gain through the hybrid of Figure 23 (G HT ) is nominally 0.4 (-- 8.0 dB). Therefore a gain of 357 V/V is required of the microphone and hybrid amplifiers. It is desirable to have the majority of that gain in the microphone amplifier for three reasons: 1) the low level signals from the microphone should be MOTOROLA MOTOROLA 25 Ω SPEAKER (300 mV) 8 7 -+ 4 RLI2 20 R 200 pF 5.1 k -+ 6 110 k 15 VB 14 1 3 V B 10 k 0.05 0.1 22 RXO VOLUME CONTROL 9.1 k 0.1 0.05 26 RLI1 TLI1 23 VB 51 k -+ 10 k BNM R -+ R FO 1 56 k +1 FILTER MC34118 VB 0.1 RHF HTI 7 6 HTO-- 0.1 5.1 k FI 2 VCC 0.01 4700 pF 4700 pF VB 2.0 µF 47 µF 100 k 2.0 µF 0.05 220 k GND 28 CPR 27 RLO1 25 TLO1 24 HTO+ 5 VCC 4 CD 3 820 300 k 1. These two resistors depend on the specific microphone selected. 2. Component values shown are recommendations only, and will vary in different applications. 3. This capacitor must be physically adjacent to Pin 4 of the MC34118. 4. BNM = Background Noise Monitor; DTD = Dial Tone Detector. 5. Recommended transformer: Stancor TTPC--13, Microtran T5115. 6. Capacitor values are in µF except where noted. 20 k 21 RXI VB + -- + -- VB TXO 8 RHI 10 k VB Rx ATTENUATOR DTD -+ ATTENUATOR CONTROL AGC VCC Tx ATTENUATOR VLC 13 VB -- CT 19 2.0 µF CPT 16 RLO2 18 VB -- MUT TLO2 0.1 5.1 k TXI TLI2 17 9 VB BNM MCO 10 0.1 12 220 µF 120 k MC34119 R 5.0 µF 47 µF 100 k 2.0 µF MUTE VB VCC -+ 180 k R MF 270 pF MCI 11 RMI 5.1 k VB 0.02 0.2 20 µF + 5 VCC 620 MIKE (NOTE 1) 1.0 k VCC + Figure 23. MC34118 Application Circuit (Basic Line Powered Speakerphone MC34118 15 DISABLE 1000 µF (NOTE 3) 1N4733 5.1 V HOOK SWITCH RING TIP VM MIKE AMP V 1 I1 TXI TXO -+ C2 + -- RLI2 I3 SPEAKER AMP. HYBRID TIP RING RLI1 GHR R3 V3 GST + -- C1 GHT TLI1 -+ CONTROL (GAC) I2 R2 R1 TLI2 ACOUSTIC COUPLING HYBRID AMP V 2 Tx ATTENUATOR R4 RXO RXI I4 V4 FILTER Rx ATTENUATOR Figure 24. Basic Block Diagram for Design Purposes amplified as soon as possible to minimize signal/noise problems; 2) to provide a reasonable signal level to the TLI2 level detector; and 3) to minimize any gain applied to broadband noise generated within the attenuator. However, to cover the normal voiceband, the microphone amplifier’s gain should not exceed 48 dB (see Figure 16). For the circuit of Figure 23, the gain of the microphone amplifier was set at 35 V/V (31 dB), and the differential gain of the hybrid amplifiers was set at 10.2 V/V (20.1 dB). II) Receive Gain The overall receive gain depends on the incoming signal level, and the desired output power at the speaker. Nominal receive levels (independent of the peaks) at Tip/Ring can be 35 mVrms (-- 27 dBm), although on long lines that level can be down to 8.0 mVrms (-- 40 dBm). The speaker power is: dBm∕10 x 0.6 (Equation 1) R S where RS is the speaker impedance, and the dBm term is the incoming signal level increased by the gain of the receive path. Experience has shown that ≈30 dB gain is a satisfactory amount for the majority of applications. Using the above numbers and Equation 1, it would appear that the resulting power to the speaker is extremely low. However, Equation 1 does not consider the peaks in normal speech, which can be 10 to 15 times the rms value. Considering the peaks, the overall average power approaches 20 -- 30 mW on long lines, and much more on short lines. Referring to Figure 23, the gain from Tip/Ring to the filter input was measured at 0.833 V/V (-- 1.6 dB), the filter’s gain P SPK = 10 MC34118 16 is unity, and the receive attenuator ’s gain is 2.0 V/V (+ 6.0 dB) at maximum volume. The speaker amplifier’s gain is set at 22 V/V (26.8 dB), which puts the overall gain at ≈31.2 dB. III) Loop Gain The total loop gain (of Figure 24) must add up to less than zero dB to obtain a stable circuit. This can be expressed as: GMA + GTX + GHA + GST + GFO + GRX + GSA + GAC < 0 (Equation 2) Using the typical numbers mentioned above, and knowing that GTX + GRX = -- 40 dB, the required acoustic coupling can be determined: GAC < -- [31 + 20.1 + (-- 15) + 0 + (-- 40) + 26.8] = -- 22.9 dB. (Equation 3) An acoustic loss of at least 23 dB is necessary to prevent instability and oscillations, commonly referred to as “singing.” However, the following equations show that greater acoustic loss is necessary to obtain proper level detection and switching. IV) Switching Thresholds To switch comparator C1, currents I1 and I3 need to be determined. Referring to Figure 24, with a receive signal VL applied to Tip/Ring, a current I3 will flow through R3 into RLI2 according to the following equation: V I3 = L R3  G HR xG FO xG RX x G  SA 2 (Equation 4) MOTOROLA where the terms in the brackets are the V/V gain terms. The speaker amplifier gain is divided by two since GSA is the differential gain of the amplifier, and V3 is obtained from one side of that output. The current I1, coming from the microphone circuit, is defined by: Equations 7, 11, and 12 define the thresholds for switching, and are represented in the following graph: VM MRX V xG MA (Equation 5) I1 = M R1 where VM is the microphone voltage. Since the switching threshold occurs when I1 = I3, combining the above two equations yields: V = V x R1 M L R3 MI GHRx GFOx GRXx GSA (Equation 6) x2 MA This is the general equation defining the microphone voltage necessary to switch comparator C1 when a receive signal VL is present. The highest VM occurs when the receive attenuator is at maximum gain (+ 6.0 dB). Using the typical numbers for Equation 6 yields: G VM = 0.52 VL  G MA xG TX x G  HA 2 (Equation 8) Since GHA is the differential gain of the hybrid amplifiers, it is divided by two to obtain the voltage V 2 applied to R2. Comparator C2 switches when I4 = I2. I4 is defined by: V xG I4 = L G HR FO R4   (Equation 9) Setting I4 = I2, and combining the above equations results in: V = V x R4 L M R2 GMA x GTX x GHA GHR x GFO x 2 (Equation 10) This equation defines the line voltage at Tip/Ring necessary to switch comparator C2 in the presence of a microphone voltage. The highest VL occurs when the circuit is in the transmit mode (GTX = + 6.0 dB). Using the typical numbers for Equation 10 yields: VL = 840 VM (or VM = 0.0019 VL) (Equation 11) At idle, where the gain of the two attenuators is -- 20 dB (0.1 V/V), Equations 6 and 10 yield the same result: VM = 0.024 VL MOTOROLA VL Figure 25. Switching Thresholds (Equation 7) To switch comparator C2, currents I2 and I4 need to be determined. With sound applied to the microphone, a voltage VM is created by the microphone, resulting in a current I2 into TLI1: V I2 = M R2 MTX (Equation 12) The “M” terms are the slopes of the lines (0.52, 0.024, and 0.0019) which are the coefficients of the three equations. The MRX line represents the receive to transmit threshold in that it defines the microphone signal level necessary to switch to transmit in the presence of a given receive signal level. The MTX line represents the transmit to receive threshold. The MI line represents the idle condition, and defines the threshold level on one side (transmit or receive) necessary to overcome noise on the other. Some comments on the above graph: — Acoustic coupling and sidetone coupling were not included in Equations 7 and 12. Those couplings will affect the actual performance of the final speakerphone due to their interaction with speech at the microphone, and the receive signal coming in at Tip/Ring. The effects of those couplings are difficult to predict due to their associated phase shifts and frequency response. In some cases the coupling signal will add, and other times subtract from the incoming signal. The physical design of the speakerphone enclosure, as well as the specific phone line to which it is connected, will affect the acoustic and sidetone couplings, respectively. — The MRX line helps define the maximum acoustic coupling allowed in a system, which can be found from the following equation: G = R1 2 x R3 x G (Equation 13) MA Equation 13 is independent of the volume control setting. Conversely, the acoustic coupling of a designed system helps determine the minimum slope of that line. Using the component values of Figure 23 in Equation 13 yields a AC–MAX MC34118 17 GAC--MAX of -- 37 dB. Experience has shown, however, that an acoustic coupling loss of > 40 dB is desirable. — The MTX line helps define the maximum sidetone coupling (GST) allowed in the system, which can be found from the following equation: G = R4 2 x R2 x G (Equation 14) FO Using the component values of Figure 23 in Equation 14 yields a maximum sidetone of 0 dB. Experience has shown, however, that a minimum of 6.0 dB loss is preferable. The above equations can be used to determine the resistor values for the level detector inputs. Equation 6 can be used to determine the R1/R3 ratio, and Equation 10 can be used to determine the R4/R2 ratio. In Figure 24, R1 -- R4 each represent the combined impedance of the resistor and coupling capacitor at each level detector input. The magnitude of each RC’s impedance should be kept within the range of 2.0 k -- 15 kΩ in the voiceband (due to the typical signal levels present) to obtain the best performance from the level detectors. The specific R and C at each location will determine the frequency response of that level detector. ST APPLICATION INFORMATION DIAL TONE DETECTOR The threshold for the dial tone detector is internally set at 15 mV (10 mVrms) below VB (see Figure 5). That threshold can be reduced by connecting a resistor from RXI to ground. The resistor value is calculated from: R = 10 k  V  B –1 ∆V where VB is the voltage at Pin 15, and ∆V is the amount of threshold reduction. By connecting a resistor from VCC to RXI, the threshold can be increased. The resistor value is calculated from: R = 10 k  V trol of the circuit from idle will be easier for that side towards which it is biased since that path will have less attenuation at idle. By connecting a resistor from CT (Pin 14) to ground, the circuit will be biased towards the transmit side. The resistor value is calculated from: R=R   V B –1 T ∆V where R is the added resistor, RT is the resistor normally between Pins 14 and 15 (typically 120 kΩ), and ∆V is the difference between VB and the voltage at CT at idle (refer to Figure 10). By connecting a resistor from CT (Pin 14) to VCC, the circuit will be biased towards the receive side. The resistor value is calculated from: R=R  V T –V B –1 CC ∆V  R, RT, and ∆V are the same as above. Switching time will be somewhat affected in each case due to the different voltage excursions required to get to transmit and receive from idle. For practical considerations, the ∆V shift should not exceed 100 mV. VOLUME CONTROL If a potentiometer with a standard linear taper is used for the volume control, the graph of Figure 14 indicates that the receive gain will not vary in a linear manner with respect to the pot’s position. In situations where this may be objectionable, a potentiometer with an audio taper (commonly used in radio volume controls) will provide a more linear relationship as indicated in Figure 26. The slight non--linearity at each end of the graph is due to the physical construction of the potentiometer, and will vary among different manufacturers. +10  –V B –1 CC ∆V 0 BACKGROUND NOISE MONITORS For testing or circuit analysis purposes, the transmit or receive attenuators can be set to the “on” position, by disabling the background noise monitors, and applying a signal so as to activate the level detectors. Grounding the CPR pin will disable the receive background noise monitor, thereby indicating the “presence of speech” to the attenuator control block. Grounding CPT does the same for the transmit path. Additionally, the receive background noise monitor is automatically disabled by the dial tone detector whenever the receive signal exceeds the detector’s threshold. TRANSMIT/RECEIVE DETECTION PRIORITY Although the MC34118 was designed to have an idle mode such that the attenuators are halfway between their full on and full off positions, the idle mode can be biased towards the transmit or the receive side. With this done, gaining con- MC34118 18 GAIN (dB) where ∆V is the amount of the threshold increase. --10 --20 --30 -- 40 0 40 80 120 160 200 240 280 320 DEGREES OF ROTATION Figure 26. Receive Attenuator Gain versus Potentiometer Position Using Audio Taper APPLICATION CIRCUIT The circuit of Figure 23 is a basic speakerphone, to be used in parallel with any other telephone which contains the MOTOROLA ringer, dialer, and handset functions. The circuit is powered entirely by the telephone line’s loop current, and its characteristics are shown in Figures 27 -- 30. 28 26 GAIN (dB) 24 VOLTAGE AT TIP/RING (VOLTS) 20 16 24 22 600 T VL R FIGURE 23 20 12 GAIN = VSPKR/VL 8.0 100 VSPKR 500 1.0K FREQUENCY (Hz) 4.0 5.0K Figure 30. Receive Gain 0 20 40 60 80 100 120 LOOP CURRENT (mA) ZAC (OHMS) Figure 27. DC V--1 Characteristics Figure 31 shows how the same circuit can be configured to be powered from a 3.5 -- 6.0 volt power supply rather than the phone line. 1000 TO HTO-- 800 TO FILTER 600 TO HTO+ TO VCC 400 600 FIGURE 23 Zac R 200 0 T 300 820 1.0K FREQUENCY (Hz) TIP RING 0.01 100 µF (NOTE 1) TO CD TO CD (MC34119) 100 HOOK SWITCH 0.05 + -- FROM POWER SUPPLY DISABLE 1. This capacitor must be physically adjacent to Pin 4 of the MC34118. 5.0K Figure 31. Operating from a Power Supply Figure 28. AC Termination Impedance ADDING A DIALER 50 GAIN (dB) 48 46 44 T 0.2 5.1 k VM 40 MCI FIGURE 23 GAIN = VLINE/VM 100 1.0K 600 VLINE R 5.0K FREQUENCY (Hz) Figure 29. Transmit Gain — Microphone to Tip/Ring MOTOROLA Figure 32 shows the addition of a dialer to the circuit of Figure 23, with the additional components shown in bold. The MC145412 pulse/tone dialer is shown configured for DTMF operation. The DTMF output (Pin 18) is fed to the hybrid amplifiers at HTI, and the DTMF levels at Tip/Ring are adjusted by varying the 39 kΩ resistor. The Mute Output (active low at Pin 11) mutes the microphone amplifier, and attenuates the DTMF signals in the receive path (by means of the 10 k/3.0 k divider). The MC34118 is forced into the fast idle mode during dialing. The 3.0 volt battery provides for memory retention of the dialer’s 10 number storage when the circuit is unpowered. RFI INTERFERENCE Potential radio frequency interference problems should be addressed early in the electrical and mechanical design of the speakerphone. RFI may enter the circuit through Tip and MC34118 19 Ring, through the microphone wiring to the microphone amplifier, or through any of the PC board traces. The most sensitive pins on the MC34118 are the inputs to the level detectors (RLI1, RLI2, TLI1, TLI2) since, when there is no speech present, the inputs are high impedance and these op amps are in a near open loop condition. The board traces to these pins should be kept short, and the resistor and capacitor for each of these pins should be physically close to the pins. Any other high impedance input pin (MCI, HTI, FI, VLC) should be considered sensitive to RFI signals. IN THE FINAL ANALYSIS . . . Proper operation of a speakerphone is a combination of proper mechanical (acoustic) design as well as proper electronic design. The acoustics of the enclosure must be considered early in the design of a speakerphone. In general, electronics cannot compensate for poor acoustics, low speaker quality, or any combination of the two. Proper acoustic separation of the speaker and microphone, as described in the Design Equations, is essential. The physical location of the microphone, along with the characteristics of the selected microphone, will play a large role in the quality of the transmitted sound. The microphone and speaker vendors can usually provide additional information on the use of their products. In the final analysis, the circuits shown in this data sheet will have to be “fine tuned” to match the acoustics of the enclosure, the specific hybrid, and the specific microphone and speaker selected. The component values shown in this data sheet should be considered as starting points only. The gains of the transmit and receive paths are easily adjusted at the microphone and speaker amplifiers, respectively. The switching response can then be fine tuned by varying (in small steps) the components at the level detector inputs until satisfactory operation is obtained for both long and short lines. SUGGESTED VENDORS Microphones Primo Microphones Inc. Bensenville, IL 60106 312--595--1022 Model EM--60 Hosiden America Corp. Elk Grove Village, IL 60007 312--981--1144 Model KUC2123 25 Ω Speakers Panasonic Industrial Co. Seacaucus, N.J. 07094 201--348--5233 Model EAS--45P19S Telecom Transformers Microtran Co., Inc. Valley Stream, N.Y. 11528 516--561--6050 Various models — ask for catalog and Application Bulletin F232 39 k 18 16 Motorola Inc. does not endorse or warrant the suppliers referenced. 15 300 13 2 5 9 11 10 12 6 4 3 2 3 A 4 5 6 7 8 0 9 C # D * B 0.05 820 HTO-- R VB HTO+ -+ 5 FILTER +1 VCC MUT 12 TO TIP & RING 0.01 R 3.58 MHz MUTE 1 -+ MC34118 8 MC145412 6 4 0.1 14 10 k HTI 7 VB VCC 1 Onan Power/Electronics Minneapolis, MN 55437 612--921--5600 Model TC 38--6 0.1 8 TXO 0.1 2.0 k Stancor Products Logansport, IN 46947 219--722--2244 Various models — ask for catalog PREM Magnetics, Inc. McHenry, IL 60050 815--385--2700 Various models — ask for catalog 51 k 3.0 V MURA Corp. Westbury, N.Y. 11590 516--935--3640 Model EC--983--7 4700 pF 2 FI 10 k 3.0 k FO 1 56 k 27 k 240 k 3300 180 k 2N2222A OR EQUIVALENT Figure 32. Adding a Dialer to the Speakerphone MC34118 20 MOTOROLA PACKAGE DIMENSIONS P SUFFIX PLASTIC PACKAGE CASE 710--02 ISSUE B 28 PDIP 28 NOTES: 1. POSITIONAL TOLERANCE OF LEADS (D), SHALL BE WITHIN 0.25 (0.010) AT MAXIMUM MATERIAL CONDITION, IN RELATION TO SEATING PLANE AND EACH OTHER. 2. DIMENSION L TO CENTER OF LEADS WHEN FORMED PARALLEL. 3. DIMENSION B DOES NOT INCLUDE MOLD FLASH. 15 B DIM A B C D F G H J K L M N 14 1 L C A N G H F M K D J SEATING PLANE MILLIMETERS MIN MAX 36.45 37.21 13.72 14.22 3.94 5.08 0.36 0.56 1.02 1.52 2.54 BSC 1.65 2.16 0.20 0.38 2.92 3.43 15.24 BSC 0_ 15_ 0.51 1.02 INCHES MIN MAX 1.435 1.465 0.540 0.560 0.155 0.200 0.014 0.022 0.040 0.060 0.100 BSC 0.065 0.085 0.008 0.015 0.115 0.135 0.600 BSC 0_ 15_ 0.020 0.040 EG (Pb--free) SUFFIX DW SUFFIX PLASTIC PACKAGE CASE 751F--05 ISSUE F 28 SOICW D A NOTES: 1. DIMENSIONS ARE IN MILLIMETERS. 2. INTERPRET DIMENSIONS AND TOLERANCES PER ASME Y14.5M, 1994. 3. DIMENSIONS D AND E DO NOT INCLUDE MOLD PROTRUSIONS. 4. MAXIMUM MOLD PROTRUSION 0.015 PER SIDE. 5. DIMENSION B DOES NOT INCLUDE DAMBAR PROTRUSION. ALLOWABLE DAMBAR PROTRUSION SHALL BE 0.13 TOTAL IN EXCESS OF B DIMENSION AT MAXIMUM MATERIAL CONDITION. 15 M 0.25 E H B M 28 1 14 PIN 1 IDENT A1 A B e B 0.025 MOTOROLA M C A S B C S L 0.10 SEATING PLANE C θ DIM A A1 B C D E e H L θ MILLIMETERS MIN MAX 2.35 2.65 0.13 0.29 0.35 0.49 0.23 0.32 17.80 18.05 7.40 7.60 1.27 BSC 10.05 10.55 0.41 0.90 0_ 8_ MC34118 21 Motorola reserves the right to make changes without further notice to any products herein. Motorola makes no warranty, representation or guarantee regarding the suitability of its products for any particular purpose, nor does Motorola assume any liability arising out of the application or use of any product or circuit, and specifically disclaims any and all liability, including without limitation consequential or incidental damages. “Typical” parameters can and do vary in different applications. All operating parameters, including “Typicals” must be validated for each customer application by customer’s technical experts. Motorola does not convey any license under its patent rights nor the rights of others. Motorola products are not designed, intended, or authorized for use as components in systems intended for surgical implant into the body, or other applications intended to support or sustain life, or for any other application in which the failure of the Motorola product could create a situation where personal injury or death may occur. Should Buyer purchase or use Motorola products for any such unintended or unauthorized application, Buyer shall indemnify and hold Motorola and its officers, employees, subsidiaries, affiliates, and distributors harmless against all claims, costs, damages, and expenses, and reasonable attorney fees arising out of, directly or indirectly, any claim of personal injury or death associated with such unintended or unauthorized use, even if such claim alleges that Motorola was negligent regarding the design or manufacture of the part. Motorola and are registered trademarks of Motorola, Inc. Motorola, Inc. is an Equal Opportunity/Affirmative Action Employer. How to reach us: USA/EUROPE: Motorola Literature Distribution; P.O. Box 20912; Phoenix, Arizona 85036. 1--800--441--2447 JAPAN: Nippon Motorola Ltd.; Tatsumi--SPD--JLDC, Toshikatsu Otsuki, 6F Seibu--Butsuryu--Center, 3--14--2 Tatsumi Koto--Ku, Tokyo 135, Japan. 03--3521--8315 MFAX: RMFAX0@email.sps.mot.com -- TOUCHTONE (602) 244--6609 INTERNET: http://Design--NET.com HONG KONG: Motorola Semiconductors H.K. Ltd.; 8B Tai Ping Industrial Park, 51 Ting Kok Road, Tai Po, N.T., Hong Kong. 852--26629298 MC34118 22 ◊ MC34118/D MOTOROLA *MC34118/D*