CS5161

CS5161, CS5161H CPU 5−Bit Synchronous Buck Controller The CS5161/5161H are 5−bit synchronous dual N−Channel buck controllers designed to provide unprecedented transient response for today’s demanding high−density, high−speed logic. They operate using a proprietary control method which allows a 100 ns response time to load transients. The CS5161 is designed to operate over a 9−16 V range (VCC) using 12 V to power the IC and 5.0 V as the main supply for conversion. The CS5161H operates from a 12 V input as the main supply for conversion using a discrete charge pump circuit to provide up to 20 V for VCC2 and high side gate drive. The CS5161/5161H are specifically designed to power Pentium® III processors and other high performance core logic. They include the following features: on board 5−bit DAC, short circuit protection, 1.0% output tolerance, VCC monitor, and programmable Soft Start capability. The CS5161/5161H are available in 16 pin surface mount packages. Features • Dual N−Channel Design • Excess of 1.0 MHz Operation • 100 ns Transient Response • 5−Bit DAC • Backward Compatible with CS515X Family • 30 ns Gate Rise/Fall Times • 1.0% DAC Accuracy • 5.0 V & 12 V Operation • Remote Sense • Programmable Soft Start • Lossless Short Circuit Protection • VCC Monitor • 50 ns FET Nonoverlap Time • V2™ Control Topology • Current Sharing • Overvoltage Protection http://onsemi.com MARKING DIAGRAMS 16 SO−16 D SUFFIX CASE 751B 1 A WL, L YY, Y WW, W = Assembly Location = Wafer Lot = Year = Work Week CS5161 AWLYWW PIN CONNECTIONS VID0 VID1 VID2 VID3 SS VID4 COFF VFFB 1 VFB COMP LGnd VCC1 VGATE(L) PGnd VGATE(H) VCC2 ORDERING INFORMATION Device CS5161GD16 CS5161GDR16 CS5161HGD16 CS5161HGDR16 Package SO−16 SO−16 SO−16 SO−16 Shipping 48 Units/Rail 2500 Tape & Reel 48 Units/Rail 2500 Tape & Reel © Semiconductor Components Industries, LLC, 2006 July, 2006 − Rev. 12 1 Publication Order Number: CS5161/D CS5161, CS5161H 5V 12 V 1.0 μH 0.1 μF 1200 μF/10 V × 6.0 Sanyo GX SILICONIX SUD50NO3−10P 1.8 μH VOUT VCC1 VCC2 VGATE(H) VID0 VID1 VID2 VID3 VID4 VID0 VID1 VID2 VID3 VID4 COFF VGATE(L) MBRS140T3 PGnd SILICONIX SUD50NO3−07 CS5161 680 pF SS 0.1 μF COMP LGnd 0.33 μF 1.3 k VFB 1200 μF/10 V × 8.0 Sanyo GX VFFB Figure 1. Application Diagram,5.0 V to 1.5 V/15 A Core Logic Converter with 12 V Bias MAXIMUM RATINGS* Rating Operating Junction Temperature, TJ Lead Temperature Soldering: Storage Temperature Range, TS ESD Susceptibility (Human Body Model) 1. 60 second maximum above 183°C. *The maximum package power dissipation must be observed. Reflow: (SMD styles only) (Note 1) Value 0 to 150 230 peak −65 to +150 2.0 Unit °C °C °C kV MAXIMUM RATINGS Pin Name VCC1 VCC2 (CS5161) VCC2 (CS5161H) SS Max Operating Voltage 16 V/−0.3 V 18 V/−0.3 V 20 V/−0.3 V 6.0 V/−0.3 V Max Current 100 mA DC/1.5 A peak 100 mA DC/1.5 A peak 100 mA DC/1.5 A peak −100 μA http://onsemi.com 2 CS5161, CS5161H MAXIMUM RATINGS (continued) Pin Name COMP VFB COFF VFFB VID0 − VID4 VGATE(H) (CS5161) VGATE(H) (CS5161H) VGATE(L) LGnd PGnd Max Operating Voltage 6.0 V/−0.3 V 6.0 V/−0.3 V 6.0 V/−0.3 V 6.0 V/−0.3 V 6.0 V/−0.3 V 18 V/−0.3 V 20 V/−0.3 V 16 V/−0.3 V 0V 0V Max Current 200 μA −0.2 μA −0.2 μA −0.2 μA −50 μA 100 mA DC/1.5 A peak 100 mA DC/1.5 A peak 100 mA DC/1.5 A peak 25 mA 100 mA DC/1.5 A peak ELECTRICAL CHARACTERISTICS (0°C < TA < +70°C; CS5161: 0°C < TJ < +85°C; CS5161H: 0°C < TJ < +125°C; 9.5 V < VCC1 < 14 V; CS5161: 5.0 V < VCC2 < 16 V; CS5161H: 5.0 V < VCC2 < 20 V; DAC Code: VID4 = VID2 = VID1 = VID0 =1; VID3 = 0; CVGATE(L) and CVGATE(H) = 1.0 nF; COFF = 330 pF; CSS = 0.1 μF, unless otherwise specified.) Characteristic Error Amplifier VFB Bias Current Open Loop Gain Unity Gain Bandwidth COMP SINK Current COMP SOURCE Current COMP CLAMP Current COMP High Voltage COMP Low Voltage PSRR Transconductance VCC1 Monitor Start Threshold Stop Threshold Hysteresis Soft Start (SS) Charge Time Pulse Period Duty Cycle COMP Clamp Voltage VFFB SS Fault Disable High Threshold 2. Guaranteed by design, not 100% tested in production. − − (Charge Time /Pulse Period) × 100 VFB = 0 V; VSS = 0 VGATE(H) = Low; VGATE(L) = Low − 1.6 25 1.0 0.50 0.9 − 3.3 100 3.3 0.95 1.0 2.5 5.0 200 6.0 1.10 1.1 3.0 ms ms % V V V Output switching Output not switching Start−Stop 8.70 8.55 − 9.05 8.90 150 9.40 9.25 − V V mV VFB = 0 V 1.25 V < VCOMP , 4.0 V; CCOMP = 0.1 μF; Note 2 CCOMP = 0.1 μF; Note 2 VCOMP = 1.5 V; VFB = 3.0 V; VSS > 2.0 V VCOMP = 1.2 V; VFB = 2.7 V; VSS = 5.0 V VCOMP = 0 V; VFB = 2.7 V VFB = 2.7 V; VSS = 5.0 V VFB = 3.0 V 8.0 V < VCC1 < 14 V @ 1.0 kHz; CCOMP = 0.1 μF; Note 2 − − − − 30 15 0.4 4.0 − − − 0.3 80 50 60 30 1.0 4.3 1.0 70 33 1.0 − − 120 60 1.6 5.0 1.15 − − μA dB kHz μA μA mA V V dB mmho Test Conditions Min Typ Max Unit http://onsemi.com 3 CS5161, CS5161H ELECTRICAL CHARACTERISTICS (continued) (0°C < TA < +70°C; CS5161: 0°C < TJ < +85°C; CS5161H: 0°C < TJ < +125°C; 9.5 V < VCC1 < 14 V; CS5161: 5.0 V < VCC2 < 16 V; CS5161H: 5.0 V < VCC2 < 20 V; DAC Code: VID4 = VID2 = VID1 = VID0 =1; VID3 = 0; CVGATE(L) and CVGATE(H) = 1.0 nF; COFF = 330 pF; CSS = 0.1 μF, unless otherwise specified.) Characteristic PWM Comparator Transient Response VFFB Bias Current DAC Input Threshold Input Pull Up Resistance Pull Up Voltage Accuracy (all codes except 11111, 10110, 10101, 10100, 10011, 10010, 10001, 10000) VID4 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 1 1 1 1 1 1 1 1 1 1 1 VID3 1 1 1 1 1 1 1 1 0 0 0 0 0 0 0 0 1 1 1 1 1 1 1 1 0 0 0 VID2 1 1 1 1 0 0 0 0 1 1 1 1 0 0 0 0 1 1 1 1 0 0 0 0 1 1 1 VID1 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 VID0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 − − − − − − − − − − − − − − − − − − − − − − − − − − − 1.2870 1.3365 1.3860 1.4355 1.4850 1.5345 1.5840 1.6335 1.6830 1.7325 1.7820 1.8315 1.8810 1.9305 1.9800 2.0295 1.2191 2.0790 2.1780 2.2770 2.3760 2.4750 2.5740 2.6730 2.7720 2.8420 2.9400 1.3000 1.3500 1.4000 1.4500 1.5000 1.5500 1.6000 1.6500 1.7000 1.7500 1.8000 1.8500 1.9000 1.9500 2.0000 2.0500 1.2440 2.1000 2.2000 2.3000 2.4000 2.5000 2.6000 2.7000 2.8000 2.9000 3.0000 1.3130 1.3635 1.4140 1.4645 1.5150 1.5655 1.6160 1.6665 1.7170 1.7675 1.8180 1.8685 1.9190 1.9695 2.0200 2.0705 1.2689 2.1210 2.2220 2.3230 2.4240 2.5250 2.6260 2.7270 2.8280 2.9580 3.0600 V V V V V V V V V V V V V V V V V V V V V V V V V V V VID0, VID1, VID2, VID3, VID4 VID0, VID1, VID2, VID3, VID4 − Measure VFB = COMP, CS5161: 25°C ≤ TJ ≤ 85°C CS5161H: 25°C ≤ TJ ≤ 125°C 1.00 25 4.85 − 1.25 50 5.00 − 2.40 110 5.15 ± 1.0 V kΩ V % VFFB = 0 to 5.0 V to VGATE(H) = 9.0 V to 1.0 V; VCC1 = VCC2 = 12 V VFFB = 0 V − − 100 0.3 125 − ns μA Test Conditions Min Typ Max Unit http://onsemi.com 4 CS5161, CS5161H ELECTRICAL CHARACTERISTICS (continued) (0°C < TA < +70°C; CS5161: 0°C < TJ < +85°C; CS5161H: 0°C < TJ < +125°C; 9.5 V < VCC1 < 14 V; CS5161: 5.0 V < VCC2 < 16 V; CS5161H: 5.0 V < VCC2 < 20 V; DAC Code: VID4 = VID2 = VID1 = VID0 =1; VID3 = 0; CVGATE(L) and CVGATE(H) = 1.0 nF; COFF = 330 pF; CSS = 0.1 μF, unless otherwise specified.) Characteristic DAC 1 1 1 1 1 0 0 0 0 0 1 0 0 0 0 0 1 1 0 0 0 1 0 1 0 − − − − − 3.0380 3.1360 3.2340 3.3320 3.4300 3.1000 3.2000 3.3000 3.4000 3.5000 3.1620 3.2640 3.3660 3.4680 3.5700 V V V V V Test Conditions Min Typ Max Unit VGATE(H) and VGATE(L) Out SOURCE Sat at 100 mA Out SINK Sat at 100 mA Out Rise Time Out Fall Time Delay VGATE(H) to VGATE(L) Delay VGATE(L) to VGATE(H) VGATE(H), VGATE(L) Resistance VGATE(H), VGATE(L) Schottky Supply Current ICC1 No Switching ICC2 No Switching Operating ICC1 Operating ICC2 COFF Normal Charge Time Discharge Current Time Out Timer Time Out Time Fault Mode Duty Cycle VFB = VCOMP; VFFB = 2.0 V; Record VGATE(H) Pulse High Duration VFFB = 0V 10 35 30 50 65 70 μs % VFFB = 1.5 V; VSS = 5.0 V COFF to 5.0 V; VFB > 1.0 V 1.0 5.0 1.6 − 2.2 − μs mA VFB = COMP = VFFB VFB = COMP = VFFB − − − − − − 9.5 2.0 9.0 2.5 14.5 3.5 14 5.5 mA mA mA mA Measure VCC1 − VGATE(L); VCC2 − VGATE(H) Measure VGATE(H) − VPGnd; VGATE(L) − VPGnd 1.0 V < VGATE(H) < 9.0 V; 1.0 V < VGATE(L) < 9.0 V; VCC1 = VCC2 = 12 V 9.0 V < VGATE(H) > 1.0 V; 9.0 V > VGATE(L) > 1.0 V; VCC1 = VCC2 = 12 V VGATE(H) falling to 2.0 V; VCC1 = VCC2 = 8.0 V VGATE(L) rising to 2.0 V VGATE(L) falling to 2.0 V; VCC1 = VCC2 = 8.0 V VGATE(H) rising to 2.0 V Resistor to LGnd. Note 3 LGnd to VGATE(H) @ 10 mA; LGnd to VGATE(L) @ 10 mA − − − − 20 20 20 − 1.2 1.0 30 30 50 50 50 600 2.0 1.5 50 50 90 90 100 800 V V ns ns ns ns kΩ mV 3. Guaranteed by design, not 100% tested in production. http://onsemi.com 5 CS5161, CS5161H PACKAGE PIN DESCRIPTION PACKAGE PIN # 1, 2, 3, 4, 6 PIN SYMBOL VID0−VID4 FUNCTION Voltage ID DAC input pins. These pins are internally pulled up to 5.0 V providing logic ones if left open. VID4 selects the DAC range. When VID4 is High (logic one), the DAC range is 2.10 V to 3.50 V with 100 mV increments. When VID4 is Low (logic zero), the DAC range is 1.30 V to 2.05 V with 50 mV increments. VID0 − VID4 select the desired DAC output voltage. Leaving all 5 DAC input pins open results in a DAC output voltage of 1.2440 V, allowing for adjustable output voltage, using a traditional resistor divider. Soft Start Pin. A capacitor from this pin to LGnd in conjunction with internal 60 μA current source provides Soft Start function for the controller. This pin disables fault detect function during Soft Start. When a fault is detected, the Soft Start capacitor is slowly discharged by internal 2.0 μA current source setting the time out before trying to restart the IC. Charge/discharge current ratio of 30 sets the duty cycle for the IC when the regulator output is shorted. A capacitor from this pin to ground sets the time duration for the on board one shot, which is used for the constant off time architecture. Fast feedback connection to the PWM comparator. This pin is connected to the regulator output. The inner feedback loop terminates on time. Boosted power for the high side gate driver. High FET driver pin capable of 1.5 A peak switching current. Internal circuit prevents VGATE(H) and VGATE(L) from being in high state simultaneously. High current ground for the IC. The MOSFET drivers are referenced to this pin. Input capacitor ground and the source of lower FET should be tied to this pin. Low FET driver pin capable of 1.5 A peak switching current. Input power for the IC and low side gate driver. Signal ground for the IC. All control circuits are referenced to this pin. Error amplifier compensation pin. A capacitor to ground should be provided externally to compensate the amplifier. Error amplifier DC feedback input. This is the master voltage feedback which sets the output voltage. This pin can be connected directly to the output or a remote sense trace. 5 SS 7 COFF 8 VFFB 9 10 VCC2 VGATE(H) 11 PGnd 12 13 14 15 16 VGATE(L) VCC1 LGnd COMP VFB http://onsemi.com 6 CS5161, CS5161H VCC1 Monitor − Comparator + 9.05 V 8.90V SS 2.0 μA VID0 VID1 VID2 VID3 VID4 5 BIT DAC Error Amplifier + − PWM Comparator − VFB COMP VFFB Fast Feedback − + LGnd 1.0 V VFFB Low Comparator Time−Out Timer (30 μs) Edge Triggered Slow Feedback + Maximum On−Time Timeout Normal Off−Time Timeout Extended Off−Time Timeout R S Q Q PMW Latch Off−Time Timeout GATE(H) = ON GATE(H) = OFF COFF One Shot R S Q COFF 60 μA 0.7 V + − 2.5 V VGATE(L) SS High Comparator 5.0 V − + SS Low Comparator FAULT FAULT PGnd VCC2 VCC1 VGATE(H) R S Q Q FAULT Latch VCC1 PGnd Figure 2. Block Diagram APPLICATIONS INFORMATION THEORY OF OPERATION V2 Control Method PWM Comparator + VGATE(H) C VGATE(L) − The V2 method of control uses a ramp signal that is generated by the ESR of the output capacitors. This ramp is proportional to the AC current through the main inductor and is offset by the value of the DC output voltage. This control scheme inherently compensates for variation in either line or load conditions, since the ramp signal is generated from the output voltage itself. This control scheme differs from traditional techniques such as voltage mode, which generates an artificial ramp, and current mode, which generates a ramp from inductor current. Ramp Signal VFFB Error Amplifier COMP Error Signal E Output Voltage Feedback VFB Reference Voltage − + Figure 3. V2 Control Diagram http://onsemi.com 7 CS5161, CS5161H The V2 control method is illustrated in Figure 3. The output voltage is used to generate both the error signal and the ramp signal. Since the ramp signal is simply the output voltage, it is affected by any change in the output regardless of the origin of that change. The ramp signal also contains the DC portion of the output voltage, which allows the control circuit to drive the main switch to 0% or 100% duty cycle as required. A change in line voltage changes the current ramp in the inductor, affecting the ramp signal, which causes the V2 control scheme to compensate the duty cycle. Since the change in inductor current modifies the ramp signal, as in current mode control, the V2 control scheme has the same advantages in line transient response. A change in load current will have an affect on the output voltage, altering the ramp signal. A load step immediately changes the state of the comparator output, which controls the main switch. Load transient response is determined only by the comparator response time and the transition speed of the main switch. The reaction time to an output load step has no relation to the crossover frequency of the error signal loop, as in traditional control methods. The error signal loop can have a low crossover frequency, since transient response is handled by the ramp signal loop. The main purpose of this ‘slow’ feedback loop is to provide DC accuracy. Noise immunity is significantly improved, since the error amplifier bandwidth can be rolled off at a low frequency. Enhanced noise immunity improves remote sensing of the output voltage, since the noise associated with long feedback traces can be effectively filtered. Line and load regulation are drastically improved because there are two independent voltage loops. A voltage mode controller relies on a change in the error signal to compensate for a deviation in either line or load voltage. This change in the error signal causes the output voltage to change corresponding to the gain of the error amplifier, which is normally specified as line and load regulation. A current mode controller maintains fixed error signal under deviation in the line voltage, since the slope of the ramp signal changes, but still relies on a change in the error signal for a deviation in load. The V2 method of control maintains a fixed error signal for both line and load variation, since the ramp signal is affected by both line and load. Constant Off Time Constant off time provides a number of advantages. Switch duty cycle can be adjusted from 0 to 100% on a pulse by pulse basis when responding to transient conditions. Both 0% and 100% duty cycle operation can be maintained for extended periods of time in response to load or line transients. PWM slope compensation to avoid sub−harmonic oscillations at high duty cycles is avoided. Switch on time is limited by an internal 30 μs timer, minimizing stress to the power components. Programmable Output The CS5161/5161H is designed to provide two methods for programming the output voltage of the power supply. A five bit on board digital to analog converter (DAC) is used to program the output voltage within two different ranges. The first range is 2.10 V to 3.50 V in 100 mV steps, the second is 1.30 V to 2.05 V in 50 mV steps, depending on the digital input code. If all five bits are left open, the CS5161/5161H enters adjust mode. In adjust mode, the designer can choose any output voltage by using resistor divider feedback to the VFB and VFFB pins, as in traditional controllers. Start Up To maximize transient response, the CS5161/5161H uses a constant off time method to control the rate of output pulses. During normal operation, the off time of the high side switch is terminated after a fixed period, set by the COFF capacitor. To maintain regulation, the V2 control loop varies switch on time. The PWM comparator monitors the output voltage ramp, and terminates the switch on time. Until the voltage on the VCC1 supply pin exceeds the 9.05 V monitor threshold, the Soft Start and gate pins are held low. The FAULT latch is reset (no Fault condition). The output of the error amplifier (COMP) is pulled up to 1.0 V by the comparator clamp. When the VCC1 pin exceeds the monitor threshold, the GATE(H) output is activated, and the Soft Start capacitor begins charging. The GATE(H) output will remain on, enabling the NFET switch, until terminated by either the PWM comparator, or the maximum on time timer. If the maximum on time is exceeded before the regulator output voltage achieves the 1.0 V level, the pulse is terminated. The GATE(H) pin drives low, and the GATE(L) pin drives high for the duration of the extended off time. This time is set by the time out timer and is approximately equal to the maximum on time, resulting in a 50% duty cycle. The GATE(L) pin will then drive low, the GATE(H) pin will drive high, and the cycle repeats. When regulator output voltage achieves the 1.0 V level present at the COMP pin, regulation has been achieved and normal off time will ensue. The PWM comparator terminates the switch on time, with off time set by the COFF capacitor. The V2 control loop will adjust switch duty cycle as required to ensure the regulator output voltage tracks the output of the error amplifier. The Soft Start and COMP capacitors will charge to their final levels, providing a controlled turn on of the regulator output. Regulator turn on time is determined by the COMP http://onsemi.com 8 CS5161, CS5161H capacitor charging to its final value. Its voltage is limited by the Soft Start COMP clamp and the voltage on the Soft Start pin (see Figures 4 and 5). M 10.0 μs Trace 1− Regulator Output Voltage (5.0 V/div.) M 250 μs Trace 1− Regulator Output Voltage (1.0 V/div.) Trace 2− Inductor Switching Node (2.0 V/div.) Trace 3− 12 V Input (VCC1 and VCC2) (5.0 V/div.) Trace 4− 5.0 V Input (1.0 V/div.) Trace 2− Inductor Switching Node (5.0 V/div.) Figure 6. CS5161/5161H Enable Startup Waveforms Normal Operation Figure 4. CS5161/5161H Startup in Response to Increasing 12 V and 5.0 V Input Voltages. Extended Off Time is Followed by Normal Off Time Operation when Output Voltage Achieves Regulation to the Error Amplifier Output. During normal operation, switch off time is constant and set by the COFF capacitor. Switch on time is adjusted by the V2 control loop to maintain regulation. This results in changes in regulator switching frequency, duty cycle, and output ripple in response to changes in load and line. Output voltage ripple will be determined by inductor ripple current working into the ESR of the output capacitors (see Figures 7 and 8). M 2.50 ms Trace 1− Regulator Output Voltage (1.0 V/div.) Trace 3− COMP PIn (error amplifier output) (1.0 V/div.) Trace 4− Soft Start Pin (2.0 V/div.) M 1.00 μs Trace 1− Regulator Output Voltage (10 mV/div.) Trace 2− Inductor Switching Node (5.0 V/div.) Figure 5. CS5161/5161H Startup Waveforms If the input voltage rises quickly, or the regulator output is enabled externally, output voltage will increase to the level set by the error amplifier output more rapidly, usually within a couple of cycles (see Figure 6). Figure 7. CS5161/5161H Peak−to−Peak Ripple on VOUT = 2.8 V, IOUT = 0.5 A (Light Load) http://onsemi.com 9 CS5161, CS5161H M 1.00 μs Trace 1− Regulator Output Voltage (10 mV/div.) Trace 2− Inductor Switching Node (5.0 V/div.) Trace 1− Inductor Switching Node (5.0 V/div.) Trace 2− Regulator Output Voltage (output set for 1.55 V, 20 mV/div.) 10 μs/div. Figure 8. CS5161/5161H Peak−to−Peak Ripple on VOUT = 2.8 V, IOUT = 13 A (Heavy Load) Transient Response Figure 10. CS5161/5161H Pentium® III Converter Output Voltage Response to a 0 to 12 A Load Increase The CS5161/5161H V2 control loop’s 100 ns reaction time provides unprecedented transient response to changes in input voltage or output current. Pulse by pulse adjustment of duty cycle is provided to quickly ramp the inductor current to the required level. Since the inductor current cannot be changed instantaneously, regulation is maintained by the output capacitor(s) during the time required to slew the inductor current. For best transient response, a combination of a number of high frequency and bulk output capacitors are usually used. If the maximum on time is exceeded while responding to a sudden increase in load current, a normal off time occurs to prevent saturation of the output inductor. Trace 1− Inductor Switching Node (5 V/div.) 10 μs/div. Trace 2− Regulator Output Voltage (output set for 1.55 V, 20 mV/div.) Figure 11. CS5161/5161H Pentium® III Converter Output Voltage Response to a 12 to 0 A Load Decrease PROTECTION AND MONITORING FEATURES To maintain predictable startup and shutdown characteristics an internal VCC1 monitor circuit is used to prevent the part from operating below 8.70 V minimum startup. The VCC1 monitor comparator provides hysteresis and guarantees a 8.55 V minimum shutdown threshold. Short Circuit Protection 100 μs/div. Trace 2− Regulator Output Voltage (output set for 1.55 V, 20 mV/div.) VCC1 Monitor Figure 9. CS5161/5161H Pentium® III Converter Output Voltage Response to a 12 A Load Pulse. A lossless hiccup mode short circuit protection feature is provided, requiring only the Soft Start capacitor to implement. If a short circuit condition occurs (VFFB < 1.0 V), the VFFB low comparator sets the FAULT http://onsemi.com 10 CS5161, CS5161H latch. This causes the top MOSFET to shut off, disconnecting the regulator from it’s input voltage. The Soft Start capacitor is then slowly discharged by a 2.0 μA current source until it reaches it’s lower 0.7 V threshold. The regulator will then attempt to restart normally, operating in it’s extended off time mode with a 50% duty cycle, while the Soft Start capacitor is charged with a 60 μA charge current. If the short circuit condition persists, the regulator output will not achieve the 1.0 V low VFFB comparator threshold before the Soft Start capacitor is charged to it’s upper 2.5 V threshold. If this happens the cycle will repeat itself until the short is removed. The Soft Start charge/discharge current ratio sets the duty cycle for the pulses (2.0 μA/60 μA = 3.3%), while actual duty cycle is half that due to the extended off time mode (1.65%). This protection feature results in less stress to the regulator components, input power supply, and PC board traces than occurs with constant current limit protection (see Figures 12 and 13). If the short circuit condition is removed, output voltage will rise above the 1.0 V level, preventing the FAULT latch from being set, allowing normal operation to resume. M 50.0 μs Trace 4− 5.0 V from PC Power Supply (2.0 V/div.) Trace 2− Inductor Switching Node (2.0 V/div.) Figure 13. CS5161/5161H Startup with Regulator Output Shorted Overvoltage Protection Overvoltage protection (OVP) is provided as result of the normal operation of the V2 control topology and requires no additional external components. The control loop responds to an overvoltage condition within 100 ns, causing the top MOSFET to shut off, disconnecting the regulator from it’s input voltage. The bottom MOSFET is then activated, resulting in a “crowbar” action to clamp the output voltage and prevent damage to the load (see Figures 14 and 15). The regulator will remain in this state until the overvoltage condition ceases or the input voltage is pulled low. The bottom FET and board trace must be properly designed to implement the OVP function. M 25.0 ms Trace 4− 5.0 V Supply Voltage (2.0 V/div.) Trace 3− Soft Start Timing Capacitor (1.0 V/div.) Trace 2− Inductor Switching Node (2.0 V/div.) Figure 12. CS5161/5161H Hiccup Mode Short Circuit Protection. Gate Pulses are Delivered While the Soft Start Capacitor Charges, and Cease During Discharge Trace 4− 5.0 V from PC Power Supply (5.0 V/div.) Trace 1− Regulator Output Voltage (1.0 V/div.) Trace 2− Inductor Switching Node 5.0 V/div.) M 10.0 μs Figure 14. CS5161/5161H OVP Response to an Input−to−Output Short Circuit by Immediately Providing 0% Duty Cycle, Crow−Barring the Input Voltage to Ground http://onsemi.com 11 CS5161, CS5161H 5.0 V R3 10 k VOUT R1 10 k R2 6.2 k PN3904 Power Good PN3904 CS5161 Figure 17. Implementing Power Good with the CS5161/5161H M 5.00 ms Trace 4− 5.0 V from PC Power Supply (5.0 V/div.) Trace 1− Regulator Output Voltage (1.0 V/div.) Figure 15. CS5161/5161H OVP Response to an Input−to−Output Short Circuit by Pulling the Input Voltage to Ground External Output Enable Circuit On/off control of the regulator can be implemented through the addition of two additional discrete components (see Figure 16). This circuit operates by pulling the Soft Start pin high, and the VFFB pin low, emulating a short circuit condition. M 2.50 ms 5.0 V Trace 3 − 12 V Input (VCC1) and (VCC2) (10 V/div.) Trace 4− 5.0 V Input (2.0 V/div.) Trace 1− Regulator Output Voltage (1.0 V/div.) Trace 2− Power Good Signal (2.0 V/div.) MMUN2111T1 (SOT−23) 5 SS Figure 18. CS5161/5161H During Power Up. Power Good Signal is Activated when Output Voltage Reaches 1.70 V. CS5161 8V FFB IN4148 Shutdown Input Figure 16. Implementing Shutdown with the CS5161/5161H External Power Good Circuit An optional Power Good signal can be generated through the use of four additional external components (see Figure 17). The threshold voltage of the Power Good signal can be adjusted per the following equation: VPower Good + (R1 ) R2) 0.65 V R2 This circuit provides an open collector output that drives the Power Good output to ground for regulator voltages less than VPower Good. The V2 control method uses a ramp signal, generated by the ESR of the output capacitors, that is proportional to the ripple current through the inductor. To maintain regulation, the V2 control loop monitors this ramp signal, through the PWM comparator, and terminates the switch on−time. The stringent load transient requirements of modern microprocessors require the output capacitors to have very low ESR. The resulting shallow slope presented to the PWM comparator, due to the very low ESR, can lead to pulse width jitter and variation caused by both random or synchronous noise. Adding slope compensation to the control loop, avoids erratic operation of the PWM circuit, particularly at lower duty cycles and higher frequencies, where there is not enough ramp signal, and provides a more stable switchpoint. The scheme that prevents that switching noise prematurely triggers the PWM circuit consists of adding a positive voltage slope to the output of the Error Amplifier (COMP pin) during an off−time cycle. Slope Compensation http://onsemi.com 12 CS5161, CS5161H The circuit that implements this function is shown in Figure 19. COMP 16 CCOMP CS5161 VOUT 12 R2 C1 R1 GATE(L) To Synchronous FET M 1.00 μs Trace 3 = VGATE(H) (10 V/div.) Math 1 = VGATE(H) − 5.0 VIN Trace 4 = VGATE(L) (10 V/div.) Trace 2− Inductor Switching Nodes (5.0 V/div.) Figure 19. Small RC Filter Provides the Proper Voltage Ramp at the Beginning of each On−Time Cycle The ramp waveform is generated through a small RC filter that provides the proper voltage ramp at the beginning of each on−time cycle. The resistors R1 and R2 in the circuit of Figure 14 form a voltage divider from the GATE(L) output, superimposing a small artificial ramp on the output of the error amplifier. It is important that the series combination R1/R2 is high enough in resistance not to load down and negatively affect the slew rate on the GATE(L) pin. Selecting External Components Figure 20. CS5161/5161H Gate Drive Waveforms Depicting Rail to Rail Swing The CS5161/5161H can be used with a wide range of external power components to optimize the cost and performance of a particular design. The following information can be used as general guidelines to assist in their selection. NFET Power Transistors The most important aspect of MOSFET performance is RDSON, which effects regulator efficiency and MOSFET thermal management requirements. The power dissipated by the MOSFETs may be estimated as follows; Switching MOSFET: Power + ILOAD2 RDSON duty cycle Synchronous MOSFET: Power + ILOAD2 RDSON (1 * duty cycle) Duty Cycle = VOUT ) (ILOAD RDSON OF SYNCH FET) VIN)(ILOAD RDSON OF SYNCH FET) * (ILOAD RDSON OF SWITCH FET) Off Time Capacitor (COFF) Both logic level and standard MOSFETs can be used. The reference designs derive gate drive from the 12 V supply which is generally available in most computer systems and utilize logic level MOSFETs. Multiple MOSFETs may be paralleled to reduce losses and improve efficiency and thermal management. Voltage applied to the MOSFET gates depends on the application circuit used. Both upper and lower gate driver outputs are specified to drive to within 1.5 V of ground when in the low state and to within 2.0 V of their respective bias supplies when in the high state. In practice, the MOSFET gates will be driven rail to rail due to overshoot caused by the capacitive load they present to the controller IC. For the typical application where VCC1 = VCC2 = 12 V and 5.0 V is used as the source for the regulator output current, the following gate drive is provided; VGATE(H) + 12 V * 5.0 V + 7.0 V, VGATE(L) + 12 V The COFF timing capacitor sets the regulator off time: TOFF + COFF 4848.5 The preceding equations for duty cycle can also be used to calculate the regulator switching frequency and select the COFF timing capacitor: COFF + Perioid (1 * duty cycle) 4848.5 where: Period + 1 switching frequency (see Figure 20.) http://onsemi.com 13 CS5161, CS5161H Schottky Diode for Synchronous MOSFET A Schottky diode may be placed in parallel with the synchronous MOSFET to conduct the inductor current upon turn off of the switching MOSFET to improve efficiency. For a design operating at 200 kHz or so, the low non−overlap time combined with Schottky forward recovery time may make the benefits of this device not worth the additional expense (see Figure 8, channel 2). The power dissipation in the synchronous MOSFET due to body diode conduction can be estimated by the following equation: Power + VBD ILOAD conduction time switching frequency A heatsink may be added to TO−220 components to reduce their thermal impedance. A number of PC board layout techniques such as thermal vias and additional copper foil area can be used to improve the power handling capability of surface mount components. EMI Management Where VBD = the forward drop of the MOSFET body diode. For the CS5161/5161H demonstration board as shown in Figure 8; Power + 1.6 V 13 A 100 ns 233 kHz + 0.48 W This is only 1.3% of the 36.4 W being delivered to the load. Input and Output Capacitors These components must be selected and placed carefully to yield optimal results. Capacitors should be chosen to provide acceptable ripple on the input supply lines and regulator output voltage. Key specifications for input capacitors are their ripple rating, while ESR is important for output capacitors. For best transient response, a combination of low value/high frequency and bulk capacitors placed close to the load will be required. Output Inductor As a consequence of large currents being turned on and off at high frequency, switching regulators generate noise as a consequence of their normal operation. When designing for compliance with EMI/EMC regulations, additional components may be added to reduce noise emissions. These components are not required for regulator operation and experimental results may allow them to be eliminated. The input filter inductor may not be required because bulk filter and bypass capacitors, as well as other loads located on the board will tend to reduce regulator di/dt effects on the circuit board and input power supply. Placement of the power component to minimize routing distance will also help to reduce emissions. 2.0 μH 33 Ω 1000 pF The inductor should be selected based on its inductance, current capability, and DC resistance. Increasing the inductor value will decrease output voltage ripple, but degrade transient response. THERMAL MANAGEMENT Thermal Considerations for Power MOSFETs and Diodes Figure 21. Filter Components 2.0 μH + 1200 pF × 3.0/16 V In order to maintain good reliability, the junction temperature of the semiconductor components should be kept to a maximum of 150°C or lower. The thermal impedance (junction to ambient) required to meet this requirement can be calculated as follows: Thermal Impedance + TJUNCTION(MAX) * TAMBIENT Power Figure 22. Input Filter http://onsemi.com 14 CS5161, CS5161H Layout Guidelines 1. Place 12 V filter capacitor next to the IC and connect capacitor ground to pin 11 (PGnd). 2. Connect pin 11 (PGnd) with a separate trace to the ground terminals of the 5.0 V input capacitors. 3. Place fast feedback filter capacitor next to pin 8 (VFFB) and connect it’s ground terminal with a separate, wide trace directly to pin 14 (LGnd). 4. Connect the ground terminals of the Compensation capacitor directly to the ground of the fast feedback filter capacitor to prevent common mode noise from effecting the PWM comparator. 5. Place the output filter capacitor(s) as close to the load as possible and connect the ground terminal to pin 14 (LGnd). 6. Connect the VFB pin directly to the load with a separate trace (remote sense). 7. Place 5.0 V input capacitors close to the switching MOSFET and synchronous MOSFET. Route gate drive signals VGATE(H) (pin 10) and VGATE(L) (pin 12 when used) with traces that are a minimum of 0.025 inches wide. VCC 0.1 μF 15 1.0 μF VCOMP To the negative terminal of the input capacitors 11 8 5 100 pF VFFB SOFT START OFF TIME To the negative terminal of the output capacitors Figure 23. Layout Guidelines http://onsemi.com 15 CS5161, CS5161H ADDITIONAL APPLICATION DIAGRAMS 12 V MBRS 120 0.1 μF MBRS120 1.0 μF MBRS120 1.0 μF + 100 μF/10 V × 3.0 Tantalum Remote Sense 3.3 V/10 A VCC1 VID0 VID1 VID2 VID3 VID4 COFF VCC2 VGATE(H) Si4410 3.0 μH VFB 10 Ω CS5161 VGATE(L) MBRS140T3 PGnd LGnd VFFB 3.3 k + 100 μF/10 V × 3.0 Tantalum Si9410 300 pF SS 0.1 μF COMP 033 μF 100 pF Connect to other circuits for current sharing Figure 24. 12 V to 3.3 V/10 A Converter with Remote Sense and Current Sharing 12 V 1.0 μF VCC1 VID0 VID1 VID2 VID3 VID4 330 pF COFF SS 0.1 μF COMP LGnd 0.33 μF VFFB 100 pF VFB VCC2 VGATE(H) Si9410 3.3 V 33 μF/25 V × 3.0 Tantalum 5.0 μH 2.5 V/7.0 A 100 μF/10 V × 2.0 Tantalum CS5161 Si9410 VGATE(L) MBRS140T3 PGnd 3.3 k Figure 25. 3.3 V to 2.5 V/7.0 A Converter with 12 V Bias. http://onsemi.com 16 CS5161, CS5161H ADDITIONAL APPLICATION DIAGRAMS 5.0 V 1.0 μH 12 V 0.1 μF VCC1 VID0 VID1 VID2 VID3 VID4 VCC2 VGATE(H) SILICONIX SUD50NO3−10P 1200 μF/10 V × 6.0 Sanyo GX 1.8 μH VID0 VID1 VID2 VID3 VID4 COFF VGATE(L) SILICONIX SUD50NO3−07 VOUT CS5161 PGnd 10 k MBRS140T3 30 nF 680 pF SS 0.1 μF COMP LGnd 1.0 μF VFB 500 k 1200 μF/10 V × 8.0 Sanyo GX VFFB 00.1 μF 100 10 k Figure 26. Pentium®III Converter with Slope Compensation and Adaptive Voltage Positioning http://onsemi.com 17 CS5161, CS5161H PACKAGE DIMENSIONS SO−16 D SUFFIX CASE 751B−05 ISSUE J − A− 16 9 NOTES: 1. DIMENSIONING AND TOLERANCING PER ANSI Y14.5M, 1982. 2. CONTROLLING DIMENSION: MILLIMETER. 3. DIMENSIONS A AND B DO NOT INCLUDE MOLD PROTRUSION. 4. MAXIMUM MOLD PROTRUSION 0.15 (0.006) PER SIDE. 5. DIMENSION D DOES NOT INCLUDE DAMBAR PROTRUSION. ALLOWABLE DAMBAR PROTRUSION SHALL BE 0.127 (0.005) TOTAL IN EXCESS OF THE D DIMENSION AT MAXIMUM MATERIAL CONDITION. DIM A B C D F G J K M P R MILLIMETERS MIN MAX 9.80 10.00 3.80 4.00 1.35 1.75 0.35 0.49 0.40 1.25 1.27 BSC 0.19 0.25 0.10 0.25 0_ 7_ 5.80 6.20 0.25 0.50 INCHES MIN MAX 0.386 0.393 0.150 0.157 0.054 0.068 0.014 0.019 0.016 0.049 0.050 BSC 0.008 0.009 0.004 0.009 0_ 7_ 0.229 0.244 0.010 0.019 − B− 1 8 P 8 PL 0.25 (0.010) M B S G F K C −T− SEATING PLANE R X 45 _ M D 16 PL M J 0.25 (0.010) TB S A S PACKAGE THERMAL DATA Parameter RΘJC RΘJA Typical Typical 16 Lead SO Narrow 28 115 Unit °C/W °C/W V2 is a trademark of Switch Power, Inc. Pentium is a registered trademark of Intel Corporation. ON Semiconductor and are registered trademarks of Semiconductor Components Industries, LLC (SCILLC). SCILLC reserves the right to make changes without further notice to any products herein. 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