NCP5391
2/3 Phase Buck Controller
for VR11 Pentium IV
Processor Applications
The NCP5391 is a two- or three-phase buck controller which
combines differential voltage and current sensing, and adaptive
voltage positioning to power Intel's most demanding Pentium® IV
Processors and low voltage, high current power supplies. Dual-edge
pulse-width modulation (PWM) combined with inductor current
sensing reduces system cost by providing the fastest initial response
to transient loads thereby requiring less bulk and ceramic output
capacitors to satisfy transient load-line requirements.
A high performance operational error amplifier is provided, which
allows easy compensation of the system. The proprietary method of
Dynamic Reference Injection (Patented) makes the error amplifier
compensation virtually independent of the system response to VID
changes, eliminating the need for tradeoffs between load transients
and Dynamic VID performance.
Features
•Meets Intel's VR 11.0 Specification
•Dual-Edge PWM for Fastest Initial Response to Transient Loading
•High Performance Operational Error Amplifier
•Supports VR11 Soft-Start Mode
•Dynamic Reference Injection (Patent# 7057381)
•8-Bit DAC per Intel's VR11 Specifications
•DAC Range from 0.5 V to 1.6 V
•"0.5% System Voltage Accuracy
•2 or 3-Phase Operation
•True Differential Remote Voltage Sensing Amplifier
•Phase-to-Phase Current Balancing
•“Lossless” Differential Inductor Current Sensing
•Differential Current Sense Amplifiers for each Phase
•Adaptive Voltage Positioning (AVP)
•Fixed No-Load Voltage Positioning at –19 mV
•Frequency Range: 100 kHz – 1.0 MHz
•Threshold Sensitive Enable Pin for VTT Sensing
•Power Good Output with Internal Delays
•Programmable Soft-Start Time
•Operates from 12 V
•This is a Pb-Free Device*
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MARKING
DIAGRAM
1
32
1
NCP5391
AWLYYWWG
QFN32, 5x5
MN SUFFIX
CASE 488AM
NCP5391 = Specific Device Code
A
= Assembly Location
WL
= Wafer Lot
YY
= Year
WW
= Work Week
G
= Pb-Free Package
*Pin 33 is the thermal pad on the bottom of the device.
ORDERING INFORMATION
Device
Package
Shipping†
NCP5391MNR2G
QFN32
(Pb-Free)
3000 / Tape & Reel
†For information on tape and reel specifications,
including part orientation and tape sizes, please
refer to our Tape and Reel Packaging Specification
Brochure, BRD8011/D.
*For additional information on our Pb-Free strategy
and soldering details, please download the
ON Semiconductor Soldering and Mounting
Techniques Reference Manual, SOLDERRM/D.
Applications
•Pentium IV Processors
•VRM Modules
•Graphics Cards
•Low Voltage, High Current Power Supplies
© Semiconductor Components Industries, LLC, 2007
July, 2007 - Rev. 1
1
Publication Order Number:
NCP5391/D
NCP5391
VID0
25
G3
G2
26
27
DGND
28
VCC
29
VR_RDY
31
30
ROSC2
1
ROSC
ILIM
32
PIN CONNECTIONS
G1
2
23
VID1
DRVON
VID2
CS3
NCP5391
2/3-Phase Buck Controller
(32-Pin QFN)
4
VID3
VDRP
16
VFB
COMP
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2
CS1
15
13
12
VS-
VS+
VID7
14
VID6
DIFFOUT
AGND down-bonded
to exposed flag
SS
8
CS2N
VID5
10
7
CS2
EN
6
CS3N
VID4
9
5
11
3
24
CS1N
22
21
20
19
18
17
NCP5391
VID0
VID1
VID2
VID3
VID4
VID5
VID6
VID7
VR11
DAC
SS
NCP5391
DAC
VS-
-
VS+
+
Diff Amp
DIFFOUT
Fault
1.3 V
+
VFB
Error Amp
COMP
VDRP
Droop
Amplifier
DGND
+ -
1.3 V
CS1
CS1N
+
-
+
-
ENB
+
-
ENB
+
-
ENB
G1
Gain = 6
CS2
CS2N
+
-
G2
Gain = 6
CS3
CS3N
+
-
G3
Gain = 6
4OFF
ROSC2
OVER
Oscillator
Fault
DIFFOUT
1.3 V
ROSC
+
ILIM
Current Limit
EN
VCC
+
AGND
9.0 V
UVLO
Figure 1. Simplified Block Diagram
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3
Fault Logic
3 Phase
Detect
and
Monitor
Circuits
DRVON
VR_RDY
NCP5391
12 V_FILTER
+12 V
12 V_FILTER
D1
BAT54HT1
VTT
680 PULLUPS
C4
RVCC
C3
CVCC1
NCP3418B
VCC
VCC
VID0
VID0
VID1
VID1
VID2
VID2
VID3
VID3
VID4
VID4
VID5
VID5
VID6
VID6
VID7
VID7
OD
DGND
L1
SW
DRVL
AGND
IN
R2
PGND
C1
RS1
NTD85N02RT4
C2
CS1
G1
CS1
CS1N
12 V_FILTER
12 V_FILTER
G2
EN
VR_EN
NTD60N02RT4
BST
DRVH
U1
VR_RDY
VR_RDY
CS2
CS2N
VS-
RISO1
RISO2
RT2
VCC
G3
VS+
NCP5391
OD
CS3
CS3N
SW
DRVL
IN
CFB1
BST
DRVH
PGND
RFB1
DIFFOUT
RFB
VFB
12 V_FILTER
RDRP
VDRP
CD1
RD1
CF
RF
12 V_FILTER
DRVON
ROSC2
COMP
ILIM
ROSC SS
VCC
ROSC2
OD
CH
RLIM1
BST
DRVH
CSS
SW
DRVL
IN
COSC2
PGND
RLIM2
RT2 LOCATED NEAR OUTPUT INDUCTORS
VCCP
+
VSSP
CPU GND
Figure 2. 3-Phase Application Schematic
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4
NCP5391
12 V_FILTER
+12 V
12 V_FILTER
D1
BAT54HT1
VTT
680 PULLUPS
C4
RVCC
C3
CVCC1
NCP3418B
VCC
VCC
VID0
VID0
VID1
VID1
VID2
VID2
VID3
VID3
VID4
VID4
VID5
VID5
VID6
VID6
VID7
VID7
L1
SW
DRVL
AGND
IN
R2
PGND
C1
RS1
NTD85N02RT4
C2
CS1
G1
CS1
CS1N
G2
VR_RDY
VR_RDY
OD
DGND
EN
VR_EN
NTD60N02RT4
BST
DRVH
U1
CS2
CS2N
VSG3
VS+
RISO1
RISO2
RT2
CFB1
NCP5391
CS3
CS3N
RFB1
DIFFOUT
RFB
VFB
12 V_FILTER
RDRP
VDRP
12 V_FILTER
DRVON
RD1
CD1
ROSC2
COMP
RF
CF
ILIM
ROSC SS
VCC
ROSC2
OD
CH
RLIM1
BST
DRVH
CSS
SW
DRVL
IN
COSC2
PGND
RLIM2
RT2 LOCATED NEAR OUTPUT INDUCTORS
VCCP
+
VSSP
CPU GND
Figure 3. 2-Phase Application Schematic
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5
NCP5391
PIN DESCRIPTIONS
Pin No.
Symbol
Description
1-8
VID0–VID7
9
EN
Pull this pin high to enable controller. Pull this pin low to disable controller. Either an open-collector output
(with a pull-up resistor) or a logic gate (CMOS or totem-pole output) may be used to drive this pin. A Low
to High transition on this pin will initiate a soft start. If the Enable function is not required, this pin should be
tied directly to VREF.
10
SS
A capacitor from this pin to ground programs the soft-start time.
11
VS+
Non-inverting input to the internal differential remote VCORE sense amplifier.
12
VS-
Inverting input to the internal differential remote VCORE sense amplifier.
13
DIFFOUT
14
COMP
15
VFB
Error amplifier inverting input. Connect a resistor from this pin to DIFFOUT. The value of this resistor and
the amount of current from the droop resistor (RDRP) will set the amount of output voltage droop (AVP)
during load.
16
VDRP
Current signal output for Adaptive Voltage Positioning (AVP). The voltage of this pin minus 1.3 V is
proportional to the output current. Connect a resistor from this pin to VFB to set the amount of AVP current
into the feedback resistor (RFB) to produce an output voltage droop. Leave this pin open for no AVP.
17, 19, 21
CSxN
Inverting input to current sense amplifier #x, x = 1, 2, 3
18, 20, 22
CSx
23
DRVON
Gate Driver enable output. This pin produces a logic HIGH to enable gate drivers and a logic LOW to
disable gate drivers and has an internal 70 k to ground.
24, 25, 26
G1 – G3
PWM control signal outputs to gate drivers.
27
DGND
Voltage ID DAC inputs.
Output of the differential remote sense amplifier.
Output of the error amplifier.
Non-inverting input to current sense amplifier #x, x = 1, 2, 3
Power supply return for the digital circuits. Connect to AGND.
28
VCC
29
VR_RDY
Power for the internal control circuits.
30
ROSC2
Use for Enhanced Performance
31
ROSC
A resistance from this pin to ground programs the oscillator frequency. Also, this pin supplies a regulated
2.0 V which may be used with a voltage divider to the ILIM pin to set the over current shutdown threshold
as shown in the Applications Schematics.
32
ILIM
33
THPAD/
AGND
Voltage Regulator Ready (PowerGood) output. Open drain type output with internal delays that will
transition High when VCORE is higher than 300 mV below DAC, Low when VCORE is lower than 380 mV
below DAC, and Low when VCORE is higher than DAC+185 mV. This output is latched Low if VCORE
exceeds DAC+185 mV until VCC is removed.
Over current shutdown threshold. To program the shutdown threshold, connect this pin to the ROSC pin via
a resistor divider as shown in the Applications Schematics. To disable the over current feature connect this
pin directly to the ROSC pin. To guarantee correct operation, this pin should only be connected to the
voltage generated by the ROSC pin – do not connect this pin to any externally generated voltages.
Copper pad on the bottom of the IC for heatsinking. This pin should be connected to the ground plane
under the IC. Power supply return for the analog circuits that control output voltage.
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6
NCP5391
MAXIMUM RATINGS
Rating
Value
Unit
Operating Ambient Temperature Range
0 to 70
°C
Operating Junction Temperature Range
0 to 85
°C
-55 to 150
°C
Lead Temperature Soldering, Reflow (60 to 120 seconds minimum above 237°C)
260
°C
Thermal Resistance, Junction-to-Ambient (RJA) on a thermally conductive PCB in free air
56
°C/W
JEDEC Moisture Sensitivity Level
≤1
MSL
Maximum Voltage – VCC pin with respect to AGND
15
V
Maximum Voltage – all other pins with respect to AGND
5.5
V
Minimum Voltage – all pins with respect to AGND
-0.3
V
Maximum Current into pins: COMP, VDRP, DIFFOUT
3.0
mA
Maximum Current into pins: VR_RDY, G1, G2, G3, SS, DRVON
20
mA
Maximum Current out of pins: COMP, VDRP, DIFFOUT, ROSC
3.0
mA
Maximum Current out of pins: G1, G2, G3
20
mA
Storage Temperature Range
Stresses exceeding Maximum Ratings may damage the device. Maximum Ratings are stress ratings only. Functional operation above the
Recommended Operating Conditions is not implied. Extended exposure to stresses above the Recommended Operating Conditions may affect
device reliability.
NOTE: ESD Sensitive Device
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7
NCP5391
ELECTRICAL CHARACTERISTICS
(0°C < TA < 70°C; 0°C < TJ < 85°C; 10.8 V < VCC < 13.2 V; All DAC Codes; CVCC = 0.1 F, FSW = 400 kHz, unless otherwise stated)
Parameter
Test Conditions
Min
Typ
Max
Unit
-200
-50
-10
nA
-
1.3
-
V
-1.0
-
1.0
mV
ERROR AMPLIFIER
Input Bias Current
Inverting Input Voltage
1.0 k between VFB and COMP Pins
Input Offset Voltage (Note 1)
Open Loop DC Gain (Note 1)
CL = 60 pF to GND,
RL = 10 k to GND
-
78
-
dB
Open Loop Unity Gain Bandwidth
(Note 1)
CL = 60 pF to GND,
RL = 10 k to GND
-
15
-
MHz
Open Loop Phase Margin (Note 1)
CL = 60 pF to GND,
RL = 10 k to GND
-
65
-
°
Slew Rate (Note 1)
Vin = 100 mV, G = -1.0 V/V,
1.2 V < Vout < 2.2 V,
CL = 60 pF,
DC Load = ±125 A
-
5.0
-
V/s
Maximum Output Voltage
ISOURCE = 1.0 mA
3.0
3.3
-
V
Minimum Output Voltage
ISINK = 1.0 mA
-
0.9
1.0
V
Output Source Current (Note 1)
Vout = 3.0 V
-
2.0
-
mA
Output Sink Current (Note 1)
Vout = 1.0 V
-
2.0
-
mA
REMOTE SENSE DIFFERENTIAL AMPLIFIER
VS+ Input Resistance (Note 1)
DRVON = High
DRVON = Low
-
17
0.5
-
k
VS+ Input Open Circuit Voltage
(Note 1)
DRVON = High
DRVON = Low
-
0.67
0.05
-
V
VS- Input Resistance (Note 1)
VS+ = DAC Voltage
DRVON = High
-
10
-
k
VS- Input Open Circuit Voltage
(Note 1)
DRVON = High
VS+ = DAC Voltage
= 0.333*
DAC
+ 0.433
Input Voltage Range
-0.3
Input Offset Voltage (Note 1)
V
-
3.0
V
-1.0
-
1.0
mV
-
12
-
MHz
0.982
1.0
1.018
V/V
-
10
-
V/s
3.0
-
-
V
-
0.5
V
-3dB Bandwidth (Note 1)
CL = 80 pF to GND,
RL = 10 k to GND
DC Gain
IDIFFOUT = 100 A
Slew Rate (Note 1)
Vin = 1.0 V,
Vout = 1.0 V to 2.0 V,
CL = 80 pF to GND,
Load = ±125 A
Maximum Output Voltage
ISOURCE = 1.0 mA
Minimum Output Voltage
ISINK = 1.0 mA
-
Output Source Current (Note 1)
Vout = 2.1 V
-
25
-
mA
Output Sink Current (Note 1)
Vout = 1.0 V
-
1.4
-
mA
5.7
6.0
6.3
V/V
-
7.2
-
MHz
VDRP ADAPTIVE VOLTAGE POSITIONING AMPLIFIER
Current Sense Input to VDRP Gain
-60 mV < (CSx-CSxN)
< +60 mV, TA = 25°C
Current Sense Input to VDRP Output
-3dB Bandwidth (Note 1)
CL = 330 pF to GND,
RL = 10 k to GND
1. Guaranteed by design. Not tested in production.
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8
NCP5391
ELECTRICAL CHARACTERISTICS
(0°C < TA < 70°C; 0°C < TJ < 85°C; 10.8 V < VCC < 13.2 V; All DAC Codes; CVCC = 0.1 F, FSW = 400 kHz, unless otherwise stated)
Parameter
Test Conditions
Min
Typ
Max
Unit
Current Sense Input to VDRP Output
Slew Rate (Note 1)
V(CSx-CSxN) = 25 mV (all phases), 1.3 V < Vout
< 1.9 V,
CL = 330 pF to GND,
Load = ±400 A
-
3.7
-
V/s
Current Summing Amp Output Offset
Voltage
CSx – CSxN = 0, CSx =1.0 V
-15
-
+15
mV
Maximum VDRP Output Voltage
CSx - CSxN = 0.12 V
(all phases),
ISOURCE = 1.0 mA
3.02
-
-
V
Minimum VDRP Output Voltage
CSx - CSxN = -0.12 V
(all phases),
ISINK = 1.0 mA
-
-
0.5
V
Output Source Current (Note 1)
VDRP = 2.9 V
-
9.0
-
mA
Output Sink Current (Note 1)
VDRP = 1.0 V
-
2.0
-
mA
-200
-50
-10
nA
Common Mode Input Voltage Range
(Note 1)
-0.3
-
2.0
V
Differential Mode Input Voltage Range
-120
-
120
mV
VDRP ADAPTIVE VOLTAGE POSITIONING AMPLIFIER
CURRENT SENSE AMPLIFIERS
Input Bias Current
CSx = CSxN = 1.4 V
Input Offset Voltage (Note 1)
CSx = CSxN = 1.0 V
-3.0
-
3.0
mV
Current Sense Input to
PWM Comparator Input Gain
0 mV < (CSx-CSxN) < 25 mV
TA = 25°C
5.7
6.0
6.3
V/V
100
-
1000
kHz
Switching Frequency Accuracy
(Note 1)
ROSC = 100 k, 2-phase
93.6
104
114.4
kHz
Switching Frequency Accuracy
ROSC = 49.9 k, 2-phase
184.5
205
225.5
kHz
Switching Frequency Accuracy
ROSC = 24.9 k, 2-phase
360
400
440
kHz
Switching Frequency Accuracy
ROSC = 10 k, 2-phase
829
921
1013
kHz
Switching Frequency Accuracy
(Note 1)
ROSC = 100 k, 3-phase
90
100
110
kHz
Switching Frequency Accuracy
ROSC = 49.9 k, 3-phase
178.2
198
217.8
kHz
Switching Frequency Accuracy
ROSC = 24.9 k, 3-phase
351
390
429
kHz
Switching Frequency Accuracy
ROSC = 10 k, 3-phase
818
909
1000
kHz
ROSC Output Voltage
10 k < ROSC < 49.9 k
1.92
2.00
2.08
V
ROSC Output Voltage (Note 1)
49.9 k < ROSC < 100 k
-
2.00
-
V
Fs = 400 kHz
-
30
40
ns
-
1.0
-
V
OSCILLATOR
Switching Frequency Range (Note 1)
MODULATORS (PWM COMPARATORS)
Minimum Pulse Width
Magnitude of the PWM Ramp
0% Duty Cycle
COMP voltage when the PWM
outputs remain LO
-
1.2
-
V
100% Duty Cycle
COMP voltage when the PWM
outputs remain HI
-
2.3
-
V
Minimum PWM Linear Duty Cycle
(Note 1)
FS = 400 kHz
-
90
-
%
1. Guaranteed by design. Not tested in production.
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NCP5391
ELECTRICAL CHARACTERISTICS
(0°C < TA < 70°C; 0°C < TJ < 85°C; 10.8 V < VCC < 13.2 V; All DAC Codes; CVCC = 0.1 F, FSW = 400 kHz, unless otherwise stated)
Parameter
Test Conditions
Min
Typ
Max
Unit
-
-
40
mV
-15
-
15
°
MODULATORS (PWM COMPARATORS)
PWM Comparator Offset Mismatch
(Note 1)
Between any 2 phases,
FS = 400 kHz
Phase Angle Error
Between adjacent phases,
FS = 400 kHz
Propagation Delay (Note 1)
Ramp/Comp crossing to Gx high
-
20
-
ns
Propagation Delay (Note 1)
Ramp/Comp crossing to Gx low
-
20
-
ns
3.3
4.0
4.7
V
PWM OUTPUTS
Output High Voltage
Sourcing 500 A
Output Low Voltage
Sinking 500 A
-
25
100
mV
Rise Time
CL = 20 pF, Vo = 0.3 to 2.0 V
-
10
-
ns
Fall Time
CL = 20 pF, Vo = Vmax to 0.7 V
-
10
-
ns
Output Impedance – LO State
Resistance to GND (Gx = LO)
-
50
-
G3 Gate Pin Source Current during
Phase Detect
-
70
-
A
Phase Detection Period
-
50
-
s
G3 Phase Detect Threshold
Resistance
-
-
1.0
k
4.0
5.3
5.5
V
GATE DRIVER ENABLE (DRVON)
Output High Voltage
Sourcing 500 A
Output Low Voltage
Sinking 500 A
-
50
200
mV
Rise Time
CL (PCB) = 20 pF,
Vo = 10% to 90%
-
25
-
ns
Fall Time
CL (PCB) = 20 pF,
Vo = 10% to 90%
-
25
-
ns
Internal Pulldown Resistance
VCC < UVLO Threshold
-
70
140
k
Saturation Voltage
ISINK = 10 mA
-
-
0.4
V
Rise Time
External pullup of 1.0 k to 1.25 V, CLOAD = 20 pF,
Vo = 10% to 90%
-
-
150
ns
Output Voltage at Power-up (Note 1)
External VR_RDY pullup resistor of 2.0 k to 5.0 V,
tR_VCC ≤ 3 x tR_5V,
-
-
1.0
V
VR_RDY (POWER GOOD) OUTPUT
100 s ≤ tR_VCC ≤ 20 ms
High – Output Leakage Current
VR_RDY = 5.5 V via 1.0 K
-
-
1.0
A
Upper Threshold Voltage
VCORE increasing,
DAC = 1.3 V
-
300
-
mV
below
DAC
Rising Delay
VCORE increasing
0.3
1.40
2.0
ms
Falling Delay
VCORE decreasing
-
5.0
-
s
SS Pin Source Current
ENABLE = HI, VSS PIN < 1.1 V
-
5.0
-
A
SS Pin Source Current
ENABLE = HI, VSS PIN > 1.15 V, VR11 SS Mode
Only
125
-
-
A
Soft-Start Ramp Time
CSS = 0.01 F, DRVON = HI to VSS PIN = 1.1 V
1.5
2.2
3.0
ms
SS Pin Discharge Voltage
ENABLE = LO
-
-
50
mV
SOFT-ST ART
1. Guaranteed by design. Not tested in production.
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10
NCP5391
ELECTRICAL CHARACTERISTICS
(0°C < TA < 70°C; 0°C < TJ < 85°C; 10.8 V < VCC < 13.2 V; All DAC Codes; CVCC = 0.1 F, FSW = 400 kHz, unless otherwise stated)
Parameter
Test Conditions
Min
Typ
Max
Unit
-
5.0
-
s
VR11 VBOOT Threshold Voltage
-
1.081
-
V
VR11 Dwell Time at VBOOT (Note 1)
50
225
900
s
-
-
10
A
SOFT-ST ART
Soft-Start Discharge Time
From ENABLE = LO to VSS PIN < max Discharge
Voltage, CSS = 0.01 F
ENABLE INPUT
Enable High Input Leakage Current
EN = 3.0 V
Upper Threshold
VUPPER
0.80
0.85
0.90
V
Lower Threshold
VLOWER
0.67
0.75
0.83
V
Total Hysteresis
VUPPER – VLOWER
70
100
130
mV
Enable Delay Time
Enable transitioning HI to start of SS voltage rise
0.5
1.5
3.0
ms
Disable Delay Time
Enable transitioning Low to DRVON = Low
-
-
200
ns
5.7
6.0
6.3
V/V
-
0.1
1.0
A
ILIM Pin Working Voltage Range
(Note 1)
0.3
-
2.0
V
ILIM Input Offset Voltage (Note 1)
-50
-
50
mV
DAC
+160
DAC+18
0
DAC
+200
mV
UVLO Start Threshold
8.2
9.0
9.5
V
UVLO Stop Threshold
7.2
8.0
8.5
V
-
1.0
-
V
CURRENT LIMIT
Current Sense Inputs to ILIM Gain
(Note 1)
20 mV < (CSx-CSxN) < 60 mV
TA = 25°C (all CS channels together)
ILIM Pin Input Bias Current
VILIM = 2.0 V
OVERVOLTAGE PROTECTION
Overvoltage Threshold (Note 1)
UNDERVOLTAGE PROTECTION
UVLO Hysteresis
VID INPUTS
Upper Threshold
VUPPER
-
-
800
mV
Lower Threshold
VLOWER
400
-
-
mV
Input Bias Current
VVIDX = 1.25 V
Delay before Latching VID Change
(VID De-Skewing)
Measured from the 1st edge of a VID change
-
100
500
nA
400
-
1000
ns
INTERNAL DAC SLEW RATE LIMITER
Positive Slew Rate Limit
VID step range of +10mV to +500mV
-
7.3
-
mV/s
Negative Slew Rate Limit
VID step range of -10mV to -500mV
-
7.3
-
mV/s
FSW = 400 kHz
-
20
-
mA
System Voltage Accuracy
1.0 V < DAC < 1.6 V
0.8 V < DAC < 1.0 V
0.5 V < DAC < 0.8 V
-
-
±0.5
±5.0
±8.0
%
mV
mV
No-Load Offset Voltage from
Nominal DAC Specification
With CS Input Vin = 0 V
INPUT SUPPLY CURRENT
VCC Operating Current
VR 11 DAC
1. Guaranteed by design. Not tested in production.
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11
-19
mV
NCP5391
Table 2: VR11 VID Codes
VID7
800 mV
VID6
400 mV
VID5
200 mV
VID4
100 mV
VID3
50 mV
VID2
25 mV
VID1
12.5 mV
VID0
6.25 mV
Nominal
DAC
Voltage (V)
HEX
0
0
0
0
0
0
0
0
OFF
00
0
0
0
0
0
0
0
1
OFF
01
0
0
0
0
0
0
1
0
1.60000
02
0
0
0
0
0
0
1
1
1.59375
03
0
0
0
0
0
1
0
0
1.58750
04
0
0
0
0
0
1
0
1
1.58125
05
0
0
0
0
0
1
1
0
1.57500
06
0
0
0
0
0
1
1
1
1.56875
07
0
0
0
0
1
0
0
0
1.56250
08
0
0
0
0
1
0
0
1
1.55625
09
0
0
0
0
1
0
1
0
1.55000
0A
0
0
0
0
1
0
1
1
1.54375
0B
0
0
0
0
1
1
0
0
1.53750
0C
0
0
0
0
1
1
0
1
1.53125
0D
0
0
0
0
1
1
1
0
1.52500
0E
0
0
0
0
1
1
1
1
1.51875
0F
0
0
0
1
0
0
0
0
1.51250
10
0
0
0
1
0
0
0
1
1.50625
11
0
0
0
1
0
0
1
0
1.50000
12
0
0
0
1
0
0
1
1
1.49375
13
0
0
0
1
0
1
0
0
1.48750
14
0
0
0
1
0
1
0
1
1.48125
15
0
0
0
1
0
1
1
0
1.47500
16
0
0
0
1
0
1
1
1
1.46875
17
0
0
0
1
1
0
0
0
1.46250
18
0
0
0
1
1
0
0
1
1.45625
19
0
0
0
1
1
0
1
0
1.45000
1A
0
0
0
1
1
0
1
1
1.44375
1B
0
0
0
1
1
1
0
0
1.43750
1C
0
0
0
1
1
1
0
1
1.43125
1D
0
0
0
1
1
1
1
0
1.42500
1E
0
0
0
1
1
1
1
1
1.41875
1F
0
0
1
0
0
0
0
0
1.41250
20
0
0
1
0
0
0
0
1
1.40625
21
0
0
1
0
0
0
1
0
1.40000
22
0
0
1
0
0
0
1
1
1.39375
23
0
0
1
0
0
1
0
0
1.38750
24
0
0
1
0
0
1
0
1
1.38125
25
0
0
1
0
0
1
1
0
1.37500
26
0
0
1
0
0
1
1
1
1.36875
27
0
0
1
0
1
0
0
0
1.36250
28
0
0
1
0
1
0
0
1
1.35625
29
0
0
1
0
1
0
1
0
1.35000
2A
0
0
1
0
1
0
1
1
1.34375
2B
0
0
1
0
1
1
0
0
1.33750
2C
0
0
1
0
1
1
0
1
1.33125
2D
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12
NCP5391
Table 2: VR11 VID Codes
VID7
800 mV
VID6
400 mV
VID5
200 mV
VID4
100 mV
VID3
50 mV
VID2
25 mV
VID1
12.5 mV
VID0
6.25 mV
Nominal
DAC
Voltage (V)
HEX
0
0
1
0
1
1
1
0
1.32500
2E
0
0
1
0
1
1
1
1
1.31875
2F
0
0
1
1
0
0
0
0
1.31250
30
0
0
1
1
0
0
0
1
1.30625
31
0
0
1
1
0
0
1
0
1.30000
32
0
0
1
1
0
0
1
1
1.29375
33
0
0
1
1
0
1
0
0
1.28750
34
0
0
1
1
0
1
0
1
1.28125
35
0
0
1
1
0
1
1
0
1.27500
36
0
0
1
1
0
1
1
1
1.26875
37
0
0
1
1
1
0
0
0
1.26250
38
0
0
1
1
1
0
0
1
1.25625
39
0
0
1
1
1
0
1
0
1.25000
3A
0
0
1
1
1
0
1
1
1.24375
3B
0
0
1
1
1
1
0
0
1.23750
3C
0
0
1
1
1
1
0
1
1.23125
3D
0
0
1
1
1
1
1
0
1.22500
3E
0
0
1
1
1
1
1
1
1.21875
3F
0
1
0
0
0
0
0
0
1.21250
40
0
1
0
0
0
0
0
1
1.20625
41
0
1
0
0
0
0
1
0
1.20000
42
0
1
0
0
0
0
1
1
1.19375
43
0
1
0
0
0
1
0
0
1.18750
44
0
1
0
0
0
1
0
1
1.18125
45
0
1
0
0
0
1
1
0
1.17500
46
0
1
0
0
0
1
1
1
1.16875
47
0
1
0
0
1
0
0
0
1.16250
48
0
1
0
0
1
0
0
1
1.15625
49
0
1
0
0
1
0
1
0
1.15000
4A
0
1
0
0
1
0
1
1
1.14375
4B
0
1
0
0
1
1
0
0
1.13750
4C
0
1
0
0
1
1
0
1
1.13125
4D
0
1
0
0
1
1
1
0
1.12500
4E
0
1
0
0
1
1
1
1
1.11875
4F
0
1
0
1
0
0
0
0
1.11250
50
0
1
0
1
0
0
0
1
1.10625
51
0
1
0
1
0
0
1
0
1.10000
52
0
1
0
1
0
0
1
1
1.09375
53
0
1
0
1
0
1
0
0
1.08750
54
0
1
0
1
0
1
0
1
1.08125
55
0
1
0
1
0
1
1
0
1.07500
56
0
1
0
1
0
1
1
1
1.06875
57
0
1
0
1
1
0
0
0
1.06250
58
0
1
0
1
1
0
0
1
1.05625
59
0
1
0
1
1
0
1
0
1.05000
5A
0
1
0
1
1
0
1
1
1.04375
5B
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13
NCP5391
Table 2: VR11 VID Codes
VID7
800 mV
VID6
400 mV
VID5
200 mV
VID4
100 mV
VID3
50 mV
VID2
25 mV
VID1
12.5 mV
VID0
6.25 mV
Nominal
DAC
Voltage (V)
HEX
0
1
0
1
1
1
0
0
1.03750
5C
0
1
0
1
1
1
0
1
1.03125
5D
0
1
0
1
1
1
1
0
1.02500
5E
0
1
0
1
1
1
1
1
1.01875
5F
0
1
1
0
0
0
0
0
1.01250
60
0
1
1
0
0
0
0
1
1.00625
61
0
1
1
0
0
0
1
0
1.00000
62
0
1
1
0
0
0
1
1
0.99375
63
0
1
1
0
0
1
0
0
0.98750
64
0
1
1
0
0
1
0
1
0.98125
65
0
1
1
0
0
1
1
0
0.97500
66
0
1
1
0
0
1
1
1
0.96875
67
0
1
1
0
1
0
0
0
0.96250
68
0
1
1
0
1
0
0
1
0.95625
69
0
1
1
0
1
0
1
0
0.95000
6A
0
1
1
0
1
0
1
1
0.94375
6B
0
1
1
0
1
1
0
0
0.93750
6C
0
1
1
0
1
1
0
1
0.93125
6D
0
1
1
0
1
1
1
0
0.92500
6E
0
1
1
0
1
1
1
1
0.91875
6F
0
1
1
1
0
0
0
0
0.91250
70
0
1
1
1
0
0
0
1
0.90625
71
0
1
1
1
0
0
1
0
0.90000
72
0
1
1
1
0
0
1
1
0.89375
73
0
1
1
1
0
1
0
0
0.88750
74
0
1
1
1
0
1
0
1
0.88125
75
0
1
1
1
0
1
1
0
0.87500
76
0
1
1
1
0
1
1
1
0.86875
77
0
1
1
1
1
0
0
0
0.86250
78
0
1
1
1
1
0
0
1
0.85625
79
0
1
1
1
1
0
1
0
0.85000
7A
0
1
1
1
1
0
1
1
0.84375
7B
0
1
1
1
1
1
0
0
0.83750
7C
0
1
1
1
1
1
0
1
0.83125
7D
0
1
1
1
1
1
1
0
0.82500
7E
0
1
1
1
1
1
1
1
0.81875
7F
1
0
0
0
0
0
0
0
0.81250
80
1
0
0
0
0
0
0
1
0.80625
81
1
0
0
0
0
0
1
0
0.80000
82
1
0
0
0
0
0
1
1
0.79375
83
1
0
0
0
0
1
0
0
0.78750
84
1
0
0
0
0
1
0
1
0.78125
85
1
0
0
0
0
1
1
0
0.77500
86
1
0
0
0
0
1
1
1
0.76875
87
1
0
0
0
1
0
0
0
0.76250
88
1
0
0
0
1
0
0
1
0.75625
89
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14
NCP5391
Table 2: VR11 VID Codes
VID7
800 mV
VID6
400 mV
VID5
200 mV
VID4
100 mV
VID3
50 mV
VID2
25 mV
VID1
12.5 mV
VID0
6.25 mV
Nominal
DAC
Voltage (V)
HEX
1
0
0
0
1
0
1
0
0.75000
8A
1
0
0
0
1
0
1
1
0.74375
8B
1
0
0
0
1
1
0
0
0.73750
8C
1
0
0
0
1
1
0
1
0.73125
8D
1
0
0
0
1
1
1
0
0.72500
8E
1
0
0
0
1
1
1
1
0.71875
8F
1
0
0
1
0
0
0
0
0.71250
90
1
0
0
1
0
0
0
1
0.70625
91
1
0
0
1
0
0
1
0
0.70000
92
1
0
0
1
0
0
1
1
0.69375
93
1
0
0
1
0
1
0
0
0.68750
94
1
0
0
1
0
1
0
1
0.68125
95
1
0
0
1
0
1
1
0
0.67500
96
1
0
0
1
0
1
1
1
0.66875
97
1
0
0
1
1
0
0
0
0.66250
98
1
0
0
1
1
0
0
1
0.65625
99
1
0
0
1
1
0
1
0
0.65000
9A
1
0
0
1
1
0
1
1
0.64375
9B
1
0
0
1
1
1
0
0
0.63750
9C
1
0
0
1
1
1
0
1
0.63125
9D
1
0
0
1
1
1
1
0
0.62500
9E
1
0
0
1
1
1
1
1
0.61875
9F
1
0
1
0
0
0
0
0
0.61250
A0
1
0
1
0
0
0
0
1
0.60625
A1
1
0
1
0
0
0
1
0
0.60000
A2
1
0
1
0
0
0
1
1
0.59375
A3
1
0
1
0
0
1
0
0
0.58750
A4
1
0
1
0
0
1
0
1
0.58125
A5
1
0
1
0
0
1
1
0
0.57500
A6
1
0
1
0
0
1
1
1
0.56875
A7
1
0
1
0
1
0
0
0
0.56250
A8
1
0
1
0
1
0
0
1
0.55625
A9
1
0
1
0
1
0
1
0
0.55000
AA
1
0
1
0
1
0
1
1
0.54375
AB
1
0
1
0
1
1
0
0
0.53750
AC
1
0
1
0
1
1
0
1
0.53125
AD
1
0
1
0
1
1
1
0
0.52500
AE
1
0
1
0
1
1
1
1
0.51875
AF
1
0
1
1
0
0
0
0
0.51250
B0
1
0
1
1
0
0
0
1
0.50625
B1
1
0
1
1
0
0
1
0
0.50000
B2
1
1
1
1
1
1
1
0
OFF
FE
1
1
1
1
1
1
1
1
OFF
FF
OFF
B3 to FD
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15
NCP5391
TYPICAL CHARACTERISTICS
13.6
VCC, UNDERVOLTAGE LOCKOUT
THRESHOLD VOLTAGE (V)
ICC, IC QUIESCENT CURRENT (mA)
10
13.4
13.2
13.0
12.8
9
VCC Increasing Voltage
8
VCC Decreasing Voltage
7
12.6
10
20
30
40
50
60
70
0
10
20
30
40
50
60
TA, AMBIENT TEMPERATURE (°C)
TA, AMBIENT TEMPERATURE (°C)
Figure 4. IC Quiescent Current vs. Ambient
Temperature
Figure 5. VCC Undervoltage Lockout
Threshold Voltage vs. Ambient Temperature
0.0198
0.0196
25°C
0.0194
DAC OFFSET
0
0.0192
0.0190
0.0188
0°C
0.0186
0.0184
70°C
0.0182
0.0180
0.5 0.6 0.7
0.8 0.9 1.0 1.1
1.2 1.3 1.4
1.5 1.6
VID
Figure 6. Typical DAC Voltage Offset vs.
Temperature
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16
70
NCP5391
FUNCTIONAL DESCRIPTION
Gate Output Connections
General
The NCP5391 dual edge modulated multiphase PWM
controller is specifically designed with the necessary
features for a high current VR11 CPU power system. The
IC consists of the following blocks: Precision
Programmable DAC, Differential Remote Voltage Sense
Amplifier, High Performance Voltage Error Amplifier,
Differential Current Feedback Amplifiers, Precision
Oscillator and Triangle Wave Generators, and PWM
Comparators. Protection features include Undervoltage
Lockout, Soft-Start, Overcurrent Protection, Overvoltage
Protection, and Power Good Monitor.
Mode
G1
G2
G3
2-Phase
Normal
GND
Normal
3-Phase
Normal
Normal
Normal
These are the only allowable connection schemes to
program the modes of operation.
Differential Current Sense Amplifiers
A precision programmable DAC is provided. This DAC
has 0.5% accuracy over the entire operating temperature
range of the part.
Three differential amplifiers are provided to sense the
output current of each phase. The inputs of each current
sense amplifier must be connected across the current
sensing element of the phase controlled by the
corresponding gate output (G1, G2 or G3). If 2 phase is
unused, the differential inputs to that phase's current
sense amplifier must be shorted together and connected
to VCCP as shown in the 2-phase Application
Schematics.
A voltage is generated across the current sense element
(such as an inductor or sense resistor) by the current
flowing in that phase. The output of the current sense
amplifiers are used to control three functions. First, the
output controls the adaptive voltage positioning, where the
output voltage is actively controlled according to the
output current. In this function, all of the current sense
outputs are summed so that the total output current is used
for output voltage positioning. Second, the output signal is
fed to the current limit circuit. This again is the summed
current of all phases in operation. Finally, the individual
phase current is connected to the PWM comparator. In this
way current balance is accomplished.
High Performance Voltage Error Amplifier
Oscillator and Triangle Wave Generator
The error amplifier is designed to provide high slew rate
and bandwidth. Although not required when operating as
a voltage regulator, a capacitor from COMP to VFB is
required for stable unity gain test configurations.
A programmable precision oscillator is provided. The
oscillator 's frequency is programmed by the resistance
connected from the ROSC pin to ground. The user will
usually form this resistance from two resistors in order to
create a voltage divider that uses the ROSC output voltage
as the reference for creating the current limit setpoint
voltage. The oscillator frequency range is 100 kHz/phase
to 1.0 MHz/phase. The oscillator generates up to 3 triangle
waveforms (symmetrical rising and falling slopes)
between 1.3 V and 2.3 V. The triangle waves have a phase
delay between them such that for 2-, 3-phase operation the
PWM outputs are separated by 180 and 120 angular
degrees, respectively.
Remote Output Sensing Amplifier (RSA)
A true differential amplifier allows the NCP5391 to
measure Vcore voltage feedback with respect to the Vcore
ground reference point by connecting the Vcore reference
point to VS+, and the Vcore ground reference point to VS-.
This configuration keeps ground potential differences
between the local controller ground and the Vcore ground
reference point from affecting regulation of Vcore between
Vcore and Vcore ground reference points. The RSA also
subtracts the DAC (minus VID offset) voltage, thereby
producing an unamplified output error voltage at the
DIFFOUT pin. This output also has a 1.3 V bias voltage to
allow both positive and negative error voltages.
Precision DAC
Gate Driver Outputs and 2/3 Phase Operation
The part can be configured to run in 2- or 3-phase mode.
In 2-phase mode, phases 1 and 3 should be used to drive the
external gate drivers as shown in the 2-phase Applications
Schematic. In 2-phase mode, gate output G2 must be
grounded as shown in the 2-phase Applications Schematic.
The following truth table summarizes the modes of
operation:
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17
NCP5391
PWM Comparators with Hysteresis
is the summed current information from the current sense
amplifiers. The overcurrent latch is set when the current
information exceeds the voltage at the ILIM pin. The
outputs are immediately disabled, the VR_RDY and
DRVON pins are pulled low, and the soft-start is pulled
low. The outputs will remain disabled until the VCC voltage
is removed and re-applied, or the ENABLE input is
brought low and then high.
Three PWM comparators receive the error amplifier
output signal at their noninverting input. Each comparator
receives one of the triangle waves offset by 1.3 V at it's
inverting input. The output of the comparator generates the
PWM outputs G1, G2 and G3.
During steady state operation, the duty cycle will center
on the valley of the triangle waveform, with steady state
duty cycle calculated by Vout/Vin. During a transient event,
both high and low comparator output transitions shift phase
to the points where the error amplifier output intersects the
down and up ramp of the triangle wave.
Overvoltage Protection and Power Good Monitor
An output voltage monitor is incorporated. During
normal operation, if the voltage at the DIFFOUT pin
exceeds 1.3 V, the VR_RDY pin goes low, the DRVON
signal remains high, the PWM outputs are set low. The
outputs will remain disabled until the VCC voltage is
removed and reapplied. During normal operation, if the
output voltage falls more than 300 mV below the DAC
setting, the VR_RDY pin will be set low until the output
rises.
PROTECTION FEATURES
Undervoltage Lockout
An undervoltage lockout (UVLO) senses the VCC input.
During powerup, the input voltage to the controller is
monitored, and the PWM outputs and the soft-start circuit
are disabled until the input voltage exceeds the threshold
voltage of the UVLO comparator. The UVLO comparator
incorporates hysteresis to avoid chattering, since VCC is
likely to decrease as soon as the converter initiates
soft-start.
Soft-Start
The NCP5391 incorporates an externally programmable
soft-start. The soft-start circuit works by controlling the
ramp-up of the DAC voltage during powerup. The initial
soft-start pin voltage is 0 V. The soft-start circuitry clamps
the DAC input of the Remote Sense Amplifier to the SS pin
voltage until the SS pin voltage exceeds the DAC setting
minus VID offset. The soft-start pin is pulled to 0 V if there
is an overcurrent shutdown, if the ENABLE pin is low, if
VCC is below the UVLO threshold, or if an overvoltage
condition exists.
The NCP5391 ramps Vcore to 1.1 V at the SS capacitor
charge rate, pauses at 1.1 V for 170 s, reads the VID pins
to determine the DAC setting, then ramps Vcore to the
final DAC setting at the Dynamic VID slew rate of 7.3
mV/s. Typical soft- start sequence is shown in the
following graph.
Overcurrent Shutdown
A programmable overcurrent function is incorporated
within the IC. A comparator and latch makeup this
function. The inverting input of the comparator is
connected to the ILIM pin. The voltage at this pin sets the
maximum output current the converter can produce. The
ROSC pin provides a convenient and accurate reference
voltage from which a resistor divider can create the
overcurrent setpoint voltage. Although not actually
disabled, tying the ILIM pin directly to the ROSC pin sets
the limit above useful levels – effectively disabling
overcurrent shutdown. The comparator noninverting input
2.4
2.2
2.0
VID Setting
VOLTAGE
1.8
Boot Voltage
1.6
1.4
1.2
1.0
Boot
Dwell Time
0.8
0.6
Vcore Voltage
SS Pin Voltage
0.4
0.2
0
NCP5391
Internal Dynamic
VID Rate Limit
TIME
0
Figure 7. Typical VR11 Soft-Start Sequence to Vcore = 1.3 V
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18
NCP5391
APPLICATION INFORMATION
lights should light up to match the VTT tool VID
setting.
15. Set the VR_ENABLE DIP switch up to start the
NCP5391.
16. Check that the output voltage is about 19 mV
below the VID setting.
The NCP5391 is a high performance multiphase
controller optimized to meet the Intel VR11 Specifications.
The demo board for the NCP5391 is available by request.
It is configured as a three phase solution with decoupling
designed to provide a 1.0 m load line under a 50 A step
load. A schematic is available upon request from ON
Semiconductor.
Step Load Testing
The VTT tool is used to generate the high di/dt step load.
Select the dynamic loading option in the VTT test tool
software. Set the desired step load size, frequency, duty,
and slew rate. See Figures 8 and 9.
Startup Procedure
The demo board comes with a Socket 775 and requires
an Intel dynamic load tool (VTT Tool) available through a
third party supplier, Cascade Systems. The web page is
http://www.cascadesystems.net/.
Start by installing the test tool software. It's best to power
the test tool from a separate ATX power supply. The test
tool should be set to a valid VID code of 0.5 V or above
in-order for the controller to start. Consult the VTT help
manual for more detailed instructions.
VOUT
Startup Sequence
1. Make sure the VTT software is installed.
2. Powerup the PC or Laptop do not start the VTT
software.
3. Insert the VTT Test Tool adapter into the socket
and lock it down.
4. Insert the socket saver pin field into the bottom of
the VTT test tool.
5. Carefully line up the tool with the socket in the
board and press tool into the board.
6. Connect the scope probe, or DMM to the voltage
sense lines on the test tool. When using a scope
probe it is best to isolate the scope from the AC
ground. Make the ground connection on the scope
probe as short as possible.
7. Connect the first ATX supply to the VTT tool.
8. Powerup the first ATX supply to the VTT tool.
9. Start the VTT tool software in VR11 mode with
the current limit set to 150 A.
10. Using the VTT tool software, select a VID code
that is 0.5 V or above.
11. Connect the second ATX supply to the demo
board.
12. Set the VID DIP switches. All the VID switches
should be up or open.
13. Set the VR_ENABLE DIP switch down or
closed.
14. Start the second ATX supply by turning it on and
setting the PSON DIP switch low. The green VID
Load Current
Figure 8. Typical Step Load Response
VOUT
Load Current
Figure 9. Typical Load Release Event
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19
NCP5391
Dynamic VID Testing
The VTT tool provides for VID stepping based on the
Intel Requirements. Select the Dynamic VID option.
Before enabling the test set the lowest VID to 0.5 V or
greater and set the highest VID to a value that is greater than
the lowest VID selection, then enable the test. See Figures
10 through 12.
Design Methodology
Decoupling the VCC Pin on the IC
An RC input filter is required as shown in the VCC pin to
minimize supply noise on the IC. The resistor should be
sized such that it does not generate a large voltage drop
between the 12 V supply and the IC. See the schematic
values.
Understanding Soft-Start
The controller will ramp to the 1.1 V, with a pause to
capture the VID code then resume ramping to target value
based on an internal slew rate limit. See Figure 13. The
controller is designed to regulate to the voltage on the SS
pin until it reaches the internal DAC voltage. The soft-start
cap sets the initial ramp rate using a typical 5.0 A current.
The typical value to use for the soft-start cap (SS), is
typically set to 0.01 F. This results in a ramp time to 1.1 V
of 2.2 ms based on equation 1.
dt
Css ^ iss ss
dvss
1.1·V + dvss and i + 5·A
ss
2.2·ms
dtss
Figure 10. 1.6 to 0.5 Dynamic VID Response
Css + 0.01·F
Figure 11. Dynamic VID Settling Time Rising
Figure 13. VR11 Startup
Figure 12. Dynamic VID Settling Time Falling
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20
(eq. 1)
NCP5391
Programming the Current Limit and the Oscillator
Frequency
This equation is valid for the individual phase frequency in
both three and four phase mode.
The demo board is set for an operating frequency of
approximately 300 kHz. The OSC pin provides a 2.0 V
reference voltage which is divided down with a resistor
divider and fed into the current limit pin ILIM. Calculate
the total series resistance to set the frequency and then
calculate the individual values for current limit divider.
The series resistors RLIM1 and RLIM2 sink current to
ground. This current is internally mirrored into a capacitor
to create an oscillator. The period is proportional to the
resistance and frequency is inversely proportional to the
resistance. The resistance may be estimated by equation 2.
9
32.36k ^ 10.14 10 * 1440
300·k
(eq. 3)
3 Phase Mode
9
ROSC + 9.711 10 * 1111
Frequency
100
100
90
90
80
80
70
70
FOSC (2, Measured)
60
ROSC (k)
ROSC (k)
(eq. 2)
2 Phase Mode
9
ROSC + 10.14 10 * 1440
Frequency
50
40
FOSC (3, Calculated)
60
50
40
30
30
20
20
FOSC (2, Calculated)
10
FOSC (3, Measured)
10
0
0
0
200
400
600
FOSC (kHz)
800
0
1000
200
400
600
800
1000
FOSC (kHz)
Figure 14. ROSC vs. 2-Phase Mode
Figure 15. ROSC vs. 3-Phase Mode
The current limit function is based on the total sensed
current of all phases multiplied by a gain of 5.94. DCR
sensed inductor current is function of the winding
temperature. The best approach is to set the maximum
current limit based on the expected average maximum
temperature of the inductor windings.
DCRTmax + DCR25C·
(1 ) 0.00393·C- 1(TTmax- 25·C))
Calculate the current limit voltage:
ǒ
VILIMIT ^ 5.94· IMIN_OCP·DCRTmax )
ǓǓ * 0.02
ǒ
DCR50C·Vout
· Vin- Vout * (N- 1)· Vout
L
L
2·Vin·Fs
(eq. 4)
(eq. 5)
Solve for the individual resistors:
V
·ROSC
RLIM2 + ILIMIT
2·V
RLIM1 + ROSC- RLIM2
(eq. 6)
Final Equation for the Current Limit Threshold
ILIMIT(Tinductor) ^
2·V·RLIM2 Ǔ
ǒRLIM1)RLIM2
) 0.02
5.94·(DCR25C·(1 ) 0.00393·C- 1(TInductor- 25·C)))
*
ǒ
Ǔ
Vout · Vin- Vout * (N- 1)· Vout
2·Vin·Fs
L
L
(eq. 7)
Inductor Selection
When using inductor current sensing it is recommended
that the inductor does not saturate by more than 10% at
maximum load. The inductor also must not go into hard
saturation before current limit trips. The demo board includes
a four phase output filter using the T50- 8 core from
Micrometals with 4turns and a DCR target of 0.75 m @
25°C. Smaller DCR values can be used, however, current
sharing accuracy and droop accuracy decrease as DCR
decreases. Use the excel spreadsheet for regulation accuracy
calculations for a specific value of DCR.
The inductors on the demo board have a DCR at 25°C of
0.75 m. Selecting the closest available values of 16.9 k
for RLIM1 and 15.8 k for RLIM2 yield a nominal
operating frequency of 305 kHz and an approximate
current limit of 180 A at 100°C. The total sensed current
can be observed as a scaled voltage at the VDRP pin added
to a positive, no-load offset of approximately 1.3 V.
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21
NCP5391
Inductor Current Sense Compensation
The NCP5391 uses the inductor current sensing method.
This method uses an RC filter to cancel out the inductance
of the inductor and recover the voltage that is the result of
Rsense(T) +
the current flowing through the inductor's DCR. This is
done by matching the RC time constant of the current sense
filter to the L/DCR time constant. The first cut approach is
to use a 0.47 F capacitor for C and then solve for R.
L
0.47·F·DCR25C·(1 ) 0.00393·C- 1·(T- 25·C))
(eq. 8)
inductor temperature final selection of R is best done
experimentally on the bench by monitoring the Vdroop pin
and performing a step load test on the actual solution.
It is desirable to keep the Rsense resistor value below
1.0 k whenever possible by increasing the capacitor values
in the inductor compensation network. The bias current
flowing out of the current sense pins is approximately
100 nA. This current flows through the current sense
resistor and creates an offset at the capacitor which will
appear as a load current at the Vdroop pin. A 1.0 k resistor
will keep this offset at the droop pin below 2.5 mV.
Figure 16.
Simple Average PSPICE Model
A simple state average model shown in Figure 17 can be
used to determine a stable solution and provide insight into
the control system.
The demoboard inductor measured 350 nH and 0.75 m
at room temp. The actual value used for Rsense was 953
which matches the equation for Rsense at approximately
50C. Because the inductor value is a function of load and
E1
+
+
- E
0 GAIN = 6
12
+
0
+
VRamp_min
1.3 V
+
L
1
DCR
2
(250e-9/3)
1
100 p
CBulk
(560e-6*10)
(0.85e-3/3)
Vin
12
LBRD
2
RBRD
0.75 m
CCer
(22e-6*18)
1Aac
ESRCer
0Adc
(1.5e-3/18)
2
ESRBulk
(7e-3/10)
2
Voff
0
3
RDRP
5.11 k
ESLBulk
(3.5e-9/10)
ESLCer
(1.5e-9/18)
1
1
+
-
I1 = 10
I2 = 110
TD = 10u
TR = 50n
TF = 50n
PW = 40u
PER = 80u
+
-
I2
CH
0
22 p
RF
CF
4.3 k
1.5 n
CFB1
680 p
1E3
Unity
Gain
BW = 15 MHz R6
RFB1
100
RFB
-+
Voff
-
1k
+
1k
C3
10.6 n
1.3
Voffset
-
Vout
+
+
VDAC
-
1.25 V
0
0
Figure 17.
http://onsemi.com
22
0
NCP5391
A complex switching model is available by request
which includes a more detailed board parasitic for this
demo board.
bulk capacitors have an ESR of 7.0 m. Thus the bulk ESR
plus the board impedance is 0.7 m + 0.75 m or
1.45 m. The actual output filter impedance does not drop
to 1.0 m until the ceramic breaks in at over 375 kHz. The
controller must provide some loop gain slightly less than
one out to a frequency in excess 300 kHz. At frequencies
below where the bulk capacitance ESR breaks with the
bulk capacitance, the DC-DC converter must have
sufficiently high gain to control the output impedance
completely. Standard Type-3 compensation works well
with the NCP5391. RFB1 should be kept above 50 for
amplifier stability reasons.
The goal is to compensate the system such that the
resulting gain generates constant output impedance from
DC up to the frequency where the ceramic takes over
holding the impedance below 1.0 m. See the example of
the locations of the poles and zeros that were set to optimize
the model above.
Compensation and Output Filter Design
The values shown on the demo board are a good place to
start for any similar output filter solution. The dynamic
performance can then be adjusted by swapping out various
individual components.
If the required output filter and switching frequency are
significantly different, it's best to use the available PSPICE
models to design the compensation and output filter from
scratch.
The design target for this demo board was 1.0 m out to
2.0 MHz. The phase switching frequency is currently set to
300 kHz. It can easily be seen that the board impedance of
0.75 m between the load and the bulk capacitance has a
large effect on the output filter. In this case the ten 560 F
Zout Open Loop
Zout Closed Loop
Open Loop Gain with Current loop Closed
80
Voltage Loop Compensation Gain
1/(2*PI*CFB1*(RFB1+RFB))
60
20
1/(2*PI*RF*CH)
1/(2*PI*CF*RF)
40
RF/RFB1
RF/RFB
Error Amp
Open Loop
Gain
dB
0
1/(2*PI*(RBRD+ESRBulk)*CBulk)
-20
-40
1/(2*PI*SQRT(ESL_Cer*CCer))
1mOhm
-60
-80
1/(2*PI*CCer*(RBRD+ESRBulk))
-100
100
1000
10000
100000
1000000
10000000
Frequency
Figure 18.
By matching the following equations a good set of starting compensation values can be found for a typical mixed bulk
and ceramic capacitor type output filter.
1
1
+
2·CF·RF
2·(RBRD ) ESRBulk)·CBulk
1
1
+
2·CFBI·(RFBI ) RFB)
2·CCer * (RBRD ) ESRBulk)
http://onsemi.com
23
(eq. 9)
NCP5391
RFB is always set to 1.0 k and RFB1 is usually set to
100 for maximum phase boost. The value of RF is
typically set to 4.0 k.
RRDP determines the target output impedance by the
basic equation:
Vout + Zout + RFB·DCR·5.94
RDRP
Iout
Droop Injection and Thermal Compensation
The VDRP signal is generated by summing the sensed
output currents for each phase and applying a gain of
approximately six. VDRP is externally summed into the
feedback network by the resistor RDRP. This induces an
offset which is proportional to the output current thereby
forcing the controlled resistive output impedance.
RDRP + RFB·DCR·5.94
Zout
(eq.
10)
The value of the inductor's DCR varies with temperature
according to the following equation 10:
DCRTmax + DCR25C·(1 ) 0.00393·C- 1(TTmax- 25·C))
The system can be thermally compensated to cancel this
effect out to a great degree by adding an NTC (negative
temperature coefficient resistor) in parallel with RFB to
reduce the droop gain as the temperature increases. The
NTC device is nonlinear. Putting a resistor in series with the
(eq. 11)
NTC helps make the device appear more linear with
temperature. The series resistor is split and inserted on both
sides of the NTC to reduce noise injection into the feedback
loop. The recommended value for RISO1 and RISO2 is
approximately 1.0 k.
The output impedance varies with inductor temperature by the equation:
Zout(T) +
RFB·DCR25C·(1 ) 0.00393·C- 1(T max - 25C))·5.94
Rdroop
(eq. 12)
By including the NTC RT2 and the series isolation resistors the new equation becomes:
Zout(T) +
RFB·(RISO1)RT2(T))RISO2)
·DCR25C·(1
RFB)RISO1)RT2(T))RISO2
1 Ǔƫ
ƪǒ2731) TǓ * ǒ298
(eq. 13)
Rdroop
OVP
The overvoltage protection threshold is not adjustable.
OVP protection is enabled as soon as soft-start begins and
is disabled when the part is disabled. When OVP is tripped,
the controller commands all four gate drivers to enable
their low side MOSFETs, and VR_RDY transitions low.
The OVP is non-latching and auto recovers. The OVP
circuit monitors the output of DIFFOUT. If the DIFFOUT
signal reaches 180 mV above the nominal 1.3 V offset the
OVP will trip. The DIFFOUT signal is the difference
between the output voltage and the DAC voltage plus the
1.3 V internal offset. This results in the OVP tracking the
DAC voltage even during a dynamic change in the VID
setting during operation.
The typical equation of a NTC is based on a curve fit
equation 13.
RT2(T) + RT225C·e
) 0.00393·C- 1(T max - 25C))·5.94
(eq. 14)
The demo board is populated with a 10 k NTC with a
Beta of 4300. Figure 19 shows the uncompensated and
compensated output impedance versus temperature.
Gate Driver and MOSFET Selection
ON Semiconductor provides the companion gate driver
IC (NCP3418B). The NCP3418B driver is optimized to
work with a range of MOSFETs commonly used in CPU
applications.
The NCP3418B provides special
functionality and is required for the high performance
dynamic VID operation of the part. Contact your local
ON Semiconductor applications engineer for MOSFET
recommendations.
Figure 19. Uncompensated and Compensated Output
Impedance vs. Temperature
ON Semiconductor provides an excel spreadsheet to
help with the selection of the NTC. The actual selection of
the NTC will be effected by the location of the output
inductor with respect to the NTC and airflow, and should
be verified with an actual system thermal solution.
Board Stack-Up
The demo board follows the recommended Intel
Stack-up and copper thickness as shown.
http://onsemi.com
24
NCP5391
Figure 20.
Board Layout
A complete Allegro ATX and BTX demo board layout
file and schematics are available by request at
www.onsemi.com and can be viewed using the Allegro
Free Physical Viewer 15.x from the Cadence website
http://www.cadence.com/.
Close attention should be paid to the routing of the sense
traces and control lines that propagate away from the
controller IC. Routing should follow the demo board
example. For further information or layout review contact
ON Semiconductor.
The products described herein (NCP5391/D), may be covered by one or more of the following U.S. patent; 7057381. There may be other patents pending.
Pentium is a registered trademark of Intel Corporation.
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25
MECHANICAL CASE OUTLINE
PACKAGE DIMENSIONS
QFN32 5x5, 0.5P
CASE 488AM
ISSUE A
1 32
SCALE 2:1
A
D
PIN ONE
LOCATION
ÉÉ
ÉÉ
NOTES:
1. DIMENSIONS AND TOLERANCING PER
ASME Y14.5M, 1994.
2. CONTROLLING DIMENSION: MILLIMETERS.
3. DIMENSION b APPLIES TO PLATED
TERMINAL AND IS MEASURED BETWEEN
0.15 AND 0.30MM FROM THE TERMINAL TIP.
4. COPLANARITY APPLIES TO THE EXPOSED
PAD AS WELL AS THE TERMINALS.
L
L
B
DATE 23 OCT 2013
L1
DETAIL A
ALTERNATE TERMINAL
CONSTRUCTIONS
E
DIM
A
A1
A3
b
D
D2
E
E2
e
K
L
L1
0.15 C
0.15 C
EXPOSED Cu
A
DETAIL B
0.10 C
(A3)
A1
0.08 C
DETAIL A
9
32X
L
ALTERNATE
CONSTRUCTION
GENERIC
MARKING DIAGRAM*
K
D2
1
XXXXXXXX
XXXXXXXX
AWLYYWWG
G
17
8
MOLD CMPD
DETAIL B
SEATING
PLANE
C
SIDE VIEW
NOTE 4
ÉÉ
ÉÉ
ÇÇ
TOP VIEW
MILLIMETERS
MIN
MAX
0.80
1.00
−−−
0.05
0.20 REF
0.18
0.30
5.00 BSC
2.95
3.25
5.00 BSC
2.95
3.25
0.50 BSC
0.20
−−−
0.30
0.50
−−−
0.15
E2
1
32
25
e
e/2
32X
b
0.10
M
C A B
0.05
M
C
BOTTOM VIEW
XXXXX = Specific Device Code
A
= Assembly Location
WL
= Wafer Lot
YY
= Year
WW
= Work Week
G
= Pb−Free Package
(Note: Microdot may be in either location)
*This information is generic. Please refer
to device data sheet for actual part marking.
Pb−Free indicator, “G” or microdot “ G”,
may or may not be present.
NOTE 3
RECOMMENDED
SOLDERING FOOTPRINT*
5.30
32X
0.63
3.35
3.35 5.30
0.50
PITCH
32X
0.30
DIMENSION: MILLIMETERS
*For additional information on our Pb−Free strategy and soldering
details, please download the ON Semiconductor Soldering and
Mounting Techniques Reference Manual, SOLDERRM/D.
DOCUMENT NUMBER:
DESCRIPTION:
98AON20032D
QFN32 5x5 0.5P
Electronic versions are uncontrolled except when accessed directly from the Document Repository.
Printed versions are uncontrolled except when stamped “CONTROLLED COPY” in red.
PAGE 1 OF 1
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PUBLICATION ORDERING INFORMATION
LITERATURE FULFILLMENT:
Email Requests to: orderlit@onsemi.com
onsemi Website: www.onsemi.com
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TECHNICAL SUPPORT
North American Technical Support:
Voice Mail: 1 800−282−9855 Toll Free USA/Canada
Phone: 011 421 33 790 2910
Europe, Middle East and Africa Technical Support:
Phone: 00421 33 790 2910
For additional information, please contact your local Sales Representative
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