NCP1608BOOSTGEVB
NCP1608 100 W Boost
Evaluation Board User's
Manual
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EVAL BOARD USER’S MANUAL
Introduction
The NCP1608 is a voltage mode power factor correction
(PFC) controller designed to implement converters to
comply with line current harmonic regulations. The device
operates in critical conduction mode (CrM) for optimal
performance in applications up to 350 W. Its voltage mode
scheme enables it to obtain near unity power factor (PF)
without the need for a line-sensing network. The output
voltage is accurately controlled with an integrated high
precision transconductance error amplifier. The controller
also implements a comprehensive set of safety features that
simplify system design.
This application note describes the design and
implementation of a 400 V, 100 W, CrM boost PFC
converter using the NCP1608. The converter exhibits high
PF, low standby power dissipation, high active mode
efficiency, and a variety of protection features.
Most electronic ballasts and switch−mode power supplies
(SMPS) use a diode bridge rectifier and a bulk storage
capacitor to produce a dc voltage from the utility ac line.
This causes a non-sinusoidal current consumption and
increases the stress on the power delivery infrastructure.
Government regulations and utility requirements mandate
control over line current harmonic content. Active PFC
circuits are the most popular method to comply with these
harmonic content requirements. System solutions consist of
connecting a PFC pre−converter between the rectifier bridge
and the bulk capacitor (Figure 1). The boost converter is the
most popular topology for active PF correction. It produces
a constant output voltage and consumes a sinusoidal input
current from the line.
PFC Pre−Converter
Rectifiers
AC Line
The Need for PFC
+
High
Frequency
Bypass
Capacitor
Converter
+
NCP1608
Bulk
Storage
Capacitor
Load
Figure 1. Active PFC Stage with the NCP1608
Basic Operation of a CrM Boost Converter
operation. This control method causes the frequency to vary
with the instantaneous line input voltage (Vin) and the output
load. The operation and waveforms of a CrM PFC boost
converter are illustrated in Figure 2. For detailed
information on the operation of a CrM boost converter for
PFC applications, please refer to AND8123 at
www.onsemi.com.
For medium power (< 350 W) applications, CrM is the
preferred control method. CrM operates at the boundary
between discontinuous conduction mode (DCM) and
continuous conduction mode (CCM). In CrM, the drive on
time begins when the inductor current reaches zero.
CrM combines the reduced peak current of CCM
operation with the zero current switching of DCM
© Semiconductor Components Industries, LLC, 2012
November, 2012 − Rev. 1
1
Publication Order Number:
EVBUM2162/D
NCP1608BOOSTGEVB
Diode Bridge
+
Vin
IL
+
L
+
AC Line
Diode Bridge
IL
Vdrain
Vin
Vdrain
L
+
AC Line
+
Vout
−
−
The power switch is ON
The power switch is OFF
With the power switch voltage being about zero, the
input voltage is applied across the inductor. The inductor current linearly increases with a (Vin/L) slope.
Inductor
Current
The inductor current flows through the diode. The inductor
voltage is (Vout − Vin) and the inductor current linearly decays
with a (Vout − Vin)/L slope.
(Vout − Vin)/L
Vin/L
Critical Conduction Mode:
Next current cycle starts
when the core is reset.
IL(peak)
Vdrain
Vout
Vin
If next cycle does not start
then Vdrain rings towards Vin
Figure 2. Schematic and Waveforms of an Ideal CrM Boost Converter
Features of the NCP1608
The NCP1608 is an excellent controller for robust
medium power CrM boost PFC applications due to its
integrated safety features, low impedance driver, high
precision error amplifier, and low standby current
consumption.
Vin
For detailed information on the operation of the
NCP1608, please refer to NCP1608/D at www.onsemi.com.
A CrM boost pre-converter featuring the NCP1608 is
shown in Figure 3.
L
Vout
D
NB:NZCD
LOAD
(Ballast,
SMPS, etc.)
RZCD
+
AC Line
EMI
Filter
Cin
Rout1
1
2
3
Rout2
CCOMP
4
Ct
NCP1608
FB
VCC
Control DRV
Ct
GND
CS
ZCD
VCC
8
7
+
M
Cbulk
6
5
Rsense
Figure 3. CrM Boost PFC Stage Featuring the NCP1608
A combination of resistors and capacitors connected
between the Control and ground pins forms a compensation
network that limits the bandwidth of the converter. For high
PF, the bandwidth is set to less than 20 Hz. A capacitor
connected to the Ct pin sets the maximum on time. The CS
pin provides cycle−by−cycle overcurrent protection. The
The FB pin senses the boost output voltage through the
resistor divider formed by Rout1 and Rout2. The FB pin
includes overvoltage protection (OVP), undervoltage
protection (UVP), and floating pin protection (FPP). This
pin is the input to the error amplifier. The output of the error
amplifier is the Control pin.
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NCP1608BOOSTGEVB
The VCC pin is the supply pin of the controller. When VCC
is less than the turn on voltage (VCC(on)), the current
consumption of the device is less than 35 mA. This results in
fast startup times and reduced standby power losses.
internal comparator compares the voltage developed across
Rsense (VCS) to an internal reference (VILIM). The driver
turns off when VCS reaches VILIM. The ZCD pin senses the
demagnetization of the boost inductor to turn on the drive.
The drive on time begins after the ZCD pin voltage (VZCD)
exceeds VZCD(ARM) and then decreases to less than
VZCD(TRIG). A resistor in series with the ZCD winding
limits the ZCD pin current.
The NCP1608 features a powerful output driver on the
DRV pin. The driver is capable of switching the gates of
large MOSFETs efficiently because of its low source and
sink impedances. The driver includes active and passive
pull−down circuits to prevent the output from floating high
when the NCP1608 is disabled.
Design Procedure
The design of a CrM boost PFC converter is discussed in
many ON Semiconductor application notes. Table 1 lists
some examples.
This application note describes the design procedure for
a 400 V, 100 W converter using the features of the NCP1608.
A dedicated NCP1608 design tool that enables users to
determine component values quickly is available at
www.onsemi.com.
Table 1. Additional Resources for the Design and Understanding of CrM Boost PFC Circuits Available at
www.onsemi.com.
AND8123
Power Factor Correction Stages Operating in Critical Conduction Mode
AND8016
Design of Power Factor Correction Circuits Using the MC33260
AND8154
NCP1230 90 W, Universal Input Adapter Power Supply with Active PFC
HBD853
Power Factor Correction Handbook
DESIGN STEP 1: Define the Required Parameters
The converter parameters are shown in Table 2.
Table 2. CONVERTER PARAMETERS
Parameter Name
Symbol
Value
Units
Minimum Line Input Voltage
VacLL
85
Vac
Maximum Line Input Voltage
VacHL
265
Vac
Minimum Line Frequency
fline(MIN)
47
Hz
Maximum Line Frequency
fline(MAX)
63
Hz
Output Voltage
Vout
400
V
Full Load Output Current
Iout
250
mA
Full Load Output Power
Pout
100
W
Maximum Output Voltage
Vout(MAX)
440
V
Minimum Switching Frequency
fSW(MIN)
40
kHz
Minimum Full Load Efficiency
h
92
%
PF
0.9
−
Minimum Full Load Power Factor
DESIGN STEP 2: Calculate the Boost Inductor
Where L LL is the inductor value calculated at Vac LL.
The value of the boost inductor (L) is calculated using
Equation 1:
Lv
ǒ
Ǔ
V out
Vac 2 @
* Vac @ h
Ǹ2
Ǹ2 @ V @ P @ f
out
out
SW(MIN)
ǒ
(eq. 1)
Where L HL is the inductor value calculated at Vac HL.
A value of 400 mH is selected. The inductance tolerance
is ±15%. The maximum inductance (LMAX) value is
460 mH. Equation 2 is used to calculate the minimum
frequency at full load.
To ensure that the switching frequency exceeds the
minimum frequency, L is calculated at both the minimum
and maximum rms input line voltage:
ǒ
Ǔ
265 2 @ 400 * 265 @ 0.92
Ǹ2
L HL v
+ 509 mH
Ǹ2 @ 400 @ 100 @ 40 k
Ǔ
85 2 @ 400 * 85 @ 0.92
Ǹ2
L LL v
+ 581 mH
Ǹ2 @ 400 @ 100 @ 40 k
f SW +
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3
ǒ
Ǹ2 @ Vac
Vac 2 @ h
@ 1*
2 @ L MAX @ P out
V out
Ǔ
(eq. 2)
NCP1608BOOSTGEVB
f SW(LL) +
f SW(HL) +
ǒ
ǒ
Ǔ
Ǔ
MOSFET Conduction
Diode Conduction
85 2 @ 0.92 @ 1 * Ǹ2 @ 85 + 50.5 kHz
400
2 @ 460 m @ 100
265 2 @ 0.92 @ 1 * Ǹ2 @ 265 + 44.3 kHz
400
2 @ 460 m @ 100
IL(peak)
DESIGN STEP 3: Size the Ct Capacitor
t on(MAX) +
2 @ 460 m @ 100
+ 13.8 ms
0.92 @ 85 2
(eq. 3)
Ct w
h @ Vac LL 2 @ V Ct(MAX)
VZCD(WIND),off
0V
VZCD(WIND),on
VZCD
VCL(POS)
VZCD(ARM)
VZCD(TRIG)
VCL(NEG)
(eq. 4)
ton
Where Icharge and VCt(MAX) are specified in the NCP1608
datasheet. To ensure that the controller sets the maximum on
time to a value sufficient to deliver the required output
power, the maximum Icharge and the minimum VCt(MAX)
values are used in the calculations for Ct.
From the NCP1608 datasheet:
− VCt(MAX) = 4.775 V (minimum)
To activate the ZCD detector of the NCP1608, the ZCD
turns ratio is sized such that at least VZCD(ARM) (1.55 V
maximum) is applied to the ZCD pin during all operating
conditions (see Figure 4). The boost winding to ZCD
winding turns ratio (N = NB:NZCD) is calculated using
Equation 5.
Nv
400 * ǒǸ2 @ 265Ǔ
1.55
R ZCD w
Ǹ2 @ Vac
HL
I ZCD(MAX) @ N
R ZCD w
Ǹ2 @ 265
+ 3.75 kW
10 m @ 10
(eq. 6)
The value of RZCD and the parasitic capacitance of the
ZCD pin determine when the ZCD winding signal is
detected and the drive turn on begins. A large RZCD value
creates a long delay before detecting the ZCD event. In this
case, the controller operates in DCM and the PF is reduced.
If the RZCD value is too small, the drive turns on when the
drain voltage is high and efficiency is reduced. A popular
strategy for selecting RZCD is to use the RZCD value that
achieves minimum drain voltage turn on. This value is found
experimentally.
During the delay caused by RZCD and the ZCD pin
capacitance, the equivalent drain capacitance (CEQ(drain))
discharges through the path shown in Figure 5.
DESIGN STEP 4: Determine the ZCD Turns Ratio
V ZCD(ARM)
RZCD
Delay
A turns ratio of 10 is selected for this design. RZCD is
connected between the ZCD winding and the ZCD pin to
limit the ZCD pin current. This current must be limited
below 10 mA. RZCD is calculated using Equation 6:
A normalized value of 1 nF (±10%) provides sufficient
margin. A value of 1.22 nF is selected for Total Harmonic
Distortion (THD) reduction (see the Additional THD
Reduction section of this application note for more
information).
V out * ǒǸ2 @ Vac HLǓ
toff
Figure 4. Realistic CrM Waveforms Using a ZCD
Winding with RZCD and the ZCD Pin Capacitance
2 @ 100 @ 460 m @ 297 m
+ 860 pF
0.92 @ 85 2 @ 4.775
Nv
0V
tdiode
TSW
− Icharge = 297 mA (maximum)
Ct is equal to:
Ct w
0V
Minimum Voltage Turn on
VZCD(WIND)
Sizing Ct to an excessively large value causes the
application to deliver excessive output power and reduces
the control range at VacHL or low output power. It is
recommended to size the Ct capacitor to a value slightly
larger than that calculated by Equation 4:
2 @ P out @ L MAX @ I charge
0V
Vdrain
Vout
The Ct capacitor is sized to set the maximum on time for
minimum line input voltage and maximum output power.
The maximum on time is calculated using Equation 3:
2 @ L MAX @ P out
h @ Vac LL 2
0A
IL(NEG)
DRV
fSW is equal to 50.5 kHz at VacLL and 44.3 kHz at VacHL.
t on(MAX) +
tz
IL
(eq. 5)
+ 16
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4
NCP1608BOOSTGEVB
L
IL
Iin
AC Line
+
EMI
Filter
Vout
D
+
Cin
CEQ(drain)
Cbulk
Figure 5. Equivalent Drain Capacitance Discharge Path
CEQ(drain) is the combined parasitic capacitances of the
MOSFET, the diode, and the inductor. Cin is charged by the
energy discharged by CEQ(drain). The charging of Cin reverse
biases the bridge rectifier and causes the input current (Iin)
to decrease to zero. The zero input current causes THD to
increase. To reduce THD, the ratio (tz / TSW) is minimized,
where tZ is the period from when IL = 0 A to when the drive
turns on. The ratio (tz / TSW) is inversely proportional to the
square root of L.
R out2 +
R out2 +
Ǔ
4 M @ 4.6 M
+ 25.3 kW
4.6 M @ 400 * 1 * 4 M
2.5
ǒ
ǒ
Rout1 and Rout2 form a resistor divider that scales down
Vout before it is applied to the FB pin. The error amplifier
adjusts the on time of the drive to maintain the FB pin
voltage equal to the error amplifier reference voltage
(VREF). The divider network bias current (Ibias(out))
selection is the first step in the calculation. The divider
network bias current is selected to optimize the tradeoff of
noise immunity and power dissipation. Rout1 is calculated
using the optimized bias current and output voltage using
Equation 7:
V out
I bias(out)
(eq. 8)
Vout
R FB @
* 1 * R out1
VREF
Ǔ
Rout2 is selected as 25.5 kW for this design.
Using the selected resistor, the resulting output voltage is
calculated using Equation 9:
DESIGN STEP 5: Set the FB, OVP, and UVP Levels
R out1 +
ǒ
R out1 @ R FB
V out + V REF @ R out1 @
ǒ
V out + 2.5 @ 4 M @
Ǔ
R out2 ) R FB
)1
R out2 @ R FB
(eq. 9)
Ǔ
25.5 k ) 4.6 M
) 1 + 397 V
25.5 k @ 4.6 M
The low bandwidth of the PFC stage causes overshoots
during transient loads or during startup. The NCP1608
includes an integrated OVP circuit to prevent the output
from exceeding a safe voltage. The OVP circuit compares
VFB to the internal overvoltage detect threshold voltage to
determine if an OVP fault occurs. The OVP detection
voltage is calculated using Equation 10:
(eq. 7)
(eq. 10)
ǒ
A bias current of 100 mA provides an acceptable tradeoff
of power dissipation to noise immunity.
Ǔ
V
R
) R FB
V out(OVP) + OVP @ V REF @ R out1 @ out2
)1
V REF
R out2 @ R FB
R out1 + 400 + 4 MW
100 m
ǒ
V out(OVP) + 1.06 @ 2.5 @ 4 M @
The output voltage signal is delayed before it is applied to
the FB pin due to the time constant set by Rout1 and the FB
pin capacitance. Rout1 must not be sized too large or this
delay may cause overshoots of the OVP detection voltage.
Rout2 is dependent on Vout, Rout1, and the internal
feedback resistor (RFB, shown in the NCP1608 specification
table). Rout2 is calculated using Equation 8:
Ǔ
25.5 k ) 4.6 M
) 1 + 421 V
25.5 k @ 4.6 M
The output capacitor (Cbulk) value is sized to be large
enough so that the peak-to-peak output voltage ripple
(Vripple(peak-peak)) is less than the OVP detection voltage.
Cbulk is calculated using Equation 11:
C bulk w
P out
2 @ p @ V ripple(peak−peak) @ f line @ V out
(eq. 11)
Where fline = 47 Hz is the worst case for the ripple voltage
and Vripple(peak-peak) < 42 V.
C bulk w
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5
100
+ 20 mF
2 @ p @ 42 @ 47 @ 400
NCP1608BOOSTGEVB
DESIGN STEP 6: Size the Power Components
The value of Cbulk is selected as 68 mF to reduce
Vripple(peak-peak) to less than 15 V. This results in a peak
output voltage of 406.25 V, which is less than the peak output
OVP detection voltage (421 V).
The NCP1608 includes undervoltage protection (UVP).
During startup, Cbulk charges to the peak of the ac line
voltage. If Cbulk does not charge to a minimum voltage, the
NCP1608 detects an UVP fault. The UVP detection voltage
is calculated using Equation 12:
ǒ
(eq. 12)
I L(peak) +
25.5 k ) 4.6 M
) 1 + 49 V
25.5 k @ 4.6 M
I L(peak) +
V out(UVP) + V UVP @ R out1 @
ǒ
V out(UVP) + 0.31 @ 4 M @
Ǔ
R out2 ) R FB
)1
R out2 @ R FB
The power components are sized such that there is
sufficient margin to sustain the currents and voltages applied
to them. At minimum line input voltage and maximum
output power the inductor peak current is at the maximum,
which causes the greatest stress to the power components.
The components are referenced in Figure 3.
1. The inductor peak current (IL(peak)) is calculated
using Equation 13:
Ǔ
Ǹ2 @ 2 @ P
out
h @ Vac
Ǹ2 @ 2 @ 100
0.92 @ 85
(eq. 13)
+ 3.62 A
The inductor rms current (IL(RMS)) is calculated using
Equation 14:
The UVP feature protects against open loop conditions in
the feedback loop. If the FB pin is inadvertently floating
(perhaps due to a bad solder joint), the coupling within the
system may cause VFB to be within the regulation range (i.e.
VUVP < VFB < VREF). The controller responds by delivering
maximum power. The output voltage increases and over
stresses the components. The NCP1608 includes a feature to
protect the system if FB is floating. The internal pull-down
resistor (RFB) ensures that VFB is below the UVP threshold
if the FB pin is floating.
If the FB pin floats during operation, VFB begins
decreasing from VREF. The rate of decrease depends on RFB
and the FB pin parasitic capacitance. As VFB decreases,
VControl increases, which causes the on time to increase until
VFB < VUVP. When VFB < VUVP, the UVP fault is detected
and the controller is disabled. The sequence is depicted in
Figure 6.
I L(RMS) +
2 @ P out
Ǹ3 @ Vac @ h
I L(RMS) +
2 @ 100
+ 1.48 A
Ǹ3 @ 85 @ 0.92
(eq. 14)
2. The output diode (D) rms current (ID(RMS)) is
calculated using Equation 15:
I D(RMS) + 4 @
3
ǸǸ2p@ 2 @
I D(RMS) + 4 @
3
ǸǸ2p@ 2 @
P out
(eq. 15)
h @ ǸVac @ V out
100
+ 0.75 A
0.92 @ Ǹ85 @ 400
The diode maximum voltage is equal to VOVP (421 V)
plus the overshoot caused by parasitic contributions. For this
evaluation board, the maximum voltage is 450 V. A 600 V
diode provides a 25% derating factor. The MUR460
(4 A/600 V) diode is selected for this design.
3. The MOSFET (M) rms current (IM(RMS)) is
calculated using Equation 16:
VCC
VCC(on)
VCC(off)
Ǹ ǒ
ǓǸ ǒ
ǒ
Ǔ
P out
I M(RMS) + 2 @
@
Ǹ3
h @ Vac
Vout
Vout
Loop is Opened
ǒ
VFB
100
I M(RMS) + 2 @
@
Ǹ3 0.92 @ 85
VREF
VUVP
1*
Ǹ2 @ 8 @ Vac
3 @ p @ V out
Ǔ
Ǔ
(eq. 16)
Ǹ2 @ 8 @ 85
1−
+1.27 A
3 @ p @ 400
The MOSFET maximum voltage is equal to VOVP
(421 V) plus the overshoot caused by parasitic
contributions. For this evaluation board, the maximum
voltage is 450 V. A 560 V MOSFET provides a 20% derating
factor. The SPP12N50C3 (11.6 A/560 V) MOSFET is
selected for this design.
4. The current sense resistor (Rsense) limits the
maximum inductor peak current of the MOSFET
and is calculated using Equation 17:
VControl
VEAH
Ct(offset)
UVP Fault
Figure 6. UVP Operation if Loop is Opened During
Operation
R sense +
V ILIM
I L(peak)
(eq. 17)
Where VILIM is specified in the NCP1608 datasheet.
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NCP1608BOOSTGEVB
Once VCC reaches VCC(on), the internal references and
logic of the NCP1608 turn on. The NCP1608 includes an
undervoltage lockout (UVLO) feature that ensures that the
NCP1068 remains enabled unless VCC decreases to less than
VCC(off). This hysteresis ensures sufficient time for another
supply to power VCC.
The ZCD winding is a possible solution, but the voltage
induced on the winding may be less than the required
voltage. An alternative is to implement a charge pump to
supply VCC. A schematic is illustrated in Figure 7.
R sense + 0.5 + 0.138 W
3.62
The current sense resistor is selected as 0.125 W for
decreased power dissipation. The resulting maximum
inductor peak current is 4 A. Since the MOSFET continuous
current rating is 7 A (for TC = 100°C as specified in the
manufacturer’s datasheet) and the inductor saturation
current is 4.7 A, the maximum peak inductor current of 4 A
is sufficiently low.
The power dissipated by Rsense is calculated using
Equation 18:
PR
PR
sense
sense
+ I M(RMS) 2 @ R sense
RZCD
(eq. 18)
C3
+ 1.27 2 @ 0.125 + 0.202 W
R1
5. The output capacitor (Cbulk) rms current is
calculated using Equation 19:
I C(RMS) +
I C(RMS) +
Ǹ
Ǹ2 @ 32 @ P 2
out
* I load(RMS) 2
9 @ p @ Vac @ V out @ h 2
Ǹ
+
(eq. 19)
Cin
Ǹ2@Vac
* I CC(startup)
Rstart
DV C3 +
CVcc
V out
* V CC
N
(eq. 21)
The current that charges CVcc is calculated using
Equation 22:
ǒ
(eq. 22)
Ǔ
V out
I AUX + C3 @ f SW @ DV C3 + C3 @ f SW @
* V CC
N
For off−line ac-dc applications that require PFC, a 2-stage
approach is typically used. The first stage is the CrM boost
PFC. This supplies the 2nd stage, which is traditionally an
isolated flyback or forward converter. This solution is
cost−effective and exhibits excellent performance. During
low output power conditions the PFC stage is not required
and reduces efficiency. Advanced controllers, such as the
NCP1230 and NCP1381 detect the low output power
condition and shut down the PFC stage by removing
PFC(VCC) (Figure 8).
(eq. 20)
Where ICC(startup) = 24 mA (typical).
If CVcc is selected as a 47 mF capacitor and Rstart is
selected as 660 kW, tstartup is equal to:
t startup +
+
C3 stores the energy for the charge pump. R1 limits the
current by reducing the rate of voltage change. DAUX supplies
current to C3 when its cathode is negative. When its cathode
is positive it limits the maximum voltage applied to VCC.
The voltage change across C3 over one period is
calculated using Equation 21:
The typical method to charge the VCC capacitor (CVcc) to
VCC(on) is to connect a resistor between Vin and VCC. The
low startup current consumption of the NCP1608 enables
most of the resistor current to charge CVcc during startup.
The low startup current consumption enables faster startup
times and reduces standby power dissipation. The startup
time (tstartup) is approximated with Equation 20:
@ V CC(on)
NCP1608
8
FB
VCC
2
7
Control DRV
3
6
Ct
GND
4
5
CS
ZCD
Figure 7. The ZCD Winding Supplies VCC using a
Charge Pump Circuit
DESIGN STEP 7: Supply VCC Voltage
CC
D1
1
The value of Cbulk is calculated in Step 5 to ensure a ripple
voltage that is sufficiently low to not trigger OVP. The value
of Cbulk may need to be increased so that the rms current
does not exceed the ratings of Cbulk.
The voltage rating of Cbulk is required to be greater than
Vout(OVP). Since Vout(OVP) is 421 V, Cbulk is selected to have
a voltage rating of 450 V.
CV
DAUX
Rstart
Ǹ2 @ 32 @ 100 2
* 0.25 2 + 0.7 A
9 @ p @ 85 @ 400 @ 0.92 2
t startup +
IAUX
47 m @ 12
+ 3.57 s
Ǹ2@85
* 24 m
660 k
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NCP1608BOOSTGEVB
D
+
1
8
2
7
3
4
PFC(VCC)
1
8
2
7
6
3
6
5
4
5
+
Cbulk
VCC
+
+
+
+
−
NCP1230
NCP1608
Figure 8. Using the SMPS Controller to Supply Power to the NCP1608
DESIGN STEP 8: Limit the Inrush Current
Dbypass
The sudden application of the ac line voltage to the PFC
pre−converter causes an inrush current and a resonant
voltage overshoot that is several times the normal value.
Resizing the power components to handle inrush current and
a resonant voltage overshoot is cost prohibitive.
1. External Inrush Current Limiting Resistor
A NTC (negative temperature coefficient) thermistor
connected in series with the diode limits the inrush current
(Figure 9). The resistance of the NTC decreases from a few
ohms to a few milliohms as the NTC is heated by the I2R
power dissipation. However, an NTC resistor may not be
sufficient to protect the inductor and Cbulk from inrush
current during a brief interruption of the ac line voltage, such
as during ac line dropout and recovery.
2. Startup Bypass Rectifier
A rectifier is connected from Vin to Vout (Figure 10). This
bypasses the inductor and diverts the startup current directly
to Cbulk. Cbulk is charged to the peak ac line voltage without
resonant overshoot and without excessive inductor current.
After startup, Dbypass is reverse biased and does not interfere
with the boost converter.
Vin
+
NCP1608
Figure 10. Use a Second Diode to Route the
Inrush Current Away from the Inductor
DESIGN STEP 9: Develop the Compensation Network
The pre−converter is compensated to ensure stability over
the input voltage and output power range. To compensate the
loop, a compensation network is connected between the
Control and ground pins. To ensure high PF, the bandwidth
of the loop is set below 20 Hz. A type 2 compensation
network is selected for this design to increase the phase
margin. The type 2 compensation network is shown in
Figure 11.
Vout
NTC
Vin
Rout1
Vac
E/A
FB
Vout
RFB
Rout2
Figure 9. Use a NTC to Limit the Inrush Current
Through the Inductor
−
+
+
+
NCP1608
Vout
Vac
gm
VREF
Control
CCOMP
RCOMP1
VControl
Compensation
Network
CCOMP1
Figure 11. Type 2 Compensation Network
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NCP1608BOOSTGEVB
The type 2 network is composed of CCOMP, CCOMP1, and
RCOMP1. CCOMP1 sets the crossover frequency (fCROSS) and
is calculated using Equation 23:
gm
C COMP1 +
2 @ p @ f CROSS
(eq. 23)
1
2 @ p @ f zero @ C COMP
R COMP1 +
1
+ 19.3 kW
2 @ p @ 2.5 @ 3.3 m
(eq. 24)
RCOMP1 is selected as 20 kW.
CCOMP is used to filter high frequency noise and is set to
between 1/10 and 1/5 of CCOMP1. For this design, CCOMP is
selected to be 1/5 of CCOMP1.
For this design, fCROSS is set to 5 Hz at the average input
voltage (175 Vac) to decrease THD and gm is specified in the
NCP1608 datasheet:
C COMP1 +
R COMP1 +
ǒǓ
C COMP + 1 @ 3.3 m + 0.66 mF
5
110 m
+ 3.5 mF
2@p@5
CCOMP is selected as 0.68 mF.
The phase margin and crossover frequency change with
the ac line voltage. It is critical that the gain and phase are
measured for all operating conditions. The measurement
setup using a network analyzer is shown in Figure 12.
A normalized value of 3.3 mF is selected, which sets
fCROSS to 5.3 Hz.
The addition of RCOMP1 causes a zero in the loop
response. The zero frequency (fzero) is typically set to half
the crossover frequency, which is 2.5 Hz for this case.
RCOMP1 is calculated using Equation 24:
Ch A
High−Voltage
(> 450 V)
Isolation Probe
Ch B
High−Voltage
(> 450 V)
Isolation Probe
Network Analyzer
D
L
Vout
Isolator
RZCD
+
AC Line
EMI
Filter
1 kW
Rout1
1
Cin
2
Rout2
3
Ct
4
NCP1608
FB
VCC
Control DRV
Ct
GND
CS
ZCD
Load
VCC
8
7
+
M
Cbulk
6
5
CCOMP
Rsense
Figure 12. Gain-Phase Measurement Setup for a Boost PFC Pre−Converter
1. Improve the THD/PF at Maximum Output Power by
Increasing the On Time at the Zero Crossing:
There is a tradeoff of transient response for PF and THD.
The low bandwidth of the feedback loop reduces the Control
pin ripple voltage. The reduction of the Control pin ripple
voltage increases PF and reduces THD, but increases the
magnitude of overshoots and undershoots.
One disadvantage of constant on time CrM control is that
at the zero crossing of the ac line, the instantaneous input
voltage is not large enough to store sufficient energy in the
inductor during the constant on time. Minimal energy is
processed and “zero crossing distortion” is produced as
shown in Figure 13.
Additional THD Reduction
The constant on time architecture of the NCP1608
provides flexibility in optimizing each design.
The following design guidelines provide methods to
further improve PF and THD.
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NCP1608BOOSTGEVB
Vin (100V/div)
Iin (500mA/div)
Vout (10V/div, ac coupled)
Zero
Crossing
Distortion
Figure 13. Full Load Input Current (Vin = 230 Vac 50 Hz, Iout = 250 mA)
reduces the instantaneous input voltage at which the
distortion begins.
This method is implemented by connecting a resistor from
Vin to Ct as shown in Figure 14. The resistor current (ICTUP)
is proportional to the instantaneous line voltage and is
summed with Icharge to increase the charging current of Ct.
ICTUP is maximum at the peak of Vin and is approximately
zero at the zero crossing.
The zero crossing distortion increases the THD and
decreases the PF of the pre-converter. To meet
IEC61000-3-2 requirements, this is generally not an issue as
the NCP1608 reduces input current distortion with sufficient
margin. If improved THD or PF is required, then zero
crossing distortion must be reduced. To reduce the zero
crossing distortion, the on time is increased as the
instantaneous input voltage is decreasing to zero. This
increases the time for the inductor current to build up and
L
Vin
I CTUP +
+
AC Line
Cin
V in
R CTUP
VDD
RCTUP
VControl
PWM
−
+
Icharge
Ct
ton
DRV
Ct
Ct(offset)
Figure 14. .Add RCTUP to Modulate the On Time and Reduce Zero Crossing Distortion
The increased charging current at the peak of Vin enables
the increased sizing of the Ct capacitor without reducing the
control range at VacHL or low output power. The larger value
of the Ct capacitor increases the on time near the zero
crossing and reduces the zero crossing distortion as shown
in Figure 15. This reduces the frequency variation over the
ac line cycle. The tradeoff is that the standby power
dissipation is increased by RCTUP. The designer must
balance the desired THD and PF performance with the
standby power dissipation requirements.
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NCP1608BOOSTGEVB
Vac(t)
with RCTUP
ton
no RCTUP
no RCTUP
fSW
with RCTUP
time
Figure 15. On Time and Switching Frequency With and Without RCTUP
The dependency of THD on RCTUP is illustrated in
Figure 16.
14
Vout
12
VFB
RCTUP = open
Ct = 1 nF
10
THD (%)
Vout
VREF
8
VControl
6
RCTUP = 1.5 MW
Ct = 1.22 nF
4
Ct(offset)
2
0
DRV
85
115
145
175
Vin (Vac)
205
235
265
Figure 16. Dependency of THD on RCTUP
(Iout = 250 mA)
Figure 17. Required On Time Less Than the
Minimum On Time
This sequence increases the input current distortion.
There are two solutions to improve THD/PF at maximum
input voltage or low output current:
1. Properly size the Ct capacitor. As previously
mentioned, the Ct capacitor is sized to set the
maximum on time for minimum line input voltage
and maximum output power. Sizing Ct to an
excessively large value reduces the control range
at VacHL or low output power.
2. Compensate for propagation delays. If optimizing
the Ct capacitor does not achieve the desired
performance, then it may be necessary to
compensate for the PWM propagation delay by
connecting a resistor (RCT) in series with Ct.
When the Ct voltage reaches the VControl setpoint,
the PWM comparator sends a signal to end the on
time of the driver as shown in Figure 18.
2. Improve the THD/PF at Maximum Input Voltage or
Low Output Current:
If the required on time at maximum input voltage or low
output current is less than the minimum on time (tPWM), then
DRV pulses must be skipped to prevent excessive power
delivery to the output. This results in the following
sequence:
1. The excessive on time causes VControl to decrease
to Ct(offset).
2. When VControl < Ct(offset), the drive is disabled.
3. The drive is disabled and Vout decreases.
4. As Vout decreases, VControl increases.
5. The sequence repeats. Figure 17 depicts the
sequence:
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NCP1608BOOSTGEVB
VControl
Control
Iswitch
VDD
Icharge
Ct
VCt(off)
RCT
PWM
−
+
Driver
Vgate
DRV
RDRV
DRV
Rsense
Ct(offset)
Ct
Figure 18. Block Diagram of the Propagation Delay Components
A value of RCT = 365 W compensates for the propagation
delays. Figure 20 shows the decrease of THD at VacHL and
low output power by compensating for the propagation delay.
There is a delay (tdelay) from when VCt(off) is reached to
when the MOSFET completely turns off. tdelay is caused by
the propagation delay of the PWM comparator (tPWM) and
the time for the gate voltage of the MOSFET to decrease to
zero (tgate). The delays are illustrated in Figure 19.
50
40
VCt(off)
RCT = 0 W
THD (%)
Ct
tPWM
Vgate
30
DRV Pulse Skipping Begins
20
RCT = 365 W
10
0
25
30
35
Pout (W)
Iswitch
The total delay is calculated using Equation 25:
(eq. 25)
tdelay increases the effective on time of the MOSFET.
If a resistor (RCT) is connected in series with the Ct
capacitor, then the total on time reduction is calculated using
Equation 26:
50
40
(eq. 26)
THD (%)
DV RCT
+ Ct @ R CT
DI RCT
The value of RCT to compensate for the propagation delay
is calculated using Equation 27:
R CT +
t delay
Ct
(eq. 27)
30
DRV Pulse Skipping Begins
RCTUP = open
RCT = 0 W Ct = 1 nF
20
10
The NCP1608 datasheet specifies the maximum tPWM as
130 ns. tgate is a dependent on the gate charge of the
MOSFET and RDRV. For this demo board, the gate delay is
measured as 230 ns.
R CT +
50
Both THD reduction techniques can be combined to
decrease the THD for the entire output power range.
Figure 21 shows the decreased THD at the maximum input
voltage across the output power range by decreasing zero
crossing distortion and by compensating for the propagation
delay.
Figure 19. Turn Off Propagation Delays
Dt on + Ct @
45
Figure 20. Low Output Power THD Reduction with
RCT (Vin = 265 Vac 50 Hz, RCTUP = open, Ct = 1 nF)
tgate
tdelay
t delay + t PWM ) t gate
40
0
RCTUP = 1.5 MW
RCT = 365 W Ct = 1.22 nF
25
35
45
65
55
Pout (W)
75
85
95
Figure 21. THD Reduction with RCTUP and RCT
(Vin = 265 Vac 50 Hz)
360 n
+ 360 W
1n
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NCP1608BOOSTGEVB
Design Results
The completed evaluation board schematic is shown in Figure 22.
Figure 22. NCP1608BOOSTGEVB Evaluation Board Schematic
− The input power, PF, and THD are measured using a
PM3000A power meter
− The output voltage is measured using a HP34401A
multimeter
− The output current is set using a PLZ1003WH electronic
load
− The output current is measured using a HP34401A
multimeter
− The output power is calculated by multiplying the output
voltage and output current
The bill of materials (BOM), layout, and summary of
boost equations are shown in Appendix 1, Appendix 2, and
Appendix 3 respectively. This pre−converter exhibits
excellent THD (Figure 23 and Figure 24), PF (Figure 25),
and efficiency (Figure 26). All measurements are performed
with the following conditions:
− After the board is operated at full load and minimum line
input voltage for 30 minutes
− At an ambient temperature of 25°C, open frame, and
without forced air flow
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NCP1608BOOSTGEVB
0.7
12
0.6
Pout = 50 W
10
THD (%)
HARMONIC CURRENT (A)
14
8
6
Pout = 100 W
4
2
0
80
130
180
230
0.3
IEC61000−3−2 Class D Limits
0.2
0.1
1
3 5 7 9 11 13 15 17 19 21 23 25 27 29 31 33 35 37 39
Nth HARMONIC
Figure 23. THD vs. Input Voltage
Figure 24. Individual Harmonic Current
100
1.00
Pout = 100 W
0.98
98
Pout = 50 W
EFFICIENCY (%)
0.97
0.96
PF
Pin = 75 W
0.4
Vin (Vac)
0.99
0.95
0.94
0.93
96
Pout = 100 W
94
Pout = 50 W
92
0.92
0.91
0.90
0.5
0
280
115 Vac 60 Hz
230 Vac 50 Hz
80
115
150
185
220
255
290
90
80
115
150
185
220
255
Vin (Vac)
Vin (Vac)
Figure 25. PF vs. Input Voltage
Figure 26. Efficiency vs. Input Voltage
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290
NCP1608BOOSTGEVB
Input Current and Output Voltage
The input current and output voltage ripple are shown in
Figure 27. The overvoltage protection is observed by
starting up the pre-converter with no load as shown in
Figure 28. The NCP1608 detects an OVP fault when Vout
reaches 421 V and restarts when Vout decreases to 410 V.
Vin (50V/div)
Iin (1A/div)
Vout (10V/div, ac coupled)
Figure 27. Input Current and Output Voltage Ripple (Vin = 115 Vac 60 Hz, Iout = 250 mA)
VCC (10V/div)
VDRV (10V/div)
Vout (100V/div)
Vin (100V/div)
Figure 28. Startup Transient Showing OVP Detection and Recovery (Vin = 115 Vac 60 Hz, Iout = 0 mA)
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15
NCP1608BOOSTGEVB
Frequency Response
crossover frequency is 2 Hz and the phase margin is 71°.
Figure 30 shows that at maximum input voltage, the
crossover frequency is 10 Hz and the phase margin is 53°.
100
150
80
120
60
90
Phase
GAIN (dB)
40
Phase Margin
20
30
0
0
−20
Gain
−40
−30
−60
−90
−60
−80
−100
60
PHASE (degrees)
The frequency response is measured at the minimum and
maximum input voltages and maximum output power.
Figure 29 shows that at minimum input voltage, the
1
fCROSS
10
FREQUENCY (Hz)
−120
−150
100
Figure 29. Frequency Response Vin = 85 Vac 60 Hz Iout = 250 mA
100
150
80
120
60
GAIN (dB)
40
Phase Margin
20
60
30
0
0
Gain
−20
−30
PHASE (degrees)
90
Phase
−40
−60
−60
−90
−80
−100
−120
1
10
fCROSS
−150
100
FREQUENCY (Hz)
Figure 30. Frequency Response Vin = 265 Vac 50 Hz Iout = 250 mA
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NCP1608BOOSTGEVB
Floating Pin Protection (FPP) Jumper
The evaluation board includes a jumper (J1) between the
FB pin and the feedback network to demonstrate the FPP
feature of the NCP1608. If J1 is removed before applying the
line input voltage, the drive is never enabled as shown in
Figure 31. If J1 is removed during operation, the drive is
disabled as shown in Figure 32. J1 is for FPP evaluation
purposes only and should not be included in manufactured
systems.
Vin (100V/div)
VCC (5V/div)
VDRV (5V/div)
Vout (100V/div)
No DRV Pulses
Figure 31. Startup with Jumper Removed (Vin = 265 Vac 50 Hz, Iout = 0 mA)
t(4ms/div)
Vin (100V/div)
DRV Pulses Stop
Vout (100V/div)
VCC (5V/div)
VDRV (5V/div)
t(8μs/div)
(Zoomed In)
Figure 32. Removing the Jumper During Operation (Vin = 265 Vac 50 Hz, Iout = 250 mA)
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NCP1608BOOSTGEVB
The evaluation board can be configured for THD reduction or power dissipation reduction. Table 3 shows the configuration
results.
Table 3. EVALUATION BOARD CONFIGURATION RESULTS
Efficiency (Pout = 100 W)
THD (Pout = 100 W)
Ct
(RCT = 0 W)
Shutdown Power Dissipation (VFB = 0 V)
115 Vac
230 Vac
115 Vac
RCTUP
(Vin = 265 Vac 50 Hz)
60 Hz
50 Hz
60 Hz
50 Hz
open
1 nF
224 mW
93.5%
95.7%
8.4%
12.5%
1.5 MW
1.22 nF
294 mW
93.5%
95.5%
4.4%
6.2%
Safety Precautions
230 Vac
The evaluation board includes the following unpopulated
footprints to enable user experimentation:
1. CCS to add a decoupling capacitor to the CS pin.
2. CZCD to add a decoupling capacitor to the ZCD
pin.
3. DDRV to add a diode for faster turn off of Q1.
4. DVCC to add a diode to clamp VCC.
5. ROUT2B to add a resistor for a more accurate
output voltage.
6. RS3 to add a resistor for a more accurate inductor
peak current limit or to reduce the heating of the
current sense resistors.
Since the FPP feature is only intended to protect the
system in the case of a floating FB pin, care must be taken
when removing the jumper. Do not attach any wires to the
jumper pins with the jumper removed. Connecting wires
to the FB pin couples excessive noise to the FB pin. This
prevents the correct operation of FPP and causes maximum
power to be delivered to the output. This can cause excessive
voltage to be applied to Cbulk. Always wear proper eye
protection when the jumper is removed.
The jumper is located next to high voltage components.
Do not remove the jumper during operation with bare
fingers or non-insulated metal tools.
Summary
Layout Considerations
A universal input voltage 100 W converter is designed
using the boost topology. The converter is implemented with
the NCP1608. Over the input voltage range and with an
output power of 100 W, the PF, THD, and efficiency are
measured as greater than 0.97, less than 8%, and greater than
92% respectively. The converter complies with
IEC61000−3−2 Class D limits for an input power of 75 W.
The converter is stable over the input voltage range with a
measured phase margin greater than 50 degrees. Finally, the
overvoltage protection and floating pin protection features
protect the converter from excessive output voltage.
The evaluation board is designed to showcase the features
and flexibility of the NCP1608. This design is a guideline
only and does not guarantee performance for any
manufacturing or production purposes.
Careful consideration must be given to the placement of
components during layout of switching power supplies.
Noise generated by the large voltages and currents can be
coupled to the pins of the NCP1608. The following
guidelines reduce the probability of excessive coupling:
1. Place the following components as close as
possible to the NCP1608:
a. Ct capacitor
b. VCC decoupling capacitor
c. Control pin compensation components
2. Minimize trace length, especially for high current
loops.
3. Use wide traces for high current connections.
4. Use a single point ground connection between
power ground and signal ground.
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NCP1608BOOSTGEVB
Table 4. BILL OF MATERIALS*
Manufacturer
Manufacturer Part
Number
Substitution
Allowed
Vishay
KBL06-E4/51
Yes
Radial
EPCOS
B32923C3474M
Yes
5%
1206
TDK
Corporation
C3216C0G1H822J
Yes
68 mF
20%
Radial
United
Chemi−Con
EKXG451ELL680MMN3S
Yes
Capacitor, Ceramic, SMD,
25 V
0.68 mF
10%
1206
TDK
Corporation
C3216X7R1E684K
Yes
1
Capacitor, Ceramic, SMD,
25 V
3.3 mF
10%
1206
TDK
Corporation
C3216X7R1E335K
Yes
CCS,
CZCD
2
Capacitor, Ceramic, SMD
open
−
1206
−
−
−
CIN
1
Capacitor, EMI Suppression,
305 Vac
0.1 mF
20%
Radial
EPCOS
B32921A2104M
Yes
CT1
1
Capacitor, Ceramic, SMD,
50 V
1 nF
10%
1206
Yageo
CC1206KRX7R9BB102
Yes
CT2
1
Capacitor, Ceramic, SMD,
50 V
220 pF
10%
1206
Yageo
CC1206KRX7R9BB221
Yes
CVCC
1
Capacitor, Electrolytic, 25 V
47 mF
20%
Radial
Panasonic
EEU-FC1E470
Yes
CVCC2
1
Capacitor, Ceramic, SMD,
50 V
0.1 mF
10%
1206
Yageo
CC1206KRX7R9BB104
Yes
D1
1
Diode, Switching, 100 V
−
−
SOD123
ON
Semiconductor
MMSD4148T1G
No
DAUX
1
Diode, Zener, 18 V
−
−
SOD123
ON
Semiconductor
MMSZ4705T1G
No
DBOOST
1
Diode, Ultrafast, 4 A, 600 V
−
−
Axial
ON
Semiconductor
MUR460RLG
No
DDRV
1
Diode, Switching
open
−
SOD123
−
−
−
DVCC
1
Diode, Zener
open
−
SOD123
−
−
−
F1
1
Fuse, SMD, 2 A, 600 V
−
−
SMD
Littelfuse
0461002.ER
Yes
J1
1
Header 1 Row of 2, 100 mil
−
−
2.54 mm
3M
929400-01-36-RK
Yes
J2, J3
2
Connector, 156 mil 3 pin
−
−
156 mil
MOLEX
26−60−4030
Yes
L1
1
Inductor, Radial, 4 A
180 mH
10%
Radial
Coilcraft
PCV-2-184-05L
No
L2
1
Line Filter, 2.7 A
4.7 mH
−
Through
Hole
Panasonic
ELF-20N027A
Yes
LBOOST
1
Inductor, NB:NZCD = 10:1
400 mH
−
Custom
Coilcraft
JA4224−AL
No
MECHA−
NICAL
1
Shorting Jumper on J1
−
−
−
3M
929955-06
Yes
MECHA−
NICAL
1
Heatsink
−
−
TO−220
Aavid
590302B03600
Yes
MECHA−
NICAL
1
Screw, Phillips, 4−40, ¼”,
Steel
−
−
−
Building
Fasteners
PMSSS 440 0025 PH
Yes
MECHA−
NICAL
1
Nut, Hex 4−40, Steel
−
−
−
Building
Fasteners
HNSS440
Yes
MECHA−
NICAL
1
Shoulder Washer #4, Nylon
−
−
−
Keystone
3049
Yes
MECHA−
NICAL
1
TO 220 Thermal Pad, 9 mil
−
−
−
Wakefield
173-9-240P
Yes
MECHA−
NICAL
4
Standoffs, Hex 4−40, 0.75”,
Nylon
−
−
−
Keystone
4804K
Yes
MECHA−
NICAL
4
Nut, Hex 4−40, Nylon
−
−
−
Building
Fasteners
NY HN 440
Yes
Designator
Value
Tolerance
Qty
Description
BRIDGE
1
Bridge Rectifier, 4 A, 600 V
Footprint
−
−
KBL
C1, C2
2
Capacitor, EMI Suppression,
305 Vac
0.47 mF
20%
C3
1
Capacitor, Ceramic, SMD,
50 V
8.2 nF
CBULK
1
Capacitor, Electrolytic, 450 V
CCOMP
1
CCOMP1
*All products listed are Pb−free
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NCP1608BOOSTGEVB
Table 4. BILL OF MATERIALS*
Manufacturer
Manufacturer Part
Number
Substitution
Allowed
EPCOS
B57238S479M
Yes
TO−220
Infineon
SPP12N50C3
Yes
1%
1206
Vishay
CRCW1206100RFKEA
Yes
20 kW
1%
1206
Vishay
CRCW120620K0FKEA
Yes
510 W
5%
Axial
Yageo
CFR-25JB-510R
Yes
Resistor, SMD
0W
−
1206
Vishay
CRCW12060000Z0EA
Yes
2
Resistor, 0.25 W Axial
750 kW
5%
Axial
Yageo
CFR-25JB-750K
Yes
RDRV
1
Resistor, SMD
10 W
1%
1206
Vishay
CRCW120610R0FKEA
Yes
RO1A,
RO1B
2
Resistor, SMD
2 MW
1%
1206
Vishay
CRCW12062M00FKEA
Yes
ROUT2A
1
Resistor, SMD
25.5 kW
1%
1206
Vishay
CRCW120625K5FKEA
Yes
Designator
Value
Tolerance
Qty
Description
NTC
1
Thermistor, Inrush Current
Limiter
Footprint
4.7 W
20%
Radial
Q1
1
MOSFET, N−Channel,
11.6 A, 560 V
−
−
R1
1
Resistor, SMD
100 W
RCOMP1
RCS
1
Resistor, SMD
1
Resistor, 0.25 W Axial
RCT
1
RCTUP1,
RCTUP2
ROUT2B
1
Resistor, SMD
open
−
1206
−
−
-
RS1, RS2
2
Resistor, SMD, 1 W
0.25 W
1%
2512
Vishay
WSL2512R2500FEA
Yes
RS3
1
Resistor, SMD
open
−
2512
−
−
-
RSTART1
RSTART2
2
Resistor, 0.25 W Axial
330 kW
5%
Axial
Yageo
CFR-25JB-330K
Yes
RZCD
1
Resistor, 0.25 W Axial
100 kW
5%
Axial
Yageo
CFR-25JB-100K
Yes
U1
1
CrM PFC Controller
NCP1608
−
SOIC−8
ON
Semiconductor
NCP1608BDR2G
No
*All products listed are Pb−free
Figure 33. Evaluation Board Photo
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NCP1608BOOSTGEVB
LAYOUT
Figure 34. Top View of the Layout
Figure 35. Bottom View of the Layout
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NCP1608BOOSTGEVB
TEST PROCEDURE FOR THE NCP1608BOOSTGEVB EVALUATION BOARD
Required Equipment
•
•
•
•
•
15. Repeat steps 9-13 with the ac source set to 115 Vac
/ 60 Hz, 230 Vac / 50 Hz, 265 Vac / 50 Hz. Verify
the results are within the limits of Table 5.
16. Turn off the ac source.
17. Since high voltage will be present after the voltage
is removed, wait for the dc voltmeter to show
approximately 0 V before continuing.
18. Disconnect the ac source.
19. Disconnect the oscilloscope.
20. Disconnect the electronic load.
21. Disconnect both multimeters.
22. End of test.
(*Equivalent test equipment may be substituted.)
*Chroma 61604 AC Power Source
*Voltech PM3000A Power Analyzer
2× *Agilent 34401A Multimeter
*Chroma 6314 Electronic Load with *Chroma 63108
High Voltage Module
*Tektronix TDS5034B Oscilloscope with
*Tektronix P5205 Differential Probes
Test Procedure
1. Ensure that there is a jumper on J1. If not, place a
jumper on J1 for the remainder of the test
procedure.
2. Connect the electronic load with high voltage
module to the output labeled “400 V, 100 W”.
3. Connect one of the multimeters in series with the
output and load and set it to measure current.
4. Connect the second multimeter to the output and
set it to measure voltage.
5. Connect the oscilloscope with differential probes
to the output and set it to measure output ripple
and frequency.
6. Connect the ac power source and power analyzer
to the terminals labeled “Input”. Set the current
compliance limit to 1.8 A.
7. Set the ac power source to 85 Vac / 60 Hz.
8. Set the high voltage electronic load to 250 mA.
9. Turn the AC source on.
10. Wait 10 seconds, and then check the output
voltage (VOUT) using the corresponding
multimeter. Verify it is within the limits of
Table 5.
11. Measure power factor (PF) and input power (PIN)
using the power analyzer.
12. Measure the peak-to-peak voltage and frequency
of the output ripple using the oscilloscope.
13. Measure IOUT using the corresponding multimeter.
14. Calculate efficiency (h) using the equation:
h+
I OUT
V OUT
P IN
Table 5. DESIRED RESULTS
For 85 Vac /
60 Hz input
VOUT = 397 ±15 V
PF > 0.99
Output Ripple Voltage < 20 VPP
Output Ripple Frequency = 120 Hz sine
wave
h > 90%
For 115 Vac /
60 Hz input
VOUT = 397 ±15 V
PF > 0.99
Output Ripple Voltage < 20 VPP
Output Ripple Frequency = 120 Hz sine
wave
h > 90%
For 230 Vac /
50 Hz input
VOUT = 397 ±15 V
PF > 0.95
Output Ripple Voltage < 20 VPP
Output Ripple Frequency = 100 Hz sine
wave
h > 90%
For 265 Vac /
50 Hz input
VOUT = 397 ±15 V
PF > 0.95
Output Ripple Voltage < 20 VPP
Output Ripple Frequency = 100 Hz sine
wave
100%
h > 90%
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NCP1608BOOSTGEVB
Figure 36. Test Setup
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23
NCP1608BOOSTGEVB
Table 6. SUMMARY OF BOOST EQUATIONS Components are identified in Figure 3
Input rms Current
P out
h @ Vac
h (the efficiency of only the PFC
stage) is generally in the range of 90
− 95%. Vac is the rms ac line input
voltage.
Ǹ2 @ 2 @ P
out
h @ Vac
The maximum inductor peak current
occurs at the minimum line input
voltage and maximum output power.
Iac +
Inductor Peak Current
I L(peak) +
Inductor Value
Vac 2 @
Lv
ǒ
Ǔ
Ǹ2 @ V @ P @ f
out
out
SW(MIN)
On Time
t on +
2 @ L @ P out
h @ Vac 2
The maximum on time occurs at the
minimum line input voltage and
maximum output power.
t on
The off time is a maximum at the
peak of the ac line voltage and approaches zero at the ac line zero
crossings. Theta (q) represents the
angle of the ac line voltage.
Off Time
t off +
Vout
Vac@Ťsin qŤ@Ǹ2
Switching Frequency
f SW +
ǒ
Ct w
Output Voltage and
Output Divider
V out * ǒǸ2 @ Vac HLǓ
Ǹ2 @ Vac
HL
I ZCD(MAX) @ (N B : N ZCD)
ǒ
R FB @
Output Voltage Ripple and
Output Capacitor Value
Ǔ
ǒǒ
ǒ
R out1 @ R FB
Vout
VREF
Ǔ
* 1 * R out1
ǒ
Ǔ
VOVP
V REF
Ǔǒ
Ǔ
@ V REF −V OVP(HYS) @ R out1 @
R out2 ) R FB
R out2 @ R FB
V ripple(peak−peak) t 2 @ ǒV out(OVP) * V outǓ
I C(RMS) +
Where VREF is the internal reference voltage and RFB is the pull−
down resistor used for FPP. VREF
and RFB are shown in the specification table. Ibias(out) is the bias current of the output voltage divider.
R out2 ) R FB
)1
R out2 @ R FB
V OVP
R
) R FB
@ V REF @ R out1 @ out2
)1
V REF
R out2 @ R FB
C bulk w
Output Capacitor rms
Current
Where IZCD(MAX) is maximum rated
current for the ZCD pin (10 mA).
V out
I bias(out)
R out2 +
V out(OVPL) +
Where VacHL is the maximum line
input voltage. VZCD(ARM) is shown in
the specification table.
V ZCD(ARM)
V out + V REF @ R out1 @
V out(OVP) +
Where VacLL is the minimum line input voltage and LMAX is the maximum inductor value. Icharge and
VCt(MAX) are shown in the specification table.
h @ Vac LL 2 @ V Ct(MAX)
R ZCD w
R out1 +
Output Voltage OVP
Detection and Recovery
Ǔ
2 @ P out @ L MAX @ I charge
N B : N ZCD v
Resistor from ZCD
Winding to the ZCD pin
*1
Vac 2 @ h
Vac @ |sin q| @ Ǹ2
@ 1*
2 @ L @ P out
V out
On Time Capacitor
Inductor Turns to ZCD
Turns Ratio
fSW(MIN) is the minimum desired
switching frequency. The maximum
L is calculated at both the minimum
line input voltage and maximum line
input voltage.
V out
* Vac @ h
Ǹ2
P out
2 @ p @ V ripple(peak−peak) @ f line @ V out
Ǹ
Ǹ2 @ 32 @ P 2
out
* I load(RMS) 2
9 @ p @ Vac @ V out @ h 2
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VOVP/VREF and VOVP(HYS) are
shown in the specification table.
Ǔ
)1
Where fline is the ac line frequency
and Vripple(peak−peak) is the peak−to−
peak output voltage ripple. Use fline
= 47 Hz for universal input worst
case.
Where Iload(RMS) is the rms load current.
NCP1608BOOSTGEVB
Table 6. SUMMARY OF BOOST EQUATIONS Components are identified in Figure 3 (Continued)
Output Voltage UVP
Detection
Inductor rms Current
Output Diode rms
Current
MOSFET rms Current
ǒ
V out(UVP) + V UVP @ R out1 @
I L(RMS) +
I D(RMS) + 4 @
3
ǒ
Ǹ
R sense +
PR
Type 1 Compensation
ǸǸ2p@ 2 @
Ǔ
sense
R out2 @ R FB
Ǔ
)1
VUVP is shown in the specification
table.
2 @ P out
Ǹ3 @ Vac @ h
P out
I M(RMS) + 2 @
@
Ǹ3
h @ Vac
Current Sense Resistor
R out2 ) R FB
P out
h @ ǸVac @ V out
1*
ǒ
Ǹ2 @ 8 @ Vac
3 @ p @ V out
V ILIM
I L(peak)
Ǔ
VILIM is shown in the specification
table.
+ I M(RMS) 2 @ R sense
gm
C COMP +
2 @ p @ f CROSS
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Where fCROSS is the crossover frequency and is typically less than
20 Hz. gm is shown in the specification table.
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