QProx™
™ QT110 / QT110H
CHARGE-TRANSFER TOUCH SENSOR
Less expensive than many mechanical switches
Projects a ‘touch button’ through any dielectric
Turns small objects into intrinsic touch sensors
100% autocal for life - no adjustments required
Only one external part required - a 1¢ capacitor
Piezo sounder direct drive for ‘tactile’ click feedback
LED drive for visual feedback
2.5 to 5V 20µ
µA single supply operation
Toggle mode for on/off control (strap option)
10s or 60s auto-recalibration timeout (strap option)
Pulse output mode (strap option)
Gain settings in 3 discrete levels
Simple 2-wire operation possible
HeartBeat™ health indicator on output
Active Low (QT110), Active High (QT110H) versions
Vdd
1
Out
2
Opt1
3
Opt2
4
QT110
!
!
!
!
!
!
!
!
!
!
!
!
!
!
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8
Vss
7
Sns2
6
Sns1
5
Gain
APPLICATIONS !
!
Light switches
Industrial panels
!
!
Appliance control
Security systems
!
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Access systems
Pointing devices
!
!
Elevator buttons
Toys & games
The QT110 / QT110H charge-transfer (“QT’”) touch sensor is a self-contained digital IC capable of detecting near-proximity or touch.
It will project a sense field through almost any dielectric, like glass, plastic, stone, ceramic, and most kinds of wood. It can also turn
small metal-bearing objects into intrinsic sensors, making them respond to proximity or touch. This capability coupled with its ability
to self calibrate continuously can lead to entirely new product concepts.
It is designed specifically for human interfaces, like control panels, appliances, toys, lighting controls, or anywhere a mechanical
switch or button may be found; it may also be used for some material sensing and control applications provided that the presence
duration of objects does not exceed the recalibration timeout interval.
The IC requires only a common inexpensive capacitor in order to function. A bare piezo beeper can be connected to create a ‘tactile’
feedback clicking sound; the beeper itself then doubles as the required external capacitor, and it can also become the sensing
electrode. An LED can also be added to provide visual sensing indication. With a second inexpensive capacitor the device can
operated in 2-wire mode, where both power and signal traverse the same wire pair to a host. This mode allows the sensor to be wired
to a controller with only a twisted pair over a long distances.
Power consumption is under 20µA in most applications, allowing operation from Lithium cells for many years. In most cases the
power supply need only be minimally regulated.
The IC’s RISC core employs signal processing techniques pioneered by Quantum; these are specifically designed to make the device
survive real-world challenges, such as ‘stuck sensor’ conditions and signal drift. Even sensitivity is digitally determined and remains
constant in the face of large variations in sample capacitor CS and electrode CX. No external switches, opamps, or other analog
components aside from CS are usually required.
The device includes several user-selectable built in features. One, toggle mode, permits on/off touch control, for example for light
switch replacement. Another makes the sensor output a pulse instead of a DC level, which allows the device to 'talk' over the power
rail, permitting a simple 2-wire interface. The Quantum-pioneered HeartBeat™ signal is also included, allowing a host controller to
monitor the health of the QT110 continuously if desired. By using the charge transfer principle, the IC delivers a level of performance
clearly superior to older technologies in a highly cost-effective package.
TA
00C to +700C
00C to +700C
-400C to +850C
-400C to +850C
Quantum Research Group Ltd
AVAILABLE OPTIONS
SOIC
QT110-S
QT110H-S
QT110-IS
QT110H-IS
8-PIN DIP
QT110-D
QT110H-D
Copyright © 1999 Quantum Research Group Ltd
R1.02/0109
1 - OVERVIEW
Figure 1-1 Standard mode options
The QT110 is a digital burst mode charge-transfer (QT)
sensor designed specifically for touch controls; it includes all
hardware and signal processing functions necessary to
provide stable sensing under a wide variety of changing
conditions. Only a single low cost, non-critical capacitor is
required for operation.
+2.5 to 5
2
Figure 1-1 shows the basic QT110 circuit using the device,
with a conventional output drive and power supply
connections. Figure 1-2 shows a second configuration using
a common power/signal rail which can be a long twisted pair
from a controller; this configuration uses the built-in pulse
mode to transmit output state to the host controller (QT110
only).
3
4
OU TP UT=D C
TIM EO UT= 10 S ecs
TOGG LE=OF F
GA IN= HIGH
1.1 BASIC OPERATION
The QT110 employs short, ultra-low duty cycle bursts of
charge-transfer cycles to acquire its signal. Burst mode
permits power consumption in the low microamp range,
dramatically reduces RF emissions, lowers susceptibility to
EMI, and yet permits excellent response time. Internally the
signals are digitally processed to reject impulse noise, using
a 'consensus' filter which requires four consecutive
confirmations of a detection before the output is activated.
2
3
O UT
S NS 2
O PT 1
G A IN
S E NS IN G
E LE C T RO DE
Cs
10 nF
Cx
4
O PT 2
S NS 1
S N S1
Cs
1 0nF
Cx
6
Vss
8
The internal ADC treats Cs as a floating transfer capacitor;
as a direct result, the sense electrode can be connected to
either SNS1 or SNS2 with no performance difference. In both
cases the rule Cs >> Cx must be observed for proper
operation. The polarity of the charge buildup across Cs
during a burst is the same in either case.
7
5
OP T2
5
1.2 ELECTRODE DRIVE
22 µF 10V AL
1
G A IN
Option pins allow the selection or alteration of several
special features and sensitivity.
+3V
V dd
OP T1
7
A simple circuit variation is to replace Cs with a bare piezo
sounder (Section 2), which is merely another type of
capacitor, albeit with a large thermal drift coefficient. If Cpiezo
is in the proper range, no other external component is
required. If Cpiezo is too small, it can simply be ‘topped up’
with an inexpensive ceramic capacitor connected in parallel
with it. The QT110 drives a 4kHz signal across SNS1 and
SNS2 to make the piezo (if installed) sound a short tone for
75ms immediately after detection, to act as an audible
confirmation.
+
Tw ist e d
pa ir
S N S2
Cs is thus non-critical; as it drifts with temperature, the
threshold algorithm compensates for the drift automatically.
Figure 1-2 2-wire operation, self-powered (QT110 only)
2 . 2k
Vdd
OU T
The IC is highly tolerant of changes in Cs since it computes
the threshold level ratiometrically with respect to absolute
load, and does so dynamically at all times.
The QT switches and charge measurement hardware
functions are all internal to the QT110 (Figure 1-3). A 14-bit
single-slope switched capacitor ADC includes both the
required QT charge and transfer switches in a configuration
that provides direct ADC conversion. The ADC is designed to
dynamically optimize the QT burst length according to the
rate of charge buildup on Cs, which in turn depends on the
values of Cs, Cx, and Vdd. Vdd is used as the charge
reference voltage. Larger values of Cx cause the charge
transferred into Cs to rise more rapidly, reducing available
resolution; as a minimum resolution is required for proper
operation, this can result in dramatically reduced apparent
gain. Conversely, larger values of Cs reduce the rise of
differential voltage across it, increasing available resolution
by permitting longer QT bursts. The value of Cs can thus be
increased to allow larger values of Cx to be tolerated
(Figures 4-1, 4-2, 4-3 in Specifications, rear).
CMOS
GATE
S E NS ING
E LEC TRO DE
1
6
It is possible to connect separate Cx and
Cx’ loads to SNS1 and SNS2
simultaneously, although the result is no
different than if the loads were
connected together at SNS1 (or SNS2).
It is important to limit the amount of
stray capacitance on both terminals,
especially if the load Cx is already large,
for example by minimizing trace lengths
and widths so as not to exceed the Cx
load specification and to allow for a
larger sensing electrode size if so
desired.
The PCB traces, wiring, and any
components associated with or in
contact with SNS1 and SNS2 will
become touch sensitive and should be
V ss
8
-2-
Figure 1-3 Internal Switching & Timing
ELE C TRO DE
R esult
1.3.1 ELECTRODE GEOMETRY AND SIZE
Start
There is no restriction on the shape of
the electrode; in most cases common
sense and a little experimentation can
result in a good electrode design. The
QT110 will operate equally well with
long, thin electrodes as with round or
square ones; even random shapes are
acceptable. The electrode can also be
a 3-dimensional surface or object.
Sensitivity is related to electrode
surface area, orientation with respect
to the object being sensed, object composition, and the
ground coupling quality of both the sensor circuit and the
sensed object.
If a relatively large electrode surface is desired, and if tests
Figure 1-4 Mesh Electrode Geometry
Single -Slo pe 14-bit
Switched Cap acito r AD C
1.3 ELECTRODE DESIGN
SNS2
Bu rst Controller
treated with caution to limit the touch
area to the desired location. Multiple
touch electrodes can be used, for
example to create a control button on
both sides of an object, however it is
impossible for the sensor to distinguish
between the two touch areas.
Do ne
Cs
Cx
SNS1
C ha rg e
Am p
will provide ample ground coupling, since there is
capacitance between the windings and/or the transformer
core, and from the power wiring itself directly to 'local earth'.
Even when battery powered, just the physical size of the
PCB and the object into which the electronics is embedded
will generally be enough to couple a few picofarads back to
local earth.
1.3.3 VIRTUAL CAPACITIVE GROUNDS
When detecting human contact (e.g. a fingertip), grounding
of the person is never required. The human body naturally
has several hundred picofarads of ‘free space’ capacitance
to the local environment (Cx3 in Figure 1-5), which is more
than two orders of magnitude greater than that required to
create a return path to the QT110 via earth. The QT110's
PCB however can be physically quite small, so there may be
little ‘free space’ coupling (Cx1 in Figure 1-5) between it and
the environment to complete the return path. If the QT110
circuit ground cannot be earth grounded by wire, for example
via the supply connections, then a ‘virtual capacitive ground’
may be required to increase return coupling.
show that the electrode has more capacitance than the
QT110 can tolerate, the electrode can be made into a sparse
mesh (Figure 1-4) having lower Cx than a solid plane.
Sensitivity may even remain the same, as the sensor will be
operating in a lower region of the gain curves.
Figure 1-5 Kirchoff's Current Law
1.3.2 KIRCHOFF’S CURRENT LAW
Like all capacitance sensors, the QT110 relies on Kirchoff’s
Current Law (Figure 1-5) to detect the change in capacitance
of the electrode. This law as applied to capacitive sensing
requires that the sensor’s field current must complete a loop,
returning back to its source in order for capacitance to be
sensed. Although most designers relate to Kirchoff’s law with
regard to hardwired circuits, it applies equally to capacitive
field flows. By implication it requires that the signal ground
and the target object must both be coupled together in some
manner for a capacitive sensor to operate properly. Note that
there is no need to provide actual hardwired ground
connections; capacitive coupling to ground (Cx1) is always
sufficient, even if the coupling might seem very tenuous. For
example, powering the sensor via an isolated transformer
-3-
CX2
S e n se E le ctro de
S EN SO R
CX1
Su rro und in g e nv iro nm e n t
C X3
millimeters of internal air gap; if the product is very thin and
contact with the product's back is a concern, then some form
of rear shielding may be required.
Figure 1-6 Shielding Against Fringe Fields
1.3.5 SENSITIVITY
Sen se
w ire
The QT110 can be set for one of 3 gain levels using option
pin 5 (Table 1-1). If left open, the gain setting is high. The
sensitivity change is made by altering the numerical
threshold level required for a detection. It is also a function
of other things: electrode size, shape, and orientation, the
composition and aspect of the object to be sensed, the
thickness and composition of any overlaying panel material,
and the degree of ground coupling of both sensor and object
are all influences.
Sens e
w ire
1.3.5.1 Increasing Sensitivity
In some cases it may be desirable to increase sensitivity
further, for example when using the sensor with very thick
panels having a low dielectric constant.
U ns hielded
Electrod e
S h ield ed
E lec trod e
A ‘virtual capacitive ground’ can be created by connecting
the QT110’s own circuit ground to:
(1) A nearby piece of metal or metallized housing;
(2) A floating conductive ground plane;
(3) A nail driven into a wall when used with small
electrodes;
(4) A larger electronic device (to which its output might be
connected anyway).
Free-floating ground planes such as metal foils should
maximize exposed surface area in a flat plane if possible. A
square of metal foil will have little effect if it is rolled up or
crumpled into a ball. Virtual ground planes are more
effective and can be made smaller if they are physically
bonded to other surfaces, for example a wall or floor.
1.3.4 FIELD SHAPING
The electrode can be prevented from sensing in undesired
directions with the assistance of metal shielding connected
to circuit ground (Figure 1-6). For example, on flat surfaces,
the field can spread laterally and create a larger touch area
than desired. To stop field spreading, it is only necessary to
surround the touch electrode on all sides with a ring of metal
connected to circuit ground; the ring can be on the same or
opposite side from the electrode. The ring will kill
field spreading from that point outwards.
If one side of the panel to which the electrode is
fixed has moving traffic near it, these objects can
cause inadvertent detections. This is called
‘walk-by’ and is caused by the fact that the fields
radiate from either surface of the electrode
equally well. Again, shielding in the form of a
metal sheet or foil connected to circuit ground
will prevent walk-by; putting a small air gap
between the grounded shield and the electrode
will keep the value of Cx lower and is
encouraged. In the case of the QT110, the
sensitivity is low enough that 'walk-by' should not
be a concern if the product has more than a few
Sensitivity can often be increased by using a bigger
electrode, reducing panel thickness, or altering panel
composition. Increasing electrode size can have diminishing
returns, as high values of Cx will reduce sensor gain
(Figures 4-1 ~ 4-3). Also, increasing the electrode's surface
area will not substantially increase touch sensitivity if its
Table 1-1 Gain Setting Strap Options
Gain
High
Medium
Low
Tie Pin 5 to:
None
Pin 6
Pin 7
diameter is already much larger in surface area than the
object being detected. The panel or other intervening
material can be made thinner, but again there are
diminishing rewards for doing so. Panel material can also be
changed to one having a higher dielectric constant, which
will help propagate the field through to the front. Locally
adding some conductive material to the panel (conductive
materials essentially have an infinite dielectric constant) will
also help dramatically; for example, adding carbon or metal
fibers to a plastic panel will greatly increase frontal field
strength, even if the fiber density is too low to make the
plastic bulk-conductive.
1.3.5.2 Decreasing Sensitivity
In some cases the QT110 may be too sensitive, even on low
gain. In this case gain can be lowered further by any of a
Figure 2-1 Drift Compensation
S ign a l
H yste resis
T hr es ho ld
R e fe re nce
O u tpu t
-4-
number of strategies: making the electrode smaller,
connecting a very small capacitor in series with the sense
lead, or making the electrode into a sparse mesh using a
high space-to-conductor ratio (Figure 1-4). A deliberately
added Cx capacitor can also be used to reduce sensitivity
according to the gain curves (see Section 4).
Intermediate levels of gain (e.g. between 'medium' and 'low'
can be obtained by a combination of jumper settings with
one or more of the above strategies.
2 - QT110 SPECIFICS
2.1 SIGNAL PROCESSING
The QT110 processes all signals using 16 bit math, using a
number of algorithms pioneered by Quantum. The
algorithms are specifically designed to provide for high
'survivability' in the face of all kinds of adverse
environmental changes.
sensor will compensate for the object's removal very quickly,
usually in only a few seconds.
2.1.2 THRESHOLD CALCULATION
Sensitivity is dependent on the threshold level as well as
ADC gain; threshold in turn is based on the internal signal
reference level plus a small differential value. The threshold
value is established as a percentage of the absolute signal
level. Thus, sensitivity remains constant even if Cs is altered
dramatically, so long as electrode coupling to the user
remains constant. Furthermore, as Cx and Cs drift, the
threshold level is automatically recomputed in real time so
that it is never in error.
The QT110 employs a hysteresis dropout below the
threshold level of 50% of the delta between the reference
and threshold levels.
2.1.3 MAX ON-DURATION
If an object or material obstructs the sense pad the signal
2.1.1 DRIFT COMPENSATION ALGORITHM
Table 2-1 Output Mode Strap Options
Signal drift can occur because of changes in Cx and Cs over
time. It is crucial that drift be compensated for, otherwise
false detections, non-detections, and sensitivity shifts will
follow.
Drift compensation (Figure 2-1) is performed by making the
reference level track the raw signal at a slow rate, but only
while there is no detection in effect. The rate of adjustment
must be performed slowly, otherwise legitimate detections
could be ignored. The QT110 drift compensates using a
slew-rate limited change to the reference level; the threshold
and hysteresis values are slaved to this reference.
Once an object is sensed, the drift compensation
mechanism ceases since the signal is legitimately high, and
therefore should not cause the reference level to change.
The QT110's drift compensation is 'asymmetric': the
reference level drift-compensates in one direction faster than
it does in the other. Specifically, it compensates faster for
decreasing signals than for increasing signals. Increasing
signals should not be compensated for quickly, since an
approaching finger could be compensated for partially or
entirely before even touching the sense pad. However, an
obstruction over the sense pad, for which the sensor has
already made full allowance for, could suddenly be removed
leaving the sensor with an artificially elevated reference level
and thus become insensitive to touch. In this latter case, the
Figure 2-2 Powering From a CMOS Port Pin
P O RT X .m
0.01µF
C MO S
m icro controller
V dd
P O RT X .n
O UT
Q T11 0
V ss
Tie
Pin 3 to:
Tie
Pin 4 to:
Max OnDuration
DC Out
Vdd
Vdd
10s
DC Out
Vdd
Gnd
60s
Toggle
Gnd
Gnd
10s
Pulse
Gnd
Vdd
10s
may rise enough to create a detection, preventing further
operation. To prevent this, the sensor includes a timer which
monitors detections. If a detection exceeds the timer setting,
the timer causes the sensor to perform a full recalibration.
This is known as the Max On-Duration feature.
After the Max On-Duration interval, the sensor will once
again function normally, even if partially or fully obstructed,
to the best of its ability given electrode conditions. There are
two timeout durations available via strap option: 10 and 60
seconds.
2.1.4 DETECTION INTEGRATOR
It is desirable to suppress detections generated by electrical
noise or from quick brushes with an object. To accomplish
this, the QT110 incorporates a detect integration counter that
increments with each detection until a limit is reached, after
which the output is activated. If no detection is sensed prior
to the final count, the counter is reset immediately to zero.
In the QT110, the required count is 4.
The Detection Integrator can also be viewed as a
'consensus' filter, that requires four detections in four
successive bursts to create an output. As the basic burst
spacing is 75ms, if this spacing was maintained throughout
all 4 counts the sensor would react very slowly. In the
QT110, after an initial detection is sensed, the remaining
three bursts are spaced about 18ms apart, so that the
slowest reaction time possible is 75+18+18+18 or 129ms
and the fastest possible is 54ms, depending on where in the
initial burst interval the contact first occurred. The response
time will thus average 92ms.
-5-
2.1.5 FORCED SENSOR RECALIBRATION
The QT110 has no recalibration pin; a forced recalibration is
accomplished only when the device is powered up. However,
supply drain is so low it is a simple matter to treat the entire
IC as a controllable load; simply driving the QT110's Vdd pin
directly from another logic gate or a microprocessor port
(Figure 2-2) will serve as both power and 'forced recal'. The
source resistance of most CMOS gates and microprocessors
is low enough to provide direct power without any problems.
Note that most 8051-based micros have only a weak pullup
drive capability and will require true CMOS buffering. Any
74HC or 74AC series gate can directly power the QT110, as
can most other microprocessors.
Option strap configurations are read by the QT110 only on
powerup. Configurations can only be changed by powering
the QT110 down and back up again; again, a microcontroller
can directly alter most of the configurations and cycle power
to put them in effect.
2.2 OUTPUT FEATURES
The QT110 / QT110H are designed for maximum flexibility
and can accommodate most popular sensing requirements.
These are selectable using strap options on pins OPT1 and
OPT2. All options are shown in Table 2-1.
2.2.1 DC MODE OUTPUT
The output of the device can respond in a DC mode, where
the output is active-low (QT110) or active-high (QT110H)
upon detection. The output will remain active for the duration
of the detection, or until the Max On-Duration expires,
whichever occurs first. If the latter occurs first, the sensor
performs a full recalibration and the output becomes inactive
until the next detection.
In this mode, two Max On-Duration timeouts are available:
10 and 60 seconds.
2.2.2 TOGGLE MODE OUTPUT
This makes the sensor respond in an on/off mode like a flip
flop. It is most useful for controlling power loads, for
example in kitchen appliances, power tools, light switches,
etc.
Max On-Duration in Toggle mode is fixed at 10 seconds.
When a timeout occurs, the sensor recalibrates but leaves
the output state unchanged.
2.2.3 PULSE MODE OUTPUT
This generates a pulse of 75ms duration (QT110 negative-going; QT110H - positive-going) with every new
detection. It is most useful for 2-wire operation, but can also
be used when bussing together several devices onto a
common output line with the help of steering diodes or logic
gates, in order to control a common load from several
places.
Max On-Duration is fixed at 10 seconds if in Pulse output
mode.
2.2.4 HEARTBEAT™ OUTPUT
The output has a full-time HeartBeat™ ‘health’ indicator
superimposed on it. This operates by taking 'Out' into a
3-state mode for 350µs once before every QT burst. This
output state can be used to determine that the sensor is
operating properly, or, it can be ignored using one of several
simple methods.
QT110: The HeartBeat indicator can be sampled by using a
pulldown resistor on Out, and feeding the resulting
negative-going pulse into a counter, flip flop, one-shot, or
other circuit. Since Out is normally high, a pulldown resistor
will create negative HeartBeat pulses (Figure 2-3) when the
sensor is not detecting an object; when detecting an object,
the output will remain active for the duration of the detection,
and no HeartBeat pulse will be evident.
QT110H: Same as QT110 but inverted logic (use a
pull-down resistor instead of a pull-up etc.)
If the sensor is wired to a microprocessor as shown in Figure
2-4, the microprocessor can reconfigure the load resistor to
either ground or Vcc depending on the output state of the
device, so that the pulses are evident in either state.
Electromechanical devices will usually ignore this short
pulse. The pulse also has too low a duty cycle to visibly
activate LED’s. It can be filtered completely if desired, by
adding an RC timeconstant to filter the output, or if
interfacing directly and only to a high-impedance CMOS
input, by doing nothing or at most adding a small non-critical
capacitor from Out to ground (Figure 2-5).
Figure 2-3
Figure 2-4
Getting HB pulses with a pull-down resistor (QT110 shown; use
pull-up resistor with QT110H)
+2 .5 to 5
H eartBeat™ P ulses
Using a micro to obtain HB pulses in either output state
(QT110 or QT110H)
1
2
V dd
O UT
S NS 2
O PT 1
GAIN
O PT 2
S NS 1
2
P O RT _M .x
7
OUT
SN S 2
O PT1
G A IN
O PT2
SN S 1
7
Ro
Ro
3
4
5
3
M icro pro ce sso r
6
P O RT _M .y
V ss
8
-6-
4
5
6
in Vdd, as happens when loads are switched on. This can
induce detection ‘cycling’, whereby an object is detected, the
load is turned on, the supply sags, the detection is no longer
sensed, the load is turned off, the supply rises and the object
is reacquired, ad infinitum. To prevent this occurrence, the
output should only be lightly loaded if the device is operated
from an unregulated supply, e.g. batteries. Detection
‘stiction’, the opposite effect, can occur if a load is shed
when Out is active.
Figure 2-5 Eliminating HB Pulses
G ATE OR
MIC RO INPU T
O UT
SN S 2
O PT1
GA IN
O PT2
SN S 1
7
Co
100p F
3
4
5
QT110: The output of the QT110 can directly drive a
resistively limited LED. The LED should be connected with
its cathode to the output and its anode towards Vcc, so that
it lights when the sensor is active-low. If desired the LED can
be connected from Out to ground, and driven on when the
sensor is inactive, but only with less drive current (1mA).
6
2.2.5 PIEZO ACOUSTIC DRIVE
A piezo drive signal is generated for use with a bare piezo
sounder immediately after a detection is made; the tone lasts
for a nominal 75ms to create a reassuring ‘tactile feedback’
sound.
The sensor will drive most common bare piezo ‘beepers’
directly using an H-bridge drive configuration for the highest
possible sound level at all supply voltages; H-bridge drive
effectively doubles the supply voltage across the piezo. The
piezo is connected across pins SNS1 and SNS2. This drive
operates at a nominal 4kHz frequency, a common resonance
point for enclosed piezo sounders. Other frequencies can be
obtained upon special request.
QT110H: This part is active-high, so it works in reverse to
that described above.
3 - CIRCUIT GUIDELINES
Figure 2-6 Damping Piezo Clicks with Rx
+ 2.5 to 5
If desired a bare piezo sounder can be directly adhered to
the rear of a control panel, provided that an acoustically
resonant cavity is also incorporated to give the desired
sound level.
2
3
Since piezo sounders are merely high-K ceramic capacitors,
the sounder will double as the Cs capacitor, and the piezo's
metal disc will act as the sensing electrode. Piezo transducer
capacitances typically range from 6nF to 30nF (0.006µF to
0.03µF) in value; at the lower end of this range an additional
capacitor should be added to bring the total Cs across SNS1
and SNS2 to at least 10nF, or more if Cx is large.
The burst acquisition process induces a small but audible
voltage step across the piezo resonator, which occurs when
SNS1 and SNS2 rapidly discharge residual voltage stored on
the resonator. The resulting slight clicking sound can be
used to provide an audible confirmation of functionality if
desired, or, it can be suppressed by placing a non-critical 1M
to 2M ohm bleed resistor in parallel with the resonator. The
resistor acts to slowly discharge the resonator, preempting
the occurrence of the harmonic-rich step (Figure 2-6).
With the resistor in place, an almost inaudible clicking sound
may still be heard, which is caused by the small charge
buildup across the piezo device during each burst.
2.2.6 OUTPUT DRIVE
The QT110’s `output is active low (QT110) or active high
(QT110H) and can source 1mA or sink 5mA of non-inductive
current. If an inductive load is used, such as a small relay,
the load should be diode clamped to prevent damage.
Care should be taken when the IC and the load are both
powered from the same supply, and the supply is minimally
regulated. The device derives its internal references from the
power supply, and sensitivity shifts can occur with changes
S ENSING
E LEC TRO DE
1
4
V dd
OU T
S N S1
OP T1
G A IN
OP T2
S N S2
7
5
Pie zo Sounde r
10-30 nF
2
C MO S
Rx
Cx
6
V ss
8
3.1 SAMPLE CAPACITOR
Charge sampler Cs can be virtually any plastic film or high-K
ceramic capacitor. Since the acceptable Cs range is
anywhere from 10nF to 30nF, the tolerance of Cs can be the
lowest grade obtainable so long as its value is guaranteed to
remain in the acceptable range under expected temperature
conditions. Only if very fast, radical temperature swings are
expected will a higher quality capacitor be required, for
example polycarbonate, PPS film, or NPO/C0G ceramic.
3.2 PIEZO SOUNDER
The use of a piezo sounder in place of Cs is described in the
previous section. Piezo sounders have very high,
uncharacterized thermal coefficients and should not be used
if fast temperature swings are anticipated.
3.3 OPTION STRAPPING
The option pins Opt1 and Opt2 should never be left floating.
If they are floated, the device will draw excess power and the
options will not be properly read on powerup. Intentionally,
-7-
there are no pullup resistors on these lines,
since pullup resistors add to power drain if tied
low.
Figure 2-7 ESD Protection
+2.5 to 5
The Gain input is designed to be floated for
sensing one of the three gain settings. It
should never be connected to a pullup resistor
or tied to anything other than Sns1 or Sns2.
Table
2-1
shows
the
configurations available.
option
strap
+
2
OU T
Vdd
D1
SNS2
7
R e3
S ENSIN G
ELEC TR O DE
R e1
3
3.4 POWER SUPPLY, PCB LAYOUT
The power supply can range from 2.5 to 5.0
volts. At 3 volts current drain averages less
than 20µA in most cases, but can be higher if
Cs is large. Interestingly, large Cx values will
actually decrease power drain. Operation can
be from batteries, but be cautious about loads
causing supply droop (see Output Drive,
previous section).
10µF
R e2
1
4
C1
O PT1
G AIN
O PT2
SNS1
D2
5
Cs
6
Vss
8
As battery voltage sags with use or fluctuates slowly with
temperature, the IC will track and compensate for these
changes automatically with only minor changes in sensitivity.
If the power supply is shared with another electronic system,
care should be taken to assure that the supply is free of
digital spikes, sags, and surges which can adversely affect
the device. The IC will track slow changes in Vdd, but it can
be affected by rapid voltage steps.
if desired, the supply can be regulated using a conventional
low current regulator, for example CMOS regulators that
have nanoamp quiescent currents. Care should be taken that
the regulator does not have a minimum load specification,
which almost certainly will be violated by the QT110's low
current requirement.
Since the IC operates in a burst mode, almost all the power
is consumed during the course of each burst. During the
time between bursts the sensor is quiescent.
For proper operation a 100nF (0.1uF) ceramic bypass
capacitor should be used between Vdd and Vss; the bypass
cap should be placed very close to the device’s power pins.
3.4.1 MEASURING SUPPLY CURRENT
Measuring average power consumption is a fairly difficult
task, due to the burst nature of the device’s operation. Even
a good quality RMS DMM will have difficulty tracking the
relatively slow burst rate.
The simplest method for measuring average current is to
replace the power supply with a large value low-leakage
electrolytic capacitor, for example 2,700µF. 'Soak' the
capacitor by connecting it to a bench supply at the desired
operating voltage for 24 hours to form the electrolyte and
reduce leakage to a minimum. Connect the capacitor to the
circuit at T=0, making sure there will be no detections during
the measurement interval; at T=30 seconds measure the
capacitor's voltage with a DMM. Repeat the test without a
load to measure the capacitor's internal leakage, and
subtract the internal leakage result from the voltage droop
measured during the QT110 load test. Be sure the DMM is
connected only at the end of each test, to prevent the DMM's
impedance from contributing to the capacitor's discharge.
Supply drain can be calculated from the adjusted voltage
droop using the basic charge equation:
i=
✁VC
t
where C is the large supply cap value, t is the elapsed
measurement time in seconds, and ∆V is the adjusted
voltage droop on C.
3.4.2 ESD PROTECTION
In cases where the electrode is placed behind a dielectric
panel, the IC will be protected from direct static discharge.
However, even with a panel, transients can still flow into the
electrode via induction, or in extreme cases, via dielectric
breakdown. Porous materials may allow a spark to tunnel
right through the material; partially conducting materials like
'pink poly' will conduct the ESD right to the electrode. Testing
is required to reveal any problems. The device does have
diode protection on its terminals which can absorb and
protect the device from most induced discharges, up to
20mA; the usefulness of the internal clamping will depending
on the dielectric properties, panel thickness, and rise time of
the ESD transients.
ESD dissipation can be aided further with an added diode
protection network as shown in Figure 2-7, in extreme cases.
Because the charge and transfer times of the QT110 are
relatively long, the circuit can tolerate very large values of
Re, more than 100k ohms in most cases where electrode Cx
is small. The added diodes shown (1N4150, BAV99 or
equivalent low-C diodes) will shunt the ESD transients away
from the part, and Re1 will current limit the rest into the
QT110's own internal clamp diodes. C1 should be around
10µF if it is to absorb positive transients from a human body
model standpoint without rising in value by more than 1 volt.
If desired C1 can be replaced with an appropriate zener
diode. Directly placing semiconductor transient protection
devices or MOV's on the sense lead is not advised; these devices
have extremely large amounts of parasitic C which will swamp the
capacitance of the electrode.
Re1 should be as large as possible given the load value of
Cx and the diode capacitances of D1 and D2. Re1 should be
low enough to permit at least 6 timeconstants of RC to occur
during the charge and transfer phases.
-8-
Re2 functions to isolate the transient from the Vdd pin;
values of around 1K ohms are reasonable.
As with all ESD protection networks, it is crucial that the
transients be led away from the circuit. PCB ground layout is
crucial; the ground connections to D1, D2, and C1 should all
go back to the power supply ground or preferably, if
available, a chassis ground connected to earth. The currents
should not be allowed to traverse the area directly under the
IC.
If the device is connected to an external circuit via a cable or
long twisted pair, it is possible for ground-bounce to cause
damage to the Out pin; even though the transients are led
away from the IC itself, the connected signal or power
ground line will act as an inductor, causing a high differential
voltage to build up on the Out wire with respect to ground. If
this is a possibility, the Out pin should have a resistance Re3
in series with it to limit current; this resistor should be as
large as can be tolerated by the load.
-9-
4.1 ABSOLUTE MAXIMUM SPECIFICATIONS
Operating temp . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . as designated by suffix
Storage temp . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -55OC to +125OC
VDD . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.5 to +6.5V
Max continuous pin current, any control or drive pin . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±20mA
Short circuit duration to ground, any pin . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . infinite
Short circuit duration to VDD, any pin . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . infinite
Voltage forced onto any pin . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.6V to (Vdd + 0.6) Volts
4.2 RECOMMENDED OPERATING CONDITIONS
VDD . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +2.5 to 5.5V
Supply ripple+noise . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20mV p-p max
Load capacitance . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 0 to 20pF
Cs value . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10nF to 30nF
4.3 AC SPECIFICATIONS
Parameter
Vdd = 3.0, Ta = recommended operating range
Description
Min
Typ
Max
Units
TRC
Recalibration time
550
ms
TPC
Charge duration
2
µs
TPT
Transfer duration
2
µs
TBS
Burst spacing interval
TBL
Burst length
TR
Response time
75
0.5
Notes
ms
7
ms
129
ms
FP
Piezo drive frequency
4
kHz
TP
Piezo drive duration
75
ms
TPO
Pulse output width on Out
75
ms
THB
Heartbeat pulse width
300
µs
4.4 SIGNAL PROCESSING
Description
Min
Typ
Max
Units
Notes
Threshold differential, high gain
3.1
%
Note 1
Threshold differential, medium gain
4.7
%
Note 1
Threshold differential, low gain
6.25
%
Note 1
50
%
Note 2
Hysteresis
4
samples
Positive drift compensation rate
Consensus filter length
750
ms/level
Negative drift compensation rate
75
ms/level
Post-detection recalibration timer duration
10
Note 1: Of absolute full scale signal
Note 2: Of signal threshold
Note 3: Strap option.
- 10 -
60
secs
Note 3
4.5 DC SPECIFICATIONS
Vdd = 3.0V, Cs = 10nF, Cx = 5pF, TA = recommended range, unless otherwise noted
Parameter
VDD
IDD
Description
Min
Supply voltage
Typ
2.45
Supply current
VDDS
Supply turn-on slope
VIL
Low input logic level
VHL
High input logic level
VOL
Low output voltage
VOH
High output voltage
IIL
Input leakage current
CX
Load capacitance range
IX
Min shunt resistance
AR
Acquisition resolution
S[1]
Sensitivity - high gain
S[2]
Sensitivity - medium gain
Max
Units
5.25
V
20
µA
100
V/s
0.8
2.2
0.6
Vdd-0.7
0
OPT1, OPT2
V
OPT1, OPT2
V
OUT, 4mA sink
V
OUT, 1mA source
µA
30
pF
✡
14
Required for proper startup
V
±1
500K
S[3]
Sensitivity - low gain
Preliminary Data: All specifications subject to change.
Notes
OPT1, OPT2
Resistance from SNS1 to SNS2
bits
1
pF
Refer to Figures 4-1 through 4-3
1.5
pF
Refer to Figures 4-1 through 4-3
3
pF
Refer to Figures 4-1 through 4-3
Figure 4-1 High Gain Sensitivity
and Range @ Vdd = 3V
Figure 4-2 Medium Gain Sensitivity
and Range @ Vdd = 3V
3.0
4.0
Sensitivity, p F
Cx=30pF
Sensitivity, p F
Cx=30pF
2.5
25pF
2.0
20pF
1.5
10pF
5pF
1.0
0pF
10
20
20pF
2.0
10pF
5pF
0pF
1.0
Valid operating range
0.5
25pF
3.0
Valid operating range
30
10
20
C s, nF
30
C s, nF
Figure 4-3 Low Gain Sensitivity
and Range @ Vdd = 3V
Figure 4-4 Supply Current vs.
Voltage; Cx = 10pF
180
160
8.0
Current (microamps)
Sen sitivity, pF
Cx=30pF
25pF
6.0
20pF
4.0
2.0
10pF
5pF
0pF
140
120
100
Cs = 100nF
80
47nF
60
22nF
40
10nF
20
Valid operating range
0
10
20
3
30
3.5
4
Vss, Volts
Cs, nF
- 11 -
4.5
5
Package type: 8pin Dual-In-Line
SYMBOL
a
A
M
m
Q
P
L
L1
F
R
r
S
S1
Aa
x
Y
Min
Millimeters
Max
6.096
7.62
9.017
7.62
0.889
0.254
0.355
1.397
2.489
3.048
0.381
3.048
7.62
8.128
0.203
7.112
8.255
10.922
7.62
0.559
1.651
2.591
3.81
3.556
4.064
7.062
9.906
0.381
Notes
Typical
BSC
Typical
BSC
Min
Inches
Max
0.24
0.3
0.355
0.3
0.035
0.01
0.014
0.055
0.098
0.12
0.015
0.12
0.3
0.32
0.008
0.28
0.325
0.43
0.3
0.022
0.065
0.102
0.15
0.14
0.16
0.3
0.39
0.015
Notes
Typical
BSC
Typical
BSC
Package type: 8pin SOIC
SYMBOL
Min
Millimeters
Max
M
W
Aa
H
h
D
L
E
e
ß
Ø
4.800
5.816
3.81
1.371
0.101
1.27
0.355
0.508
0.19
0.381
0º
4.979
6.198
3.988
1.728
0.762
1.27
0.483
1.016
0.249
0.762
8º
Notes
Min
Inches
Max
BSC
0.189
0.229
0.15
0.054
0.004
0.050
0.014
0.02
0.007
0.229
0º
0.196
0.244
0.157
0.068
0.01
0.05
0.019
0.04
0.01
0.03
8º
- 12 -
Notes
BSC
5 - ORDERING INFORMATION
PART
TEMP RANGE
PACKAGE
MARKING
QT110-D
QT110-S
QT110-IS
QT110H-D
QT110H-S
QT110H-IS
0 - 70C
0 - 70C
-40 - 85C
0 - 70C
0 - 70C
-40 - 85C
PDIP
SOIC-8
SOIC-8
PDIP
SOIC-8
SOIC-8
QT1 + 10
QT1
QT1 + I
QT1 +10H
QT1 + A
QT1 + AI
Quantum Research Group Ltd
©1999 QRG Ltd.
Patented and patents pending
651 Holiday Drive Bldg. 5 / 300
Pittsburgh, PA 15220 USA
Tel: 412-391-7367 Fax: 412-291-1015
admin@qprox.com
http://www.qprox.com
In the United Kingdom
Enterprise House, Southampton, Hants SO14 3XB
Tel: +44 (0)23 8045 3934 Fax: +44 (0)23 8045 3939
Notice: This device expressly not for use in any medical or human safety related application without the express written consent of an officer of
the company.