ADP1073AR-33

a FEATURES Operates at Supply Voltages from 1.0 V to 30 V Ground Current 100 A Works in Step-Up or Step-Down Mode Very Few External Components Required Low Battery Detector On-Chip User-Adjustable Current Limit Internal 1 A Power Switch Fixed and Adjustable Output Voltage Versions 8-Lead DIP or SO-8 Package APPLICATIONS Single-Cell to 5 V Converters Laptop and Palmtop Computers Pagers Cameras Battery Backup Supplies Cellular Telephones Portable Instruments 4 mA–20 mA Loop Powered Instruments Hand-Held Inventory Computers Micropower DC–DC Converter Adjustable and Fixed 3.3 V, 5 V, 12 V ADP1073 FUNCTIONAL BLOCK DIAGRAMS SET ADP1073 A2 VIN GAIN BLOCK/ ERROR AMP 212mV REFERENCE ILIM SW1 A1 OSCILLATOR DRIVER SW2 AO COMPARATOR GND FB ADP1073 SET ADP1073-3.3 ADP1073-5 ADP1073-12 VIN A2 GAIN BLOCK/ ERROR AMP 212mV REFERENCE A1 OSCILLATOR DRIVER SW2 AO ILIM SW1 GENERAL DESCRIPTION The ADP1073 is part of a family of step-up/step-down switching regulators that operates from an input supply voltage of as little as 1.0 V. This extremely low input voltage allows the ADP1073 to be used in applications requiring use of a single cell battery as the primary power source. The ADP1073 can be configured to operate in either step-up or step-down mode but for input voltages greater than 3 V, the ADP1173 is recommended. An auxiliary gain amplifier can serve as a low battery detector or linear regulator. Quiescent current on the ADP1073-5 is only 100 µA unloaded, making it ideal for systems where long battery life is required. The ADP1073 can deliver 40 mA at 5 V from an input voltage range as low as 1.25 V, or 10 mA at 5 V from a 1.0 V input. Current limiting is available by adding an external resistor. R1 COMPARATOR R2 904k ADP1073-3.3: R1 = 62.1k ADP1073-5: R1 = 40k ADP1073-12: R1 = 16.3k SENSE GND ADP1073-3.3, 5, 12 R EV. 0 Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781/329-4700 World Wide Web Site: http://www.analog.com Fax: 781/326-8703 © Analog Devices, Inc., 1997 ADP1073–SPECIFICATIONS (@ T = 0 C to +70 C, V A IN = 1.5 V unless otherwise noted) Symbol IQ IQ Min Typ 100 100 100 100 1.15 1.0 200 212 3.30 5.00 12.00 5 90 125 300 fOSC 14 57 28 19 72 38 60 100 0.15 0.35 0.05 VCESAT 300 400 700 AV IREV ILIM 400 1000 750 400 –0.3 12.6 12.6 30 222 3.47 5.25 12.6 10 130 250 600 24 80 50 300 220 0.4 0.15 450 600 550 750 1000 1500 Max 165 Units µA µA µA µA V V V mV V V V mV mV mV mV kHz % µs nA nA V %/V %/V mV mV mV mV mV mV V/V mA mA %/°C 15 –350 µA mV Parameter QUIESCENT CURRENT QUIESCENT CURRENT, STEP-UP MODE CONFIGURATION INPUT VOLTAGE Conditions Switch Off No Load, ADP1073-3.3 ADP1073-5 ADP1073-12, TA = +25 °C Step-Up Mode Step-Up Mode, TA = +25° C Step-Down Mode ADP10731 ADP1073-3.3 ADP1073-52 ADP1073-122 ADP1073 ADP1073-3.3 ADP1073-5 ADP1073-12 2 VIN COMPARATOR TRIP POINT VOLTAGE OUTPUT SENSE VOLTAGE VOUT 3.14 4.75 11.4 COMPARATOR HYSTERESIS OUTPUT HYSTERESIS OSCILLATOR FREQUENCY MAXIMUM DUTY CYCLE SWITCH ON TIME FEEDBACK PIN BIAS CURRENT SET PIN BIAS CURRENT AO OUTPUT LOW REFERENCE LINE REGULATION SWITCH SATURATION VOLTAGE STEP-UP MODE ADP1073 VFB = 0 V VSET = VREF IAO = 100 µA 1.0 V ≤ VIN ≤ 1.5 V 1.5 V ≤ VIN ≤ 12 V VIN = 1.5 V, ISW = 400 mA, +25 °C TMIN to TMAX VIN = 1.5 V, ISW = 500 mA, +25 °C TMIN to TMAX VIN = 5 V, ISW = 1 A, +25°C TMIN to TMAX RL = 100 kΩ3 4 Full Load (VFB < VREF) DC tON IFB ISET VAO A2 ERROR AMP GAIN REVERSE BATTERY CURRENT CURRENT LIMIT CURRENT LIMIT TEMPERATURE COEFFICIENT SWITCH-OFF LEAKAGE CURRENT MAXIMUM EXCURSION BELOW GND TA = +25°C 220 Ω Between ILIM and VIN TA = +25°C Measured at SW1 Pin TA = +25°C ISW1 ≤ 10 µA, Switch Off TA = +25°C ILEAK VSW2 1 –400 NOTES 1 This specification guarantees that both the high and low trip point of the comparator fall within the 200 mV to 222 mV range. 2 This specification guarantees that the output voltage of the fixed versions will always fall within the specified range. The waveform at the sense pin will exhibit a sawtooth shape due to the comparator hysteresis. 3 100 kΩ resistor connected between a 5 V source and the AO pin. 4 The ADP1073 is guaranteed to withstand continuous application of +1.6 V applied to the GND and SW2 pins while VIN, ILIM and SW1 pins are grounded. All limits at temperature extremes are guaranteed via correlation using standard Quality Control methods. Specifications subject to change without notice. – 2– REV. 0 ADP1073 ABSOLUTE MAXIMUM RATINGS PIN FUNCTION DESCRIPTIONS Input Supply Voltage, Step-Up Mode . . . . . . . . . . . . . . . 15 V Input Supply Voltage, Step-Down Mode . . . . . . . . . . . . . 36 V SW1 Pin Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 50 V SW2 Pin Voltage . . . . . . . . . . . . . . . . . . . . . . . . .–0.4 V to VIN Feedback Pin Voltage (ADP1073) . . . . . . . . . . . . . . . . . . . 5 V Switch Current . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .1.5 A Maximum Power Dissipation . . . . . . . . . . . . . . . . . . 500 mW Operating Temperature Range (A) . . . . . . . . . . 0°C to +70°C Storage Temperature Range . . . . . . . . . . . . –65°C to +150°C Lead Temperature (Soldering, 10 sec) . . . . . . . . . . . . +300°C CADDELL-BURNS 7200-12 82 H 1N5818 ILIM 1.5V AA CELL* VIN SW1 +5V 40mA Pin 1 Mnemonic ILIM Function For normal conditions this pin is connected to VIN. When a lower current limit is required, a resistor should be connected between ILIM and V IN. Limiting the switch current to 400 mA is achieved by connecting a 220 Ω resistor. Input Voltage. Collector Node of Power Transistor. For step-down configuration, connect to VIN; for step-up configuration, connect to an inductor/diode. Emitter Node of Power Transistor. For step- down configuration, connect to inductor/diode; for step-up configuration, connect to ground. Do not allow this pin to drop more than a diode drop below ground. Ground. Auxiliary Gain (GB) Output. The open collector can sink 100 µA. Gain Amplifier Input. The amplifier’s positive input is connected to the SET pin and its negative input is connected to the 212 mV reference. On the ADP1073 (adjustable) version this pin is connected to the comparator input. On the ADP1073-3.3, ADP10735 and ADP1073-12, the pin goes directly to the internal application resistor that sets output voltage. 2 3 VIN SW1 4 SW2 ADP1073-5 SENSE GND SW2 100 F SANYO OS-CON OPERATES WITH CELL VOLTAGE 1.0V *ADD 10 F DECOUPLING CAPACITOR IF BATTERY IS MORE THAN 2 INCHES AWAY FROM ADP1073 5 6 7 GND AO SET Figure 1. Typical Application ORDERING GUIDE Model* ADP1073AN ADP1073AR ADP1073AN-3.3 ADP1073AR-3.3 ADP1073AN-5 ADP1073AR-5 ADP1073AN-12 ADP1073AR-12 Output Voltage ADJ ADJ 3.3 V 3.3 V 5V 5V 12 V 12 V Package Options** N-8 SO-8 N-8 SO-8 N-8 SO-8 N-8 SO-8 8 FB/SENSE PIN CONFIGURATIONS 8-Lead Plastic DIP (N-8) ILIM 1 VIN 2 8 FB (SENSE)* 8-Lead Small Outline Package (SO-8) ILIM 1 VIN 2 8 FB (SENSE)* NOTES **Temperature Range: 0°C to +70°C. **N = Plastic DIP; SO = Small Outline Package. ADP1073 7 SET TOP VIEW SW1 3 (Not to Scale) 6 AO SW2 4 5 GND 7 SET TOP VIEW SW1 3 (Not to Scale) 6 AO SW2 4 5 GND ADP1073 * FIXED VERSIONS * FIXED VERSIONS CAUTION ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the human body and test equipment and can discharge without detection. Although the ADP1073 features proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation or loss of functionality. WARNING! ESD SENSITIVE DEVICE REV. 0 –3– ADP1073 –Typical Performance Characteristics 1.2 VIN = 1.5V 1 VCE (SAT) – Volts VIN = 5.0V 0.8 VIN = 3.0V 0.6 0.4 VIN = 1.25V 0.2 VIN = 1.0V 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1 1.1 1.2 SWITCH CURRENT – Amps VIN = 2.0V SWITCH ON VOLTAGE – Volts 2 1.8 SWITCH CURRENT – mA 1.6 1.4 1.2 1 0.8 0.6 0.4 0.2 0 0.05 0.1 0 0.2 0.3 0.4 0.5 0.6 SWITCH CURRENT – Amps 0.7 10 30 50 70 90 200 400 600 800 1000 RLIM – SATURATION VOLTAGE 1400 1200 1000 800 600 400 200 VIN = 1.5V WITH L = 82 H VIN = 3V WITH L = 82 H VIN = 12V WITH L = 150 H Figure 2. Saturation Voltage vs. Switch Current in Step-Up Mode Figure 3. Switch ON Voltage vs. Switch Current in Step-Down Mode Figure 4. Maximum Switch Current vs. RLIM 1000 120 110 SET PIN BIAS CURRENT – nA 160 140 SUPPLY CURRENT – A VIN = 1.5V 120 100 80 60 40 40 0 25 70 TEMPERATURE – C 85 40 OUTPUT CURRENT – mA 100 90 80 70 60 50 40 100 10 FOR VIN > 1.6V, RLIM = 68 0 1 1.5 2 2.5 3 INPUT VOLTAGE – Volts 3.5 0 25 70 TEMPERATURE – C 85 Figure 5. Guaranteed Minimum Output Current at VOUT = 5 V vs. Input Voltage Figure 6. Set Pin Bias Current vs. Temperature Figure 7. Supply Current vs. Temperature 22 OSCILLATOR FREQUENCY – kHz 21 20 19 18 17 16 15 14 40 0 25 70 TEMPERATURE – C 85 DUTY CYCLE – % 70 68 66 64 62 60 58 56 SWITCH-ON TIME – s 34.5 34 33.5 33 32.5 32 31.5 31 30.5 40 0 25 70 TEMPERATURE – C 85 30 40 0 25 70 TEMPERATURE – C 85 Figure 8. Oscillator Frequency vs. Temperature Figure 9. Duty Cycle vs. Temperature Figure 10. Switch ON Time vs. Temperature –4– REV. 0 ADP1073 2300 2100 GAIN BLOCK GAIN – V/V 1900 1700 1500 1300 1100 40 VIN = 1.5V RL = 100k 0 25 70 TEMPERATURE – C 85 Figure 11. “Gain Block” Gain vs. Temperature THEORY OF OPERATION The ADP1073 is a flexible, low power switch mode power supply (SMPS) controller. The regulated output voltage can be greater than the input voltage (boost or step-up mode) or less than the input (buck or step-down mode). This device uses a gated-oscillator technique to provide very high performance with low quiescent current. A functional block diagram of the ADP1073 is shown on the front page. The internal 212 mV reference is connected to one input of the comparator, while the other input is externally connected (via the FB pin) to a feedback network connected to the regulated output. When the voltage at the FB pin falls below 212 mV, the 19 kHz oscillator turns on. A driver amplifier provides base drive to the internal power switch and the switching action raises the output voltage. When the voltage at the FB pin exceeds 212 mV, the oscillator is shut off. While the oscillator is off, the ADP1073 quiescent current is only 100 µA. The comparator includes a small amount of hysteresis, which ensures loop stability without requiring external components for frequency compensation. The maximum current in the internal power switch can be set by connecting a resistor between VIN and the ILIM pin. When the maximum current is exceeded, the switch is turned OFF. The current limit circuitry has a time delay of about 2 µs. If an external resistor is not used, connect ILIM to VIN. Further information on ILIM is included in the Limiting the Switch Current section of this data sheet. The ADP1073 internal oscillator provides 38 µs ON and 15 µs OFF times, which is ideal for applications where the ratio between VIN and VOUT is roughly a factor of three (such as generating +5 V from a single 1.5 V cell). Wider range conversions, as well as step-down converters, can also be accomplished with a slight loss in the maximum output power that can be obtained. An uncommitted gain block on the ADP1073 can be connected as a low-battery detector, linear post-regulator or undervoltage lockout detector. The inverting input of the gain block is internally connected to the 212 mV reference. The noninverting input is available at the SET pin. A resistor divider, connected between VIN and GND with the junction connected to the SET pin, causes the AO output to go LOW when the input voltage goes below the low battery set point. The AO output is an open collector NPN transistor that can sink 100 µA. The ADP1073 provides external connections for both the collector and emitter of its internal power switch, which permits both step-up and step-down modes of operation. For the stepup mode, the emitter (Pin SW2) is connected to GND and the collector (Pin SW1) drives the inductor. For step-down mode, the emitter drives the inductor while the collector is connected to VIN. The output voltage of the ADP1073 is set with two external resistors. Three fixed-voltage models are also available: ADP1073-3.3 (+3.3 V), ADP1073-5 (+5 V) and ADP1073-12 (+12 V). The fixed-voltage models are identical to the ADP1073, except that laser-trimmed voltage-setting resistors are included on the chip. Only three external components are required to form a +3.3 V, +5 V or +12 V converter. On the fixed-voltage models of the ADP1073, simply connect the feedback pin (Pin 8) directly to the output voltage. The ADP1073 oscillator only turns on when the output voltage is below the programmed voltage. When the output voltage is above the programmed voltage, the ADP1073 remains in its quiescent state to conserve power. Output ripple, which is inherent in gated oscillator converters, is typically 125 mV for a 5 V output and 300 mV for a 12 V output. This ripple voltage can be greatly reduced by inserting the gain-block between the output and the FB pin. Further information and a typical circuit are shown in the Programming the Gain Block section. REV. 0 –5– ADP1073 COMPONENT SELECTION General Notes on Inductor Selection When the ADP1073 internal power switch turns on, current begins to flow in the inductor. Energy is stored in the inductor core while the switch is on, and this stored energy is then transferred to the load when the switch turns off. Both the collector and the emitter of the switch transistor are accessible on the ADP1073, so the output voltage can be higher, lower or of opposite polarity than the input voltage. To specify an inductor for the ADP1073, the proper values of inductance, saturation current and dc resistance must be determined. This process is not difficult, and specific equations for each circuit configuration are provided in this data sheet. In general terms, however, the inductance value must be low enough to store the required amount of energy (when both input voltage and switch ON time are at a minimum) but high enough that the inductor will not saturate when both VIN and switch ON time are at their maximum values. The inductor must also store enough energy to supply the load without saturating. Finally, the dc resistance of the inductor should be low so that excessive power will not be wasted by heating the windings. For most ADP1073 applications, an 82 µH to 1000 µH inductor with a saturation current rating of 300 mA to 1 A is suitable. Ferrite core inductors that meet these specifications are available in small, surface-mount packages. To minimize Electro-Magnetic Interference (EMI), a toroid or pot core type inductor is recommended. Rod core inductors are a lower cost alternative if EMI is not a problem. Calculating the Inductor Value where L is in henrys and R' is the sum of the switch equivalent resistance (typically 0.8 Ω at +25°C) and the dc resistance of the inductor. If the voltage drop across the switch is small compared to VIN, a simpler equation can be used: I L (t ) = V IN t L (4) Replacing t in the above equation with the ON time of the ADP1073 (38 µs, typical) will define the peak current for a given inductor value and input voltage. At this point, the inductor energy can be calculated as follows: EL = 1 L × I 2 PEAK 2 (5) As previously mentioned, EL must be greater than PL/fOSC so the ADP1073 can deliver the necessary power to the load. For best efficiency, peak current should be limited to 1 A or less. Higher switch currents will reduce efficiency because of increased saturation voltage in the switch. High peak current also increases output ripple. As a general rule, keep peak current as low as possible to minimize losses in the switch, inductor and diode. In practice, the inductor value is easily selected using the equations above. For example, consider a supply that will generate 5 V at 25 mA from two alkaline batteries with a 2 V end-of-life voltage. The inductor power required is, from Equation 1: P L = (5V + 0.5V – 2 V ) × (25 mA) = 87.5 mW On each switching cycle, the inductor must supply: P L 87.5 mW = = 4.6 µJ f OSC 19 kHz Selecting the proper inductor value is a simple three-step process: 1. Define the operating parameters: minimum input voltage, maximum input voltage, output voltage and output current. 2. Select the appropriate conversion topology (step-up, stepdown or inverting). 3. Calculate the inductor value, using the equations in the following sections. Inductor Selection—Step-Up Converter Since the inductor power is low, the peak current can also be low. Assuming a peak current of 100 mA as a starting point, Equation 4 can be rearranged to recommend an inductor value: L= V IN 2V t= 38 µs = 760 µH I L( MAX ) 100 mA Substituting a standard inductor value of 470 µH, with 1.2 Ω dc resistance, will produce a peak switch current of: I PEAK = 2V 2.0 Ω –2.0 Ω × 38 µs   1 – e 470 µH  = 149 mA     In a step-up, or boost, converter (Figure 15), the inductor must store enough power to make up the difference between the input voltage and the output voltage. The power that must be stored is calculated from the equation: P L = V OUT + V D – V IN ( MIN ) × IOUT Once the peak current is known, the inductor energy can be calculated from Equation 5: 1 E L = (470 µH ) × (149 mA)2 = 5.2 µJ 2 ( )( ) (1) where VD is the diode forward voltage (≈ 0.5 V for a 1N5818 Schottky). Energy is only stored in the inductor while the ADP1073 switch is ON, so the energy stored in the inductor on each switching cycle must be must be equal to or greater than: PL f OSC (2) in order for the ADP1073 to regulate the output voltage. When the internal power switch turns ON, current flow in the inductor increases at the rate of: I L (t ) = – R ′t  V IN  1– e L  R′   The inductor energy of 5.2 µJ is greater than the PL/f OSC requirement of 4.6 µJ, so the 470 µH inductor will work in this application. The optimum inductor value can be determined by substituting other inductor values into the same equations. When selecting an inductor, the peak current must not exceed the maximum switch current of 1.5 A. The peak current must be evaluated for both minimum and maximum values of input voltage. If the switch current is high when VIN is at its minimum, then the 1.5 A limit may be exceeded at the maximum value of VIN. In this case, the ADP1073’s current (3) –6– REV. 0 ADP1073 limit feature can be used to limit switch current. Simply select a resistor (using Figure 4) that will limit the maximum switch current to the IPEAK value calculated for the minimum value of VIN. This will improve efficiency by producing a constant IPEAK as VIN increases. See the Limiting the Switch Current section of this data sheet for more information. Note that the switch current limit feature does not protect the circuit if the output is shorted to ground. In this case, current is limited only by the dc resistance of the inductor and the forward voltage of the diode. Inductor Selection—Step-Down Converter To avoid exceeding the maximum switch current when the input voltage is at +9 V, an RLIM resistor should be specified. Inductor Selection—Positive-to-Negative Converter The configuration for a positive-to-negative converter using the ADP1073 is shown in Figure 17. As with the step-up converter, all of the output power for the inverting circuit must be supplied by the inductor. The required inductor power is derived from the formula: P L = (| OUT|+V D ) × ( IOUT ) V (8) The step-down mode of operation is shown in Figure 16. Unlike the step-up mode, the ADP1073’s power switch does not saturate when operating in the step-down mode. Switch current should therefore be limited to 600 mA for best performance in this mode. If the input voltage will vary over a wide range, the ILIM pin can be used to limit the maximum switch current. The first step in selecting the step-down inductor is to calculate the peak switch current as follows: I PEAK = 2 × IOUT  V OUT +V D  DC V IN –V SW +V D    The ADP1073 power switch does not saturate in positive-tonegative mode. The voltage drop across the switch can be modeled as a 0.75 V base-emitter diode in series with a 0.65 Ω resistor. When the switch turns on, inductor current will rise at a rate determined by: I L (t ) = – R't  VL  1– e L  R'   (9) where R' = 0.65 Ω + RL(DC) VL = VIN – 0.75 V (6) where DC = duty cycle (0.72 for the ADP1073) VSW = voltage drop across the switch VD = diode drop (0.5 V for a 1N5818) IOUT = output current VOUT = the output voltage VIN = the minimum input voltage As previously mentioned, the switch voltage is higher in stepdown mode than in step-up mode. VSW is a function of switch current and is therefore a function of VIN, L, time and VOUT. For most applications, a VSW value of 1.5 V is recommended. The inductor value can now be calculated: L= V IN (MIN ) –V SW –V OUT × tON I PEAK For example, assume that a –5 V output at 75 mA is to be generated from a +4.5 V to +5.5 V source. The power in the inductor is calculated from Equation 8: P L = (|− 5V|+ 0.5V ) × (75 mA) = 413 mW During each switching cycle, the inductor must supply the following energy: PL 413 mW = = 21.7 µJ f OSC 19 kHz Using a standard inductor value of 330 µH, with 1 Ω dc resistance, will produce a peak switch current of: I PEAK = –1.65 Ω × 38 µs  4.5V – 0.75 V  330 µH 1– e  = 393 mA 0.65 Ω + 1 Ω   (7) where tON = switch ON time (38 µs) If the input voltage will vary (such as an application which must operate from a battery), an RLIM resistor should be selected from Figure 4. The RLIM resistor will keep switch current constant as the input voltage rises. Note that there are separate RLIM values for step-up and step-down modes of operation. For example, assume that +3.3 V at 150 mA is required from a 9 V battery with a 6 V end-of-life voltage. Deriving the peak current from Equation 6 yields: I PEAK = 2 × 150 mA  3.3 + 0.5   6 – 1.5 + 0.5  = 317 mA 0.72   Once the peak current is known, the inductor energy can be calculated from Equation 9: 1 E L = (330 µH ) × (393 mA)2 = 25.5 µJ 2 The inductor energy of 25.5 µJ is greater than the PL/f OSC requirement of 21.7 µJ, so the 330 µH inductor will work in this application. The input voltage varies between only 4.5 V and 5.5 V in this example. Therefore, the peak current will not change enough to require an RLIM resistor and the ILIM pin can be connected directly to VIN. Care should be taken, of course, to ensure that the peak current does not exceed 800 mA. The peak current can than be inserted into Equation 7 to calculate the inductor value: L= 6 –1.5 – 3.3 × 38 µs = 144 µH 317 mA Since 144 µH is not a standard value, the next lower standard value of 100 µH would be specified. REV. 0 –7– ADP1073 Capacitor Selection For optimum performance, the ADP1073’s output capacitor must be carefully selected. Choosing an inappropriate capacitor can result in low efficiency and/or high output ripple. Ordinary aluminum electrolytic capacitors are inexpensive, but often have poor Equivalent Series Resistance (ESR) and Equivalent Series Inductance (ESL). Low ESR aluminum capacitors, specifically designed for switch mode converter applications, are also available, and these are a better choice than general purpose devices. Even better performance can be achieved with tantalum capacitors, although their cost is higher. Very low values of ESR can be achieved by using OS-CON capacitors (Sanyo Corporation, San Diego, CA). These devices are fairly small, available with tape-and-reel packaging and have very low ESR. The effects of capacitor selection on output ripple are demonstrated in Figures 12, 13 and 14. These figures show the output of the same ADP1073 converter, which was evaluated with three different output capacitors. In each case, the peak switch current is 500 mA and the capacitor value is 100 µF. Figure 12 shows a Panasonic HF-series radial aluminum electrolytic. When the switch turns off, the output voltage jumps by about 90 mV and then decays as the inductor discharges into the capacitor. The rise in voltage indicates an ESR of about 0.18 Ω. In Figure 13, the aluminum electrolytic has been replaced by a Sprague 593D-series device. In this case the output jumps about 35 mV, which indicates an ESR of 0.07 Ω. Figure 14 shows an OS-CON SA series capacitor in the same circuit, and ESR is only 0.02 Ω. Figure 14. OS-CON Capacitor If low output ripple is important, the user should consider using the ADP3000. This device switches at 400 kHz, and the higher switching frequency simplifies the design of the output filter. Consult the ADP3000 data sheet for additional details. All potential current paths must be considered when analyzing very low power applications, and this includes capacitor leakage current. OS-CON capacitors have leakage in the 5 µA to 10 µA range, which will reduce efficiency when the load is also in the microampere range. Tantalum capacitors, with typical leakage in the 1 µA to 5 µA range, are recommended for very low power applications. Diode Selection In specifying a diode, consideration must be given to speed, forward voltage drop and reverse leakage current. When the ADP1073 switch turns off, the diode must turn on rapidly if high efficiency is to be maintained. Schottky rectifiers, as well as fast signal diodes such as the 1N4148, are appropriate. The forward voltage of the diode represents power that is not delivered to the load, so VF must also be minimized. Again, Schottky diodes are recommended. Leakage current is especially important in low current applications, where the leakage can be a significant percentage of the total quiescent current. For most circuits, the 1N5818 is a suitable companion to the ADP1073. This diode has a VF of 0.5 V at 1 A, 4 µA to 10 µA leakage and fast turn-on and turn-off times. A surface mount version, the MBRS130T3, is also available. For applications where the ADP1073 is “off” most of the time, such as when the load is intermittent, a silicon diode may provide higher overall efficiency due to lower leakage. For example, the 1N4933 has a 1 A capability, but with a leakage current of less than 1 µA. The higher forward voltage of the 1N4933 reduces efficiency when the ADP1073 delivers power, but the lower leakage may outweigh the reduction in efficiency. For switch currents of 100 mA or less, a Schottky diode such as the BAT85 provides a VF of 0.8 V at 100 mA and leakage less than 1 µA. A similar device, the BAT54, is available in an SOT-23 package. Even lower leakage, in the 1 nA to 5 nA range, can be obtained with a 1N4148 signal diode. General purpose rectifiers, such as the 1N4001, are not suitable for ADP1073 circuits. These devices, which have turn-on times of 10 µs or more, are too slow for switching power supply applications. Using such a diode “just to get started” will result in wasted time and effort. Even if an ADP1073 circuit appears to function with a 1N4001, the resulting performance will not be indicative of the circuit performance when the correct diode is used. –8– REV. 0 Figure 12. Aluminum Electrolytic Figure 13. Tantalum Electrolytic ADP1073 Circuit Operation, Step-Up (Boost) Mode In boost mode, the ADP1073 produces an output voltage that is higher than the input voltage. For example, +5 V can be derived from one alkaline cell (+1.5 V), or +12 V can be generated from a +5 V logic power supply. Figure 15 shows an ADP1073 configured for step-up operation. The collector of the internal power switch is connected to the output side of the inductor, while the emitter is connected to GND. When the switch turns on, Pin SW1 is pulled near ground. This action forces a voltage across L1 equal to VIN – VCE(SAT) and current begins to flow through L1. This current reaches a final value (ignoring second-order effects) of: VIN C2 R3 220 1 2 3 ILIM VIN SW1 FB 8 L1 VOUT R1 D1 1N5818 C1 R2 ADP1073 SW2 4 GND 5 Figure 16. Step-Down Mode Operation I PEAK ≅ V IN –V CE (SAT ) × 38 µs L where 38 µs is the ADP1073 switch’s “on” time. L1 VIN R3* 1 2 D1 VOUT When the switch turns off, the magnetic field collapses. The polarity across the inductor changes and the switch side of the inductor is driven below ground. Schottky diode D1 then turns on and current flows into the load. Notice that the Absolute Maximum Rating for the ADP1073’s SW2 pin is 0.5 V below ground. To avoid exceeding this limit, D1 must be a Schottky diode. Using a silicon diode in this application will generate forward voltages above 0.5 V, which will cause potentially damaging power dissipation within the ADP1073. The output voltage of the buck regulator is fed back to the ADP1073’s FB pin by resistors R1 and R2. When the voltage at pin FB falls below 212 mV, the internal power switch turns “on” again and the cycle repeats. The output voltage is set by the formula:  R1 V OUT = 212 mV × 1+   R2 R1 SW1 3 FB 8 R2 C1 ILIM VIN ADP1073 GND 5 SW2 4 *OPTIONAL Figure 15. Step-Up Mode Operation When the switch turns off, the magnetic field collapses. The polarity across the inductor changes, current begins to flow through D1 into the load and the output voltage is driven above the input voltage. The output voltage is fed back to the ADP1073 via resistors R1 and R2. When the voltage at pin FB falls below 212 mV, SW1 turns “on” again and the cycle repeats. The output voltage is therefore set by the formula:  R1 V OUT = 212 mV × 1+   R2 The output voltage should be limited to 6.2 V or less when using the ADP1073 in step-down mode. If the input voltage to the ADP1073 varies over a wide range, a current limiting resistor at Pin 1 may be required. If a particular circuit requires high peak inductor current with minimum input supply voltage the peak current may exceed the switch maximum rating and/or saturate the inductor when the supply voltage is at the maximum value. See the Limiting the Switch Current section of this data sheet for specific recommendations. Positive-to-Negative Conversion The circuit of Figure 15 shows a direct current path from VIN to VOUT, via the inductor and D1. Therefore, the boost converter is not protected if the output is short circuited to ground. Circuit Operation, Step-Down (Buck) Mode) The ADP1073’s step-down mode is used to produce an output voltage that is lower than the input voltage. For example, the output of four NiCd cells (+4.8 V) can be converted to a +3.3 V logic supply. A typical configuration for step-down operation of the ADP1073 is shown in Figure 16. In this case, the collector of the internal power switch is connected to VIN and the emitter drives the inductor. When the switch turns on, SW2 is pulled up toward VIN. This forces a voltage across L1 equal to (VIN – VCE ) – VOUT, and causes current to flow in L1. This current reaches a final value of: I PEAK V –V CE –V OUT ≅ IN × 38 µs L The ADP1073 can convert a positive input voltage to a negative output voltage, as shown in Figure 17. This circuit is essentially identical to the step-down application of Figure 16, except that the “output” side of the inductor is connected to power ground. When the ADP1073’s internal power switch turns off, current flowing in the inductor forces the output (–VOUT) to a negative VIN R3 ILIM C2 VIN SW1 FB L1 ADP1073 GND SW2 R2 D1 1N5818 C1 R1 VOUT where 38 µs is the ADP1073 switch’s “on” time. Figure 17. A Positive-to-Negative Converter potential. The ADP1073 will continue to turn the switch on until its FB pin is 212 mV above its GND pin, so the output voltage is determined by the formula: REV. 0 –9– ADP1073  R1 V OUT = 212 mV × 1+   R2 The design criteria for the step-down application also apply to the positive-to-negative converter. The output voltage should be limited to |6.2 V| and D1 must be a Schottky diode to prevent excessive power dissipation in the ADP1073. Negative-to-Positive Conversion The circuit of Figure 18 converts a negative input voltage to a positive output voltage. Operation of this circuit configuration is similar to the step-up topology of Figure 16, except that the current through feedback resistor R1 is level-shifted below ground by a PNP transistor. The voltage across R1 is (VOUT – VBE(Q1)). However, diode D2 level-shifts the base of Q1 about 0.6 V below ground, thereby cancelling the VBE of Q1. The addition of D2 also reduces the circuit’s output voltage sensitivity to temperature, which would otherwise be dominated by the –2 mV/°C VBE contribution of Q1. The output voltage for this circuit is determined by the formula:  R1 V OUT = 212 mV × 1+   R2 the switch turns on for the next cycle, the inductor current begins to ramp up from the residual level. If the switch ON time remains constant, the inductor current will increase to a high level (see Figure 19). This increases output ripple and can require a larger inductor and capacitor. By controlling switch current with the ILIM resistor, output ripple current can be maintained at the design values. Figure 20 illustrates the action of the ILIM circuit. Unlike the positive step-up converter, the negative-to-positive converter’s output voltage can be either higher or lower than the input voltage. L1 RLIM ILIM C2 VIN SW1 FB D1 1N5818 POSITIVE OUTPUT R1 Q1 2N3906 CL D2 1N4148 Figure 19. (I LIM Operation, RLIM = 0 Ω) ADP1073 AO SET GND SW2 R2 NEGATIVE INPUT NC NC 10k Figure 18. A Negative-to-Positive Converter Limiting the Switch Current Figure 20. (I LIM Operation, RLIM = 240 Ω) The ADP1073’s RLIM pin permits the switch current to be limited with a single resistor. This current limiting action occurs on a pulse by pulse basis. This feature allows the input voltage to vary over a wide range without saturating the inductor or exceeding the maximum switch rating. For example, a particular design may require peak switch current of 800 mA with a 2.0 V input. If VIN rises to 4 V, however, the switch current will exceed 1.6 A. The ADP1073 limits switch current to 1.5 A and thereby protects the switch, but the output ripple will increase. Selecting the proper resistor will limit the switch current to 800 mA, even if VIN increases. The relationship between RLIM and maximum switch current is shown in Figure 4. The ILIM feature is also valuable for controlling inductor current when the ADP1073 goes into continuous conduction mode. This occurs in the step-up mode when the following condition is met: V OUT +V DIODE 1 < V IN –V SW 1– DC The internal structure of the ILIM circuit is shown in Figure 21. Q1 is the ADP1073’s internal power switch, which is paralleled by sense transistor Q2. The relative sizes of Q1 and Q2 are scaled so that IQ2 is 0.5% of IQ1. Current flows to Q2 through an internal 80 Ω resistor and through the RLIM resistor. These two resistors parallel the base-emitter junction of the oscillatordisable transistor, Q3. When the voltage across R1 and RLIM exceeds 0.6 V, Q3 turns on and terminates the output pulse. If only the 80 Ω internal resistor is used (i.e., the ILIM pin is connected directly to VIN), the maximum switch current will be 1.5 A. Figure 4 gives RLIM values for lower current-limit values. The delay through the current limiting circuit is approximately 2 µs. If the switch ON time is reduced to less than 5 µs, accuracy of the current trip point is reduced. Attempting to program a switch ON time of 2 µs or less will produce spurious responses in the switch ON time. However, the ADP1073 will still provide a properly regulated output voltage. where DC is the ADP1073’s duty cycle. When this relationship exists, the inductor current does not go all the way to zero during the time that the switch is OFF. When –10– REV. 0 ADP1073 RLIM (EXTERNAL) VIN Q3 DRIVER OSCILLATOR Q2 Q1 SW2 ILIM R1 VBAT R2 33k R3 1.6M +5V ADP1073 212mV REF SET GND ADP1073 R1 80 (INTERNAL) VIN AO 47k TO PROCESSOR SW1 Figure 21. Current Limit Operation Programming the Gain Block Figure 22b. Adding Hysteresis to the Low Battery Detector The gain block of the ADP1073 can be used as a low battery detector, error amplifier or linear post regulator. The gain block consists of an op amp with PNP inputs and an open-collector NPN output. The inverting input is internally connected to the ADP1073’s 212 mV reference, while the noninverting input is available at the SET pin. The NPN output transistor will sink about 100 µA. Figure 22a shows the gain block configured as a low-battery monitor. Resistors R1 and R2 should be set to high values to reduce quiescent current, but not so high that bias current in the SET input causes large errors. A value of 100 kΩ for R2 is a good compromise. The value for R1 is then calculated from the formula: R1 = V LOBATT − 212 mV 212 mV R2 The circuit of Figure 22a may produce multiple pulses when approaching the trip point, due to noise coupled into the SET input. To prevent multiple interrupts to the digital logic, hysteresis can be added to the circuit (Figure 22b). Resistor RHYS, with a value of 1 MΩ to 10 MΩ, provides the hysteresis. The addition of RHYS will change the trip point slightly, so the new value for R1 will be: R1 = V LOBATT – 212 mV  212 mV  V L – 212 mV   R2  –  R + R    L HYS  where VL is the logic power supply voltage, RL is the pull-up resistor and R HYS creates the hysteresis. The gain block can also be used as a control element to reduce output ripple. The ADP3000 is normally recommended for lowripple applications, but its minimum input voltage is 2 V. The gain-block technique using the ADP1073 can be useful for stepup converters operating down to 1 V. A step-up converter using this technique is shown in Figure 23. This configuration uses the gain block to sense the output voltage and control the comparator. The result is that the comparator hysteresis is reduced by the open loop gain of the gain block. Output ripple can be reduced to only a few millivolts with this technique, versus a typical value of 150 mV for a +5 V converter using just the comparator. For best results, a large output capacitor (1000 µF or more) should be specified. This technique can also be used for step-down or inverting applications, but the ADP3000 is usually a more appropriate choice. See the ADP3000 data sheet for further details. L1 R3 680k VBAT AO FB D1 VOUT ILIM VIN SW1 SET R2 R1 C1 where VLOBATT is the desired low battery trip point. Since the gain block output is an open-collector NPN, a pull-up resistor should be connected to the positive logic power supply. +5V R1 VBAT R2 ADP1073 212mV REF SET GND VIN AO 100k TO PROCESSOR R1 = R2 VLB (212mV –1) VLB = BATTERY TRIP POINT Figure 22a. Setting the Low Battery Detector Trip Point ADP1073 GND SW2 VOUT = ( R1 +1) (212mV) R2 Figure 23. Using the Gain Block to Reduce Output Ripple REV. 0 –11– ADP1073 –Typical Application Circuits L1* 120 H 1M +12V 1 F* 100 V1 V2 – V1 VSET IIN = 100 *NON-POLARIZED V2 100 F *L1 = GOWANDA GA10-123k OR CADDELL-BURNS 7300-14 ILIM VIN SW1 12V OUTPUT 5mA AT VBATTERY = 1.0V 12mA AT VBATTERY = 1.5V 47 F 1N5818 ADP1073 CIRCUIT 1.5 VOLT CELL ADP1073-12 SENSE GND SW2 Figure 24. Test Circuit Measures No Load Quiescent Current of ADP1073 Converter L1* 120 H 220 1.00M ** ILIM VIN SW1 FB 23.3k ** 47 F TWO 1.5 VOLT CELLS 1N5818 9V OUTPUT 5mA AT VBATTERY = 1.00V 12mA AT VBATTERY = 1.5V Figure 28. 1.5 V to 12 V Step-Up Converter L1* 68 H 51 1N5818 12V OUTPUT 25mA AT VBATTERY = 2.0V 1.5 VOLT CELL ADP1073 GND ILIM VIN SW1 47 F SW2 ADP1073-12 SENSE GND SW2 = GOWANDA GA10-123k OR CADDELL-BURNS 7300-14 **1% METAL FILM *L1 *L1 = GOWANDA GA10-682k OR CADDELL-BURNS 7300-11 Figure 25. 1.5 V to 9 V Step-Up Converter Figure 29. 3 V to 12 V Step-Up Converter L1* 68 H 1N5818 15V OUTPUT 20mA AT VBATTERY = 2.0V L1* 68 H 56 1N5818 5V OUTPUT 100mA AT VBATTERY = 2.0V 51 1.00M ** TWO 1.5 VOLT CELLS ILIM VIN SW1 FB SW2 14.3k ** 47 F TWO 1.5 VOLT CELLS ILIM VIN SW1 100 F ADP1073 GND ADP1073-5 SENSE GND SW2 *L1 = GOWANDA GA10-682k OR CADDELL-BURNS 7300-11 = GOWANDA GA10-682k OR CADDELL-BURNS 7300-11 **1% METAL FILM *L1 Figure 26. 3 V to 5 V Step-Up Converter L1* 120 H 220 536k ** 1.5 VOLT CELL ILIM VIN SW1 FB 40.2k ** 100 F 1N5818 3V OUTPUT 15mA AT VBATTERY = 1.00V 100 F Figure 30. 3 V to 15 V Step-Up Converter 5 VIN L1* 150 H 1N5818 15V OUTPUT 100mA AT 4.5 VIN 100 F 14.3k ** ILIM VIN SW1 FB 1M ** ADP1073 GND ADP1073 GND SW2 SW2 = GOWANDA GA10-123k OR CADDELL-BURNS 7300-14 **1% METAL FILM *L1 = GOWANDA GA10-153k OR CADDELL-BURNS 7200-15 **1% METAL FILM *L1 Figure 27. 1.5 V to 3 V Step-Up Converter Figure 31. 5 V to 15 V Step-Up Converter –12– REV. 0 ADP1073 5 VIN L1* 150 H 3V OUTPUT 1N5818 12V OUTPUT 100mA AT 4.5 VIN 100 F 220 536k ILIM 100 F ILIM 9 VOLT BATTERY GND VIN VIN SW1 SW2 L1* 100 H *L1 SW1 ADP1073 FB ADP1073-12 SENSE GND SW2 40.2k = GOWANDA GA20-153k OR CADDELL-BURNS 7200-15 1N5818 *L1 = GOWANDA GA10-103k OR CADDELL-BURNS 7300-13 100 F Figure 32. 5 V to 12 V Step-Up Converter L1* 82 H 1N5818 5V OUTPUT ILIM 1.5 VOLT CELL VIN SW1 FB 40.2k ** 1N4148 909k ** 100 F 9 VOLT BATTERY Figure 35. 9 V to 3 V Step-Down Converter 220 ADP1073 GND SW2 ILIM GND VIN SW2 SW1 SENSE ADP1073-5 SHUTDOWN *L1 OPERATE 74C04 1N5818 L1* 100 H 5V OUTPUT 100 F = GOWANDA GA10-822k OR CADDELL-BURNS 7200-12 **1% METAL FILM Figure 33. 1.5 V to 5 V Step-Up Converter with Logic Shutdown L1* 82 H 1N5818 5V OUTPUT 442k ** 1.5 VOLT CELL 100k ** 7 *L1 = GOWANDA GA10-103k OR CADDELL-BURNS 7300-13 Figure 36. 9 V to 5 V Step-Down Converter L1* 47 H 1N5818 5V OUTPUT 25mA ILIM VIN SET GND SW1 SENSE AO 100k LO BATT GOES LOW AT VBATTERY = 1.15V 2N3906 ADP1073-5 SW2 2.2 1.5 VOLT CELL 56 100 F ILIM GND VIN SW1 SW2 100 F *L1 ADP1073-5 SENSE = GOWANDA GA10-822k OR CADDELL-BURNS 7300-12 **1% METAL FILM Figure 34. 1.5 V to 5 V Step-Up Converter with Low Battery Detector *L1 = GOWANDA GA10-472k OR CADDELL-BURNS 7300-14 MINIMUM START-UP VOLTAGE = 1.1V Figure 37. 1.5 V to 5 V Bootstrapped Step-Up Converter REV. 0 –13– ADP1073 5V MAIN SUPPLY L1* 82 H 5V TO MEMORY 4.5V WHEN MAIN SUPPLY OPEN 680k 1N5818 ILIM ILIM VIN 806k ** SW1 FB 40.2k ** = GOWANDA GA10-473k OR CADDELL-BURNS 7300-21 **1% METAL FILM EFFICIENCY = 83% AT 5mA LOAD *L1 L1* 470 H 1N5818 5V OUTPUT 5mA AT VBATTERY = 1.00V 10mV p-p RIPPLE 100 F OS-CON 40.2k ** 909k ** VIN SW1 1.5 VOLT CELL 100 F*** FB 1.5 VOLT CELL ADP1073 SET AO GND SW2 ADP1073 GND SW2 *L1 = GOWANDA GA10-822k OR CADDELL-BURNS 7300-12 **1% METAL FILM ***OPTIONAL Figure 38. Memory Backup Supply L1* 68 H 1N5818 51 909k ** 3 VOLT CELL 100k 100k 2N3906 2.2M ILIM AO VIN SW1 100 F 1M ** 5V OUTPUT 100mA LOCKOUT AT 1.6V Figure 41. 1.5 V to 5 V Very Low Noise Step-Up Converter 6.5V TO 12V 680k ILIM FB VIN SW1 L1* 47 H ADP1073 SET AO GND SW2 1N5818 100 F OS-CON ADP1073 SET FB GND SW2 40.3k ** 5VOUT 90mA AT 6.5VIN 900k ** 100k = GOWANDA GA10-682k OR CADDELL-BURNS 7300-11 **1% METAL FILM *L1 = GOWANDA GA10-472k OR CADDELL-BURNS 7300-09 **1% METAL FILM EFFICIENCY = 80% IQ = 130µA OUTPUT RIPPLE = 100mV p-p *L1 40.2k ** Figure 39. 3 V to 5 V Step-Up Converter with Undervoltage Lockout L1* 82 H 680k ILIM 1.5 VOLT CELL FB 909k ** VIN SW1 100 F OS-CON 40.2k ** 1N5818 5V OUTPUT 20mV p-p RIPPLE Figure 42. 9 V to 5 V Reduced Noise Step-Down Converter L1* 25 H 1000 F 10V OUTPUT +6V, 1A AT VIN = 3V 560k ILIM VIN FB 549k ** SW1 2200 F 10V 1N5820 INPUT 3V TO 6V (2 LITHIUM CELLS) ADP1073 SET AO GND SW2 ADP1073 AO SET GND SW2 = GOWANDA GA10-822k OR CADDELL-BURNS 7300-12 **1% METAL FILM *L1 20k ** 1N5818 51 MTP3055EL 2N3906 *L1 Figure 40. 1.5 V to 5 V Low Noise Step-Up Converter 5.1k = COILTRONICS CTX25-5-52 **1% METAL FILM Figure 43. 3 V to 6 V @ 1 A Step-Up Converter –14– REV. 0 ADP1073 OUTLINE DIMENSIONS Dimensions shown in inches and (mm). 8-Lead Plastic DIP (N-8) 0.430 (10.92) 0.348 (8.84) 8 5 0.280 (7.11) 0.240 (6.10) 1 4 PIN 1 0.210 (5.33) MAX 0.160 (4.06) 0.115 (2.93) 0.060 (1.52) 0.015 (0.38) 0.130 (3.30) MIN SEATING PLANE 0.325 (8.25) 0.300 (7.62) 0.195 (4.95) 0.115 (2.93) 0.022 (0.558) 0.100 0.070 (1.77) 0.014 (0.356) (2.54) 0.045 (1.15) BSC 0.015 (0.381) 0.008 (0.204) 8-Lead Small Outline Package (SO-8) 0.1968 (5.00) 0.1890 (4.80) 8 1 5 4 0.1574 (4.00) 0.1497 (3.80) 0.2440 (6.20) 0.2284 (5.80) PIN 1 0.0098 (0.25) 0.0040 (0.10) 0.0688 (1.75) 0.0532 (1.35) 0.0196 (0.50) x 45° 0.0099 (0.25) SEATING PLANE 0.0500 0.0192 (0.49) (1.27) 0.0138 (0.35) BSC 0.0098 (0.25) 0.0075 (0.19) 8° 0° 0.0500 (1.27) 0.0160 (0.41) REV. 0 –15– – 16– C2965–8–10/97 PRINTED IN U.S.A.