LTC3833IFE#PBF

LTC3833 Fast Accurate Step-Down DC/DC Controller with Differential Output Sensing DESCRIPTION FEATURES Wide VIN Range: 4.5V to 38V n V OUT Range: 0.6V to 5.5V n Output Accuracy: ±0.25% at 25°C and ±0.67% over Temperature n Differential Output Sensing Allowing Up to 500mV Line Loss n Fast Load Transient Response n t ON(MIN) = 20ns, tOFF(MIN) = 90ns n Controlled On-Time Valley Current Mode Architecture n Frequency Programmable from 200kHz to 2MHz and Synchronizable to External Clock n R SENSE or Inductor DCR Current Sensing n Overvoltage Protection and Current Limit Foldback n Power Good Output Voltage Monitor n Output Tracking or Adjustable Soft-Start n External V CC Input for Bypassing Internal LDO n 20-Pin QFN (3mm × 4mm) and TSSOP Packages The LTC®3833 is a synchronous step-down DC/DC switching regulator controller targeted for high power applications. It drives all N-channel power MOSFETs. The controlled on-time valley current mode architecture allows for both fast transient response and constant frequency switching in steady-state operation, independent of VIN, VOUT and load current. n Differential output voltage sensing along with a precision internal reference combine to offer ±0.67% output regulation and the ability to correct for up to ±500mV variations in the output terminals due to line losses. The operating frequency can be programmed from 200kHz to 2MHz with an external resistor and can be synchronized to an external clock for noise and EMI sensitive applications. Very low tON and tOFF times allow for near 0% and near 100% duty cycles, respectively. Programmable soft-start or output voltage tracking is available. Safety features include output overvoltage protection, programmable current limit with foldback, and a power good output signal. APPLICATIONS n n n n Distributed Power Systems Point-of-Load Converters Computing Systems Datacomm Systems L, LT, LTC, LTM, OPTI-LOOP, PolyPhase, µModule, Linear Technology and the Linear logo are registered trademarks and Hot Swap, No RSENSE and UltraFast are trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. Protected by U.S. Patents, including 5481178, 5487554, 6580258, 6304066, 6476589, 6774611. TYPICAL APPLICATION 1.5V, 20A, 300kHz High Current Step-Down Converter Efficiency INTVCC 100k VIN PGOOD RUN 0.1µF LTC3833 TRACK/SS 470pF 10k 137k ITH SGND MODE/PLLIN 3.24k SW 0.47µH 0.1µF INTVCC 4.7µF 15k VOUT 1.5V 20A 330µF ×2 10k FORCED CONTINUOUS MODE 80 70 60 50 BG PGND VOSNS+ VOSNS– PULSE-SKIPPING MODE 0.1µF BOOST INTVCC 100 90 TG RT EXTVCC 300kHz 180µF VOUT SENSE– SENSE+ VIN 4.5V TO 14V EFFICIENCY (%) VRNG 40 3833 TA01a VIN = 12V VOUT = 1.5V 0.1 1 10 LOAD CURRENT (A) 100 3833 TA01b 3833f 1 LTC3833 ABSOLUTE MAXIMUM RATINGS (Note 1) VIN Voltage ................................................. –0.3V to 40V BOOST Voltage........................................... –0.3V to 46V SW Voltage.................................................... –5V to 40V INTVCC, EXTVCC, (BOOST-SW), PGOOD, RUN, MODE/PLLIN, VRNG Voltages........................ –0.3V to 6V VOUT, SENSE+, SENSE– Voltages................... –0.6V to 6V VOSNS+, VOSNS – Voltages......... –0.6V to (INTVCC + 0.3V) RT, ITH Voltages...................... –0.3V to (INTVCC + 0.3V) TRACK/SS Voltages...................................... –0.3V to 5V Operating Junction Temperature Range (Notes 2, 3, 4)......................................... –40°C to 125°C Storage Temperature Range................... –65°C to 150°C Lead Temperature (Soldering, 10 sec) FE Package........................................................ 300°C PIN CONFIGURATION PGOOD SENSE+ SENSE– VOUT TOP VIEW TOP VIEW 20 19 18 17 VOSNS– 1 16 BOOST VOSNS+ 2 15 TG TRACK/SS 3 14 SW 21 SGND ITH 4 13 BG 12 PGND VRNG 5 9 10 VIN 8 MODE/PLLIN 7 RUN 11 INTVCC EXTVCC RT 6 PGOOD 1 20 BOOST SENSE+ 2 19 TG SENSE– 3 18 SW VOUT 4 17 BG VOSNS– 5 VOSNS+ 6 TRACK/SS 7 14 VIN ITH 8 13 MODE/PLLIN VRNG 9 12 EXTVCC 21 SGND RT 10 16 PGND 15 INTVCC 11 RUN FE PACKAGE 20-LEAD PLASTIC TSSOP UDC PACKAGE 20-LEAD (3mm × 4mm) PLASTIC QFN TJMAX = 125°C, θJA = 43°C/W EXPOSED PAD (PIN 21) IS SGND, MUST BE SOLDERED TO PCB TJMAX = 125°C, θJA = 38°C/W EXPOSED PAD (PIN 21) IS SGND, MUST BE SOLDERED TO PCB ORDER INFORMATION LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE LTC3833EUDC#PBF LTC3833EUDC#TRPBF LFGT 20-Lead (3mm × 4mm) Plastic QFN –40°C to 125°C LTC3833IUDC#PBF LTC3833IUDC#TRPBF LFGT 20-Lead (3mm × 4mm) Plastic QFN –40°C to 125°C LTC3833EFE#PBF LTC3833EFE#TRPBF LTC3833FE 20-Lead Plastic TSSOP –40°C to 125°C LTC3833IFE#PBF LTC3833IFE#TRPBF LTC3833FE 20-Lead Plastic TSSOP –40°C to 125°C Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. Consult LTC Marketing for information on non-standard lead based finish parts. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ 3833f 2 LTC3833 ELECTRICAL CHARACTERISTICS l denotes the specifications The which apply over the full operating junction + – temperature range, otherwise specifications are at TA = 25°C. VIN = 15V, VFB = VOSNS – VOSNS , unless otherwise noted. (Note 4) SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS General VIN Input Voltage Operating Range 4.5 38 V VOUT Output Voltage Operating Range 0.6 5.5 V IQ Input DC Supply Current Normal Shutdown Supply Current MODE/PLLIN = INTVCC RUN = 0V 2 15 4 25 mA µA tON(MIN) Minimum On-Time VIN = 38V, VOUT = 0.6V 20 ns tOFF(MIN) Minimum Off-Time 90 ns Output Sensing Regulated Differential Feedback Voltage (VOSNS+ – VOSNS–) ITH = 1.2V (Note 5) TA = 25°C TA = 0°C to 85°C TA = –40°C to 125°C l l 0.5985 0.596 0.594 0.6 0.6 0.6 0.6015 0.604 0.606 V V V Regulated Differential Feedback Voltage Over Line, Load and Common Mode (VOSNS+ – VOSNS–) VIN = 4.5V to 38V, ITH = 0.5V to 1.9V, VOSNS– = ±500mV (Note 5) TA = 0°C to 85°C TA = –40°C to 125°C l l 0.594 0.591 0.6 0.6 0.606 0.609 V V gm(EA) Error Amplifier Transconductance ITH = 1.2V (Note 5) l 1.4 1.7 2 mS IVOSNS+ IVOSNS– VOSNS+ Input Bias Current VOSNS– Input Bias Current VFB = 0.6V ±5 ±25 nA VFB = 0.6V –35 –50 µA 100 30 50 120 38 61 mV mV mV VREG Current Sensing VSENSE(MAX) Valley Current Sense Threshold, VSENSE+ – VSENSE–, Peak Current = Valley + Ripple VRNG = 2V, VFB = 0.57V VRNG = 0V, VFB = 0.57V VRNG = INTVCC, VFB = 0.57V VSENSE(MIN) Minimum Current Sense Threshold, VSENSE+ – VSENSE–, Forced Continuous Mode VRNG = 2V, VFB = 0.63V VRNG = 0V, VFB = 0.63V VRNG = INTVCC, VFB = 0.63V VSENSE(CM) SENSE+, SENSE– Voltage Range (Common Mode) Referenced to Signal Ground (SGND) ISENSE SENSE+, SENSE– Input Bias Current VSENSE(CM) = 0.6V VSENSE(CM) = 5V l l l 80 22 39 –50 –15 –25 l –0.5 mV mV mV 5.5 V ±5 1 ±50 4 nA µA 1.2 1.3 V Start-Up and Shutdown VRUN(TH) RUN Pin On Threshold VRUN Rising VRUN(HYS) RUN Pin Hysteresis ISS Soft-Start Charging Current VTRACK/SS = 0V UVLOLOCK INTVCC Undervoltage Lockout l INTVCC Falling l UVLORELEASE INTVCC Undervoltage Lockout Release Switching Frequency and Clock Synchronization INTVCC Rising l f Free Running Switching Frequency RT = 205k RT = 80.6k RT = 18.2k VCLK(IH) Clock Input High Level into MODE/PLLIN VCLK(IL) Clock Input Low Level into MODE/PLLIN 1.1 3.4 175 450 1800 70 mV 1.0 µA 3.65 4.0 V 4.2 4.5 V 200 500 2000 225 550 2200 2 kHz kHz kHz V 0.5 V 3833f 3 LTC3833 ELECTRICAL CHARACTERISTICS l denotes the specifications The which apply over the full operating junction + – temperature range, otherwise specifications are at TA = 25°C. VIN = 15V, VFB = VOSNS – VOSNS , unless otherwise noted. (Note 4) SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS RTG(HI) TG Driver Pull-Up On-Resistance TG High 2.5 Ω RTG(LO) TG Driver Pull-Down On-Resistance TG Low 1.2 Ω RBG(HI) BG Driver Pull-Up On-Resistance BG High 2.5 Ω RBG(LO) BG Driver Pull-Down On-Resistance BG Low 0.8 Ω tDLY(OFF) Top Gate Off to Bottom Gate On Delay Time (Note 6) 20 ns tDLY(ON) Bottom Gate Off to Top Gate On Delay Time (Note 6) 15 ns Gate Drivers Internal VCC Regulator and External VCC INTVCC Internal VCC Voltage 6V < VIN < 38V 5.1 INTVCC (%) Internal VCC Load Regulation ICC = 0mA to 50mA EXTVCC(TH) EXTVCC Switchover Voltage EXTVCC Rising EXTVCC(HYS) EXTVCC Switchover Hysteresis ∆EXTVCC EXTVCC Voltage Drop VEXTVCC = 5V, ICC = 50mA PGDOV PGOOD Upper Threshold VFB Rising (with Respect to Regulated Feedback Voltage VREG) 5 7.5 10 % PGDUV PGOOD Lower Threshold VFB Falling (with Respect to Regulated Feedback Voltage VREG) –10 –7.5 –5 % PGDHYS PGOOD Hysteresis VFB Returning 2 VPGD(LO) PGOOD Low Voltage IPGOOD = 5mA 0.15 tPGD(FALL) Delay from OV/UV Fault to PGOOD Falling 20 µs tPGD(RISE) Delay from OV/UV Recovery to PGOOD Rising 10 µs 4.4 5.3 5.55 V –1 –2 % 4.6 4.75 V 200 mV 200 mV PGOOD Output Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The junction temperature (TJ in °C) is calculated from the ambient temperature (TA in °C) and power dissipation (PD in Watts) as follows: TJ = TA + (PD • θJA) where θJA (in °C/W) is the package thermal impedance provided in the Pin Configuration section for the corresponding package. Note 3: This IC includes overtemperature protection that is intended to protect the device during momentary overload conditions. The maximum rated junction temperature will be exceeded when this protection is active. Continuous operation above the specified absolute maximum operating junction temperature may impair device reliability or permanently damage the device. % 0.4 V Note 4: The LTC3833 is tested under pulsed loading conditions such that TJ ≈ TA. The LTC3833E is guaranteed to meet specifications from 0°C to 85°C junction temperature; specifications over the –40°C to 125°C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. The LTC3833I is guaranteed to meet specifications over the full –40°C to 125°C operating junction temperature range. Note that the maximum ambient temperature consistent with these specifications is determined by specific operating conditions in conjunction with board layout, the rated package thermal impedance and other environmental factors. Note 5: The LTC3833 is tested in a feedback loop that adjusts VFB = VOSNS+ – VOSNS– to achieve a specified error amplifier output voltage (on ITH pin). The specification at 85°C is not tested in production. This specification is assured by design, characterization and correlation to production testing at 125°C. Note 6: Delay times are measured using 50% levels. 3833f 4 LTC3833 TYPICAL PERFORMANCE CHARACTERISTICS Transient Response: Forced Continuous Mode TA = 25°C unless otherwise noted Load Release: Forced Continuous Mode Load Step: Forced Continuous Mode ILOAD 20A/DIV ILOAD 20A/DIV VOUT 50mV/DIV ILOAD 20A/DIV VOUT 50mV/DIV VOUT 50mV/DIV IL 20A/DIV IL 20A/DIV 50µs/DIV LOAD TRANSIENT = 0A TO 20A VIN = 12V, VOUT = 1.5V FIGURE 10 CIRCUIT IL 20A/DIV 5µs/DIV LOAD STEP = 0A TO 20A VIN = 12V, VOUT = 1.5V FIGURE 10 CIRCUIT 3833 G01 Transient Response: PulseSkipping Mode ILOAD 20A/DIV VOUT 50mV/DIV ILOAD 20A/DIV VOUT 50mV/DIV VOUT 50mV/DIV IL 20A/DIV IL 20A/DIV 5µs/DIV LOAD RELEASE = 20A TO 500mA VIN = 12V, VOUT = 1.5V FIGURE 10 CIRCUIT 3833 G05 Soft Start-Up into a Pre-Biased Output Normal Soft Start-Up VIN 5V/DIV RUN 2V/DIV TRACK/SS 200mV/DIV VOUT 500mV/DIV TRACK/SS 200mV/DIV VOUT 500mV/DIV 10ms/DIV VIN = 12V VOUT = 1.5V FIGURE 10 CIRCUIT IL 20A/DIV 5µs/DIV LOAD STEP = 500mA TO 20A VIN = 12V, VOUT = 1.5V FIGURE 10 CIRCUIT 3833 G04 3833 G07 3833 G03 Load Release: Pulse-Skipping Mode Load Step: Pulse-Skipping Mode ILOAD 20A/DIV 50µs/DIV LOAD TRANSIENT = 500mA TO 20A VIN = 12V, VOUT = 1.5V FIGURE 10 CIRCUIT 5µs/DIV LOAD RELEASE = 20A TO 0A VIN = 12V, VOUT = 1.5V FIGURE 10 CIRCUIT 3833 G02 Output Tracking TRACK/SS 200mV/DIV VOUT 500mV/DIV VOUT PRE-BIASED TO 0.75V VIN = 12V 10ms/DIV VOUT = 1.5V FIGURE 10 CIRCUIT 3833 G06 3833 G08 10ms/DIV VIN = 12V VOUT = 1.5V FIGURE 10 CIRCUIT 3833 G09 3833f 5 LTC3833 TYPICAL PERFORMANCE CHARACTERISTICS Overcurrent Protection Short-Circuit Protection CURRENT LIMIT (25A) 7.5A ILOAD 12A ILOAD 12A NOTE 7 BG 5V/DIV Output Regulation vs Load Current 0.2 NORMALIZED ∆VOUT (%) 0 –0.1 0 5 10 15 20 25 VIN (V) 30 35 0.2 VIN = 15V VOUT = 0.6V VOUT NORMALIZED AT ILOAD = 4A 0.1 0 0 2 6 4 ILOAD (A) 8 3833 G13 0.2 2.0 NORMALIZED ∆f (%) NORMALIZED ∆f (%) 0 VOUT = 0.6V ILOAD = 5A f = 500kHz FREQUENCY NORMALIZED AT VIN = 15V –0.5 0 5 10 15 20 25 VIN (V) 30 35 –0.2 –50 –25 10 0 25 50 55 100 125 150 TEMPERATURE (°C) 3833 G15 Non-Synchronized Switching Frequency vs Load Current Non-Synchronized Switching Frequency vs Temperature 1.0 VIN = 15V VOUT = 0.6A f = 500kHz FREQUENCY NORMALIZED AT ILOAD = 4A 1.5 0.5 0 3833 G14 Non-Synchronized Switching Frequency vs Input Voltage 1.0 VIN = 15V VOUT = 0.6V ILOAD = 0A 0.1 VOUT NORMALIZED AT TA = 25°C –0.1 –0.1 –0.2 40 Output Regulation vs Temperature NORMALIZED ∆VOUT (%) VOUT = 0.6V ILOAD = 5A VOUT NORMALIZED AT VIN = 15V 0.1 3833 G12 VIN = 12V 20µs/DIV VOUT = 1.5V FIGURE 10 CIRCUIT NOTE 8: BG IS FORCED HIGH FOR EXTENDED PERIODS TO REMOVE OVERVOLTAGE 0.1 0.5 NORMALIZED ∆f (%) 0.2 NOTE 8 3833 G11 VIN = 12V 1ms/DIV VOUT = 1.5V FIGURE 10 CIRCUIT NOTE 7: INDUCTOR CURRENT REACHES CURRENT LIMIT BEFORE FOLDBACK AND DURING SHORT-CIRCUIT RECOVERY 3833 G10 Output Regulation vs Input Voltage –1.0 IL 20A/DIV INDUCTOR CURRENT FOLDBACK DURING SHORT-CIRCUIT VIN = 12V 10ms/DIV VOUT = 1.5V FIGURE 10 CIRCUIT NORMALIZED ∆VOUT (%) VOUT 200mV/DIV VOUT 1V/DIV VOUT 200mV/DIV –0.2 OVERVOLTAGE TRIGGER SHORT-CIRCUIT REGION IL 10A/DIV VOUT DROOPS DUE TO REACHING CURRENT LIMIT Overvoltage Protection OVERVOLTAGE REGION SHORTCIRCUIT TRIGGER ILOAD 7.5A 10A/DIV IL 10A/DIV TA = 25°C unless otherwise noted 0 –0.1 VIN = 15V, VOUT = 0.6V ILOAD = 0A, f = 500kHz FREQUENCY NORMALIZED AT TA = 25°C 0 –0.5 –1.0 –1.5 40 3833 G16 –0.2 0 2 6 4 ILOAD (A) 8 10 3833 G17 –2.0 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 3833 G18 3833f 6 LTC3833 TYPICAL PERFORMANCE CHARACTERISTICS tON(MIN) and tOFF(MIN) vs Voltage on VOUT Pin 100 90 90 tOFF(MIN) tON(MIN) and tOFF(MIN) vs Switching Frequency 100 tOFF(MIN) 80 70 60 60 60 tON(MIN) 40 50 40 30 30 20 20 VIN = 38V f ≈ 2000kHz 10 0 1 tON(MIN) 2 3 VOUT (V) 4 6 5 0 5 10 10 15 20 25 VIN (V) 35 30 MAXIMUM CURRENT SENSE VOLTAGE (mV) 100 CURRENT SENSE VOLTAGE (mV) 1.60 1.55 1.50 –50 –25 0 80 60 40 20 0 –40 –60 25 50 75 100 125 150 TEMPERATURE (°C) VRNG = 0.6V VRNG = 0.9V VRNG = 1.3V VRNG = 1.6V VRNG = 2.0V –20 0 0.5 1.5 1 ITH VOLTAGE (V) 2 3833 G22 SWITCHING REGION UVLO THRESHOLDS (V) 1.0 STANDBY REGION 0.8 0.6 SHUTDOWN REGION 0.4 0 25 50 75 100 125 150 TEMPERATURE (°C) 3833 G25 VRNG = 1V 40 VRNG = 0.6V 20 0 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) RUN and TRACK/SS Pull-Up Currents vs Temperature UVLO RELEASE (INTVCC RISING) 1.6 RUN 4.1 3.9 UVLO LOCK (INTVCC FALLING) 3.7 3.3 –50 –25 1.4 1.2 TRACK/SS 1.0 0.8 3.5 0.2 60 1.8 4.3 1.2 80 3833 G24 4.5 1.4 VRNG = 2V 100 Input Undervoltage Lockout Thresholds vs Temperature 1.6 0 –50 –25 2.5 120 3833 G23 RUN Thresholds vs Temperature 2000 Maximum Current Sense Voltage vs Temperature 120 1.65 1700 3833 G21 Current Sense Voltage vs ITH Voltage 1.80 TRANSCONDUCTANCE (mS) 40 3833 G20 Error Amplifier Transconductance vs Temperature 1.70 VIN = 38V VOUT = 0.6V 0 500 200 800 1100 1400 FREQUENCY (kHz) VOUT = 0.6V f ≈ 2000kHz 3833 G19 1.75 tON(MIN) 20 10 0 50 40 30 CURRENT (µA) 0 TIME (ns) 80 70 50 tOFF(MIN) 90 70 TIME (ns) TIME (ns) tON(MIN) and tOFF(MIN) vs Voltage on VIN Pin 100 80 RUN PIN THRESHOLDS (V) TA = 25°C unless otherwise noted 0 25 50 75 100 125 150 TEMPERATURE (°C) 3833 G26 0.6 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 3833 G27 3833f 7 LTC3833 PIN FUNCTIONS (FE/UDC) PGOOD (Pin 1/Pin 17): Power Good Indicator Output. This open-drain logic output is pulled to ground when the output voltage is outside of a ±7.5% window around the regulation point. ITH (Pin 8/Pin 4): Current Control Voltage and Switching Regulator Compensation Point. The current sense threshold increases with this control voltage which ranges from 0V to 2.4V. SENSE+ (Pin 2/Pin 18): Differential Current Sensing (+) Input. For RSENSE current sensing, Kelvin (4-wire) connect SENSE+ and SENSE– pins across the sense resistor. For DCR sensing, Kelvin connect SENSE+ and SENSE– pins across the sense filter capacitor. VRNG (Pin 9/Pin 5): Current Sense Voltage Range Input. The maximum allowed sense voltage between SENSE+ and SENSE– is equal to 0.05 • VRNG. If VRNG is tied to SGND, the device operates with a maximum sense voltage of 30mV. If VRNG is tied to INTVCC, the device operates with a maximum sense voltage of 50mV. SENSE– (Pin 3/Pin 19): Differential Current Sensing (–) Input. For RSENSE current sensing, Kelvin (4-wire) connect SENSE+ and SENSE– pins across the sense resistor. For DCR sensing, Kelvin connect SENSE+ and SENSE– pins across the sense filter capacitor. VOUT (Pin 4/Pin 20): Output voltage sense for adjusting the TG on-time for constant frequency operation. Tying this pin to the local output (instead of remote output) is recommended for most applications. This pin can be programmed as needed for achieving the steady-state on-time required for constant frequency operation. VOSNS– (Pin 5/Pin 1): Differential Output Sensing (–) Input. Connect this pin to the negative terminal of the output capacitor. There is a bias current of 35µA (typical) flowing out of this pin. VOSNS+ (Pin 6/Pin 2): Differential Output Sensing (+) Input. Connect this pin to the feedback resistor divider between the positive and negative output capacitor terminals. In nominal operation the LTC3833 will regulate the differential output voltage which is divided down to 0.6V by the feedback resistor divider. TRACK/SS (Pin 7/Pin 3): External Tracking and Soft-Start Input. The LTC3833 regulates the differential feedback voltage (VOSNS+ − VOSNS–) to the smaller of 0.6V or the voltage on the TRACK/SS pin. An internal 1.0μA pull-up current source is connected to this pin. A capacitor to ground at this pin sets the ramp time to the final regulated output voltage. Alternatively, another voltage supply connected through a resistor divider to this pin allows the output to track the other supply during start-up. RT (Pin 10/Pin 6): Switching Frequency Programming Pin. Connect an external resistor from RT to signal ground to program the switching frequency between 200kHz and 2MHz. An external clock applied to MODE/PLLIN must be within ±30% of this free-running frequency to ensure frequency lock. RUN (Pin 11/Pin 7): Digital Run Control Input. RUN self biases high with an internal 1.3µA pull-up. Forcing RUN below 1.2V turns off TG and BG. Taking RUN below 0.75V shuts down all bias and places the LTC3833 into micropower shutdown mode of approximately 15μA. EXTVCC (Pin 12/Pin 8): External VCC Input. When EXTVCC exceeds 4.6V, an internal switch connects this pin to INTVCC and shuts down the internal regulator so that the controller and gate drive power is drawn from EXTVCC. EXTVCC should not exceed VIN. MODE/PLLIN (Pin 13/Pin 9): External Clock Synchronization Input and/or Forced Continuous Mode Input. When an external clock is applied to this pin, the rising TG signal will be synchronized with the rising edge of the external clock. Additionally, this pin determines operation under light load conditions. When either a clock input is detected or MODE/ PLLIN is tied to INTVCC, forced continuous mode operation is selected. Tying this pin to SGND allows discontinuous pulse-skipping mode operation at light loads. 3833f 8 LTC3833 PIN FUNCTIONS VIN (Pin 14/Pin 10): Main Supply Input. The supply voltage can range from 4.5V to 38V. For increased noise immunity decouple this pin to signal ground with an RC filter. The voltage on this pin is also used to adjust the TG on-time in order to maintain constant frequency operation. INTVCC (Pin 15/Pin 11): Internal 5.3V Regulator Output. The driver and control circuits are powered from this voltage. Decouple this pin to power ground with a minimum of 4.7μF ceramic capacitor (CVCC). The anode of the Schottky diode, DB, connects to this pin. PGND (Pin 16/Pin 12): Power Ground Connection. Connect this pin as close as practical to the source of the bottom N-channel power MOSFET, the (–) terminal of CVCC and the (–) terminal of CIN. BG (Pin 17/Pin 13): Bottom Gate Drive Output. This pin drives the gate of the bottom N-channel power MOSFET between INTVCC and power ground. SW (Pin 18/Pin 14): Switch Node Connection. The (–) terminal of the bootstrap capacitor, CB, connects to this node. This pin swings from a diode voltage below ground up to VIN. TG (Pin 19/Pin 15): Top Gate Drive Output. This pin drives the gate of the top N-Channel power MOSFET between VSW and VBOOST . BOOST (Pin 20/Pin 16): Boosted Driver Supply Connection. The (+) terminal of the bootstrap capacitor, CB, as well as the cathode of the Schottky diode, DB, connects to this node. This node swings from INTVCC – VSCHOTTKY to VIN + INTVCC – VSCHOTTKY. SGND (Exposed Pad Pin 21/Exposed Pad Pin 21): Signal Ground Connection. The SGND exposed pad must be soldered to the circuit board for electrical contact and rated thermal performance. All small-signal components should be connected to the signal ground. Connect signal ground to power ground only at one point using a single PCB trace. 3833f 9 LTC3833 FUNCTIONAL DIAGRAM VIN CIN VIN UVLO RUN IN LDO OUT EN + – – BO0ST 3.65V 4.2V VOUT MT L – START LOGIC CONTROL 4.6V VOUT COUT INTVCC INTVCC STOP ONE-SHOT TIMER RSENSE EXTVCC + + – – CB DB SW 1.3µA 0.75V 1.2V TG TG DRV RFB2 CVCC BG DRV TIME ADJUST BG MB RFB1 PGND CLOCK MODE/PLLIN CLOCK DETECT – PLL SYSTEM ICMP – + IREV + SENSE+ SENSE– RT OSCILLATOR RT INTVCC RPGD PGOOD 1µA + – OV + – UV + + – 0.645V EA (gm(EA) = 1.7mS) 0.555V VRNG ITH INTVCC R1 TRACK/SS CSS 0.6V + DA (A = 1) – VOSNS+ VOSNS– 3833 FD SGND RITH CITH1 R2 OPERATION (Refer to Functional Diagram) Main Control Loop The LTC3833 uses valley current mode control to regulate the output voltage in an all N-channel MOSFET DC/DC stepdown converter. Current control is achieved by sensing the inductor current across SENSE+ and SENSE–, either by using an explicit resistor connected in series with the inductor or by implicitly sensing the inductor’s resistive (DCR) voltage drop through an RC filter connected across the inductor. In normal steady-state operation, the top MOSFET is turned on for a fixed time interval proportional to the delay in the one-shot timer. The PLL system adjusts the delay in the one-shot timer until the top MOSFET turn-on is synchronized either to the internal oscillator or the external clock input if provided. As the top MOSFET turns off, the bottom MOSFET turns on with a small time delay (dead time) to avoid shoot-through current. The next switching cycle is initiated when the current comparator, ICMP , senses that inductor current has reached the valley threshold point and turns the bottom MOSFET off immediately and the top MOSFET on. Again in order to avoid shoot-through current there is a small dead-time delay before the top MOSFET turns on. 3833f 10 LTC3833 OPERATION (Refer to Functional Diagram) The voltage on the ITH pin sets the ICMP valley threshold point. The error amplifier, EA, adjusts this ITH voltage by comparing the differential feedback signal, VOSNS+ − VOSNS–, to a 0.6V internal reference voltage. Consequently, the LTC3833 regulates the output voltage by forcing the differential feedback voltage to be equal to the 0.6V internal reference. The difference amplifier, DA, converts the differential feedback signal to a single-ended input for the EA. If the load current increases, it causes a drop in the differential feedback voltage relative to the reference. The EA forces ITH voltage to rise until the average inductor current again matches the load current. Differential Output Sensing The output voltage is resistively divided externally to create a feedback voltage for the controller. The internal difference amplifier, DA, senses this feedback voltage along with the output’s remote ground reference to create a differential feedback voltage. This scheme overcomes any ground offsets between local ground and remote output ground, resulting in a more accurate output voltage. The LTC3833 allows for remote output ground deviations as much as ±500mV with respect to local ground. INTVCC/EXTVCC Power Power for the top and bottom MOSFET drivers and most other internal circuitry is derived from the INTVCC pin. Power on the INTVCC pin is derived in two ways: if the EXTVCC pin is below 4.6V, then an internal 5.3V low dropout linear regulator, LDO, supplies INTVCC power from VIN; if the EXTVCC pin is tied to an external source larger than 4.6V, then the LDO is shut down and an internal switch shorts the EXTVCC pin to the INTVCC pin, thereby powering the INTVCC pin with the external source and helping to increase overall efficiency and decrease internal self heating through power dissipated in the LDO. This external power source could be the output of the step-down switching regulator itself if the output is programmed to higher than 4.6V. The top MOSFET driver is biased from the floating bootstrap capacitor, CB, which normally recharges during each off cycle through an external Schottky diode when the top MOSFET turns off. If the VIN voltage is low and INTVCC drops below 3.65V, undervoltage lockout circuitry disables the external MOSFET driver and prevents the power switches from turning on. Shutdown and Start-Up The LTC3833 can be shut down using the RUN pin. Pulling this pin below 1.2V prevents the controller from switching, and less than 0.75V disables most of the internal bias circuitry, including the INTVCC regulator. When RUN is less than 0.75V, the shutdown IQ is about 15μA. Pulling the RUN pin between 0.75V and 1.2V enables the controller into a standby mode where all internal circuitry is powered-up but the external MOSFET driver is disabled. The standby IQ is about 2mA. Releasing the RUN pin from ground allows an internal 1.3μA current to pull the pin above 1.2V and fully enable the controller including the external MOSFET driver. Alternatively, the RUN pin may be externally pulled up or driven directly by logic. Be careful not to exceed the absolute maximum rating of 6V on this pin. When pulled up by a resistor to an external voltage, the RUN pin will sink up to 35µA of current before reaching 6V. If the external voltage is above 6V (e.g., VIN), select a large enough resistor value so that the voltage on RUN will not exceed 6V. The start-up of the controller’s output voltage, VOUT , is controlled by the voltage on the TRACK/SS pin. When the voltage on the TRACK/SS pin is less than the 0.6V internal reference, the LTC3833 regulates the differential feedback voltage to the TRACK/SS voltage instead of the 0.6V reference. This allows the TRACK/SS pin to be used for programming a ramp-up time for VOUT by connecting an external capacitor from the TRACK/SS pin to SGND. An internal 1μA pull-up current charges this capacitor, creating a voltage ramp on the TRACK/SS pin. As the TRACK/SS voltage rises from 0V to 0.6V (and beyond), the LTC3833 forces the output voltage, VOUT , to ramp up smoothly to its final value. Alternatively, the TRACK/SS pin can be used to track the start-up of VOUT to another external supply as in a master slave configuration. Typically, this requires connecting a resistor divider from the master supply to the TRACK/SS pin (see Soft-Start and Tracking). 3833f 11 LTC3833 OPERATION (Refer to Functional Diagram) When the RUN pin is pulled low to disable the controller or when INTVCC drops below its undervoltage lockout threshold of 3.65V, the TRACK/SS pin is pulled low internally. Light Load Current Operation When the DC load current is less than 1/2 of the peakto-peak inductor current ripple, the inductor current can drop to zero or become negative. If the MODE/PLLIN pin is connected to SGND, the LTC3833 will transition into discontinuous mode operation (also called pulse-skipping mode), where a current reversal comparator, IREV , detects and prevents negative inductor current by shutting off the bottom MOSFET, MB. In this mode, both switches remain off with the output capacitor supplying the load current. As the output capacitor discharges and the output voltage droops lower, the EA will eventually move the ITH voltage above the zero current level to initiate another switching cycle. If the MODE/PLLIN pin is tied to INTVCC or an external clock is applied to MODE/PLLIN, the LTC3833 will be forced to operate in continuous mode (called forced continuous mode) and not transition into discontinuous mode. In this case the current reversal comparator, IREV , is disabled, allowing the inductor current to become negative and thus maintain constant frequency operation. Frequency Selection and External Clock Synchronization The steady-state switching frequency of the LTC3833 is set by an internal oscillator. The frequency of this internal oscillator can be programmed from 200kHz to 2MHz by connecting a resistor from the RT pin to SGND. The RT pin is forced to 1.2V internally. A phase-locked loop (PLL) system synchronizes the TG turn-on to this internal oscillator when no external clock is provided. For applications with stringent frequency or interference requirements, an external clock source connected to the MODE/PLLIN pin can be used to synchronize the TG turn-on to the rising edge of the clock. The LTC3833 operates in forced continuous mode when it is synchronized to the external clock. The external clock frequency has to be within ±30% of the internal oscillator frequency for successful synchronization and the clock input levels should be greater than 2V for HI and less than 0.5V for LO. The MODE/PLLIN pin has an internal 600k pull-down resistor. Power Good and Fault Protection The power good pin, PGOOD, is connected internally to an open-drain N-channel MOSFET. An external pull-up resistor to a voltage supply of up to 6V (or INTVCC) completes the power good detection scheme. Overvoltage and undervoltage comparators OV and UV turn on the MOSFET and pull the PGOOD pin low when the differential feedback voltage is outside a ±7.5% window of the 0.6V reference voltage. The PGOOD pin is also pulled low when the LTC3833 is in the soft-start or tracking phase, when in undervoltage lockout, or when the RUN pin is low (shut down). When the differential feedback voltage is within the ±7.5% requirement, the open-drain NMOS is turned off and the pin is pulled up by an external resistor. There is an internal delay of 10µs before the PGOOD pin will indicate power good once the differential feedback voltage is within the ±7.5% window. When the feedback voltage goes out of the ±7.5% window, there is an internal 20μs delay before PGOOD is pulled low. In an overvoltage condition, MT is turned off and MB is turned on immediately without any delay and held on until the overvoltage condition clears. Foldback current limiting is provided if the output is shorted to ground. As the differential feedback voltage drops, the current threshold voltage on the ITH pin is pulled down and clamped to 1.2V. This reduces the inductor valley current level to 1/4th of its maximum value as the differential feedback approaches 0V. Foldback current limiting is disabled at start-up. 3833f 12 LTC3833 APPLICATIONS INFORMATION The Typical Application on the first page of this data sheet is a basic LTC3833 application circuit. The LTC3833 can be configured to sense the inductor current either through a series sense resistor, RSENSE, or through an RC filter across the inductor (DCR). The choice between the two current sensing schemes is largely a design trade-off between cost, power consumption and accuracy. DCR sensing is becoming popular because it saves expensive current sensing resistors and is more power efficient, especially in high current applications. However, current sensing resistors provide the most accurate current limits for the controller. Once the required output voltage and operating frequency have been determined, external component selection is driven by load requirements, and begins with the selection of inductor and current sensing components. Next, the power MOSFETs are selected. Finally, input and output capacitors are selected. Output Voltage Programming and Differential Output Sensing The LTC3833 integrates differential output sensing with output voltage programming, allowing for simple and seamless design. As shown in Figure 1, the output voltage is programmed by an external resistor divider from the regulated output point to its ground reference. The resistive divider is tapped by the VOSNS+ pin, and the ground reference is sensed by VOSNS–. An optional feed-forward capacitor, CFF , can be used to improve the transient performance of the regulator system as discussed under OPTI-LOOP® Compensation. The resulting output voltage is given according to the following equation:  R  VOUT = 0.6V •  1+ FB2   R  FB1 VOUT LTC3833 VOSNS+ VOSNS– CFF (OPT) RFB2 COUT RFB1 Figure 1. Setting Output Voltage 3833 F01 More precisely, the VOUT value programmed in the previous equation is with respect to the output’s ground reference, and thus is a differential quantity. For example, if VOUT is programmed to 5V and the output ground reference is at –0.5V, then the output will be 4.5V with respect to signal ground. The minimum differential output voltage is limited to the internal reference, 0.6V, and the maximum differential output voltage is 5.5V. The VOSNS+ pin is high impedance with no input bias current. The VOSNS– pin has about 35μA of current flowing out of the pin. Differential output sensing allows for more accurate output regulation in high power distributed systems having large line losses. Figure 2 illustrates the potential variations in the power and ground lines due to parasitic elements. These variations are exacerbated in multi-application systems with shared ground planes. Without differential output sensing, these variations directly reflect as an error in the regulated output voltage. The LTC3833’s differential output sensing can correct for up to ±500mV of variation in the output’s power and ground lines. The LTC3833’s differential output sensing scheme is distinct from conventional schemes where the regulated output and its ground reference are directly sensed with a difference amplifier whose output is then divided down with an external resistive divider and fed into the error amplifier input. This conventional scheme is limited by the common mode input range of the difference amplifier and typically limits differential sensing to the lower range of output voltages. The LTC3833 allows for seamless differential output sensing by sensing the resistively divided feedback voltage differentially. This allows for differential sensing in the full output range from 0.6V to 5.5V. The difference amplifier of the LTC3833 has a –3dB bandwidth of 8MHz, high enough to not affect main loop compensation and transient behavior. To avoid noise coupling into VOSNS+, the resistor divider should be placed near the VOSNS+ and VOSNS– pins and physically close to the LTC3833. The remote output and ground traces should be routed together as a differential pair to the remote output. These traces should be terminated as close as physically possible to the remote output 3833f 13 LTC3833 APPLICATIONS INFORMATION CIN MT LTC3833 VOSNS+ RFB2 VOSNS– + – VIN POWER TRACE PARASITICS L ±VDROP(PWR) MB ILOAD COUT1 RFB1 COUT2 I LOAD GROUND TRACE PARASITICS ±VDROP(GND) OTHER CURRENTS FLOWING IN SHARED GROUND PLANE 3833 F02 Figure 2: Differential Output Sensing Used to Correct Line Loss Variations in a High Power Distributed System with a Shared Ground Plane point that is to be accurately regulated through remote differential sensing. for maximum synchronization margin. Refer to Phase and Frequency Synchronization for further details. Switching Frequency Programming Inductor Selection The choice of operating frequency is a trade-off between efficiency and component size. Lowering the operating frequency improves efficiency by reducing MOSFET switching losses but requires larger inductance and/or capacitance to maintain low output ripple voltage. Conversely, raising the operating frequency degrades efficiency but reduces component size. The operating frequency and inductor selection are interrelated in that higher operating frequencies allow the use of smaller inductor and capacitor values. A higher frequency generally results in lower efficiency because of MOSFET gate charge losses and top MOSFET transition losses. In addition to this basic trade-off, the effect of inductor value on ripple current and low current operation must also be considered. The switching frequency of the LTC3833 can be programmed from 200kHz to 2MHz by connecting a resistor from the RT pin to signal ground. The value of this resistor is given by the following empirical formula: R T [kΩ ] = 41550 – 2.2 f [kHz ] Not counting resistor tolerances, the switching frequency could still have a ±10% deviation from the ideal programmed value. The internal PLL has a synchronization range of ±30% around this programmed frequency. Therefore, during external clock synchronization be sure that the external clock frequency is within this ±30% range of the RT programmed frequency. It is advisable that the RT programmed frequency be equal to the external clock The inductor value has a direct effect on ripple current. The inductor ripple current, ∆IL, decreases with higher inductance or frequency and increases with higher VIN: ∆IL = VOUT f •L  V  •  1– OUT  VIN   Accepting larger values of ∆IL allows the use of low inductances, but results in higher output voltage ripple, higher ESR losses in the output capacitor, and greater core losses. A reasonable starting point for setting ripple current is ∆IL = 0.4 • IOUT(MAX) where IOUT(MAX) is the maximum output current for the application. The maximum ∆IL occurs at the maximum input voltage. To guarantee that 3833f 14 LTC3833 APPLICATIONS INFORMATION ripple current does not exceed a specified maximum, the inductance should be chosen according to: L=   VOUT V •  1− OUT  f • ∆IL(MAX)  VIN(MAX)  Once the value for L is known, the type of inductor must be selected. High efficiency converters generally cannot tolerate the core loss of low cost powdered iron cores, forcing the use of more expensive ferrite, molypermalloy or Kool Mμ cores. Ferrite core material saturates hard, meaning that inductance collapses abruptly when the peak design current is exceeded. This results in an abrupt increase in inductor ripple current and consequent output voltage ripple. Do not allow the core to saturate! A variety of inductors designed for high current, low voltage applications are available from manufacturers such as Sumida, Panasonic, Coiltronics, Coilcraft, Toko, Vishay, Pulse and Wurth. in the LTC3833 and external component values. Note that ITH is close to 2.4V when in current limit. An external resistive divider from INTVCC can be used to set the voltage on the VRNG pin between 0.6V and 2V, resulting in maximum sense voltages between 30mV and 100mV. The wide voltage sense range allows for a variety of applications. The VRNG pin can also be tied to either SGND or INTVCC to force internal defaults. When VRNG is tied to SGND, the device operates with a maximum sense voltage of 30mV. When the VRNG pin is tied to INTVCC, the device operates with a maximum sense voltage of 50mV. RSENSE Inductor Current Sensing A typical RSENSE inductor current sensing scheme is shown in Figure 3. RSENSE is chosen based on the required maximum output current. Given the maximum current, IOUT(MAX), maximum sense voltage, VSENSE(MAX), set by the VRNG pin, and maximum inductor ripple current, ∆IL(MAX), the value of RSENSE can be chosen as: Current Sense Pins and Current Limit Programming Inductor current is sensed through the SENSE+ and SENSE– pins and fed into the internal current comparators. The common mode input voltage range of the current comparators is –0.5V to 5.5V. Both SENSE pins are high impedance inputs. When the common mode range is between –0.5V to 1.1V, there is no input bias current, and when between 1.4V and 5.5V, there is less than 1μA of current flowing into the pins. Between 1.1V and 1.4V, the input bias current will be zero if the common mode voltage is ramped up from 1.1V and less than 1μA if the common mode voltage is ramped down from 1.4V. The high impedance inputs to the current comparator allow accurate DCR sensing. However, care must be taken not to float these pins during normal operation. The maximum allowed sense voltage VSENSE(MAX) between SENSE+ and SENSE– is set by the voltage applied to the VRNG pin and is given by: VSENSE(MAX) = 0.05 • VRNG The current mode control loop does not allow the inductor current valleys to exceed 0.05 • VRNG. In practice, one should allow sufficient margin to account for variations RSENSE = VSENSE(MAX) ∆IL(MAX) IOUT(MAX) – 2 Conversely, given RSENSE and IOUT(MAX), VSENSE(MAX) and thus the VRNG voltage could be determined from the above equation. To assure that the maximum rated output current can be supplied for different operating conditions and component variations, sufficient design margin should be built into these calculations. RSENSE RESISTOR AND PARASITIC INDUCTANCE R ESL VOUT LTC3833 RF SENSE+ SENSE– CF RF 3833 F03 FILTER COMPONENTS PLACED NEAR SENSE PINS Figure 3. RSENSE Current Sensing 3833f 15 LTC3833 APPLICATIONS INFORMATION Because of possible PCB noise in the current sensing loop, the current ripple of ∆VSENSE = ∆IL • RSENSE also needs to be checked in the design to get a good signal-to-noise ratio. In general, for a reasonably good PCB layout, a 10mV ∆VSENSE voltage is recommended as a conservative number to start with, either for RSENSE or DCR sensing applications. For today’s highest current density solutions the value of the sense resistor can be less than 1mΩ and the maximum sense voltage can be as low as 30mV. In addition, inductor ripple currents greater than 50% with operation up to 2MHz are becoming more common. Under these conditions, the voltage drop across the sense resistor’s parasitic inductance becomes more relevant. A small RC filter placed near the IC has been traditionally used to reduce the effects of capacitive and inductive noise coupled in the sense traces on the PCB. A typical filter consists of two series 10Ω resistors connected to a parallel 1000pF capacitor, resulting in a time constant of 20ns. The filter components need to be placed close to the IC. The positive and negative sense traces need to be routed as a differential pair and Kelvin (4-wire) connected to the sense resistor. DCR Inductor Current Sensing For applications requiring higher efficiency at high load currents, the LTC3833 is capable of sensing the voltage drop across the inductor DCR, as shown in Figure 4. The DCR of the inductor represents the small amount of DC winding resistance, which can be less than 1mΩ for today’s low value, high current inductors. In a high current application requiring such an inductor, conduction INDUCTOR L DCR VOUT COUT L/DCR = (R1||R2) C1 LTC3833 R1 SENSE+ C1 SENSE– R2 (OPT) 3833 F04 loss through a sense resistor would cost several points of efficiency compared to DCR sensing. The inductor DCR is sensed by connecting an RC filter across the inductor. This filter typically consists of one or two resistors (R1 and R2) and one capacitor (C1) as shown in Figure 4. If the external R1||R2 • C1 time constant is chosen to be exactly equal to the L/DCR time constant, the voltage drop across the external capacitor is equal to the voltage drop across the inductor DCR multiplied by R2/(R1 + R2). Therefore, R2 may be used to scale the voltage across the sense terminals when the DCR is greater than the target sense resistance. With the ability to program current limit through the VRNG pin, R2 may be optional. C1 is usually selected to be in the range of 0.01μF to 0.47μF. This forces R1|| R2 to around 2k to 4k, reducing error that might have been caused by the SENSE pins’ input bias currents. The first step in designing DCR current sensing is to determine the DCR of the inductor. Where provided, use the manufacturer’s maximum value, usually given at 25°C. Increase this value to account for the temperature coefficient of resistance, which is approximately 0.4%/°C. A conservative value for inductor temperature TL is 100°C. The DCR of the inductor can also be measured using a good RLC meter, but the DCR tolerance is not always the same and varies with temperature; consult the manufacturers’ datasheets for detailed information. From the DCR value, VSENSE(MAX) is calculated as: ( If VSENSE(MAX) is within the maximum sense voltage of the LTC3833 as programmed by the VRNG pin (30mV to 100mV), then the RC filter only needs R1. If VSENSE(MAX) is higher, then R2 may be used to scale down the maximum sense voltage so that it falls within range. The maximum power loss in R1 is related to duty cycle, and will occur in continuous mode at the maximum input voltage: C1 NEAR SENSE PINS Figure 4. DCR Current Sensing 16 ) VSENSE(MAX) = DCRMAX at 25°C • 1+ 0.4% TL(MAX) – 25°C    • IOUT(MAX) – ∆IL /2 PLOSS (R1) = (V IN(MAX) – VOUT R1 )• V OUT 3833f LTC3833 APPLICATIONS INFORMATION Ensure that R1 has a power rating higher than this value. If high efficiency is necessary at light loads, consider this power loss when deciding whether to use DCR sensing or RSENSE sensing. Light load power loss can be modestly higher with a DCR network than with a sense resistor due to the extra switching losses incurred through R1. However, DCR sensing eliminates a sense resistor, reduces conduction losses and provides higher efficiency at heavy loads. Peak efficiency is about the same with either method. To maintain a good signal-to-noise ratio for the current sense signal, use a minimum ∆VSENSE of 10mV. For a DCR sensing application, the actual ripple voltage will be determined by: ∆VSENSE = VIN – VOUT VOUT • R1• C1 VIN • f Power MOSFET Selection Two external power MOSFETs must be selected for the LTC3833 controller: one N-channel MOSFET for the top (main) switch and one N-channel MOSFET for the bottom (synchronous) switch. The peak-to-peak drive levels are set by the INTVCC voltage. This voltage is typically 5.3V. Consequently, logic-level threshold MOSFETs must be used in most applications. Pay close attention to the BVDSS specification for the MOSFETs as well; most of the logic-level MOSFETs are limited to 30V or less. Selection criteria for the power MOSFETs include the on-resistance, RDS(ON), Miller capacitance, CMILLER, input voltage and maximum output current. Miller capacitance, CMILLER, can be approximated from the gate charge curve usually provided on the MOSFET manufacturers’ data sheet. CMILLER is equal to the increase in gate charge along the horizontal axis where the curve is approximately flat, divided by the specified change in VDS. This result is then multiplied by the ratio of the application VDS to the gate charge curve specified VDS. When the IC is operating in continuous mode, the duty cycles for the top and bottom MOSFETs are given by: Main Switch Duty Cycle (DTOP ) = VOUT VIN Synchronous Switch Duty Cycle (DBOT ) = 1– The MOSFET power dissipations at maximum output current are given by: PTOP = DTOP •IOUT(MAX)2 •RDS(ON)(MAX) (1+ δ ) + VIN2 R TG(LO)  R TG(HI)  IOUT(MAX)   • CMILLER  • +   •f 2    VINTVCC – VMILLER VMILLER  PBOT = DBOT • IOUT(MAX)2 • RDS(ON)(MAX) (1 + δ) where DTOP and DBOT are the duty cycles of the top MOSFET and bottom MOSFET respectively, δ is the temperature dependency of RDS(ON), RTG(HI) is the TG pull-up resistance, and RTG(LO) is the TG pull-down resistance. VMILLER is the Miller effect VGS voltage and is taken graphically from the MOSFET’s data sheet. Both MOSFETs have I2R losses while the topside N-channel equation includes an additional term for transition losses, which are highest at high input voltages. For VIN < 20V, the high current efficiency generally improves with larger MOSFETs, while for VIN > 20V, the transition losses rapidly increase to the point that the use of a higher RDS(ON) device with lower CMILLER actually provides higher efficiency. The synchronous MOSFET losses are greatest at high input voltage when the top switch duty factor is low or during short-circuit when the synchronous switch is on close to 100% of the period. The term (1 + δ) is generally given for a MOSFET in the form of a normalized RDS(ON) vs temperature curve, but δ = 0.005/°C • (TJ – TA) can be used as an approximation for low voltage MOSFETs (TJ is estimated junction temperature of the MOSFET and TA is ambient temperature). CIN and COUT Selection In continuous mode, the source current of the top N-channel MOSFET is a square wave of duty cycle VOUT/VIN. To prevent large voltage transients, a low ESR input capacitor sized for the maximum RMS current must be used. The maximum RMS capacitor current is given by: VOUT VIN IRMS ≅IOUT(MAX) • VOUT • VIN VIN –1 VOUT This formula has a maximum at VIN = 2VOUT , where IRMS = IOUT(MAX)/2. This simple worst-case condition is commonly used for design because even significant deviations 3833f 17 LTC3833 APPLICATIONS INFORMATION do not offer much relief. Note that capacitor manufacturers’ ripple current ratings for electrolytic and conductive polymer capacitors are often based on only 2000 hours of life. This makes it advisable to further derate the capacitor or to choose a capacitor rated at a higher temperature than required. The selection of COUT is primarily determined by the effective series resistance, ESR, to minimize voltage ripple. The output ripple, ∆VOUT , in continuous mode is determined by:   1 ∆VOUT ≤ ∆IL  RESR + 8 • f • COUT   The output ripple is highest at maximum input voltage since ∆IL increases with input voltage. Typically, once the ESR requirement for COUT has been met, the RMS current rating generally far exceeds the peak-to-peak current ripple requirement. The choice of using smaller output capacitance increases the ripple voltage due to the discharging term but can be compensated for by using capacitors of very low ESR to maintain the ripple voltage. Multiple capacitors placed in parallel may be needed to meet the ESR and RMS current handling requirements. Dry tantalum, special polymer, aluminum electrolytic and ceramic capacitors are all available in surface mount packages. Special polymer capacitors offer very low ESR but have lower capacitance density than other types. Tantalum capacitors have the highest capacitance density but it is important to only use types that have been surge tested for use in switching power supplies. Aluminum electrolytic capacitors have significantly higher ESR, but can be used in cost-sensitive applications provided that consideration is given to ripple current ratings and long-term reliability. Ceramic capacitors have excellent low ESR characteristics but can have a high voltage coefficient and audible piezoelectric effects. The high Q of ceramic capacitors with trace inductance can also lead to significant ringing. When used as input capacitors, care must be taken to ensure that ringing from inrush currents and switching does not pose an overvoltage hazard to the power switches and controller. For high switching frequencies, reducing output ripple and better EMI filtering may require small-value capacitors that have low ESL (and correspondingly higher self resonant frequencies) to be placed in parallel with larger value capacitors that have higher ESL. This will ensure good noise and EMI filtering in the entire frequency spectrum of interest. Even though ceramic capacitors generally have good high frequency performance, small ceramic capacitors may still have to be parallel connected with large ones to optimize performance. Top MOSFET Driver Supply (CB, DB) An external bootstrap capacitor, CB, connected to the BOOST pin supplies the gate drive voltage for the topside MOSFET. This capacitor is charged through diode DB from INTVCC when the switch node is low. When the top MOSFET turns on, the switch node rises to VIN and the BOOST pin rises to approximately VIN + INTVCC. The boost capacitor needs to store approximately 100 times the gate charge required by the top MOSFET. In most applications a 0.1μF to 0.47μF, X5R or X7R dielectric capacitor is adequate. It is recommended that the BOOST capacitor be no larger than 10% of the INTVCC capacitor, CVCC, to ensure that the CVCC can supply the upper MOSFET gate charge and BOOST capacitor under all operating conditions. Variable frequency in response to load steps offers superior transient performance but requires higher instantaneous gate drive. Gate charge demands are greatest in high frequency low duty factor applications under high dI/dt load steps and at start-up. In order to minimize SW node ringing and EMI, connect a 5Ω to 10Ω resistor in series with the BOOST pin. Make the CB and DB connections on the other side of the resistor. This series resistor helps to slow down the TG rise time, limiting the high dI/dt current through the top MOSFET that causes SW node ringing. INTVCC Regulator and EXTVCC Power The LTC3833 features a PMOS low dropout linear regulator (LDO) that supplies power to INTVCC from the VIN supply. INTVCC powers the gate drivers and much of the LTC3833’s internal circuitry. The LDO regulates the voltage at the INTVCC pin to 5.3V. The LDO can supply a maximum current of 50mARMS and must be bypassed to ground with a minimum of 4.7μF ceramic capacitor. Good bypassing is needed to supply 3833f 18 LTC3833 APPLICATIONS INFORMATION the high transient currents required by the MOSFET gate drivers. High input voltage applications in which large MOSFETs are being driven at high frequencies may cause the maximum junction temperature rating for the LTC3833 to be exceeded, especially if the LDO is active and provides INTVCC. Power dissipation for the IC in this case is highest and is approximately equal to VIN • IINTVCC. The gate charge current is dependent on operating frequency as discussed in the Efficiency Considerations section. The junction temperature can be estimated by using the equations given in Note 2 of the Electrical Characteristics. For example, when using the LDO, LTC3833’s INTVCC current is limited to less than 38mA from a 38V supply at TA =70°C in the FE package: TJ = 70°C + (38mA)(38V)(38°C/W) ≈ 125°C To prevent the maximum junction temperature from being exceeded, the input supply current must be checked while operating in continuous conduction mode at maximum VIN. When the voltage applied to EXTVCC pin rises above 4.6V, the INTVCC LDO is turned off and the EXTVCC is connected to INTVCC with an internal switch. This switch remains on as long as the voltage applied to EXTVCC remains above 4.4V. Using the EXTVCC allows the MOSFET driver and control power to be derived from the LTC3833’s switching regulator output during normal operation and from the LDO when the output is out of regulation (e.g., start-up, short circuit). If more than 50mARMS current is required through EXTVCC, then an external Schottky diode can be added between the EXTVCC and INTVCC pins. Do not apply more than 6V to the EXTVCC pin and make sure that this external voltage source is less than VIN. Significant efficiency and thermal gains can be realized by powering INTVCC from the switching regulator output, since the VIN current resulting from the driver and control currents will be scaled by a factor of (Duty Cycle)/(Switcher Efficiency). Tying the EXTVCC pin to a 5V supply reduces the junction temperature in the previous example from 125°C to: TJ = 70°C + (38mA)(5V)(38°C/W) ≈ 77°C However, for 3.3V and other low voltage outputs, additional circuitry is required to derive EXTVCC power from the regulator output. The following list summarizes the four possible connections for EXTVCC: 1. EXTVCC left open (or grounded). This will cause INTVCC to be powered from the internal 5.3V LDO resulting in an efficiency penalty of up to 10% at high input voltages. 2. EXTVCC connected directly to switching regulator output VOUT > 4.6V. This provides the highest efficiency. 3. EXTVCC connected to an external supply. If a 4.6V or greater external supply is available, it may be used to power EXTVCC providing that the external supply is sufficient enough for MOSFET gate drive requirements. 4. EXTVCC connected to an output-derived boost network. For 3.3V and other low voltage converters, efficiency gains can still be realized by connecting EXTVCC to an output-derived voltage that has been boosted to greater than 4.6V. For applications where the main input power is less than 5.3V, tie the VIN and INTVCC pins together and tie the combined pins to the VIN input with an optional 1Ω or 2.2Ω resistor as shown in Figure 5 to minimize the voltage drop caused by the gate charge current. This will override the INTVCC LDO and will prevent INTVCC from dropping too low due to the dropout voltage. Make sure the INTVCC voltage exceeds the RDS(ON) test voltage for the external MOSFET which is typically at 4.5V for logic-level devices. LTC3833 INTVCC VIN RVIN VIN CVCC CIN 3833 F05 Figure 5. Setup for VIN ≤ 5V 3833f 19 LTC3833 APPLICATIONS INFORMATION VIN Undervoltage Lockout (UVLO) The LTC3833 has two functions that help protect the controller in case of input undervoltage conditions. A precision UVLO comparator constantly monitors the INTVCC voltage to ensure that an adequate gate-drive voltage is present. The comparator enables UVLO and locks out the switching action until INTVCC rises above 4.2V. Once UVLO is released, the comparator does not retrigger UVLO until INTVCC falls below 3.65V. This hysteresis prevents oscillations when there are disturbances on INTVCC. Another way to detect an undervoltage condition is to monitor the VIN supply. Because the RUN pin has a precision turn-on voltage of 1.2V, one can use a resistor divider from VIN to turn on the IC when VIN is high enough. The RUN pin has bias currents that depend on the RUN voltage as well as VIN voltage. These bias currents should be taken into account when designing the voltage divider and UVLO circuit to prevent faulty conditions. Generally for RUN < 3V a bias current of 1.3μA flows out of the RUN pin, and for RUN > 3V, correspondingly increasing current flows into the pin, reaching a maximum of about 35μA for RUN = 6V. Soft-Start and Tracking The LTC3833 has the ability to either soft-start by itself with a capacitor or track the output of an external supply. Soft-start or tracking features are achieved not by limiting the maximum output current of the switching regulator but by controlling the regulator’s output voltage according to the ramp rate on the TRACK/SS pin. When configured to soft-start by itself, a capacitor should be connected to the TRACK/SS pin. TRACK/SS is pulled low until the RUN pin voltage exceeds 1.2V and UVLO is released, at which point an internal current of 1μA charges the soft-start capacitor, CSS, connected to TRACK/SS. Current foldback is disabled during this phase to ensure smooth soft-start or tracking. The soft-start or tracking range is defined to be the voltage range from 0V to 0.6V on the TRACK/SS pin. The total soft-start time can be calculated as: C t SOFTSTART = 0.6V • SS 1µA When the LTC3833 is configured to track another supply, a voltage divider can be used from the tracking supply to the TRACK/SS pin to scale the ramp rate appropriately. Two common implementations of tracking as shown in Figure 6a are coincident and ratiometric. For coincident tracking, make the divider ratio from the external supply the same as the divider ratio for the differential feedback voltage. Ratiometric tracking could be achieved by using a different ratio than the differential feedback (Figure 6b). Note that the small soft-start capacitor charging current is always flowing, producing a small offset error. To minimize this error, select the tracking resistive divider values to be small enough to make this offset error negligible. Phase and Frequency Synchronization For applications that require better control of EMI and switching noise or have special synchronization needs, the LTC3833 can phase and frequency synchronize the turn-on of the top MOSFET to an external clock signal applied to the MODE/PLLIN pin. The applied clock signal needs to be within ±30% of the RT pin programmed free-running frequency to assure proper frequency and phase lock. The clock signal levels should generally comply to VIH > 2V and VIL < 0.5V. The MODE/PLLIN pin has an internal 600k pull-down resistor to ensure pulse-skipping mode if the pin is left floating. The LTC3833 uses the voltages on VIN and VOUT pins as well as the RT programmed frequency to determine the steady-state on-time as follows: tON ≈ VOUT VIN • f An internal PLL system adjusts this on-time dynamically in order to maintain phase and frequency lock with the external clock. The LTC3833 will maintain phase and frequency lock under steady-state conditions for VIN, VOUT and load current. As shown in the previous equation, the top MOSFET ontime is a function of the switching regulator’s output. This output is measured by the VOUT pin and is used to calculate the required on-time. Therefore, simply connecting VOUT to the regulator’s local output point is preferable for most applications. However, there could be applications where 3833f 20 LTC3833 APPLICATIONS INFORMATION VOUT EXTERNAL SUPPLY VOLTAGE VOLTAGE EXTERNAL SUPPLY VOUT TIME TIME Coincident Tracking Ratiometric Tracking 3833 F06 Figure 6a. Two Different Modes of Output Tracking EXT. V TO TRACK/SS VOUT RFB2 RFB2 TO VOSNS+ RFB1 RFB1 VOUT EXT. V TO TRACK/SS R1 R2 0.6V ≥ R1+ R2 EXT. V R2 TO VOSNS– RFB2 TO VOSNS+ RFB1 TO VOSNS– 3833 F06b Coincident Tracking Setup Ratiometric Tracking Setup Figure 6b. Setup for Coincident and Ratiometric Tracking the internally calculated on-time differs significantly from the real on-time required by the application. For example, if there are differences between the local output point and the remotely regulated output point due to line losses, then the internally calculated on-time will be inaccurate. Lower efficiencies in the switching regulator can also cause the real on-time to be significantly different from the internally calculated on-time (see Efficiency Considerations). For these circumstances, the voltage on the VOUT pin can be programmed with a resistive divider from INTVCC or from the regulator’s output itself. Note that there is a 500k nominal resistance looking into the VOUT pin. The PLL adjusted on-time achieved after phase locking is the steady-state on-time required by the switching regulator, and if the VOUT programmed on-time is substantially equal to this steady-state on-time, then the PLL system does not have to use its ±30% frequency lock range for systematic corrections. Instead the lock range can be used to correct for component variations or other operating point conditions. If needed, the VOUT pin can be programmed to achieve the steady-state on-time as required by the application and therefore maintain constant frequency operation. If the application requires very low on-times approaching minimum on-time, the PLL system may not be able to maintain a ±30% synchronization range. In fact, there is a possibility of losing phase/frequency lock at minimum on-time, and definitely losing phase/frequency lock for applications requiring less than minimum on-time. This is discussed further under Minimum On-Time, Minimum Off-Time and Dropout Operation. During dynamic transient conditions either in the line or load (e.g., load step or release), the LTC3833 may lose phase and frequency lock in the process of achieving faster transient response. For large slew rates (e.g., 10A/µs), phase and frequency lock will be lost (see Figure 7) until the system returns back to a steady-state condition at which point the device will resume frequency lock and eventually achieve phase lock to the external clock. For relatively small slew rates (10A/s), phase and frequency lock can still be maintained. 3833f 21 LTC3833 APPLICATIONS INFORMATION ILOAD CLOCK INPUT PHASE LOCKED LOSES PHASE LOCK DUE TO FAST LOAD STEP ESTABLISHES FREQUENCY LOCK SOON ESTABLISHES PHASE LOCK AFTER ~600µs LOSES PHASE LOCK DUE TO FAST LOAD RELEASE ESTABLISHES FREQUENCY LOCK SOON SW VOUT 3833 F07 Figure 7. Phase and Frequency Locking Behavior During Transient Load Conditions For light loading conditions, the phase and frequency synchronization will be active if there is a clock input applied. If there is no clock input during light loading, then the switching frequency is based on what the MODE/PLLIN pin is tied to. When MODE/PLLIN is tied to INTVCC, the LTC3833 will operate in forced continuous mode at the RT programmed free-running frequency. When MODE/PLLIN pin is tied to signal ground, the LTC3833 will operate in pulse-skipping discontinuous conduction mode for light loading and will switch to continuous conduction (at the free-running frequency) for normal and heavy loads. Minimum On-Time, Minimum Off-Time and Dropout Operation The minimum on-time is the smallest duration of time in which the LTC3833 can keep the top power MOSFET ’s gate (TG) in its on state. This minimum on-time is 20ns for the LTC3833 and is achieved when the VOUT pin is tied to its minimum value of 0.6V while the VIN is tied to its maximum value of 38V. For larger values of VOUT or smaller values of VIN, the minimum on-time achievable will be longer than 20ns. The minimum on-time will have a dependency on the operating conditions of the switching regulator, but is intended to be smaller for high step-down ratio applications that will require low on-times. The effective minimum on-time of the switching regulator, however, will depend also on external components (especially the characteristics of the power MOSFETs) as well as operating conditions of the switching regulator. The effective on-time is defined as the time period that the SW node stays high and this period can be different from the time period that the top MOSFET ’s gate stays high. One of the factors that contributes to this discrepancy is the on/off switching characteristics of the power MOSFETs. If, for example, the power MOSFET ’s turn-on delay is much smaller than the turn-off delay, then the effective on-time will be longer than the MOSFET ’s gate turn-on-time, thereby limiting the minimum on-time to a longer value than that forced by the LTC3833. Light loading operation in forced continuous mode will also elongate the effective minimum on-time, as shown in Figure 8. At light loading, the dead times between the top MOSFET switching on/off and the bottom MOSFET switching on/off add to the intrinsic on-time of the top MOSFET. In forced continuous light loading, when the inductor current flows in the reverse direction, the SW node is pre-biased high during the dead time from the bottom FET turning off to the top FET turning on. On the other edge, when the top MOSFET turns off and before the bottom MOSFET turns on, the SW node lingers high for a longer duration of time due to a smaller magnitude of inductor current available in light loading to pull the SW node low. In continuous mode operation, the minimum on-time limit imposes a minimum duty cycle of: DMIN = f • tON(MIN) 3833f 22 LTC3833 APPLICATIONS INFORMATION TG-SW (VGS OF TOP MOSFET) then immediately turned back on. This minimum off-time includes the time to turn on the bottom power MOSFET ’s gate and turn it back off along with the dead time delays from top MOSFET off to bottom MOSFET on and bottom MOSFET off to top MOSFET on. The minimum off-time that the LTC3833 can achieve is 90ns. DEAD-TIME DELAYS BG (VGS OF BOTTOM MOSFET) IL 0 NEGATIVE INDUCTOR CURRENT IN FCM VIN SW 3833 F08 DURING BG-TG DEAD TIME, NEGATIVE INDUCTOR CURRENT WILL FLOW THROUGH TOP MOSFET’S BODY DIODE TO PRECHARGE SW NODE IL SW + – DURING TG-BG DEAD TIME, THE RATE OF SW NODE DISCHARGE WILL DEPEND ON THE CAPACITANCE ON THE SW NODE AND INDUCTOR CURRENT MAGNITUDE VIN L L IL TOTAL CAPACITANCE ON THE SW NODE Figure 8. Light Loading On-Time Extension with Forced Continuous Mode Operation where tON(MIN) is the effective minimum on-time for the switching regulator. As the equation shows, reducing the operating frequency will alleviate the minimum duty cycle constraint. If the application requires a smaller than minimum duty cycle, the output voltage will still remain in regulation, but the switching frequency will decrease from its programmed value or lose frequency synchronization if using an external clock. Depending on the application, this may not be of critical importance. For applications that require relatively low on-times, proper caution has to be taken when choosing the top power MOSFET. If a high Qg MOSFET is chosen such that the on-time is not sufficient to fully turn the MOSFET on, there will be significant losses in efficiency as a result of larger RDS(ON) resistance and possibly failure of the MOSFET due to significant heat dissipation. The minimum off-time is the smallest duration of time that the top power MOSFET ’s gate can be turned off and The effective minimum off-time of the switching regulator is defined as the shortest period of time that the SW node can stay low. This effective minimum off-time can vary from the LTC3833’s 90ns of minimum off-time. The main factor impacting the effective minimum off-time is the top and bottom power MOSFETs’ gate charging characteristics, including Qg and turn-on/off delays. These characteristics can either extend or shorten the SW node’s minimum off-time as compared to the LTC3833’s minimum off-time. Large size (high Qg) power MOSFETs generally tend to increase the effective minimum off-time due to longer gate charging and discharging times. On the other hand, imbalances in turn-on and turn-off delays could reduce the effective minimum off-time. The minimum off-time limit imposes a maximum duty cycle of: DMAX = 1 – f • tOFF(MIN) where tOFF(MIN) is the effective minimum off-time of the switching regulator. Reducing the operating frequency alleviates the maximum duty cycle constraint. If the maximum duty cycle is reached, due to a drooping input voltage for example, then the output will drop out of regulation. The minimum input voltage to avoid dropout is: VIN(MIN) = VOUT DMAX At the onset of dropout, there is a region of VIN about 500mV that generates two discrete off-times, one being the minimum off-time and the other being an off-time that is about 40ns to 60ns larger than the minimum off-time. This secondary off-time is due to the longer delay in tripping the internal current comparator. The two off-times average out to the required duty cycle to keep the output in regulation with the output ripple remaining the same. However, there is higher SW node jitter, especially apparent when synchronized to an external clock. Depending on the application, this may not be of critical importance. 3833f 23 LTC3833 APPLICATIONS INFORMATION Fault Conditions: Current Limiting and Overvoltage The maximum inductor current is inherently limited in a current mode controller by the maximum sense voltage. In the LTC3833, the maximum sense voltage is controlled by the voltage on the VRNG pin. With valley current mode control, the maximum sense voltage and the sense resistance determine the maximum allowed inductor valley current. The corresponding output current limit is: ILIMIT = VSENSE(MAX) RSENSE 1 + • ∆IL 2 The current limit value should be checked to ensure that ILIMIT(MIN) > IOUT(MAX). The current limit value should be greater than the inductor current required to produce maximum output power at the worst-case efficiency. Worst-case efficiency typically occurs at the highest VIN and highest ambient temperature. It is important to check for consistency between the assumed MOSFET junction temperatures and the resulting value of ILIMIT which heats the MOSFET switches. To further limit current in the event of a short circuit to ground, the LTC3833 includes foldback current limiting. If the output fails by more than 50%, then the maximum sense voltage is progressively lowered to about one-fourth of its full value. If the output exceeds 7.5% of the programmed value, then it is considered as an overvoltage (OV) condition. In such a case, the top MOSFET is immediately turned off and the bottom MOSFET is turned on indefinitely until the OV condition is removed. Current limiting is not active during an OV. If the output returns to a nominal level, then normal operation resumes. If the OV persists a long time, the current through the bottom MOSFET and inductor could exceed their maximum ratings. OPTI-LOOP Compensation OPTI-LOOP compensation, through the availability of the ITH pin, allows the transient response to be optimized for a wide range of loads and output capacitors. The ITH pin not only allows optimization of the control loop behavior but also provides a test point for the step-down regulator ’s DC-coupled and AC-filtered closed-loop response. The DC step, rise time and settling at this test point truly reflects the closed-loop response. Assuming a predominantly second order system, phase margin and/or damping factor can be estimated using the percentage of overshoot seen at this pin. The bandwidth can also be estimated by examining the rise time at this pin. The ITH series RITH-CITH1 filter sets the dominant pole-zero loop compensation. Additionally, a small capacitor placed from the ITH pin to SGND, CITH2, may be required to attenuate high frequency noise. The values can be modified to optimize transient response once the final PCB layout is done and the particular output capacitor type and value have been determined. The output capacitors need to be selected because their various types and values determine the loop feedback factor gain and phase. An output current pulse of 20% to 100% of full load current having a rise time of 1μs to 10μs will produce output voltage and ITH pin waveforms that will give a sense of the overall loop stability without breaking the feedback loop. The general goal of OPTI-LOOP compensation is to realize a fast but stable ITH response with minimal output droop due to the load step. For a detailed explanation of OPTI-LOOP compensation, refer to Application Note 76. Switching regulators take several cycles to respond to a step in load current. When a load step occurs, VOUT immediately shifts by an amount equal to ∆ILOAD • ESR, where ESR is the effective series resistance of COUT . ∆ILOAD also begins to charge or discharge COUT , generating a feedback error signal used by the regulator to return VOUT to its steady-state value. During this recovery time, VOUT can be monitored for overshoot or ringing that would indicate a stability problem. Connecting a resistive load in series with a power MOSFET, then placing the two directly across the output capacitor and driving the gate with an appropriate signal generator is a practical way to produce a realistic load-step condition. The initial output voltage step resulting from the step change in output current may not be within the bandwidth of the feedback loop, so this signal cannot be used to determine phase margin. This is why it is better to look at the ITH pin signal which is in the feedback loop and is the filtered and compensated feedback loop response. 3833f 24 LTC3833 APPLICATIONS INFORMATION The gain of the loop increases with RITH and the bandwidth of the loop increases with decreasing CITH1. If RITH is increased by the same factor that CITH1 is decreased, the zero frequency will be kept the same, thereby keeping the phase the same in the most critical frequency range of the feedback loop. In addition, a feedforward capacitor, CFF , can be added to improve the high frequency response, as shown in Figure 1. Capacitor CFF provides phase lead by creating a high frequency zero with RFB2 which improves the phase margin. The output voltage settling behavior is related to the stability of the closed-loop system and will demonstrate overall performance of the step-down regulator. In some applications, a more severe transient can be caused by switching in loads with large (>10μF) input capacitors. If the switch connecting the load has low resistance and is driven quickly, then the discharged input capacitors are effectively put in parallel with COUT , causing a rapid drop in VOUT . No regulator can deliver enough current to prevent this problem. The solution is to limit the turn-on speed of the load switch driver. A Hot Swap™ controller is designed specifically for this purpose and usually incorporates current limiting, short-circuit protection and soft starting. Efficiency Considerations The percent efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Percent efficiency can be expressed as: %Efficiency = 100% – (L1 + L2 + L3 + ...) where L1, L2, etc. are the individual losses as a percentage of input power. Although all dissipative elements in the circuit produce losses, four main sources account for most of the losses: 1. I2R losses. These arise from the resistances of the MOSFETs, inductor and PC board traces and cause the efficiency to drop at high output currents. In continuous mode the average output current flows though the inductor L, but is chopped between the top and bottom MOSFETs. If the two MOSFETs have approximately the same RDS(ON), then the resistance of one MOSFET can simply by summed with the resistances of L and the board traces to obtain the DC I2R loss. For example, if RDS(ON) = 0.01Ω and RL = 0.005Ω, the loss will range from 15mW to 1.5W as the output current varies from 1A to 10A. 2. Transition loss. This loss arises from the brief amount of time the top MOSFET spends in the saturated region during switch node transitions. It depends upon the input voltage, load current, driver strength and MOSFET capacitance, among other factors. The loss is significant at input voltages above 20V. 3. INTVCC current. This is the sum of the MOSFET driver and control currents. The MOSFET driver current results from switching the gate capacitance of the power MOSFETs. Each time a MOSFET gate is switched from low to high to low again, a packet of charge, dQ, moves from INTVCC to ground. The resulting dQ/dt is a current out of INTVCC that is typically much larger than the controller IQ current. In continuous mode, IGATECHG = f • (Qg(TOP) + Qg(BOT)), where Qg(TOP) and Qg(BOT) are the gate charges of the topside and bottom side MOSFETs, respectively. Supplying INTVCC power through EXTVCC could save several points of efficiency, especially for high VIN applications. Connecting EXTVCC to an output-derived source will scale the VIN current required for the driver and controller circuits by a factor of Duty Cycle/Efficiency. For example, in a 20V to 5V application, 10mA of INTVCC current results in approximately 2.5mA of VIN current. This reduces the mid-current loss from 10% or more (if the driver was powered directly from VIN) to only a few percent. 4. CIN loss. The input capacitor has the difficult job of filtering the large RMS input current to the regulator. It must have a very low ESR to minimize the AC I2R loss and sufficient capacitance to prevent the RMS current from causing additional upstream losses in cabling, fuses or batteries. Other losses, which include the COUT ESR loss, bottom MOSFET reverse-recovery loss and inductor core loss generally account for less than 2% additional loss. 3833f 25 LTC3833 APPLICATIONS INFORMATION When making adjustments to improve efficiency, the input current is the best indicator of changes in efficiency. If you make a change and the input current decreases, then the efficiency has increased. If there is no change in input current there is no change in efficiency. The frequency is programmed by: The minimum on-time occurs for maximum VIN and should be greater than 20ns which is the best that the LTC3833 can achieve. The minimum on-time for this application is: tON(IDEAL) Efficiency Design Example tON(MIN) = VOUT VIN(MAX) • f = 1.2V ≈ 143ns 24V • 350kHz Set the inductor value to give 40% ripple current at maximum VIN: Consider a step-down converter with VIN = 6V to 24V, VOUT = 1.2V, IOUT(MAX) = 15A, and f = 350kHz (see Figure 9). The regulated output voltage is determined by: 41550 41550 – 2.2 = – 2.2 ≈ 116.5k f [kHz ] 350 Select the nearest standard value of 115k. Power losses in the switching regulator will reflect as a longer than ideal on-time. This efficiency accounted ontime in continuous mode can be calculated as: tON(REAL) ≈ R T [kΩ ] =  R  VOUT = 0.6V •  1+ FB2   R  L= 1.2V  1.2V  •  1– ≈ 0.54µH 350kHz • 40% • 15A  24V  Select 0.56μH which is the nearest standard value. FB1 Using a 20k resistor from VOSNS+ to VOSNS–, the top feedback resistor is also 20k. RPGD 100k RDIV1 52.3k VIN VRNG RDIV2 10k 350kHz CSS 0.1µF CITH1 470pF RITH 47.5k RUN SENSE– SENSE+ SW VIN CIN1 6V TO 24V 82µF 25V + 100 90 L1 0.56µH CB 0.1µF RFB2 20k DB INTVCC ITH BG VOUT 1.2V 15A INTVCC CVCC 4.7µF MB RFB1 20k COUT2 100µF ×2 + COUT1 330µF 2.5V ×2 60 FORCED CONTINUOUS MODE 50 40 20 0.1 VOSNS+ VOSNS– CIN1: SANYO 25SVPD82M COUT1: SANYO 2R5TPE330M9 DB: CENTRAL CMDSH-3 70 30 PGND RT SGND PULSE-SKIPPING MODE 80 MT BOOST TRACK/SS Efficiency CDCR RDCR 0.1µF 3.09k TG MODE/PLLIN EXTVCC CITH2 47pF RT 115k CIN2 10µF VOUT PGOOD LTC3833 EFFICIENCY (%) INTVCC 3833 F09 VIN = 12V VOUT = 1.2V 1 10 LOAD CURRENT (A) 100 3833 F09b L1: VISHAY IHLP4040DZ-056µH MB: RENESAS RJK0330DPB MT: RENESAS RJK0305DPB Figure 9. 1.2V, 15A, 350kHz Step-Down Converter 3833f 26 LTC3833 APPLICATIONS INFORMATION The resulting maximum ripple current is: ∆IL = 1.2V  1.2V  •  1– ≈ 5.8A 350kHz • 0.56µH  24V  Often in high power applications, DCR current sensing is preferred over RSENSE in order to maximize efficiency. In order to determine the DCR filter values, first the inductor manufacturer has to be chosen. For this design, the Vishay IHLP-4040DZ-01 model is chosen with a value of 0.56μH and DCRMAX =1.8mΩ. This implies that: top MOSFET (main switch), and RJK0330DBP (RDS(ON) = 3.9mΩ max, VGS = 4.5V, θJA = 40°C/W, TJ(MAX) = 150°C) is chosen for the bottom MOSFET (synchronous switch). The power dissipation and the resulting junction temperature for each MOSFET can be calculated for VIN = 24V and TA = 75°C:  1.2V  2 2 PTOP =  • (15A ) (13mΩ ) (1+ 0.4) + ( 24V )  24V  1.2Ω  2.5Ω  15A  • + 350kHz (150pF )  5.3V  2  – 3V 3V  ≈ 0.54W VSENSE(MAX) = DRCMAX at 25°C • [1 + 0.4% (TL(MAX) – 25°C)] • [IOUT(MAX) – ∆IL/2] = 1.8mΩ • [1 + 0.4% (100°C – 25°C)] • [15A – 5.8A/2] ≈ 28.3mV The maximum sense voltage is within the range that LTC3833 can handle without any additional scaling. Therefore, the DCR filter consists of a simple RC filter across the inductor. If the C is chosen to be 0.1µF, then the R can be calculated as: RDCR = L DCRMAX • CDCR = 0.56µH ≈ 3.11k 1.8mΩ • 0.1µF The closest standard value is 3.09k. The resulting value of VRNG with a 50% design margin factor is: VRNG = VSENSE(MAX)/0.05 • MF = 28.3mV/0.05 • 1.5 ≈ 850mV To generate the VRNG voltage, connect a resistive divider from INTVCC to SGND with RDIV1 = 52.3k and RDIV2 = 10k. For the external N-channel MOSFETs, Renesas RJK0305DBP (RDS(ON) = 13mΩ max, CMILLER = 150pF, VGS = 4.5V, θJA = 40°C/W, TJ(MAX) = 150°C) is chosen for the  40°C  TJ(TOP) = 75°C + ( 0.54W )  ≈ 97°C  W   24V – 1.2V  PBOT =  (15A )2 (3.9mΩ)(1+ 0.4) ≈ 1.2W  24V   40°C  TJ(BOT) = 75°C + (1.2W )  = 123°C  W  These numbers show that careful attention should be paid to proper heat sinking when operating at higher ambient temperatures. Select CIN to give an RMS current rating greater than 7A at 75°C. The output capacitor COUT is chosen for a low ESR of 4.5mΩ to minimize output voltage changes due to inductor ripple current and load steps. The output voltage ripple is given as: ∆VOUT(RIPPLE) = ∆IL(MAX) • ESR = (5.8A)(4.5mΩ) ≈ 26mV However, a 0A to 10A load step will cause an output change of up to: ∆VOUT(STEP) = ∆ILOAD • ESR = (10A)(4.5mΩ) = 45mV Optional 100μF ceramic output capacitors are included to minimize the effect of ESR and ESL in the output ripple and to improve load step response. 3833f 27 LTC3833 APPLICATIONS INFORMATION PC Board Layout Checklist When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC3833. • Multilayer boards with dedicated ground layers are preferable for reduced noise and for heat sinking purposes. Use wide rails and/or entire planes for VIN, VOUT and PGND nodes for good filtering and minimal copper loss. If a ground layer is used, then it should be immediately below (and/or above) the routing layer for the power train components which consist of CIN, power MOSFETs, inductor, sense resistor (if used) and COUT . Flood unused areas of all layers with copper for better heat sinking. • Keep signal and power grounds separate except at the point where they are shorted together. Short signal and power ground together only at a single point with a narrow PCB trace (or single via in a multilayer board). All power train components should be referenced to power ground and all small-signal components (e.g., CITH1, RT , CSS etc.) should be referenced to signal ground. • Place CIN, power MOSFETs, inductor, sense resistor (if used), and primary COUT capacitors close together in one compact area. The SW node should be compact but be large enough to handle the inductor currents without large copper losses. Connect the drain of the topside MOSFET as close as possible to the (+) plate of CIN capacitor(s) that provides the bulk of the AC current (these are normally the ceramic capacitors), and connect the source of the bottom side MOSFET as close as possible to the (–) terminal of the same CIN capacitor(s). The high dI/dt loop formed by CIN, the top MOSFET, and the bottom MOSFET should have short leads and PCB trace lengths to minimize high frequency EMI and voltage stress from inductive ringing. The (–) terminal of the primary COUT capacitor(s) which filter the bulk of the inductor ripple current (these are normally the ceramic capacitors) should also be connected close to the (–) terminal of CIN. • Place BOOST, TG, SW, BG and PGND pins facing the power train components. Keep high dV/dt signals on BOOST, TG, SW and BG away from sensitive small-signal traces and components. • For RSENSE current sensing, place the sense resistor close to the inductor on the output side. Use a Kelvin (4-wire) connection across the sense resistor and route the traces together as a differential pair. RC filter the differential sense signal close to SENSE+/SENSE– pins, placing the filter capacitor as close as possible to the pins. For DCR sensing, Kelvin connect across the inductor and place the DCR sensing resistor closer to the SW node and further away from the SENSE+/ SENSE– pins. Place the DCR capacitor close to the SENSE+/SENSE– pins. • Place the resistive feedback divider RFB1/2 as close as possible to VOSNS+/VOSNS– pins and route the remote output and ground traces together as a differential pair and terminate as close to the regulation point as possible (preferably Kelvin connect across the capacitor at the remote output point). • Place the ceramic CVCC capacitor as close as possible to INTVCC and PGND pins. Likewise, the CB capacitor should be as close as possible to BOOST and SW pins. These capacitors provide the gate charging currents for the power MOSFETs. • Place small-signal components as close to their respective pins as possible. This minimizes the possibility of PCB noise coupling into these pins. Give priority to VOSNS+/VOSNS–, SENSE+/SENSE–, ITH, RT and VRNG pins. Use sufficient isolation when routing a clock signal into MODE/PLLIN pin so that the clock does not couple into sensitive small-signal pins. • Filter the VIN input to the LTC3833 with a simple RC filter close to the pin. The RC filter should be referenced to signal ground. 3833f 28 LTC3833 APPLICATIONS INFORMATION RVIN 2.2Ω INTVCC VIN RPGD 100k LTC3833 VOUT PGOOD TG MODE/PLLIN SW EXTVCC L1 0.47µH DB RITH 84.5k RT 137k TRACK/SS RSENSE 1.5mΩ RFB2 15k CB 0.1µF INTVCC INTVCC RFB1 10k CVCC 4.7µF MB BG ITH COUT2 100µF ×2 + VOUT 1.5V 20A COUT1 330µF 2.5V ×2 PGND VOSNS+ VOSNS– RT SGND 3833 F10a CIN1: SANYO 16SVP180M COUT1: SANYO 2R5TPE330M9 DB: CENTRAL CMDSH-3 L1: PULSE PA0515.471NLT MB: RENESAS RJK0330DPB MT: RENESAS RJK0305DPB Efficiency 100 PULSE-SKIPPING MODE 90 EFFICIENCY (%) CITH1 220pF MT BOOST CSS 0.1µF CITH2 47pF VIN 4.5V TO 14V CIN1 180µF 16V CF RF1 1000pF 10Ω SENSE+ VRNG + RF2 10Ω SENSE– RUN CIN2 22µF ×2 CVIN 0.1µF FORCED CONTINUOUS MODE 80 70 60 50 40 VIN = 12V VOUT = 1.5V 0.1 1 10 LOAD CURRENT (A) 100 3833 F10b Figure 10. 1.5V, 20A, 300kHz High Current Step-Down Converter 3833f 29 LTC3833 TYPICAL APPLICATIONS 5V, 8A, 200kHz High Efficient Step-Down Converter RVIN 2.2Ω VIN INTVCC RPGD 100k EXTVCC PGOOD VOUT SENSE– SENSE+ VRNG LTC3833 TRACK/SS CITH1 220pF RITH 86.6k RT 205k ITH RT VIN 7V TO 38V L1 6µH SW INTVCC CIN1 100µF 50V MT TG BOOST + CDCR RDCR 0.22µF 5.9k RUN CSS 0.1µF CIN2 10µF ×3 CVIN 0.1µF RB 10Ω CVCC 4.7µF RFB2 147k DB CB INTVCC 0.1µF RFB1 20k MB BG VOUT 5V 8A COUT2 100µF ×2 + COUT1 330µF 6.3V ×2 MODE/PLLIN PGND SGND VOSNS+ VOSNS– 3833 TA02 CIN1: NICHICON UCJ1H101MCL1GS COUT1: SANYO 6TPE330MIL DB: DIODES INC. SDM10K45 L1: COOPER HC2LP-6R0 MB: INFINEON BSC035N04LS MT: INFINEON BSC035N04LS Efficiency 100 PULSE-SKIPPING MODE EFFICIENCY (%) 95 FORCED CONTINUOUS MODE 90 85 80 75 70 VIN = 12V VOUT = 5V 0.1 1 LOAD CURRENT (A) 10 3833 TA02b 3833f 30 LTC3833 TYPICAL APPLICATIONS 0.6V, 10A, 200kHz Low Output Step-Down Converter INTVCC RPGD 100k RVIN 2.2Ω VIN CVIN 0.1µF LTC3833 PGOOD VOUT CITH1 220pF RUN RITH 51k MT TG SW BOOST RSENSE 3mΩ + DB ITH RT INTVCC CVCC 4.7µF VOUT 0.6V COUT1 10A 330µF 2.5V ×2 COUT2 100µF ×2 MB BG EXTVCC SGND L1 1µH CB 0.1µF INTVCC RT 205k VIN CIN1 4.5V TO 14V 100µF 50V RF1 CF 1000pF 10Ω SENSE+ TRACK/SS + RF2 10Ω MODE/PLLIN VRNG SENSE– CSS 2200pF CIN2 10µF ×3 PGND VOSNS+ VOSNS– 3833 TA03 L1: WURTH 7443320100 MT: INFINEON BSC093N04LS MB: INFINEON BSC035N04LS CIN1: NICHICON UCJ1H101MCL1GS COUT: SANYO 2R5TPE330M9 DB: DIODES INC. SDM10K45 Efficiency 90 80 PULSE-SKIPPING MODE EFFICIENCY (%) 70 60 FORCED CONTINUOUS MODE 50 40 30 20 0.1 VIN = 12V VOUT = 0.6V 1 LOAD CURRENT (A) 10 3833 TA03b 3833f 31 LTC3833 TYPICAL APPLICATIONS Area Compact 2.5V, 5A, 1.2MHz Step-Down Converter RVIN 2.2Ω INTVCC VIN VRNG RPGD 100k LTC3833 VOUT SENSE– SENSE+ PGOOD RUN EXTVCC RT 33.2k SW TRACK/SS CIN2 10µF CIN1 47µF 35V L1 1µH CB 0.1µF RFB2 31.6k DB INTVCC CVCC 4.7µF RT COUT1 100µF RFB1 10k MB BG VIN 6V TO 28V VOUT 2.5V 5A BOOST INTVCC ITH + CDCR 0.1µF RDCR MT 1.1k TG MODE/PLLIN CSS 0.01µF CITH1 220pF RITH 20k CVIN 0.1µF PGND VOSNS+ VOSNS– SGND 3833 TA04 CIN1: KEMET T521X476M035ATE070 DB: DIODES INC. SDM10K45 L1: VISHAY IHLP2525CZ-1µH MT, MB: VISHAY/SILICONIX Si4816BDY Efficiency 90 80 EFFICIENCY (%) 70 PULSESKIPPING MODE 60 50 FORCED CONTINUOUS MODE 40 30 20 0.1 VIN = 12V VOUT = 2.5V 1 LOAD CURRENT (A) 10 3833 TA04b 3833f 32 LTC3833 TYPICAL APPLICATIONS 3.3V, 15A, 200kHz High Power Step-Down Converter VIN CVIN RVIN 0.1µF 2.2Ω INTVCC VOUT RUN PGOOD VRNG SENSE– CSS 0.1µF TRACK/SS CIN1 4.5V TO 24V 100µF 35V RDCR2 RDCR1 17.4k 3.92k MT TG L1 2µH SW VOUT 3.3V 15A BOOST DB INTVCC INTVCC RFB2 90.9k CB 0.1µF CVCC 4.7µF ITH RFB1 20k MB BG CITH2 47pF RT 205k CDCR 0.22µF SENSE+ MODE/PLLIN EXTVCC CITH1 680pF RITH 18.2k VIN + LTC3833 COUT2 100µF ×2 + COUT1 220µF 4V ×2 PGND RT SGND VOSNS+ VOSNS– 3833 TA06 L1: WURTH 7443551200 MB: RENESAS RJK0330DPB MT: RENESAS RJK0305DPB CIN1: SUNCON 35HVP100M COUT1: SANYO 4TPE220MFC2 DB: CENTRAL CMDSH-3 Efficiency 100 90 EFFICIENCY (%) RPGD 100k CIN2 10µF ×5 PULSE-SKIPPING MODE 80 FORCED CONTINUOUS MODE 70 60 50 40 VIN = 12V VOUT = 3.3V 0.1 1 10 LOAD CURRENT (A) 100 3833 F10b 3833f 33 LTC3833 PACKAGE DESCRIPTION UDC Package 20-Lead Plastic QFN (3mm × 4mm) (Reference LTC DWG # 05-08-1742 Rev Ø) 0.70 ±0.05 3.50 ± 0.05 2.10 ± 0.05 1.50 REF 2.65 ± 0.05 1.65 ± 0.05 PACKAGE OUTLINE 0.25 ±0.05 0.50 BSC 2.50 REF 3.10 ± 0.05 4.50 ± 0.05 RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED 3.00 ± 0.10 0.75 ± 0.05 1.50 REF 19 R = 0.05 TYP PIN 1 NOTCH R = 0.20 OR 0.25 × 45° CHAMFER 20 0.40 ± 0.10 1 PIN 1 TOP MARK (NOTE 6) 4.00 ± 0.10 2 2.65 ± 0.10 2.50 REF 1.65 ± 0.10 (UDC20) QFN 1106 REV Ø 0.200 REF 0.00 – 0.05 R = 0.115 TYP 0.25 ± 0.05 0.50 BSC BOTTOM VIEW—EXPOSED PAD NOTE: 1. DRAWING IS NOT A JEDEC PACKAGE OUTLINE 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE 3833f 34 LTC3833 PACKAGE DESCRIPTION FE Package 20-Lead Plastic TSSOP (4.4mm) (Reference LTC DWG # 05-08-1663 Rev H) Exposed Pad Variation CB 6.40 – 6.60* (.252 – .260) 3.86 (.152) 3.86 (.152) 20 1918 17 16 15 14 13 12 11 6.60 ±0.10 2.74 (.108) 4.50 ±0.10 6.40 2.74 (.252) (.108) BSC SEE NOTE 4 0.45 ±0.05 1.05 ±0.10 0.65 BSC 1 2 3 4 5 6 7 8 9 10 RECOMMENDED SOLDER PAD LAYOUT 4.30 – 4.50* (.169 – .177) 0.09 – 0.20 (.0035 – .0079) 0.25 REF 0.50 – 0.75 (.020 – .030) NOTE: 1. CONTROLLING DIMENSION: MILLIMETERS MILLIMETERS 2. DIMENSIONS ARE IN (INCHES) 3. DRAWING NOT TO SCALE 1.20 (.047) MAX 0° – 8° 0.65 (.0256) BSC 0.195 – 0.30 (.0077 – .0118) TYP 0.05 – 0.15 (.002 – .006) FE20 (CB) TSSOP REV H 0910 4. RECOMMENDED MINIMUM PCB METAL SIZE FOR EXPOSED PAD ATTACHMENT *DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.150mm (.006") PER SIDE 3833f Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 35 LTC3833 TYPICAL APPLICATION High Frequency 5.5V, 4A, 2MHz Step-Down Converter RDIV1 100k VIN PGOOD MODE/PLLIN LTC3833 VRNG RDIV2 26.1k SENSE– CSS 0.1µF TRACK/SS SENSE+ CIN2 4.7µF ×2 CVIN 0.1µF EXTVCC VOUT RUN CITH1 220pF RITH 20k RVIN 2.2Ω SW BOOST RT INTVCC SGND BG Efficiency 90 CF RF1 1000pF 10Ω 70 MT L1 1.2µH CB 0.1µF INTVCC CVCC 4.7µF RSENSE 10mΩ CFF 22pF MB PGND RFB2 165k RFB1 20k VOSNS+ VOSNS– CIN1: KEMET T521X476M035ATE070 DB: DIODES, INC. SDM10K45 VIN 7V TO 14V 80 DB RT 18.2k CIN1 47µF 35V RF2 10Ω TG ITH + VOUT 5.5V 4A COUT1 22µF ×2 EFFICIENCY (%) RPGD 100k INTVCC PULSESKIPPING MODE 60 50 FORCED CONTINUOUS MODE 40 30 20 0.1 VIN = 12V VOUT = 5.5V 10 1 LOAD CURRENT (A) 3833 TA05b 3833 TA05 L1: WURTH 744313120 MT, MB: INFINEON BSC093N04LS RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LTC3878/LTC3879 No RSENSE™ Constant On-Time Synchronous Step-Down DC/DC Controller Very Fast Transient Response, tON(MIN) = 43ns, 4V ≤ VIN ≤ 38V, 0.6V ≤ VOUT ≤ 0.9VIN, SSOP-16, MSOP-16E, 3mm × 3mm QFN-16 LTC3775 High Frequency Synchronous Voltage Mode Step-Down DC/DC Controller Very Fast Transient Response, tON(MIN) = 30ns, 4V ≤ VIN ≤ 38V, 0.6V ≤ VOUT ≤ 0.8VIN, MSOP-16E, 3mm × 3mm QFN-16 LTC3854 Small Footprint Synchronous Step-Down DC/DC Controller Fixed 400kHz Operating Frequency 4.5V ≤ VIN ≤ 38V, 0.8V ≤ VOUT ≤ 5.25V, 2mm × 3mm QFN-12 LTC3851A/LTC3851A-1 No RSENSE Wide VIN Range Synchronous Step-Down DC/DC Controller Phase-Lockable Fixed Frequency 250kHz to 750kHz, 4V ≤ VIN ≤ 38V, 0.8V ≤ VOUT ≤ 5.25V, MSOP-16E, 3mm × 3mm QFN-16, SSOP-16 LTC3891 60V, Low IQ Synchronous Step-Down DC/DC Controller PLL Capable Fixed Frequency 50kHz to 900kHz, 4V ≤ VIN ≤ 60V, 0.8V ≤ VOUT ≤ 24V, IQ = 50µA LTC3856 2-Phase, Single Output Synchronous Step-Down DC/DC Controller with Diff Amp and DCR Temperature Compensation Phase-Lockable Fixed 250kHz to 770kHz Frequency, 4.5V ≤ VIN ≤ 38V, 0.6V ≤ VOUT ≤ 5.25V LTC3829 3-Phase, Single Output Synchronous Step-Down DC/DC Controller with Diff Amp and DCR Temperature Compensation Phase-Lockable Fixed 250kHz to 770kHz Frequency, 4.5V ≤ VIN ≤ 38V, 0.6V ≤ VOUT ≤ 5.25V LTC3855 2-Phase, Dual Output Synchronous Step-Down DC/DC Controller with Differential Remote Sense Phase-Lockable Fixed Frequency 250kHz to 770kHz, 4.5V ≤ VIN ≤ 38V, 0.6V ≤ VOUT ≤ 12.5V LTC3850/LTC3850-1 2-Phase, Dual Output Synchronous Step-Down DC/DC Controllers, RSENSE or DCR Current Sensing Phase-Lockable Fixed Frequency 250kHz to 780kHz, 4V ≤ VIN ≤ 30V, 0.8V ≤ VOUT ≤ 5.25V, 4mm × 4mm QFN-28, 4mm × 5mm QFN-28, SSOP-28 LTC3853 Triple Output, Multiphase Synchronous Step-Down DC/DC Controller, RSENSE or DCR Current Sensing Phase-Lockable Fixed Frequency 250kHz to 750kHz, 4V ≤ VIN ≤ 24V, VOUT Up to 13.5V 3833f 36 Linear Technology Corporation LT 1010 • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com  LINEAR TECHNOLOGY CORPORATION 2010