TPA0162
www.ti.com
SLOS249E – JUNE 1999 – REVISED SEPTEMBER 2004
2.8-W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
FEATURES
•
•
•
•
•
•
•
•
•
•
•
Compatible With PC 99 Desktop Line-Out Into
10-kΩ Load
Compatible With PC 99 Portable Into 8-Ω Load
Internal Gain Control, Which Eliminates
External Gain-Setting Resistors
Digital Volume Control From 20 dB to -40 dB
2.8-W/Ch Output Power Into a 3-Ω Load
PC-Beep Input
Depop Circuitry
Stereo Input MUX
Fully Differential Input
Low Supply Current and Shutdown Current
Surface-Mount Power Packaging 24-Pin
TSSOP PowerPAD™
PWP PACKAGE
(TOP VIEW)
GND
UP
DOWN
LOUT+
LLINEIN
LHPIN
PVDD
RIN
LOUT–
LIN
BYPASS
GND
1
2
3
4
5
6
7
8
9
10
11
12
24
23
22
21
20
19
18
17
16
15
14
13
GND
RLINEIN
SHUTDOWN
ROUT+
RHPIN
VDD
PVDD
CLK
ROUT–
SE/BTL
PC-BEEP
GND
DESCRIPTION
The TPA0162 is a stereo audio power amplifier in a 24-pin TSSOP thermally enhanced package capable of
delivering 2.8 W of continuous RMS power per channel into 3-Ω loads.
This device minimizes the number of external components needed, which simplifies the design and frees up
board space for other features. When driving 1 W into 8-Ω speakers, the TPA0162 has less than 0.22% THD+N
across its specified frequency range. The integrated depop circuitry virtually eliminates transients that cause
noise in the speakers.
The overall gain of the amplifier is controlled digitally by the UP and DOWN terminals. At power up, the gain is
set at the lowest level, -85 dB. It can then be adjusted to any of 31 discrete steps by pulling the desired
volume-control pin to logic low. The gain is adjusted in the initial stage of the amplifier as opposed to the power
output stage. As a result, the THD changes little over all volume levels.
An internal input MUX allows two sets of stereo inputs to the amplifier. In notebook applications, where internal
speakers are driven as BTL and the line outputs (often headphone drive) are required to be SE, the TPA0162
automatically switches into SE mode when the SE/BTL input is activated. This effectively reduces the gain by 6
dB.
The TPA0162 consumes only 20 mA of supply current during normal operation. A shutdown mode is included
that reduces the supply current to 150 µA.
The PowerPAD package (PWP) delivers a level of thermal performance that was previously achievable only in
TO-220-type packages. Thermal impedances of approximately 35°C/W are truly realized in multilayer PCB
applications. This allows the TPA0162 to operate at full power into 8-Ω loads at ambient temperatures of 85°C.
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas
Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PowerPAD is a trademark of Texas Instruments.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 1999–2004, Texas Instruments Incorporated
TPA0162
www.ti.com
SLOS249E – JUNE 1999 – REVISED SEPTEMBER 2004
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
ORDERING INFORMATION
PACKAGED DEVICE
TA
TSSOP (1) (PWP)
40°C to 85°C
(1)
TPA0162PWP
The PWP package is available taped and reeled. To order a taped
and reeled part, add the suffix R to the part number (e.g.,
TPA0162PWPR).
FUNCTIONAL BLOCK DIAGRAM
RHPIN
R
MUX
RLINEIN
VDD
32-Step
Volume
Control
VDD
−
40 k
40 k
ROUT+
+
UP
DOWN
32-Step
Volume
Control
RIN
−
PC-BEEP
SE/BTL
ROUT−
PC
Beep
+
MUX
Control
Depop
Circuitry
LHPIN
LLINEIN
L
MUX
32-Step
Volume
Control
Power
Management
PVDD
VDD
BYPASS
SHUTDOWN
GND
−
LOUT+
+
LIN
32-Step
Volume
Control
−
LOUT−
+
2
TPA0162
www.ti.com
SLOS249E – JUNE 1999 – REVISED SEPTEMBER 2004
Terminal Functions
TERMINAL
NAME
NO.
I/O
DESCRIPTION
BYPASS
11
Tap to voltage divider for internal mid-supply bias generator
CLK
17
I
If a 47-nF capacitor is attached, the TPA0162 generates an internal clock. An external clock can
override the internal clock input to this terminal.
DOWN
3
I
A momentary pulse on this terminal decreases the volume level by 2 dB. Holding the terminal low for a
period of time will step the amplifier through the volume levels at a rate determined by the capacitor on
the CLK terminal.
GND
1, 12
13, 24
I
Ground connection for circuitry. Connected to thermal pad
LHPIN
6
I
Left-channel headphone input, selected when SE/BTL is held high
LIN
10
I
Common left input for fully differential input. AC ground for single-ended inputs
LLINEIN
5
I
Left-channel line negative input, selected when SE/BTL is held low
LOUT+
4
O
Left-channel positive output in BTL mode and positive in SE mode
LOUT-
9
O
Left-channel negative output in BTL mode and high impedance in SE mode
PC-BEEP
14
I
The input for PC-Beep mode. PC-BEEP is enabled when a > 1.5-V (peak-to-peak) square wave is
input to PC-BEEP or PCB ENABLE is high.
PVDD
7, 18
I
Power supply for output stage
RHPIN
20
I
Right-channel headphone input, selected when SE/BTL is held high
RIN
8
I
Common right input for fully differential input. AC ground for single-ended inputs
RLINEIN
23
I
Right-channel line input, selected when SE/BTL is held low.
ROUT+
21
O
Right-channel positive output in BTL mode and positive in SE mode
ROUT-
16
O
Right-channel negative output in BTL mode and high impedance in SE mode
SE/BTL
15
I
Input- and output-MUX control. When this terminal is held high, the LHPIN or RHPIN and SE output is
selected. When this terminal is held low, the LLINEIN or RLINEIN and BTL output are selected.
SHUTDOWN
22
I
When held low, this terminal places the entire device, except PC-BEEP detect circuitry, in shutdown
mode.
UP
2
I
A momentary pulse on this terminal increases the volume level by 2 dB. Holding the terminal low for a
period of time steps the amplifier through the volume levels at a rate determined by the capacitor on
the CLK terminal.
VDD
19
I
Analog VDD input supply. This terminal must be isolated from PVDD to achieve highest performance.
Thermal Pad
Connect to ground. Must be soldered down in all applications to properly secure device on PC board.
ABSOLUTE MAXIMUM RATINGS
over operating free-air temperature range (unless otherwise noted) (1)(1)
UNIT
Supply voltage, VDD
Input voltage, VI
Continuous total power dissipation
6V
–0.3 V to VDD 0.3 V
Internally limited (see Dissipation Rating Table)
Operating free-air temperature range, TA
–40°C to 85°C
Operating junction temperature range, TJ
–40°C to 150°C
Storage temperature range, Tstg
–65°C to 85°C
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds
(1)
260°C
Stresses beyond those listed under "absolute maximum ratings” may cause permanent damage to the device. These are stress ratings
only, and functional operation of the device at these or any other conditions beyond those indicated under "recommended operating
conditions” is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
3
TPA0162
www.ti.com
SLOS249E – JUNE 1999 – REVISED SEPTEMBER 2004
DISSIPATION RATING TABLE
(1)
PACKAGE
TA≤ 25°C
DERATING FACTOR
TA = 70°C
TA = 85°C
PWP
2.7 W (1)
21.8 mW/°C
1.7 W
1.4 W
See the Texas Instruments document, PowerPAD Thermally Enhanced Package Application Report
(literature number SLMA002), for more information on the PowerPAD™ package. The thermal data
was measured on a PCB layout based on the information in the section entitled Texas Instruments
Recommended Board for PowerPAD on page 33 of the before mentioned document.
RECOMMENDED OPERATING CONDITIONS
Supply voltage, VDD
MAX
4.5
5.5
SHUTDOWN
2
UP, DOWN
4
V
V
0.6 × VDD
SE/BTL
Low-level input voltage, VIL
UNIT
0.8 × VDD
SE/BTL
High-level input voltage, VIH
MIN
SHUTDOWN
0.8
UP, DOWN
0.5
Operating free-air temperature, TA
–40
V
°C
85
ELECTRICAL CHARACTERISTICS
at specified free-air temperature, VDD = 5 V, TA = 25°C (unless otherwise noted)
PARAMETER
TEST CONDITIONS
|VOO|
Output offset voltage (measured differentially)
VI = 0 V, AV = 2 V/V
PSRR
Power supply rejection ratio
VDD = 4.9 V to 5.1 V
|IIH|
High-level input current - SHUTDOWN, SE/BTL, UP,
DOWN
VDD = 5.5 V, VI = VDD
|IIL|
IDD
IDD(SD)
Low-level input current - SHUTDOWN, SE/BTL
Low-level input current - UP, DOWN
MIN
35
Supply current, shutdown mode
mV
67
VDD = 5.5 V, VI = 0 V
BTL mode - SHUTDOWN = 2 V,
SE/BTL = 0.6 × VDD
Supply current
TYP MAX UNIT
dB
900
nA
900
nA
125
µA
25
mA
SE mode - SHUTDOWN = 2 V,
SE/BTL = 0.8 × VDD
12.5
SHUTDOWN = 2 V,
SE/BTL = 0 v,
150
300
µA
OPERATING CHARACTERISTICS
VDD = 5 V, TA = 25°C, RL = 4Ω , Gain = 20 dB, BTL mode (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
UNIT
W
THD = 1%
2.2
W
Output power
RL = 3 Ω, f = 1 kHz
THD+N
Total harmonic distortion plus noise
PO = 1 W, f = 20 Hz to 15 kHz
BOM
Maximum output power bandwidth
THD = 5%
4
MAX
2.8
PO
Vn
TYP
THD = 10%
0.22%
>15
Supply ripple rejection ratio
f = 1 kHz,
C(BYP) = 0.47 µF
BTL mode
65
SE mode
60
Noise output voltage
C(BYP) = 0.47 µF,
f = 20 Hz to 20 kHz
BTL mode
17
SE mode
44
kHz
dB
µVRMS
TPA0162
www.ti.com
SLOS249E – JUNE 1999 – REVISED SEPTEMBER 2004
TYPICAL CHARACTERISTICS
TABLE OF GRAPHS
FIGURE
THD+N
Total harmonic distortion plus noise
vs Output power
1, 4, 6, 8, 10
vs Voltage gain
2
vs Frequency
3, 5, 7, 9, 11
vs Output voltage
Vn
SNR
12
Output noise voltage
vs Bandwidth
13
Supply ripple rejection ratio
vs Frequency
14, 15
Crosstalk
vs Frequency
16, 17, 18
Shutdown attenuation
vs Frequency
19
Signal-to-noise ratio
vs Bandwidth
20
Closed loop response
PO
21, 22
Output power
PD
Power dissipation
Zi
Input impedance
vs Load resistance
23, 24
vs Output power
25, 26
vs Ambient temperature
27
vs Gain
28
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
VOLTAGE GAIN
1%
THD+N -Total Harmonic Distortion + Noise
THD+N -Total Harmonic Distortion + Noise
10%
RL = 4 Ω
1%
RL = 8 Ω
RL = 3 Ω
0.1%
AV = 20 to 4 dB
f = 1 kHz
BTL
0.01%
0.5 0.75
1
1.25 1.5 1.75
2
2.25 2.5 2.75
3
PO = 1 W for AV ≥ 6 dB
VO = 1 VRMS for AV ≤ 4 dB
RL = 8 Ω
BTL
0.1%
0.01%
-40
-30
-20
-10
0
PO - Output Power - W
AV - Voltage Gain - dB
Figure 1.
Figure 2.
10
20
5
TPA0162
www.ti.com
SLOS249E – JUNE 1999 – REVISED SEPTEMBER 2004
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
10%
RL = 3 Ω
AV = 20 to 0 dB
BTL
1%
PO = 0.5 W
PO = 1 W
0.1%
PO = 1.75 W
0.01%
20
THD+N -Total Harmonic Distortion + Noise
THD+N -Total Harmonic Distortion + Noise
10%
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
f = 1 kHz
0.1%
f = 20 Hz
0.01%
0.01
f - Frequency - Hz
Figure 3.
Figure 4.
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
1k
10k 20k
10
10%
RL = 4 Ω
AV = 20 to 4 dB
BTL
1%
PO = 0.25 W
0.1%
PO = 1.5 W
PO = 1 W
0.01%
20
100
1k
f - Frequency - Hz
Figure 5.
10k 20k
THD+N -Total Harmonic Distortion + Noise
THD+N -Total Harmonic Distortion + Noise
f = 20 kHz
1%
0.1
1
PO - Output Power - W
100
10%
6
RL = 3 Ω
AV = 20 to 4 dB
BTL
RL = 4 Ω
AV = 20 to 4 dB
BTL
1%
f = 20 kHz
f = 1 kHz
0.1%
f = 20 Hz
0.01%
0.01
0.1
1
PO - Output Power - W
Figure 6.
10
TPA0162
www.ti.com
SLOS249E – JUNE 1999 – REVISED SEPTEMBER 2004
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
10%
RL = 8 Ω
AV = 20 to 4 dB
BTL
THD+N -Total Harmonic Distortion + Noise
THD+N -Total Harmonic Distortion + Noise
10%
1%
P0 = 0.25 W
0.1%
PO = 0.5 W
P0 = 1 W
0.01%
1%
f = 20 kHz
0.1%
f = 1 kHz
f = 20 Hz
0.01%
0.01
f - Frequency - Hz
0.1
1
PO - Output Power - W
Figure 7.
Figure 8.
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
20
100
1k
10k 20k
10
10%
10%
RL = 32 Ω
AV = 14 to 4 dB
SE
THD+N -Total Harmonic Distortion + Noise
THD+N -Total Harmonic Distortion + Noise
RL = 8 Ω
AV = 20 to 4 dB
BTL
1%
0.1%
PO = 25 mW
0.01%
PO = 50 mW
PO = 75 mW
0.001%
20
100
1k
f - Frequency - Hz
Figure 9.
10k 20k
RL = 32 Ω
AV = 14 to 4 dB
SE
1%
f = 20 kHz
0.1%
f = 1 kHz
0.01%
0.01
f = 20 Hz
0.1
PO - Output Power - W
1
Figure 10.
7
TPA0162
www.ti.com
SLOS249E – JUNE 1999 – REVISED SEPTEMBER 2004
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT VOLTAGE
10%
10%
THD+N -Total Harmonic Distortion + Noise
THD+N -Total Harmonic Distortion + Noise
RL = 10 kΩ
AV = 14 to 0 dB
SE
1%
0.1%
VO = 1 VRMS
0.01%
0.001%
20
100
1k
f = 1 kHz
0.01%
RL = 10 kΩ
AV = 14 to 4 dB
SE
0.4
0.6
f = 20 Hz
0.8
1
1.2
1.4
1.6
f - Frequency - Hz
VO - Output Voltage - VRMS
Figure 11.
Figure 12.
OUTPUT NOISE VOLTAGE
vs
BANDWIDTH
SUPPLY RIPPLE REJECTION RATIO
vs
FREQUENCY
1.8
2
0
RL = 8 Ω
C(BYP) = 0.47 µF
BTL
VDD = 5 V
RL = 4 Ω
140
Supply Ripple Rejection Ratio - dB
Vn - Output Noise Voltage - µV RMS
f = 20 kHz
0.1%
0.001%
0.2
10k 20k
160
120
100
80
60
AV = 20 dB
40
AV = 6 dB
20
0
-20
AV = 20 dB
-40
-60
-80
AV = 6 dB
-100
-120
0
100
1k
BW - Bandwidth - Hz
Figure 13.
8
1%
10k 20k
20
100
1k
f - Frequency - Hz
Figure 14.
10k 20k
TPA0162
www.ti.com
SLOS249E – JUNE 1999 – REVISED SEPTEMBER 2004
SUPPLY RIPPLE REJECTION RATIO
vs
FREQUENCY
CROSSTALK
vs
FREQUENCY
0
-40
RL = 32 Ω
C(BYP) = 0.47 µF
SE
-60
-40
Crosstalk - dB
Supply Ripple Rejection Ratio - dB
-20
AV = 0 dB
-60
-70
Left to Right
-80
-90
-80
Right to Left
AV = 14 dB
-100
-100
-110
-120
-120
20
100
1k
f - Frequency - Hz
20
10k 20k
100
1k
10k 20k
f - Frequency - Hz
Figure 15.
Figure 16.
CROSSTALK
vs
FREQUENCY
CROSSTALK
vs
FREQUENCY
-40
-40
PO = 1 W
RL = 8 Ω
AV = 6 dB
BTL
-60
-60
-70
Left to Right
-80
-90
VO = 1 VRMS
RL = 10 kΩ
AV = 6 dB
SE
-50
Crosstalk - dB
-50
Crosstalk - dB
PO = 1 W
RL = 8 Ω
AV = 20 dB
BTL
-50
Right to Left
-70
Left to Right
-80
-90
Right to Left
-100
-100
-110
-110
-120
-120
20
100
1k
10k 20k
20
100
1k
f - Frequency - Hz
f - Frequency - Hz
Figure 17.
Figure 18.
10k 20k
9
TPA0162
www.ti.com
SLOS249E – JUNE 1999 – REVISED SEPTEMBER 2004
SHUTDOWN ATTENUATION
vs
FREQUENCY
SIGNAL-TO-NOISE RATIO
vs
BANDWIDTH
120
0
PO = 1 W
RL = 8 Ω
BTL
VI = 1 VRMS
115
SNR - Signal-To-Noise Ratio - dB
RL = 10 kΩ, SE
-40
-60
RL = 32 Ω, SE
-80
-100
110
AV = 20 dB
105
100
95
AV = 6 dB
90
RL = 8 Ω, BTL
85
80
-120
20
100
1k
0
10k 20k
100
Figure 19.
Figure 20.
CLOSED LOOP RESPONSE
CLOSED LOOP RESPONSE
180°
RL = 8 Ω
AV = 20 dB
BTL
25
90°
5
Gain - dB
Phase
0°
10
RL = 8 Ω
AV = 6 dB
BTL
90°
20
15
Phase
180°
30
Gain
20
Gain - dB
10k 20k
BW - Bandwidth - Hz
30
25
1k
f - Frequency - Hz
15
Phase
0°
10
5
Gain
-90°
0
-5
-5
-10
-180°
10
100
1k
10k
f - Frequency - Hz
Figure 21.
10
-90°
0
100k
1M
-10
-180°
10
100
1k
10k
f - Frequency - Hz
Figure 22.
100k
1M
Phase
Shutdown Attenuation - dB
-20
TPA0162
www.ti.com
SLOS249E – JUNE 1999 – REVISED SEPTEMBER 2004
OUTPUT POWER
vs
LOAD RESISTANCE
OUTPUT POWER
vs
LOAD RESISTANCE
3.5
1500
AV = 20 to 0 dB
BTL
1250
PO - Output Power - mW
PO - Output Power - W
3
AV = 14 to 0 dB
SE
2.5
2
10% THD+N
1.5
1
1000
750
500
10% THD+N
250
0.5
1% THD+N
1% THD+N
0
0
0
8
16
24
32
40
48
RL - Load Resistance - Ω
56
64
0
Figure 24.
POWER DISSIPATION
vs
OUTPUT POWER
POWER DISSIPATION
vs
OUTPUT POWER
56
64
0.4
3Ω
1.6
0.35
1.4
PD - Power Dissipation - W
PD - Power Dissipation - W
16
24
32
40
48
RL - Load Resistance - Ω
Figure 23.
1.8
4Ω
1.2
1
0.8
0.6
8Ω
0.4
0.5
1
1.5
PO - Output Power - W
Figure 25.
2
4Ω
0.3
0.25
0.2
8Ω
0.15
0.1
32 Ω
f = 1 kHz
BTL
Each Channel
0.2
0
0
8
f = 1 kHz
SE
Each Channel
0.05
2.5
0
0
0.1
0.2
0.3
0.4
0.5
0.6
PO - Output Power - W
0.7
0.8
Figure 26.
11
TPA0162
www.ti.com
SLOS249E – JUNE 1999 – REVISED SEPTEMBER 2004
POWER DISSIPATION
vs
AMBIENT TEMPERATURE
INPUT IMPEDANCE
vs
GAIN
90
7
ΘJA4
PD - Power Dissipation - W
6
5
4
ΘJA3
3
ΘJA1,2
2
1
0
-40 -20
70
60
50
40
30
20
0
20 40 60 80 100 120 140 160
TA - Ambient Temperature - °C
Figure 27.
12
80
Z I - Input Impedance - k Ω
ΘJA1 = 45.9°C/W
ΘJA2 = 45.2°C/W
ΘJA3 = 31.2°C/W
ΘJA4 = 18.6°C/W
10
-40
-30
-20
-10
0
AV - Gain - dB
Figure 28.
10
20
TPA0162
www.ti.com
SLOS249E – JUNE 1999 – REVISED SEPTEMBER 2004
APPLICATION INFORMATION
VDD
0.47 µF
UP
100 kΩ
1
100 kΩ
2
LOUT+
3
DOWN
4
0.47 µF
LLINE
0.47 µF
LHP
5
6
7
0.47 µF
8
9
0.47 µF
10
LOUT–
11
0.47 µF
12
GND
UP
DOWN
LOUT+
LLINEIN
LHPIN
PVDD
RIN
LOUT–
LIN
BYPASS
GND
GND
RLINEIN
SHUTDOWN
ROUT+
RHPIN
VDD
PVDD
CLK
ROUT–
SE/BTL
PC-BEEP
GND
RLINE
24
23
22
SHUTDOWN
21
20
ROUT+
0.47 µF
RHP
19
0.1 µF
18
17
47 nF
0.1 µF
VDD
10 µF
16
ROUT–
15
SE/BTL
14
13
GND
PC-BEEP
0.47 µF
Figure 29. Typical TPA0162 Application Circuit
13
TPA0162
www.ti.com
SLOS249E – JUNE 1999 – REVISED SEPTEMBER 2004
APPLICATION INFORMATION (continued)
COMPONENT SELECTION
Figure 30 and Figure 31 are schematic diagrams of typical notebook computer application circuits.
Right CIRHP
Head- 0.47 µF
phone
Input
20
Signal
CIRLINE
Right 0.47 µF
23
Line
Input
Signal
8
CRIN
0.47 µF
Up
PC-BEEP
14
Input
Signal CPCB
0.47 µF
17
CCLK
47 nF
2
100 kΩ
3
15
VDD
RHPIN
RLINEIN
RIN
PC-BEEP
R
MUX
32-Step
Volume
Control
+
32-Step
Volume
Control
SE/BTL
COUTR
330 µF
+
Down
Depop
Circuitry
Power
Management
6
5
10
CLIN
0.47 µF
LHPIN
LLINEIN
LIN
ROUT-
PVDD
18
VDD
19
BYPASS
SHUTDOWN
11
GND
L
MUX
16
VDD
1 kΩ
100 kΩ
Gain/
MUX
Control
100 kΩ
Left CILHP
Head- 0.47 µF
phone
Input
Signal
CILLINE
Left 0.47 µF
Line
Input
Signal
21
PCBeep
CLK
UP
DOWN
ROUT+
32-Step
Volume
Control
See Note A
VDD
CSR
0.1 µF
VDD
CSR
0.1 µF
22
CBYP
0.47 µF
To
System
Control
+
LOUT+
4
+
LOUT-
9
1 kΩ
1, 12,
13, 24
COUTL
330 µF
32-Step
Volume
Control
100 kΩ
A.
A 0.1-µF ceramic capacitor should be placed as close as possible to the IC. For filtering lower-frequency noise
signals, a larger electrolytic capacitor of 10 µF or greater should be placed near the audio power amplifier.
Figure 30. Typical TPA0162 Application Circuit Using Single-Ended Inputs and Input MUX
14
TPA0162
www.ti.com
SLOS249E – JUNE 1999 – REVISED SEPTEMBER 2004
APPLICATION INFORMATION (continued)
CRHP0.47 µF
20
Right
Negative
Differential
Input Signal
CIRIN0.47 µF
23
CIRIN+
Right 0.47
µF
Positive
8
Differential
Input Signal
PC-BEEP
Input
Signal
CPCB
0.47 µF
14
17
RHPIN
RLINEIN
R
MUX
32-Step
Volume
Control
RIN
32-Step
Volume
Control
PC-BEEP
PCBeep
21
COUTR
330 µF
CLK
-
CCLK
47 nF
ROUT-
16
VDD
+
Up
2
3
DOWN
15
SE/BTL
Gain/
MUX
Control
Depop
Circuitry
VDD
Power
Management
Down
100 kΩ
CLHP0.47 µF
6
5
PVDD
18
VDD
19
BYPASS
SHUTDOWN
11
GND
LHPIN
LLINEIN
L
MUX
CILIN0.47 µF
32-Step
Volume
Control
-
LOUT+
See Note A
VDD
CSR
0.1 µF
VDD
CSR
0.1 µF
22
CBYP
0.47 µF
To
System
Control
4
1 kΩ
1, 12,
13, 24
+
CILIN+
Left 0.47
µF
Positive
10
Differential
Input Signal
LIN
1 kΩ
100 kΩ
UP
100 kΩ
Left
Negative
Differential
Input Signal
ROUT+
+
COUTL
330 µF
32-Step
Volume
Control
-
LOUT-
9
+
100 kΩ
A.
A 0.1-µF ceramic capacitor should be placed as close as possible to the IC. For filtering lower-frequency noise
signals, a larger electrolytic capacitor of 10 µF or greater should be placed near the audio power amplifier.
Figure 31. Typical TPA0162 Application Circuit Using Differential Inputs
15
TPA0162
www.ti.com
SLOS249E – JUNE 1999 – REVISED SEPTEMBER 2004
APPLICATION INFORMATION (continued)
UP/DOWN VOLUME CONTROL
Changing Volume
The default volume is set at mute mode. The volume is increased in 2-dB steps by pulling the UP terminal low.
The volume is decreased in 2-dB steps by pulling the DOWN terminal low. If power is removed, the device resets
to mute mode.
Volume Settings
VOLUME CONTROL
16
BTL (dB)
SE (dB)
20
14
18
12
16
10
14
8
12
6
10
4
8
2
6
0
4
-2
2
-4
0
-6
-2
-8
-4
-10
-6
-12
-8
-14
-10
-16
-12
-18
-14
-20
-16
-22
-18
-24
-20
-26
-22
-28
-24
-30
-26
-32
-28
-34
-30
-36
-32
-38
-34
-40
-36
-42
-38
-44
-40
-46
-85
-91
TPA0162
www.ti.com
SLOS249E – JUNE 1999 – REVISED SEPTEMBER 2004
Changing Volume When Using the Internal Clock
If using the internal clock, the maximum clock frequency is 500 Hz and the recommended frequency is 100 Hz
using a 47-nF capacitor. Use Equation 1 to calculate the clock frequency if using a capacitor to generate the
clock.
f
CLK
4.7 10
C
CLK
–6
(1)
When the desired volume-control signal is pulled low for four clock cycles, the volume increments by one step,
followed by a short delay. This delay decreases the longer the line is held low, eventually reaching a delay of
zero. The delay allows the user to pull the UP or DOWN terminal low once for one volume change, or hold down
to ramp several volume changes. The delay is optimally configured for push button volume control.
Holding either UP or DOWN low continuously causes the volume to change at an exponentially increasing rate.
When fCLK = 100 Hz, the first change in the volume occurs approximately 40 ms after either pin is initially pulled
low. If the pin stays low for approximately 400 more ms, the volume changes again. The next change occurs 200
ms after this change. The fourth change occurs 120 ms after the third change. The fifth volume change occurs
80 ms after the fourth change. Thereafter, the volume changes at 1/4 the rate of the clock (every 40 ms).
Each cycle is registered on the rising clock edge and the volume is changed after the rising edge.
Figure 32 shows increasing volume using UP, however, the volume is decreased using DOWN with the same
timing.
UP
CLK
VOLUME
40 cycles
20 cycles
12 cycles
8 cycles
4 cycles per step
4 cycles
Figure 32. Internal Clock Timing Diagram
Changing Volume When Using the External Clock (Microprocessor Mode)
The user may remove the capacitor and run the external clock directly into the clock pin to override the internal
clock generator. The maximum clock frequency is 10 kHz if using an external clock; however, a clock frequency
less than 200 Hz is recommended in normal operation so the gain does not change too quickly causing a pop at
the output. A 5-V, 50% duty-cycle clock must be used because the trip levels are 0.5 V and 4.5 V. The
recommended way to adjust the volume is to use a gated clock and hold UP or DOWN low and cycle the clock
pin four times to adjust the volume. The volume change is clocked in at the rising edge, so CLK should be held
low when not changing volume. No delay is added when using an external clock, so it is very important to input
only four clock cycles per volume change. Additional clock cycles per volume change are added to the next
volume change. For example, if five clock cycles are input while UP is held low the first volume change, the
volume change occurs after the third clock cycle the next time UP is held low. The figure below shows how
volume increases with UP when an external clock is used. The sample and hold times for UP and DOWN are
100 ns. The same timing applies if using an external clock and decreasing the volume with DOWN.
17
TPA0162
www.ti.com
SLOS249E – JUNE 1999 – REVISED SEPTEMBER 2004
UP
CLK
VOLUME
4 cycles per step
Figure 33. External Clock (4 Cycles Per Volume Change)
INPUT RESISTANCE
The gain is set by varying the input resistance of the amplifier, which can range from its smallest value to over
six times that value. As a result, if a single capacitor is used in the input high pass filter, the –3 dB or cut-off
frequency also changes by over six times. Connecting an additional resistor from the input pin of the amplifier to
ground, as shown in Figure 34, reduces the cutoff-frequency variation.
Rf
C
IN
Input Signal
Ri
R
Figure 34. Resistor on Input for Cut-Off Frequency
The input resistance at each gain setting is given in the graph for Input Impedance vs Gain in the Typical
Characteristics section.
The –3-dB frequency can be calculated using Equation 2.
ƒ–3 dB
1
2 CR R
i
(2)
To increase filter accuracy, increase the value of the capacitor and decrease the value of the resistor to ground.
In addition, the order of the filter can be increased.
18
TPA0162
www.ti.com
SLOS249E – JUNE 1999 – REVISED SEPTEMBER 2004
INPUT CAPACITOR, Ci
In a typical application, an input capacitor (Ci) is required to allow the amplifier to bias the input signal to the
proper dc level for optimum operation. In this case, Ci and the input impedance of the amplifier (Zi) form a
high-pass filter with the corner frequency determined by Equation 3.
−3 dB
fc(highpass)
1
2 ZIN C i
fc
(3)
The value of Ci directly affects the bass (low frequency) performance of the circuit. Consider the example where
Zi is 55 kΩ and the specification calls for a flat bass response down to 30 Hz. Equation 3 is reconfigured as
Equation 4.
Ci
1
2 Zi f c
(4)
In this example, Ci is 72 nF, so one would likely choose a value in the range of 0.1 µF to 1 µF. A further
consideration for this capacitor is the leakage path from the input source through the input network (Ci) and the
feedback network to the load. This leakage current creates a dc offset voltage at the input to the amplifier that
reduces useful headroom, especially in high-gain applications. For this reason a low-leakage tantalum or ceramic
capacitor is the best choice. When polarized capacitors are used, connect the positive lead of the capacitor to
the amplifier input in most applications, as the dc level there is held at VDD/2, typically higher than the source dc
level. Note that it is important to confirm the capacitor polarity in the application.
POWER SUPPLY DECOUPLING, C(S)
This high-performance CMOS audio amplifier requires adequate power-supply decoupling to minimize output
total harmonic distortion (THD). Power-supply decoupling also prevents oscillations with long lead lengths
between the amplifier and the speaker. Optimum decoupling is achieved by using two capacitors of different
types that target different types of noise on the power-supply leads. To filter high-frequency transients, spikes, or
digital hash on the line, a good low equivalent-series-resistance (ESR) ceramic capacitor, typically 0.1 µF, placed
as close as possible to the device VDD lead, works best. For filtering low-frequency noise signals, an aluminum
electrolytic capacitor of 10 µF or greater placed near the audio power amplifier is recommended.
MIDRAIL BYPASS CAPACITOR, C(BYP)
The midrail bypass capacitor, C(BYP), is the most critical capacitor and serves several important functions. During
startup or recovery from shutdown mode, C(BYP) determines the rate at which the amplifier starts up. The second
function is to reduce power-supply noise coupling into the output drive signal. This noise is from the midrail
generation circuit internal to the amplifier, and appears as degraded PSRR and THD+N.
Bypass capacitor (C(BYP)) values of 0.47-µF to 1-µF, and ceramic or tantalum low-ESR capacitors are
recommended for best THD and noise performance.
19
TPA0162
www.ti.com
SLOS249E – JUNE 1999 – REVISED SEPTEMBER 2004
OUTPUT COUPLING CAPACITOR, C(C)
In a typical single-supply SE configuration, an output coupling capacitor (C(C)) is required to block the dc bias at
the output of the amplifier to prevent dc currents in the load. As with the input coupling capacitor, the output
coupling capacitor and impedance of the load form a high-pass filter governed by Equation 5.
−3 dB
fc(high)
1
2 RL C (C)
fc
(5)
The main disadvantage, from a performance standpoint, is that load impedances are typically small, driving the
low-frequency corner higher, degrading the bass response. Large values of C(C) are required to pass low
frequencies into the load. Consider the example where a C(C) of 330 µF is chosen and loads include 3 Ω, 4 Ω, 8
Ω, 32 Ω, 10 kΩ, and 47 kΩ. Table 1 summarizes the frequency response characteristics of each configuration.
Table 1. Common Load Impedances Vs Low Frequency
Output Characteristics in SE Mode
RL
C(C)
LOWEST FREQUENCY
3Ω
330 µF
161 Hz
4Ω
330 µF
120 Hz
8Ω
330 µF
60 Hz
32 Ω
330 µF
15 Hz
10,000 Ω
330 µF
0.05 Hz
47,000 Ω
330 µF
0.01 Hz
As Table 1 indicates, most of the bass response is attenuated into a 4-Ω load, an 8-Ω load is adequate,
headphone response is good, and drive into line level inputs (a home stereo for example) is exceptional.
USING LOW-ESR CAPACITORS
Low-ESR capacitors are recommended throughout this applications section. A real (as opposed to ideal)
capacitor can be modeled simply as a resistor in series with an ideal capacitor. The voltage drop across this
resistor minimizes the beneficial effects of the capacitor in the circuit. The lower the equivalent value of this
resistance the more the real capacitor behaves like an ideal capacitor.
20
TPA0162
www.ti.com
SLOS249E – JUNE 1999 – REVISED SEPTEMBER 2004
BRIDGED-TIED LOAD VS SINGLE-ENDED MODE
Figure 35 shows a Class-AB audio power amplifier (APA) in a BTL configuration. The TPA0162 amplifier consists
of two Class-AB amplifiers driving both ends of the load. There are several potential benefits to this differential
drive configuration, but, initially consider power to the load. The differential drive to the speaker means that as
one side is slewing up, the other side is slewing down, and vice versa. This in effect doubles the voltage swing
on the load as compared to a ground referenced load. Substituting 2 × VO(PP) into the power equation, where
voltage is squared, yields 4× the output power from the same supply rail and load impedance (see Equation 6).
V(rms)
Power
V O(PP)
2 2
V(rms)
2
RL
(6)
VDD
VO(PP)
RL
2x VO(PP)
VDD
−VO(PP)
Figure 35. Bridge-Tied Load Configuration
In a typical computer sound channel operating at 5 V, bridging raises the power into an 8-Ω speaker from a
singled-ended (SE, ground reference) limit of 250 mW to 1 W. In sound power, this is a 6-dB improvement —
loudness that can be heard. In addition to increased power there are frequency-response concerns. Consider the
single-supply SE configuration shown in Figure 36. A coupling capacitor is required to block the dc offset voltage
from reaching the load. These capacitors can be quite large (approximately 33 µF to 1000 µF), so they tend to be
expensive, heavy, occupy valuable PCB area, and have the additional drawback of limiting the low-frequency
performance of the system. This frequency-limiting effect is due to the high-pass filter network created with the
speaker impedance and the coupling capacitance, and is calculated with Equation 7.
f(c)
1
2 RL C (C)
(7)
For example, a 68-µF capacitor with an 8-Ω speaker would attenuate low frequencies below 293 Hz. The BTL
configuration cancels the dc offsets, eliminating the need for blocking capacitors. Low-frequency performance is
then limited only by the input network and speaker response. Cost and PCB space are also minimized by
eliminating the bulky coupling capacitor.
21
TPA0162
www.ti.com
SLOS249E – JUNE 1999 – REVISED SEPTEMBER 2004
VDD
−3 dB
VO(PP)
C(C)
RL
VO(PP)
fc
Figure 36. Single-Ended Configuration and Frequency Response
Increasing power to the load does carry a penalty of increased internal power dissipation. The increased
dissipation is understandable, since the BTL configuration produces 4× the output power of the SE configuration.
Internal dissipation versus output power is discussed further in the Crest Factor and Thermal Considerations
section.
Single-Ended Operation
In SE mode (see Figure 36), the load is driven from the primary amplifier output for each channel (LOUT+ and
ROUT+).
The amplifier switches to single-ended operation when the SE/BTL terminal is held high. This puts the negative
outputs in a high-impedance state, and reduces the amplifier's gain by 6 dB.
BTL AMPLIFIER EFFICIENCY
Class-AB amplifiers are inefficient, primarily because of voltage drop across the output-stage transistors. The two
components of the internal voltage drop are the headroom or dc voltage drop that varies inversely to output
power, and the sine wave nature of the output. The total voltage drop can be calculated by subtracting the RMS
value of the output voltage from VDD. The internal voltage drop multiplied by the RMS value of the supply current
(IDDrms) determines the internal power dissipation of the amplifier.
An easy-to-use equation to calculate efficiency begins as the ratio of power from the power supply to the power
delivered to the load. To accurately calculate the RMS and average values of power in the load and in the
amplifier, the current and voltage waveforms must be understood (see Figure 37).
VO
IDD
IDD(avg)
V(LRMS)
Figure 37. Voltage and Current Waveforms for BTL Amplifiers
Although the voltages and currents for SE and BTL are sinusoidal in the load, currents from the supply are very
different between SE and BTL configurations. In an SE application, the current waveform is a half-wave rectified
shape, whereas in BTL it is a full-wave rectified waveform. Therefore, RMS conversion factors are different. Keep
in mind that for most of the waveform both the push and pull transistors are not on at the same time, which
supports the fact that each amplifier in the BTL device only draws current from the supply for half the waveform.
Equation 8 and Equation 9 are the basis for calculating amplifier efficiency.
22
TPA0162
www.ti.com
SLOS249E – JUNE 1999 – REVISED SEPTEMBER 2004
Efficiency of a BTL amplifier
PL
PSUP
Where:
2
V Lrms 2
V
V
, and VLRMS P , therefore, P L P
RL
2 RL
2
PL
and PSUP V DD IDDavg
IDDavg 1
and
2V P
VP
VP
1
[
cos(t)] 0
sin(t) dt
RL
RL
RL
0
Therefore,
PSUP
2 V DD VP
RL
substituting PL and PSUP into equation 7,
2
Efficiency of a BTL amplifier
Where:
VP
VP
2 RL
2 VDD V P
RL
VP
4 V DD
2 P L RL
(8)
Therefore,
BTL
2 P L RL
4 VDD
VP = Peak voltage on BTL load
IDDavg = Average current drawn from the power supply
VDD = Power supply voltage
ηBTL = Efficiency of a BTL amplifier
PL = Power delivered to load
PSUP = Power drawn from power supply
VLRMS = RMS voltage on BTL load
RL = Load resistance
(9)
Table 2 employs Equation 9 to calculate efficiencies for four different output-power levels. Note that the efficiency
of the amplifier is quite low for lower power levels and rises sharply as power to the load is increased resulting in
a nearly flat internal power dissipation over the normal operating range. Note that the internal dissipation at full
output power is less than in the half-power range. Calculating the efficiency for a specific system is the key to
proper power supply design. For a stereo 1-W audio system with 8-Ω loads and a 5-V supply, the maximum draw
on the power supply is almost 3.25 W.
Table 2. Efficiency vs Output Power in 5-V, 8-Ω BTL Systems
OUTPUT POWER (W)
EFFICIENCY
(%)
PEAK VOLTAGE (V)
INTERNAL DISSIPATION (W)
0.25
31.4
2.00
0.55
0.50
44.4
2.83
0.62
1.00
62.8
4.00
0.59
1.25
70.2
4.47 (1)
0.53
(1)
High peak voltages cause the THD to increase.
A final point to remember about Class-AB amplifiers (either SE or BTL) is how to manipulate the terms in the
efficiency equation to utmost advantage when possible. Note that in Equation 9, VDD is in the denominator. This
indicates that as VDD goes down, efficiency goes up.
23
TPA0162
www.ti.com
SLOS249E – JUNE 1999 – REVISED SEPTEMBER 2004
CREST FACTOR AND THERMAL CONSIDERATIONS
Class-AB power amplifiers dissipate a significant amount of heat in the package under normal operating
conditions. A typical music CD requires 12 dB to 15 dB of dynamic range, or headroom, above the average
power output, to pass the loudest portions of the signal without distortion. In other words, music typically has a
crest factor between 12 dB and 15 dB. When determining the optimal ambient operating temperature, the internal
dissipated power at the average output power level must be used. From the data sheet, one can see that when
the device is operating from a 5-V supply into a 3-Ω speaker that 4-W peaks are available. Use Equation 10 to
convert watts to dB.
P
P dB 10Log W 10Log 4 W 6 dB
1W
P ref
(10)
Subtracting the headroom restriction to obtain the average listening level without distortion yields:
6 dB - 15 dB = -9 dB (15-dB crest factor)
6 dB - 12 dB = -6 dB (12-dB crest factor)
6 dB - 9 dB = -3 dB (9-dB crest factor)
6 dB - 6 dB = 0 dB (6-dB crest factor)
6 dB - 3 dB = 3 dB (3-dB crest factor)
Converting dB back into watts:
PW = 10PdB/10× Pref
= 63 mW (18-dB crest factor)
= 125 mW (15-dB crest factor)
= 250 mW (9-dB crest factor)
= 500 mW (6-dB crest factor)
= 1000 mW (3-dB crest factor)
= 2000 mW (0-dB crest factor)
This is valuable information to consider when estimating the heat-dissipation requirements for the amplifier
system. Comparing the worst case, 2 W of continuous power output with a 3-dB crest factor, against 12-dB and
15-dB applications, drastically affects maximum ambient temperature ratings for the system. Using the power
dissipation curves for a 5-V, 3-Ω system, the internal dissipation and maximum ambient temperatures are shown
in the table below.
Table 3. TPA0162 Power Rating, 5-V, 3-Ω Stereo
PEAK OUTPUT POWER
(W)
(1)
24
AVERAGE OUTPUT POWER
POWER DISSIPATION
(W/Channel)
MAXIMUM AMBIENT
TEMPERATURE
4
2 W (3 dB)
1.7
-3°C
4
1000 mW (6 dB)
1.6
6°C
4
500 mW (9 dB)
1.3
24°C
4
250 mW (12 dB)
1.0
51°C
4
125 mW (15 dB)
0.9
78°C
4
63 mW (18 dB)
0.6
85°C (1)
Package limited to 85°C ambient.
TPA0162
www.ti.com
SLOS249E – JUNE 1999 – REVISED SEPTEMBER 2004
Table 4. TPA0162 Power Rating, 5-V, 8-Ω Stereo
(1)
PEAK OUTPUT POWER
(W)
AVERAGE OUTPUT POWER
POWER DISSIPATION
(W/Channel)
MAXIMUM AMBIENT
TEMPERATURE
2.5
1250 mW (3-dB crest factor)
0.53
85°C (1)
2.5
1000 mW (4-dB crest factor)
0.59
85°C (1)
2.5
500 mW (7-dB crest factor)
0.62
85°C (1)
2.5
250 mW (10-dB crest factor)
0.58
85°C (1)
Package limited to 85°C ambient.
The maximum dissipated power (PDmax) is reached at a much lower output power level for a 3-Ω load than for an
8-Ω load. As a result, the formula in Equation 11for calculating PDmax may be used for a 3-Ω application:
2V2
DD
P Dmax
2R L
(11)
However, in the case of an 8-Ω load, the PDmax occurs at a point well above the normal operating power level.
The amplifier may therefore be operated at a higher ambient temperature than required by the PDmax formula for
an 8-Ω load, but do not exceed the maximum ambient temperature of 85°.
The maximum ambient temperature depends on the heatsinking ability of the PCB system. The derating factor
for the PWP package is shown in the dissipation rating table. Converting this to θJA:
1
1
θ JA
45°CW
0.022
Derating Factor
(12)
To calculate maximum ambient temperatures, first consider that the numbers from the dissipation graphs are
per-channel, so the dissipated heat is doubled for two-channel operation. Given θJA, the maximum allowable
junction temperature, and the total internal dissipation, the maximum ambient temperature can be calculated
using Equation 13. The maximum recommended junction temperature for the device is 150°C. The internal
dissipation figures are taken from the Power Dissipation vs Output Power graphs.
T A Max T J Max θJA P D
150 45(0.6 2) 96°C (15-dB crest factor)
(13)
NOTE:
Internal dissipation of 0.6 W is estimated for a 2-W system with 15-dB crest factor per
channel.
Due to package limitiations, the actual TAMAX is 85°C.
The power rating tables show that for some applications, no airflow is required to keep junction temperatures in
the specified range. The internal thermal protection turns the device off at junction temperatures higher than
150°C to prevent damage to the IC. The power rating tables in this section were calculated for maximum
listening volume without distortion. When the output level is reduced the numbers in the table change
significantly. Also, using 8-Ω speakers dramatically increases the thermal performance by increasing amplifier
efficiency.
SE/BTL OPERATION
The ability of the TPA0162 to easily switch between BTL and SE modes is one of its most important cost-saving
features. This feature eliminates the requirement for an additional headphone amplifier in applications where
internal stereo speakers are driven in BTL mode but external headphone or speakers must be accommodated.
Two separate internal amplifiers drive OUT+ and OUT–. The SE/BTL input controls the operation of the follower
amplifier that drives LOUT– and ROUT–. When SE/BTL is held low, the amplifier is on and the device is in the
BTL mode. When SE/BTL is held high, the OUT– amplifiers are in a high output-impedance state, which
configures the device outputs as SE drivers from LOUT+ and ROUT+. IDD is reduced by approximately one-half
in SE mode. Control of the SE/BTL input can be from a logic-level CMOS source or, more typically, from a
resistor-divider network as shown in Figure 38.
25
TPA0162
www.ti.com
SLOS249E – JUNE 1999 – REVISED SEPTEMBER 2004
20
23
RHPIN
RLINEIN
32-Step
Volume
Control
R
MUX
−
+
8
RIN
ROUT+
21
32-Step
Volume
Control
VDD
−
+
ROUT−
16
100 kΩ
SE/BTL
COUTR
330 µF
15
1 kΩ
100 kΩ
Figure 38. TPA0162 Resistor Divider Network Circuit
Using a readily-available 1/8-in. (3,5 mm) stereo headphone jack, the control switch is closed when no plug is
inserted. When closed, the 100-kΩ/1-kΩ divider pulls the SE/BTL input low. When a plug is inserted, the 1-kΩ
resistor is disconnected and the SE/BTL input is pulled high. When the input goes high, the OUT– amplifier is
shut down, muting the speaker (virtually open-circuits the speaker). The OUT+ amplifier then drives through the
output capacitor (CO) into the headphone jack.
PC-BEEP OPERATION
The PC-BEEP input allows a system beep to be sent directly from a computer through the amplifier to the
speakers with few external components. The input is activated automatically. When the PC-BEEP input is active,
both LINEIN and HPIN inputs are deselected, and both the left and right channels are driven in BTL mode with
the signal from PC-BEEP. The gain from the PC-BEEP input to the speakers is fixed at 0.3 V/V and is
independent of the volume setting. When the PC-BEEP input is deselected, the amplifier returns to the previous
operating mode and volume setting. Furthermore, if the amplifier is in shutdown mode, activating PC-BEEP takes
the device out of shutdown, outputs the PC-BEEP signal, then returns the amplifier to shutdown mode.
When PCB ENABLE is held low, the amplifier automatically switches to PC-BEEP mode after detecting a valid
signal at the PC-BEEP input. The preferred input signal is a square wave or pulse train. To be accurately
detected, the signal must have a minimum of 1.5-Vpp amplitude, rise and fall times of less than 0.1 µs and a
minimum of eight rising edges. When the signal is no longer detected, the amplifier returns to its previous
operating mode and volume setting.
To ac-couple the PC-BEEP input, choose a coupling-capacitor value to satisfy Equation 14.
C
PCB
2 ƒ
1
(100 k)
PCB
(14)
The PC-BEEP input can also be dc-coupled to avoid using this coupling capacitor. The pin normally rests at
midrail when no signal is present.
26
TPA0162
www.ti.com
SLOS249E – JUNE 1999 – REVISED SEPTEMBER 2004
INPUT MUX OPERATION
CIRHP
0.47 µF
Right
Headphone
Input Signal
20
Right Line
Input Signal
23
CIRLINE
0.47 µF
8
CRIN
0.47 µF
RHPIN
RLINEIN
RIN
32-Step
Volume
Control
R
MUX
−
+
ROUT+
21
−
+
ROUT−
16
32-Step
Volume
Control
Figure 39. TPA0162 Example Input MUX Circuit
The input MUX provides the user with a means to select from two different audio sources. In BTL mode, the
LINE input is selected. In SE mode, the HP inputs are selected. RIN and LIN must be AC-grounded in SE mode.
SHUTDOWN MODES
The TPA0162 provides a shutdown mode to minimize supply current (IDD) during periods of nonuse to conserve
battery power. Hold the SHUTDOWN input terminal high during normal operation when the amplifier is in use.
Pulling SHUTDOWN low mutes the outputs and causes the amplifier to enter a low-current state, IDD = 150 µA.
Do not leave SHUTDOWN unconnected because amplifier operation would be unpredictable.
Table 5. Shutdown and Mute Mode Functions
INPUTS (1)
(1)
AMPLIFIER STATE
SE/BTL
SHUTDOWN
INPUT
OUTPUT
Low
High
Line
BTL
X
Low
X
Mute
High
High
HP
SE
Do not leave inputs unconnected.
27
PACKAGE OPTION ADDENDUM
www.ti.com
14-Oct-2022
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
Samples
(4/5)
(6)
TPA0162PWP
ACTIVE
HTSSOP
PWP
24
60
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
TPA0162
Samples
TPA0162PWPR
ACTIVE
HTSSOP
PWP
24
2000
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
TPA0162
Samples
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
工商网监
湘ICP备2023018690号
