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TPS40210-Q1, TPS40211-Q1
SLVS861F – AUGUST 2008 – REVISED JUNE 2020
TPS4021x-Q1 4.5-V to 52-V Input, Current-Mode Boost Controllers
1 Features
2 Applications
•
•
•
•
•
1
•
•
•
•
•
•
•
•
•
•
•
•
Qualified for automotive applications
AEC-Q100 qualified with the following results:
– Device temperature grade 1: –40°C to 125°C
ambient operating temperature range
– Device HBM ESD classification level 2
– Device CDM ESD classification level C4B
Functional Safety-Capable
– Documentation available to aid functional
safety system design
For boost, flyback, SEPIC, LED driver applications
Wide input operating voltage: 4.5 V to 52 V
Adjustable oscillator frequency
Fixed-frequency current-mode control
Internal slope compensation
Integrated low-side driver
Programmable closed-loop soft start
Overcurrent protection
External synchronization capable
Reference voltage: 700 mV (TPS40210-Q1),
260 mV (TPS40211-Q1)
Low-current disable function
Infotainment and cluster applications
Automotive body electronics (lighting)
HEV/EV and powertrain
3 Description
The TPS40210-Q1 and TPS40211-Q1 devices are
wide-input-voltage (4.5 V to 52 V) non-synchronous
boost controllers. They are suitable for topologies that
require a grounded source N-channel FET, including
boost, flyback, SEPIC, and various LED driver
applications. Device features include programmable
soft start, overcurrent protection with automatic retry,
and programmable oscillator frequency. Currentmode control provides improved transient response
and simplified loop compensation. The main
difference between the two parts is the reference
voltage to which the error amplifier regulates the FB
pin.
Device Information(1)
PART NUMBER
TPS40210-Q1
TPS40211-Q1
PACKAGE
PDSO (10)
BODY SIZE (NOM)
3.00 mm × 3.00 mm
(1) For all available packages, see the orderable addendum at
the end of the data sheet.
Simplified Schematic
VIN
TPS40210-Q1
1
RC
2
SS
BP
9
3
DIS/EN GDRV
8
4
COMP
ISNS
7
5
FB
GND
6
VOUT
VDD 10
R SENSE
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
TPS40210-Q1, TPS40211-Q1
SLVS861F – AUGUST 2008 – REVISED JUNE 2020
www.ti.com
Table of Contents
1
2
3
4
5
6
7
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Pin Configuration and Functions .........................
Specifications.........................................................
1
1
1
2
3
4
6.1
6.2
6.3
6.4
6.5
6.6
6.7
6.8
4
4
4
4
5
6
6
7
Absolute Maximum Ratings ......................................
ESD Ratings..............................................................
Recommended Operating Conditions.......................
Thermal Information ..................................................
Electrical Characteristics...........................................
Timing Requirements ................................................
Switching Characteristics ..........................................
Typical Characteristics ..............................................
Detailed Description ............................................ 11
7.1 Overview ................................................................. 11
7.2 Functional Block Diagram ....................................... 11
7.3 Feature Description................................................. 11
7.4 Device Functional Modes........................................ 23
8
Application and Implementation ........................ 26
8.1 Application Information............................................ 26
8.2 Typical Application .................................................. 26
9 Power Supply Recommendations...................... 33
10 Layout................................................................... 34
10.1 Layout Guidelines ................................................. 34
10.2 Layout Example .................................................... 35
11 Device and Documentation Support ................. 37
11.1
11.2
11.3
11.4
11.5
11.6
Device Support ....................................................
Documentation Support .......................................
Related Links ........................................................
Trademarks ...........................................................
Electrostatic Discharge Caution ............................
Glossary ................................................................
37
37
37
37
37
37
12 Mechanical, Packaging, and Orderable
Information ........................................................... 38
4 Revision History
Changes from Revision E (December 2014) to Revision F
•
Page
Added functional safety bullet to the Features ...................................................................................................................... 1
Changes from Revision D (April 2010) to Revision E
•
2
Page
Added ESD Ratings table, Feature Description section, Device Functional Modes, Application and Implementation
section, Power Supply Recommendations section, Layout section, Device and Documentation Support section, and
Mechanical, Packaging, and Orderable Information section ................................................................................................. 4
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Copyright © 2008–2020, Texas Instruments Incorporated
Product Folder Links: TPS40210-Q1 TPS40211-Q1
TPS40210-Q1, TPS40211-Q1
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SLVS861F – AUGUST 2008 – REVISED JUNE 2020
5 Pin Configuration and Functions
DGQ Package
10-Pin PDSO PowerPAD™ Package
(Top View)
RC
1
10
VDD
SS
2
9
BP
DIS/EN
3
8
GDRV
COMP
4
7
ISNS
FB
5
6
GND
Pin Functions
PIN
NAME
NO.
I/O
DESCRIPTION
BP
9
O
Regulator output. Connect a 1-μF bypass capacitor from this pin to GND.
COMP
4
O
Error amplifier output. Connect a control-loop compensation network between the COMP pin and the FB pin.
DIS/EN
3
I
Disable or enable. Pulling this pin high places the part into a shutdown mode. The prime characteristic of
shutdown mode is a very low quiescent current. Shutdown mode disables the functionality of all blocks and
shuts down the BP regulator. This pin has an internal 1-MΩ pulldown resistor to GND. Leaving this pin
unconnected enables the device.
FB
5
I
Error amplifier inverting input. Connect a voltage divider from the output to this pin to set the output voltage.
Connect a compensation network between this pin and COMP.
GDRV
8
O
Connect the gate of the power N-channel MOSFET to this pin.
GND
6
—
Device ground
ISNS
7
I
Current sense. Connect an external current-sensing resistor between this pin and GND. The voltage on this
pin provides current feedback in the control loop for detecting an overcurrent condition. Declaration of an
overcurrent condition occurs when ISNS pin voltage exceeds the overcurrent threshold voltage, 150 mV
typical.
RC
1
I
Switching-frequency setting. Connect a capacitor from the RC pin to GND. Connect a resistor from the RC
pin to VDD of the IC power supply and a capacitor from RC to GND.
SS
2
I
Soft-start time programming. Connect a capacitor from the SS pin to GND to program the converter softstart time. This pin also functions as a time-out timer when the power supply is in an overcurrent condition.
VDD
10
I
System input voltage. Connect a local bypass capacitor from this pin to GND. Depending on the amount of
required slope compensation, connection of this pin to the converter output might be desirable. See the
Application and Implementation section for additional details.
Copyright © 2008–2020, Texas Instruments Incorporated
Product Folder Links: TPS40210-Q1 TPS40211-Q1
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6 Specifications
6.1 Absolute Maximum Ratings
over operating free-air temperature range unless otherwise noted (1)
Input voltage range
Output voltage range
TJ
MIN
MAX
UNIT
VDD
–0.3
52
V
RC, SS, FB, DIS/EN
–0.3
10
V
ISNS
–0.3
8
V
COMP, BP, GDRV
–0.3
9
V
–40°C
150
°C
–55°C
150
°C
Operating junction temperature
Tstg Storage temperature
(1)
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, and do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended
Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
6.2 ESD Ratings
VALUE
Human-body model (HBM), per AEC Q100-002 (1)
V(ESD)
(1)
Electrostatic discharge
Charged-device model (CDM), per AEC
Q100-011
UNIT
±2000
All pins - Classification
level C4B for both All pins
and Corner pins
±750
Corner pins (1, 5, 6, and
10)
±750
V
AEC Q100-002 indicates that HBM stressing shall be in accordance with the ANSI/ESDA/JEDEC JS-001 specification.
6.3 Recommended Operating Conditions
MIN
NOM MAX UNIT
VDD
Input voltage
4.5
52
V
TJ
Operating junction temperature
–40
125
°C
6.4 Thermal Information
TPS40210-Q1,
TPS40211-Q1
THERMAL METRIC (1)
DGQ
UNIT
10 PINS
RθJA
Junction-to-ambient thermal resistance
67.2
°C/W
RθJC(top)
Junction-to-case (top) thermal resistance
50.5
°C/W
RθJB
Junction-to-board thermal resistance
41
°C/W
ψJT
Junction-to-top characterization parameter
2.4
°C/W
ψJB
Junction-to-board characterization parameter
40.7
°C/W
RθJC(bot)
Junction-to-case (bottom) thermal resistance
15.6
°C/W
(1)
4
For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report.
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SLVS861F – AUGUST 2008 – REVISED JUNE 2020
6.5 Electrical Characteristics
TJ = –40°C to 125°C, VDD= 12 Vdc, all parameters at zero power dissipation (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
COMP = FB,
4.5 ≤ VDD ≤ 52 V
TJ = 25°C
693
700
707
–40°C ≤ TJ ≤ 125°C
686
700
714
COMP = FB,
4.5 ≤ VDD ≤ 52 V
TJ = 25°C
254
260
266
–40°C ≤ TJ ≤ 125°C
250
260
270
4.5 ≤ VDD ≤ 52 V, no switching, VDIS < 0.8
1.5
2.5
2.5 ≤ VDIS ≤ 7 V
10
UNIT
VOLTAGE REFERENCE
TPS40210
-Q1
VFB
Feedback voltage range
TPS40211
-Q1
mV
INPUT SUPPLY
IDD
Operating current
VDD < VUVLO(on), VDIS < 0.8
20
530
mA
μA
UNDERVOLTAGE LOCKOUT (UVLO)
VUVLO(on)
Turnon threshold voltage
VUVLO(hyst)
UVLO hysteresis
Frequency line regulation
VSLP
4
4.25
4.5
V
140
195
240
mV
4.5 ≤ VDD ≤ 52 V
–20%
7%
7 ≤ VDD ≤ 52 V
–10%
7%
Slope compensation ramp
520
620
720
mV
PWM
VVLY
Valley voltage
1.2
V
1
V
SOFT-START
VSS(ofst)
Offset voltage from SS pin to error
amplifier input
RSS(chg)
Soft-start charge resistance
320
430
600
RSS(dchg)
Soft-start discharge resistance
840
1200
1600
1.5
3.0
MHz
60
80
dB
kΩ
ERROR AMPLIFIER
GBWP
Unity gain bandwidth product (1)
(1)
AOL
Open loop gain
IIB(FB)
Input bias current (current out of FB
pin)
ICOMP(src)
Output source current
VFB = 0.6 V, VCOMP = 1 V
100
250
μA
ICOMP(snk)
Output sink current
VFB = 1.2 V, VCOMP = 1 V
1.2
2.5
mA
4.5 ≤ VDD < 52 V, –40°C ≤ TJ ≤ 125°C
120
150
100
300
nA
OVERCURRENT PROTECTION
VISNS(oc)
Overcurrent detection threshold (at
ISNS pin)
(1)
DOC
Overcurrent duty cycle
VSS(rst)
Overcurrent reset threshold voltage
(at SS pin)
(1)
180
mV
2%
100
150
350
mV
Specified by design
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Electrical Characteristics (continued)
TJ = –40°C to 125°C, VDD= 12 Vdc, all parameters at zero power dissipation (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
4.2
5.6
7.2
V/V
1
3
μA
CURRENT-SENSE AMPLIFIER
ACS
Current sense amplifier gain
IB(ISNS)
Input bias current
DRIVER
IGDRV(src)
Gate driver source current
VGDRV = 4 V, TJ = 25°C
375
400
IGDRV(snk)
Gate driver sink current
VGDRV = 4 V, TJ = 25°C
330
400
7
8
mA
LINEAR REGULATOR
VBP
Bypass voltage output
0 mA < IBP < 15 mA
9
V
1.3
V
DISABLE AND ENABLE
VDIS(en)
Turn-on voltage
0.7
VDIS(hys)
Hysteresis voltage
25
130
220
mV
RDIS
DIS pin pulldown resistance
0.7
1.1
1.5
MΩ
6.6 Timing Requirements
MIN
TYP
MAX
275
400
90
200
170
200
UNIT
PWM
tON(min)
Minimum pulse duration
tOFF(min)
Minimum off-time
VDD = 12 V (1)
VDD = 30 V
ns
OVERCURRENT PROTECTION
tBLNK
(1)
Leading edge blanking
75
ns
Specified by design
6.7 Switching Characteristics
over operating free-air temperature range (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
RRC = 182 kΩ, CRC = 330 pF
260
TYP
MAX
UNIT
1000
KHz
OSCILLATOR
fOSC
Oscillator frequency range (1)
Oscillator frequency
(1)
6
35
300
340
Specified by design
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SLVS861F – AUGUST 2008 – REVISED JUNE 2020
6.8 Typical Characteristics
68 pF
CT(pF)
33pF
470
220
100
68
33
fSW - Frequency - kHz
1000
800
100pF
600
220 pF
400
1200
1000
fSW - Frequency - kHz
1200
800
600
400
200
200
470 pF
0
100
200
0
300 400 500 600 700 800
RT - Timing Resistance - kW
0
900 1000
Figure 1. Frequency vs Timing Resistance
0.2
0.4
0.6
0.8
D - Duty Cycle
1.0
1.2
Figure 2. Switching Frequency vs Duty Cycle
1.4
6
52 V
4.5 V
1.0
12 V
0.8
0.6
0.4
VVDD
12 V
4.5 V
52 V
0.2
5
IVDD – Shutdown Current – mA
IVDD – Quiescent Current – mA
1.2
Figure 4. Shutdown Current vs Junction Temperature
0.5
52 V
0.4
0.2
0.0
-0.2
-0.4
4.5 V
12 V
VVDD
12 V
4.5 V
52 V
-0.8
-40 -25 -10 5 20 35 50 65 80 95 110 125
TJ – Junction Temperature – ° C
Figure 5. Reference Voltage Change vs Junction
Temperature
VFB – Reference Voltage Change – %
VFB – Reference Voltage Change – %
2
0
-40 -25 -10 5 20 35 50 65 80 95 110 125
TJ – Junction Temperature – ° C
0.4
-0.6
3
1
0
-40 -25 -10 5 20 35 50 65 80 95 110 125
TJ – Junction Temperature – ° C
Figure 3. Quiescent Current vs Junction Temperature
4
0.3
0.2
0.1
0.0
-0.1
-0.2
-0.3
-0.4
-0.5
0
10
20
30
40
VVDD – Input Voltage – V
50
60
Figure 6. Reference Voltage Change vs Input Voltage
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Typical Characteristics (continued)
155
4.30
VUVLO – Undervoltage Lockout Threshold – V
UVLO
4.20
VISNS(OC) – Overcurrent Threshold – mV
4.25
UVLO On
4.15
4.10
4.05
Figure 7. Undervoltage Lockout Threshold vs Junction
Temperature
153
7.5 V
152
30 V
151
150
12 V & 20 V
149
148
Figure 8. Overcurrent Threshold vs Junction Temperature
5
154
fOSC – Switching Frequency Change – %
VISNS(OC) – Overcurrent Threshold – mV
4.5 V
7.5 V
12 V & 20 V
30 V
147
-40 -25 -10 5 20 35 50 65 80 95 110 125
TJ – Junction Temperature – ° C
155
153
152
151
150
149
148
147
146
145
5
10
15
20
25
30 35
VVDD – Input Voltage – V
40
Figure 9. Overcurrent Threshold vs Input Voltage
2
4.5 V
1
12 V
0
-1
30 V
-2
VVDD (V)
4.5 V
12 V
30 V
-3
-4
Figure 10. Switching Frequency Change vs Junction
Temperature
1400
29
27
4
3
-5
-40 -25 -10 5 20 35 50 65 80 95 110 125
TJ – Junction Temperature – ° C
45
4.5 V
RSS – Soft Start Charge/Discharge Resistance - kW
0
Slope Compensation Ratio (VVDD/VSLP)
154
UVLO Off
4.00
-40 -25 -10 5 20 35 50 65 80 95 110 125
TJ – Junction Temperature – ° C
RSS(DSCH) Discharge
1200
25
1000
23
24 V
12 V
21
19
VVDD (V)
36 V
17
12 V
24 V
36 V
4.5 V
15
-40 -25 -10 5 20 35 50 65 80 95 110 125
TJ – Junction Temperature – ° C
Figure 11. Oscillator Amplitude vs Junction Temperature
8
4.5 V
VVDD
Off
On
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800
600
400
200
RSS(CHG) Charge
0
-40 -25 -10 5 20 35 50 65 80 95 110 125
TJ – Junction Temperature – ° C
Figure 12. Soft-Start Charge and Discharge Resistance vs
Junction Temperature
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SLVS861F – AUGUST 2008 – REVISED JUNE 2020
Typical Characteristics (continued)
180
ICOMP(SRC) – Compensation Source Current – mA
300
IIB(FB) – Feedback Bias Current – nA
160
140
120
100
80
60
40
20
0
-40 -25 -10 5 20 35 50 65 80 95 110 125
TJ – Junction Temperature – ° C
Figure 13. FB Bias Current vs Junction Temperature
200
150
100
50
0
-40 -25 -10 5 20 35 50 65 80 95 110 125
TJ – Junction Temperature – ° C
Figure 14. Compensation Source Current vs Junction
Temperature
5
300
4
250
VVLY – Valley Voltage Change – %
ICOMP(SNK) – Compensation Sink Current – mA
250
200
150
100
50
3
2
1
0
-1
-2
-3
-4
0
-40 -25 -10 5 20 35 50 65 80 95 110 125
TJ – Junction Temperature – ° C
Figure 15. Compensation Sink Current vs Junction
Temperature
-5
-40 -25 -10 5 20 35 50 65 80 95 110 125
TJ – Junction Temperature – ° C
Figure 16. Valley Voltage Change vs Junction Temperature
1.10
8.8
VBP – Regulator Voltage – V
ILOAD = 0 mA
8.4
VDIS(EN) – DIS/EN Turn-On Threshold – mV
1.09
8.6
1.08
1.07
1.06
8.2
1.05
8.0
7.8
1.06
ILOAD = 5 mA
1.03
1.02
7.6
1.01
7.4
-40 -25 -10 5 20 35 50 65 80 95 110 125
TJ – Junction Temperature – ° C
Figure 17. Regulator Voltage vs Junction Temperature
1.00
-40 -25 -10 5 20 35 50 65 80 95 110 125
TJ – Junction Temperature – ° C
Figure 18. DIS/EN Turn-On Threshold vs Junction
Temperature
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Typical Characteristics (continued)
ACS – Current Sense Amplifier Gain – V/V
7
6
5
4
3
2
1
0
-40 -25 -10 5 20 35 50 65 80 95 110 125
TJ – Junction Temperature – ° C
Figure 19. Current-Sense Amplifier Gain vs Junction Temperature
10
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SLVS861F – AUGUST 2008 – REVISED JUNE 2020
7 Detailed Description
7.1 Overview
The TPS40210-Q1 and TPS40211-Q1 devices are wide-input voltage non-sync boost controllers. These devices
can be used in various topologies such as boost, flyback, SEPIC, and various LED driver applications because of
its grounded source N-channel FET. The device also features programmable soft start, overcurrent protection,
and programmable oscillator frequency. Current mode control provides improved transient response and
simplified loop compensation. The TPS40210-Q1 and TPS40211-Q1 devices differ in the reference voltage to
which the error amplifier regulates the FB pin.
7.2 Functional Block Diagram
DIS/EN
3
COMP
4
FB
5
10 VDD
+
+
SS
2
OC Fault
Soft Start
and
Overcurrent
E/A
SS Ref
700 mV
PWM
Logic
Oscillator
and
Slope
Compensation
1
9
BP
8
GDRV
6
GND
7
ISNS
Driver
Enable E/A
RC
LDO
+
Gain = 6
OC Fault
150 mV
UVLO
+
LEB
UDG-07107
7.3 Feature Description
7.3.1 Minimum On-Time and Off-Time Considerations
The TPS40210-Q1 device has a minimum off-time of approximately 200 ns and a minimum on-time of 300 ns.
These two constraints place limitations on the operating frequency that can be used for a given input-to-output
conversion ratio. See Figure 2 for the maximum frequency that can be used for a given duty cycle.
The duty cycle at which the converter operates is dependent on the mode in which the converter is running. If the
converter is running in discontinuous-conduction mode, the duty cycle varies with changes to the load much
more than it does when running in continuous-conduction mode.
In continuous-conduction mode, the duty cycle is related primarily to the input and output voltages.
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Feature Description (continued)
VOUT + VD
1
=
VIN
1- D
(1)
æ æ
VIN
D = ç1 - ç
ç
è è VOUT + VD
öö
÷ ÷÷
øø
(2)
In discontinuous-conduction mode, the duty cycle is a function of the load, input and output voltages, inductance,
and switching frequency.
D=
(
)
2 ´ VOUT + VD ´ IOUT ´ L ´ f SW
2
(VIN )
(3)
All converters using a diode as the freewheeling or catch component have a load current level at which they
transition from discontinuous conduction to continuous conduction. This is the point at which the inductor current
falls to zero. At higher load currents, the inductor current does not fall to zero but remains flowing in a positive
direction and assumes a trapezoidal wave shape as opposed to a triangular wave shape. This load boundary
between discontinuous conduction and continuous conduction can be found for a set of converter parameters as
shown in Equation 4.
2
(VOUT + VD - VIN ) ´ (VIN )
IOUT(crit) =
2
2 ´ (VOUT + VD ) ´ f SW ´ L
(4)
For loads higher than the result of Equation 4, the duty cycle is given by Equation 2, and for loads less than the
results of Equation 4, the duty cycle is given Equation 3. For Equation 1 through Equation 4, the variable
definitions are as follows:
• VOUT is the output voltage of the converter in V.
• VD is the forward conduction voltage drop across the rectifier or catch diode in V.
• VIN is the input voltage to the converter in V.
• IOUT is the output current of the converter in A.
• L is the inductor value in H.
• fSW is the switching frequency in Hz.
7.3.2 Current Sense and Overcurrent
The TPS40210-Q1 and TPS40211-Q1 devices are current-mode controllers and use a resistor in series with the
source terminal power FET to sense current for both the current-mode control and overcurrent protection. The
device enters a current-limit state if the voltage on the ISNS pin exceeds the current-limit threshold voltage
VISNS(oc) from the Electrical Characteristics. When this happens, the controller discharges the SS capacitor
through a relatively high impedance and then attempts to restart. The amount of output current that causes this
to happen is dependent on several variables in the converter.
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Feature Description (continued)
VIN
TPS40210 -Q1 or
TPS40211-Q1
TPS40210 -Q1 or
TPS40211-Q1
10 VDD
L
RT
VOU T
VDD 10
1
RC
GDRV
8
ISNS
7
CT
RIFL T
6
GND
CIFL T
GND
Figure 20. Oscillator Components
RISN S
6
Figure 21. Current Sense Components
The load current overcurrent threshold is set by proper choice of RISNS. If the converter is operating in
discontinuous mode, the current sense resistor is found in Equation 5.
f SW ´ L ´ VISNS(oc)
R ISNS =
2 ´ L ´ f SW ´ I OUT(oc) ´ VOUT + VD - VIN
(
)
(5)
If the converter is operating in continuous conduction mode, RISNS can be found in Equation 6.
VISNS
VISNS
=
R ISNS =
æ I OUT ö æ IRIPPLE ö æ I OUT ö æ D ´ VIN ö
çç
÷÷ + çç
÷÷ ç
÷
÷+ç
è 1 - D ø è 2 ø èç (1 - D ) ø÷ èç 2 ´ f SW ´ L ø÷
where
•
•
•
•
•
•
•
•
RISNS is the value of the current sense resistor in Ω.
VISNS(oc) is the overcurrent threshold voltage at the ISNS pin (from the Electrical Characteristics)
D is the duty cycle (from Equation 2)
f SW is the switching frequency in Hz
VIN is the input voltage to the power stage in V (see text)
L is the value of the inductor in H
IOUT(oc) is the desired overcurrent trip point in A
VD is the drop across the diode in Figure 21
(6)
The TPS40210-Q1 and TPS40211-Q1 devices have a fixed undervoltage lockout (UVLO) that allows the
controller to start at a typical input voltage of 4.25 V. If the input voltage is slowly rising, the converter might have
less than its designed nominal input voltage available when it has reached regulation. As a result, this can
decrease the apparent current-limit load current value and must be taken into consideration when selecting
RISNS. The value of VIN used to calculate RISNS must be the value at which the converter finishes start-up. The
total converter output current at start-up is the sum of the external load current and the current required to charge
the output capacitor(s). See the Soft Start section of this data sheet for information on calculating the required
output capacitor charging current.
The topology of the standard boost converter has no method to limit current from the input to the output in the
event of a short circuit fault on the output of the converter. If protection from this type of event is desired, it is
necessary to use some secondary protection scheme such as a fuse or rely on the current limit of the upstream
power source.
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Feature Description (continued)
7.3.3 Current Sense and Subharmonic Instability
A characteristic of peak current-mode control results in a condition where the current control loop can exhibit
instability. This results in alternating long and short pulses from the pulse-width modulator. The voltage loop
maintains regulation and does not oscillate, but the output ripple voltage increases. The condition occurs only
when the converter is operating in continuous conduction mode, and the duty cycle is 50% or greater. The cause
of this condition is described in the Modeling, Analysis and Compensation of the Current-Mode Converter
Application Report. The remedy for this condition is to apply a compensating ramp from the oscillator to the
signal going to the pulse-width modulator. In the TPS40210-Q1 and TPS40211-Q1 devices, the oscillator ramp is
applied in a fixed amount to the pulse-width modulator. The slope of the ramp is given in Equation 7.
æV
ö
se = f SW ´ ç VDD ÷
è 20 ø
(7)
To ensure that the converter does not enter into sub-harmonic instability, the slope of the compensating ramp
signal must be at least half of the down slope of the current ramp signal. Because the compensating ramp is
fixed in the TPS40210-Q1 and TPS40211-Q1 devices, this places a constraint on the selection of the current
sense resistor.
The down slope of the current sense wave form at the pulse-width modulator is described in Equation 8.
A CS ´ RISNS ´ (VOUT + VD - VIN )
L
m2 =
(8)
Because the slope compensation ramp must be at least half, and preferably equal to, the down slope of the
current sense waveform seen at the pulse-width modulator, a maximum value is placed on the current sense
resistor when operating in continuous mode at 50% duty cycle or greater. For design purposes, some margin
should be applied to the actual value of the current sense resistor. As a starting point, the actual resistor chosen
should be 80% or less that the value calculated in Equation 9. This equation calculates the resistor value that
makes the slope compensation ramp equal to one half of the current ramp downslope. Values no more than 80%
of this result are acceptable.
VVDD ´ L ´ f SW
R ISNS(max) =
60 ´ VOUT + VD - VIN
(
)
where
•
•
•
•
•
•
•
Se is the slope of the voltage compensating ramp applied to the pulse-width modulator in V/s
f SW is the switching frequency in Hz
VDD is the voltage at the VDD pin in V
m2 is the down slope of the current sense waveform seen at the pulse-width modulator in V/s
RISNS is the value of the current sense resistor in Ω
VOUT is the converter output voltage VIN is the converter power stage input voltage
VD is the drop across the diode in Figure 21
(9)
It is possible to increase the voltage compensation ramp slope by connecting the VDD pin to the output voltage of
the converter instead of the input voltage as shown in Figure 21. This can help in situations where the converter
design calls for a large ripple current value in relation to the desired output current limit setting.
NOTE
Connecting the VDD pin to the output voltage of the converter affects the start-up voltage
of the converter since the controller undervoltage lockout (UVLO) circuit monitors the VDD
pin and senses the input voltage less the diode drop before start-up. The effect is to
increase the start-up voltage by the value of the diode voltage drop.
If an acceptable RISNS value is not available, the next higher value can be used and the signal from the resistor
divided down to an acceptable level by placing another resistor in parallel with CISNS.
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Feature Description (continued)
7.3.4 Current Sense Filtering
In most cases, a small filter placed on the ISNS pin improves performance of the converter. These are the
components RIFLT and CIFLT in Figure 21. The time constant of this filter should be approximately 10% of the
nominal pulse width of the converter. The pulse width can be found using Equation 10.
D
t ON =
f SW
(10)
The suggested time constant is then
RIFLT ´ CIFLT = 0.1´ tON
(11)
The range of RIFLT should be from about 1 kΩ to 5 kΩ for best results. Higher values can be used, but this raises
the impedance of the ISNS pin connection more than necessary and can lead to noise-pickup issues in some
layouts. CISNS should be located as close as possible to the ISNS pin as well to provide noise immunity.
7.3.5 Soft Start
The soft-start feature of the TPS40210-Q1 and TPS40211-Q1 devices is a closed-loop soft start, meaning that
the output voltage follows a linear ramp that is proportional to the ramp generated at the SS pin. This ramp is
generated by an internal resistor connected from the BP pin to the SS pin and an external capacitor connected
from the SS pin to GND. The SS pin voltage (VSS) is level shifted down by approximately VSS(ofst) (approximately
1 V) and sent to one of the + inputs (the + input with the lowest voltage dominates) of the error amplifier. When
this level-shifted voltage (VSSE) starts to rise at time t1 (see Figure 22), the output voltage that the controller
expects rises as well. Since VSSE starts at near 0 V, the controller attempts to regulate the output voltage from a
starting point of zero volts. It cannot do this, due to the converter architecture. The output voltage starts from the
input voltage less the drop across the diode (VIN – VD) and rises from there. The point at which the output
voltage starts to rise (t2) is when the VSSE ramp passes the point where it is commanding more output voltage
than (VIN – VD). This voltage level is labeled VSSE(1). The time required for the output voltage to ramp from a
theoretical zero to the final regulated value (from t1 to t3) is determined by the time it takes for the capacitor
connected to the SS pin (CSS) to rise through a 700-mV range, beginning at VSS(ofst) above GND.
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Feature Description (continued)
TPS40210 -Q1 or
TPS40211-Q1
VSS
RSS(chg)
700 mV REF
SS
VSS(ofst)+700 mV
Error Amplifier
2
+
+
VSSE
VSS(ofst)
RSS(dchg)
VSSE(1)
t0
t1
VIN - VD
VOUT
t2
t3
DIS
UVLO
OC Fault
FB
5
COMP
4
Figure 22. SS Pin Voltage and Output Voltage
Figure 23. SS Pin Functional Circuit
The required capacitance for a given soft-start time, t3 – t1 in Figure 22, is calculated in Equation 12.
CSS =
tSS
æ
VBP - VSS(ofst)
RSS ´ ln ç
çV - V
SS(ofst) + VFB
è BP
(
ö
÷
÷
ø
)
where
•
•
•
•
•
•
tSS is the soft-start time
RSS(chg) is the SS charging resistance in Ω, typically 500 kΩ
CSS is the value of the capacitor on the SS pin, in F
VBP is the value of the voltage on the BP pin in V
VSS(ofst) is the approximate level shift from the SS pin to the error amplifier (~1 V)
VFB is the error amplifier reference voltage, 700 mV typical
(12)
Note that tSS is the time it takes for the output voltage to rise from 0 V to the final output voltage. Also note the
tolerance on RSS(chg) given in the Electrical Characteristics. This contributes to some variability in the output
voltage rise time, and margin must be applied to account for it in design.
Also take note of VBP. Its value varies depending on input conditions. For example, a converter operating from a
slowly rising input initializes VBP at a fairly low value and increases during the entire start-up sequence. If the
controller has a voltage above 8 V at the input and the DIS pin is used to stop and then restart the converter, VBP
is approximately 8 V for the entire start-up sequence. The higher the voltage on BP, the shorter the start-up time
is and conversely, the lower the voltage on BP, the longer the start-up time is.
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Feature Description (continued)
The soft-start time (tSS) must be chosen long enough so that the converter can start up without going into an
overcurrent state. Since the overcurrent state is triggered by sensing the peak voltage on the ISNS pin, that
voltage must be kept below the overcurrent threshold voltage, VISNS(oc). The voltage on the ISNS pin is a function
of the load current of the converter, the rate of rise of the output voltage and the output capacitance, and the
current sensing resistor. The total output current that must be supported by the converter is the sum of the
charging current required by the output capacitor and any external load that must be supplied during start-up.
This current must be less than the IOUT(oc) value used in Equation 5 or Equation 6 (depending on the operating
mode of the converter) to determine the current sense resistor value.
In these equations, the actual input voltage at the time that the controller reaches the final output voltage is the
important input voltage to use in the calculations. If the input voltage is slowly rising and is at less than the
nominal input voltage when the startup time ends, the output current limit is less than IOUT(oc) at the nominal input
voltage. The output capacitor charging current must be reduced (decrease COUT or increase the tSS) or IOUT(oc)
must be increased and a new value for RISNS calculated.
IC(chg) =
COUT ´ VOUT
tSS
(13)
COUT ´ VOUT
(IOUT(oc) - IEXT)
tSS >
where
•
•
•
•
•
•
IC(chg) is the output capacitor charging current in A
COUT is the total output capacitance in F
VOUT is the output voltage in V
tSS is the soft-start time from Equation 12
IOUT(oc) is the desired over current trip point in A
IEXT is any external load current in A
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Feature Description (continued)
The capacitor on the SS pin (CSS) also plays a role in overcurrent functionality. It is used as the timer between
restart attempts. The SS pin is connected to GND through a resistor, RSS(dchg), when the controller senses an
overcurrent condition. Switching stops and nothing else happens until the SS pin discharges to the soft-start
reset threshold, VSS(rst). At this point, the SS pin capacitor is allowed to charge again through the charging
resistor RSS(chg), and the controller restarts from that point. The shortest time between restart attempts occurs
when the SS pin discharges from VSS(ofst) (approximately 1 V) to VSS(rst) (150 mV) and then back to VSS(ofst) and
switching resumes. In actuality, this is a conservative estimate since switching does not resume until the VSSE
ramp rises to a point where it is commanding more output voltage than exists at the output of the controller. This
occurs at some SS pin voltage greater than VSS(ofst) and depends on the voltage that remains on the output
overvoltage the converter while switching has been halted. The fastest restart time can be calculated by using
Equation 15, Equation 16, and Equation 17.
æ VSS(ofst)
tDCHG = RSS(dchg) ´ CSS ´ ln ç
ç VSS(rst)
è
(
(
ö
÷
÷
ø
(15)
) ö÷
)÷ø
æ V -V
BP
SS(rst)
tCHG = RSS(chg) ´ CSS ´ ln ç
ç V -V
SS(ofst)
è BP
(16)
tRSTRT(min ) = tCHG + tDCHG
(17)
VBP
VSS
tRSTR(min)
VSS(ofst)
VSS(rst)
T - Time
Figure 24. Soft Start During Overcurrent
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Feature Description (continued)
7.3.6 BP Regulator
The TPS40210-Q1 and TPS40211-Q1 devices have an on-board linear regulator that supplies power for the
internal circuitry of the controller, including the gate driver. This regulator has a nominal output voltage of 8 V and
must be bypassed with a 1-μF capacitor. If the voltage at the VDD pin is less than 8 V, the voltage on the BP pin
is also less, and the gate drive voltage to the external FET is reduced from the nominal 8 V. This should be
considered when choosing a FET for the converter.
Connecting external loads to this regulator can be done, but care must be taken to ensure that the thermal rating
of the device is observed, because there is no thermal shutdown feature in this controller. Exceeding the thermal
ratings causes out-of-specification behavior and can lead to reduced reliability. The controller dissipates more
power when there is an external load on the BP pin and is tested for dropout voltage for up to 5-mA load. When
the controller is in the disabled state, the BP pin regulator also shuts off so loads connected there power down
as well. When the controller is disabled with the DIS/EN pin, this regulator is turned off.
The total power dissipation in the controller can be calculated as follows. The total power is the sum of PQ, PG,
and PE.
PQ = VVDD ´ IVDD(en)
(18)
PG = VVDD ´ Qg ´ f SW
(19)
PE = VVDD ´ IEXT
where
•
•
•
•
•
•
•
•
PQ is the quiescent power of the device in W
VDD is the VDD pin voltage in V
IDD(en) is the quiescent current of the controller when enabled but not switching in A
PG is the power dissipated by driving the gate of the FET in W
Qg is the total gate charge of the FET at the voltage on the BP pin in C
fSW is the switching frequency in Hz
PE is the dissipation caused be external loading of the BP pin in W
IEXT is the external load current in A
(20)
7.3.7 Shutdown (DIS/EN Pin)
The DIS/EN pin is an active-high shutdown command for the controller. Pulling this pin above 1.2 V causes the
controller to completely shut down and enter a low current consumption state. In this state, the regulator
connected to the BP pin is turned off. There is an internal 1.1-MΩ pulldown resistor connected to this pin that
keeps the pin at GND level when left floating. If this function is not used in an application, it is best to connect
this pin to GND
7.3.8 Control Loop Considerations
There are two methods to design a suitable control loop for the TPS4021x device. The first (and preferred, if
equipment is available) is to use a frequency-response analyzer to measure the open-loop modulator and power
stage gain and to then design compensation to fit that. The usage of these tools for this purpose is welldocumented with the literature that accompanies the tool and is not discussed here.
The second option is to make an initial guess at compensation, and then evaluate the transient response of the
system to see if the compensation is acceptable to the application or not. For most systems, an adequate
response can be obtained by simply placing a series resistor and capacitor (RFB and CFB) from the COMP pin to
the FB pin as shown in Figure 25.
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Feature Description (continued)
VIN
TPS40210-Q1
L
1
RC
2
SS
BP
9
3
DIS/EN GDRV
8
VOUT
VDD 10
CHF
CFB
COUT
RIFLT
RFB
4
ISNS
COMP
7
CIFLT
5
FB
GND
ROUT
RSENSE
6
R1
R2
Figure 25. Basic Compensation Network
The natural phase characteristics of most capacitors used for boost outputs combined with the current mode
control provide adequate phase margin when using this type of compensation. To determine an initial starting
point for the compensation, the desired crossover frequency must be considered when estimating the control to
output gain. The model used is a current source into the output capacitor and load.
When using these equations, the loop bandwidth should be no more than 20% of the switching frequency, fSW. A
more reasonable loop bandwidth would be 10% of the switching frequency. Be sure to evaluate the transient
response of the converter over the expected load range to ensure acceptable operation.
( ) = 19.1 S ´ 0.146 W = 2.80
K CO = gm ´ ZOUT f CO
0.13 ´ L ´
gm =
f SW
R OUT
(21)
0.13 ´ 10 mH ´
=
600 kHz
240 W
2
2
(R ISNS ) ´ (120 ´ R ISNS + L ´ fSW ) (12 mW ) ´ (120 ´ 12 mW + 10 mH ´ 600 kHz )
Z OUT = R OUT ´
2ö
æ
ç 1 + 2p ´ fL ´ R ESR ´ COUT ÷
è
ø
2ö
+ 2 ´ R OUT ´ R ESR + R ESR ÷ ´ 2p ´ fL ´ COUT
ø
(
æ
1 + ç R OUT
è
(
)
2
= 19.1 S
(22)
)
(
)
(
)
2
where
•
•
•
•
•
•
•
20
KCO is the control to output gain of the converter, in V/V
gM is the transconductance of the power stage and modulator, in S
ROUT is the output load equivalent resistance, in Ω
ZOUT is the output impedance, including the output capacitor, in Ω
RISNS is the value of the current sense resistor, in Ω
L is the value of the inductor, in H
COUT is the value of the output capacitance, in μF
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Feature Description (continued)
•
•
•
RESR is the equivalent series resistance of COUT, in Ω
f SW is the switching frequency, in Hz
f L is the desired crossover frequency for the control loop, in Hz
(23)
These equations assume that the operation is discontinuous and that the load is purely resistive. The gain in
continuous conduction can be found by evaluating Equation 22 at the resistance that gives the critical conduction
current for the converter. Loads that are more like current sources give slightly higher gains than predicted here.
To find the gain of the compensation network required for a control loop of bandwidth f L, take the reciprocal of
Equation 21.
K COMP =
1
K CO
=
1
= 0.356
2.80
(24)
The GBWP of the error amplifier is only specified to be at least 1.5 MHz. If KCOMP multiplied by the fL is greater
than 750 kHz, reduce the desired loop crossover frequency until this condition is satisfied. This ensures that the
high-frequency pole from the error amplifier response with the compensation network in place does not cause
excessive phase lag at the f L and decrease phase margin in the loop.
The R-C network connected from COMP to FB places a zero in the compensation response. That zero should be
approximately 1/10th of the desired crossover frequency, f L. With that being the case, RFB and CFB can be found
from Equation 25 and Equation 26.
RFB =
CFB =
R1
= R1´ K COMP
K CO
(25)
10
2p ´ fL ´ RFB
where
•
•
R1 is in fL is the loop crossover frequency desired, in Hz
RFB is the feedback resistor in CFB is the feedback capacitance in μF
(26)
Though not strictly necessary, it is recommended that a capacitor be added between COMP and FB to provide
high-frequency noise attenuation in the control loop circuit. This capacitor introduces another pole in the
compensation response. The allowable location of that pole frequency determines the capacitor value. As a
starting point, the pole frequency should be 10 × fL. The value of CHF can be found from Equation 27.
CHF =
1
20p ´ fL ´ RFB
(27)
The error amplifier GBWP will usually be higher, but is ensured by design to be at least 1.5 MHz. If the gain
required in Equation 24 multiplied by 10 times the desired control loop crossover frequency, the high-frequency
pole introduced by CHF is overridden by the error amplifier capability and the effective pole is lower in frequency.
If this is the case, CHF can be made larger to provide a consistent high-frequency roll off in the control loop
design. Equation 28 calculates the required CHF in this case.
CHF =
1
6
2p ´ 1.5 ´ (10 ) ´ RFB
where
•
•
CHF is the high-frequency roll-off capacitor value in μF
RFB is the mid-band gain-setting resistor value in Ω
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Feature Description (continued)
7.3.9 Gate Drive Circuit
Some applications benefit from the addition of a resistor connected between the GDRV pin and the gate of the
switching MOSFET. In applications that have particularly stringent load regulation (under 0.75%) requirements
and operate from input voltages above 5 V, or are sensitive to pulse jitter in the discontinuous conduction region,
this resistor is recommended. The recommended starting point for the value of this resistor can be calculated
from Equation 29.
RG =
105
QG
where
•
•
QG is the MOSFET total gate charge at 8-V VGS in nC
RG is the suggested starting point gate resistance in Ω
(29)
VIN
TPS40210-Q1 or
TPS40211-Q1
L
VDD 10
VOUT
RG
GDRV
8
ISNS
7
GND
6
Figure 26. Gate Drive Resistor
7.3.10 TPS40211-Q1
The only difference between the TPS40210-Q1 and the TPS40211-Q1 devices is the reference voltage that the
error amplifier uses to regulate the output voltage. The TPS40211-Q1 device uses a 260-mV reference and is
intended for applications where the output is actually a current instead of a regulated voltage. A typical example
of an application of this type is an LED driver. Figure 27 shows an example schematic.
An example of an LED driver design using the TPS40211-Q1 device with detailed analysis is available in the
TPS40211 – SEPIC Design for MR-16 LED Application Report.
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Feature Description (continued)
V IN
IOUT
1
TPS40210-Q1 or
TPS40211-Q1
RC
VDD 10
2
SS
3
DIS/EN
4
COMP
5
FB
BP
9
GDRV
8
L
ISNS 7
RIFB
GND
6
Figure 27. Typical LED Drive Schematic
The current in the LED string is set by the choice of the resistor RISNS as shown in Equation 30.
RIFB =
VFB
IOUT
where
•
•
•
RIFB is the value of the current sense resistor for the LED string in Ω
VFB is the reference voltage for the TPS40211-Q1 device in volts (0.26 V typ)
IOUT is the desired dc current in the LED string in amperes
(30)
7.4 Device Functional Modes
7.4.1 Setting the Oscillator Frequency
The oscillator frequency is determined by a resistor and capacitor connected to the RC pin of the TPS40210-Q1
device. The capacitor is charged to a level of approximately VDD / 20 by current flowing through the resistor and
is then discharged by a transistor internal to the TPS40210-Q1 device. The required resistor for a given oscillator
frequency is found from either Figure 1 or Equation 31.
1
RT =
-8
-10
-7
2
´ f SW + 1.4 ´ 10 ´ f SW - 1.5 ´ 10-4 + 1.7 ´ 10-6 ´ CT - 4 ´ 10-9 ´ CT 2
5.8 ´ 10 ´ f SW ´ CT + 8 ´ 10
where
•
•
•
RT is the timing resistance in kΩ
fSW is the switching frequency in kHz
CT is the timing capacitance in pF
(31)
For most applications, a capacitor in the range of 68 pF to 120 pF gives the best results. Resistor values should
be limited to between 100 kΩ and 1 MΩ as well. If the resistor value falls below 100 kΩ, decrease the capacitor
size and recalculate the resistor value for the desired frequency. As the capacitor size decreases below 47 pF,
the accuracy of Equation 31 degrades, and empirical means may be needed to fine tune the timing component
values to achieve the desired switching frequency.
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Device Functional Modes (continued)
7.4.2 Synchronizing the Oscillator
The TPS40210-Q1 and TPS40211-Q1 devices can be synchronized to an external clock source. Figure 28
shows the functional diagram of the oscillator. When synchronizing the oscillator to an external clock, the RC pin
must be pulled below 150 mV for 20 ns or more. The external clock frequency must be higher than the free
running frequency of the converter as well. When synchronizing the controller, if the RC pin is held low for an
excessive amount of time, erratic operation can occur. The maximum amount of time that the RC pin should be
held low is 50% of a nominal output pulse, or 10% of the period of the synchronization frequency.
Under circumstances where the duty cycle is less than 50%, a Schottky diode connected from the RC pin to an
external clock can be used to synchronize the oscillator. The cathode of the diode is connected to the RC pin.
The trip point of the oscillator is set by an internal voltage divider to be 1/20 of the input voltage. The clock signal
must have an amplitude higher than this trip point. When the clock goes low, it allows the reset current to restart
the RC ramp, synchronizing the oscillator to the external clock. This provides a simple single-component method
for clock synchronization.
VIN
IOUT
1
TPS40210-Q1 or
TPS40211-Q1
RC
V DD 10
2
SS
3
BP
9
DIS/EN
GDRV
8
4
COMP
ISNS
7
5
FB
GND
6
L
RIFB
Figure 28. Oscillator Functional Diagram
24
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Device Functional Modes (continued)
VDD
Amplitude >
VIN
20
VIN
8
+
RRC
S
Q
R
Q
CLK
Duty Cycle < 50%
RC
+
1
Frequency > Controller
Frequency
+
CRC
150 mV
GND
5
TPS40210 -Q1 or TPS40211 -Q1
Figure 29. Diode Connected Synchronization
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8 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
8.1 Application Information
The TPS40210-Q1 and TPS40211-Q1 devices support a wide range of input voltages from 4.5 V to 52 V in a
non-synchronous boost topology. The applications can also be expanded to flyback, SEPIC, and various LED
driver applications. The current-mode control provides the advantages of improved transient response and ease
of selecting compensation components. Other features of the device such as programmable soft start,
overcurrent protection with automatic retry, and adjustable oscillator frequency using external components
increase the versatility of TPS4021x-Q1 devices. The main difference between the TPS40210-Q1 and
TPS40211-Q1 devices is the reference voltage to which the error amplifier regulates the FB pin.
8.2 Typical Application
Figure 30 illustrates the design process and component selection for a 12-V to 24-V non-synchronous boost
regulator using the TPS40210-Q1 controller.
+
+
Figure 30. TPS40210-Q1 Design Example – 12 V (Typical) to 24 V at 2 A
8.2.1 Design Requirements
Table 1. TPS40210-Q1 Design Example Requirements
PARAMETER
CONDITIONS
MIN NOM MAX
UNIT
INPUT CHARACTERISTICS
VIN
26
Input voltage
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8
12
14
V
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Typical Application (continued)
Table 1. TPS40210-Q1 Design Example Requirements (continued)
PARAMETER
IIN
CONDITIONS
MIN NOM MAX
Input current
4.4
No load input current
VIN(UVLO)
0.05
Input undervoltage lockout
4.5
UNIT
A
V
OUTPUT CHARACTERISTICS
VOUT
Output voltage
23.5
24.0
Line regulation
24.5
V
1%
Load regulation
1%
VOUT(ripple)
Output voltage ripple
500
IOUT
Output current
IOCP
Output overcurrent inception point
8 V ≤ VIN ≤ 14 V
0.2
1
2
3.5
mVPP
A
Transient response
ΔI
Load step
1
A
Load slew rate
1
A/μs
500
mV
5
ms
600
kHz
Overshoot threshold voltage
Settling time
SYSTEM CHARACTERISTICS
fSW
Switching frequency
ηPK
Peak efficiency
VIN = 12 V, 0.2 A ≤ IOUT ≤ 2 A
95%
η
Full load efficiency
VIN = 12 V, IOUT = 2 A
94%
TOP
Operating temperature range
10 V ≤ VIN ≤ 14 V, 0.2 A ≤ IOUT ≤ 2 A
25
°C
MECHANICAL DIMENSIONS
W
Width
1.5
L
Length
1.5
h
Height
0.5
in
8.2.2 Detailed Design Procedure
8.2.2.1 Duty Cycle Estimation
The duty cycle of the main switching MOSFET is estimated using Equation 32 and Equation 33.
DMIN »
DMAX »
VOUT - VIN(max) + VFD 24 V - 14 V + 0.5 V
=
= 42.8%
VOUT + VFD
24 V + 0.5 V
(32)
VOUT - VIN(m in) + VFD
24 V - 8 V + 0.5 V
=
= 67.3%
VOUT + VFD
24 V + 0.5 V
(33)
Using and estimated forward drop of 0.5 V for a Schottky rectifier diode, the approximate duty cycle is 42.8%
(minimum) to 67.3% (maximum).
8.2.2.2 Inductor Selection
The peak-to-peak ripple is limited to 30% of the maximum output current.
ILrip(m ax) = 0.3 ´
IOUT(m ax)
1 - DMIN
= 0.3 ´
2
= 1.05 A
1 - 0.428
(34)
The minimum inductor size can be estimated using Equation 35.
VIN(max)
1
14 V
1
L MIN >>
´ D MIN ´
=
´ 0.428 ´
= 9.5 mH
I Lrip(max)
f SW 1.05 A
600 kHz
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The next higher standard inductor value of 10 μH is selected. The ripple current is estimated by Equation 36.
VIN
1
12 V
1
IRIPPLE »
´D´
=
´ 0.50 ´
= 1.02 A
L
f SW 10 mH
600 kHz
(36)
VIN
1
8V
1
IRIPPLE(Vinmin) »
´D´
=
´ 0.673 ´
= 0.89 A
L
f SW 10 mH
600 kHz
(37)
The worst-case peak-to-peak ripple current occurs at 50% duty cycle and is estimated as 1.02 A. Worst-case rms
current through the inductor is approximated by Equation 38.
ILrms =
(I ( ) ) + (
2
L avg
1 I
12 RIPPLE
)
2
æ IOUT(max)
» ç
ç 1- D
MAX
è
2
ö
÷÷ +
ø
(112IRIPPLE(VINmin) )
2
2
2
æ
ö
= ç
÷ +
1
0.673
è
ø
((112)´ 0.817A )
2
= 6.13 Arms
(38)
The worst case RMS inductor current is 6.13 Arms. The peak inductor current is estimated by Equation 39.
I OUT(max) æ 1 ö
2
æ 1ö
ILpeak »
+ ç ÷ IRIPPLE(Vinmin) =
+ ç ÷ 0.718 = 6.57 A
1 - DMAX è 2 ø
1 - 0.673 è 2 ø
(39)
A 10-μH inductor with a minimum RMS current rating of 6.13 A and minimum saturation current rating of 6.57 A
must be selected. A TDK RLF12560T-100M-7R5 7.5-A 10-μH inductor is selected.
This inductor power dissipation is estimated by Equation 40.
2
PL » (ILrms ) ´ DCR
(40)
The TDK RLF12560T-100M-7R5 12.4-mΩ DCR dissipates 466 mW of power.
8.2.2.3 Rectifier Diode Selection
A low-forward voltage drop Schottky diode is used as a rectifier diode to reduce its power dissipation and
improve efficiency. Using 80% derating, on VOUT for ringing on the switch node, the rectifier diode minimum
reverse break-down voltage is given by Equation 41.
V
V(BR)R(min) ³ OUT = 1.25 ´ VOUT = 1.25 ´ 24 V = 30 V
0.8
(41)
The diode must have reverse breakdown voltage greater than 30 V. The rectifier diode peak and average
currents are estimated by Equation 42 and Equation 43.
ID (avg ) » IOUT (m ax ) = 2 A
(42)
ID(peak ) = IL(peak ) = 6.57 A
(43)
For this design, 2-A average and 6.57-A peak is:
The power dissipation in the diode is estimated by Equation 44.
PD(max) » VF ´ IOUT(max) = 0.5 V ´ 2 A = 1W
(44)
For this design, the maximum power dissipation is estimated as 1 W. Reviewing 30-V and 40-V Schottky diodes,
the MBRS340T3 40-V 3-A diode in an SMC package is selected. This diode has a forward voltage drop of 0.48 V
at 6 A, so the conduction power dissipation is approximately 960 mW, less than half its rated power dissipation.
8.2.2.4 Output Capacitor Selection
Output capacitors must be selected to meet the required output ripple and transient specifications.
I OUT ´ D
æ 2 A ´ 0.673 ö
1
1
´
= 8´ç
= 35 mF
COUT = 8 ´
÷´
VOUT(ripple) f SW
è 500 mV ø 600 kHz
28
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ESR =
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VOUT(ripple )
7
7
500mV
´
= ´
= 95mW
8 IL(peak ) - IOUT 8 6.57 A - 2 A
(46)
A Panasonic EEEFC1V330P 35-V 33-μF, 120-mΩ bulk capacitor and 6.8-μF ceramic capacitor is selected to
provide the required capacitance and ESR at the switching frequency. The combined capacitances of 39.8 μF
and 60 mΩ are used in compensation calculations.
8.2.2.5 Input Capacitor Selection
Since a boost converter has continuous input current, the input capacitor senses only the inductor ripple current.
The input capacitor value can be calculated by Equation 47 and Equation 48.
IL(ripple )
1.02 A
C IN >
=
= 7 mF
4 ´ VIN(ripple ) ´ f SW
4 ´ 60 mV ´ 600 kHz
(47)
ESR <
VIN(ripple )
2 ´ IL(ripple )
=
60mV
= 30mW
2 ´ 1.02 A
(48)
For this design, to meet a maximum input ripple of 60 mV, a minimum 7.0-μF input capacitor with ESR less than
30 mΩ is needed. A 10-μF X7R ceramic capacitor is selected.
8.2.2.6 Current Sense and Current Limit
The maximum allowable current sense resistor value is limited by both the current limit and subharmonic stability.
These two limitations are given by Equation 49 and Equation 50.
RISNS <
RISNS <
VOCP(min)
(
1.1´ IL(peak ) + IDrive
=
)
110mV
= 14.2mW
1.1´ 6.57 A + 0.50 A
(49)
VDDMAX ´ L ´ fSW
14 V ´ 10 mH ´ 600kHz
=
= 133mW
60 ´ (VOUT + Vfd - VIN ) 60 ´ (24 V + 0.48 V - 14 V)
(50)
The current limit requires a resistor less than 14.2 mΩ, and stability requires a sense resistor less than 133 mΩ.
A 10-mΩ resistor is selected. Approximately 2-mΩ of routing resistance is added in compensation calculations.
8.2.2.7 Current Sense Filter
To remove switching noise from the current sense, an R-C filter is placed between the current sense resistor and
the ISNS pin. A resistor with a value between 1 kΩ and 5 kΩ is selected, and a capacitor value is calculated by
Equation 51.
CIFLT =
0.1´ DMIN
0.1´ 0.428
=
= 71pF
fSW ´ RIFLT 600kHz ´ 1kW
(51)
For a 1-kΩ filter resistor, 71 pF is calculated and a 100-pF capacitor is selected.
8.2.2.8 Switching MOSFET Selection
The TPS40210-Q1 device drives a ground referenced N-channel FET. The RDS(on) and gate charge are
estimated based on the desired efficiency target.
æ1 ö
æ1 ö
æ 1
ö
- 1÷ = 2.526 W
PDISS(total) » POUT ´ ç - 1÷ = VOUT ´ IOUT ´ ç - 1÷ = 24 V ´ 2 A ´ ç
è 0.95 ø
èh ø
èh ø
(52)
For a target of 95% efficiency with a 24-V input voltage at 2 A, maximum power dissipation is limited to 2.526 W.
The main power dissipating devices are the MOSFET, inductor, diode, current sense resistor and the integrated
circuit, the TPS40210-Q1 device.
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PFET < PDISS(total) - PL - PD - PRisns - VIN(max) ´ IVDD
(53)
This leaves 740 mW of power dissipation for the MOSFET. This can likely cause an SO-8 MOSFET to get too
hot, so power dissipation is limited to 500 mW. Allowing half for conduction and half for switching losses, you can
determine a target RDS(on) and QGS for the MOSFET by Equation 54 and Equation 55.
QGS <
3 ´ PFET ´ IDRIVE
3 ´ 0.50 W ´ 0.50 A
=
= 13.0nC
2 ´ VOUT ´ IOUT ´ fSW 2 ´ 24 V ´ 2 A ´ 600kHz
(54)
A target MOSFET gate-to-source charge of less than 13.0 nC is calculated to limit the switching losses to less
than 250 mW.
RDS(on ) <
PFET
=
2
2 ´ (IRMS ) ´ D
0.50 W
2 ´ 6.132 ´ 0.674
= 9.8mW
(55)
A target MOSFET RDS(on) of 9.8 mΩ is calculated to limit the conduction losses to less than 250 mW. Reviewing
30-V and 40-V MOSFETs, an Si4386DY 9-mΩ MOSFET is selected. A gate resistor was added per Equation 29.
The maximum gate charge at Vgs = 8 V for the Si4386DY is 33.2 nC, this implies RG = 3.3 Ω.
8.2.2.9 Feedback Divider Resistors
The primary feedback divider resistor (RFB) from VOUT to FB should be selected between 10 kΩ and 100 kΩ to
maintain a balance between power dissipation and noise sensitivity. For a 24-V output, a high feedback
resistance is desirable to limit power dissipation so RFB = 51.1 kΩ is selected.
RBIAS =
VFB ´ RFB
0.700 V ´ 51.1kW
=
= 1.53kW
VOUT - VFB
24 V - 0.700 V
(56)
RBIAS = 1.50 kΩ is selected.
8.2.2.10 Error Amplifier Compensation
While current mode control typically requires only Type II compensation, it is desirable to layout for Type III
compensation to increase flexibility during design and development.
Current mode control boost converters have higher gain with higher output impedance, so it is necessary to
calculate the control loop gain at the maximum output impedance, estimated by Equation 57.
ROUT(max ) =
VOUT
IOUT(min )
=
24 V
= 240 W
0.1A
(57)
The transconductance of the TPS40210-Q1 current-mode control can be estimated by Equation 58.
0.13 ´ L ´
gm =
f SW
0.13 ´ 10 mH ´
R OUT
=
600 kHz
240 W
2
2
(R ISNS ) ´ (120 ´ R ISNS + L ´ fSW ) (12 mW ) ´ (120 ´ 12 mW + 10 mH ´ 600 kHz )
= 19.1 S
(58)
The maximum output impedance, ZOUT, can be estimated by Equation 59.
(1+ (2p ´ f ´ R
ESR
ZOUT (f ) = ROUT ´
(
2
)
) )´ (2p ´ f ´ C
2
´ COUT )
1 + (ROUT ) + 2 ´ ROUT ´ RESR + (RESR
2
OUT
)2
(59)
(1+ (2p ´ 20kHz ´ 60mW ´ 39.8 mF) )
1 + ((240 W ) + 2 ´ 240 W ´ 60mW + (60mW ) )´ (2p ´ 20kHz ´ 39.8 mF )
2
ZOUT (fCO ) = 240 W ´
30
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2
2
2
= 0.146 W
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(60)
The modulator gain at the desired cross-over can be estimated by Equation 61.
( ) = 19.1 S ´ 0.146 W = 2.80
K CO = gm ´ ZOUT f CO
(61)
The feedback compensation network needs to be designed to provide an inverse gain at the cross-over
frequency for unit loop gain. This sets the compensation mid-band gain at a value calculated in Equation 62.
K COMP =
1
K CO
=
1
= 0.356
2.80
(62)
To set the mid-band gain of the error amplifier to KCOMP, use Equation 63.
R4 = R7 ´ K COMP =
R7
51.1kW
=
= 18.2kW
K CO
2.80
(63)
R4 = 18.7 kΩ selected.
Place the zero at one 10th of the desired cross-over frequency.
C2 =
10
10
=
= 2837pF
2p ´ fL ´ R4 2p ´ 30kHz ´ 18.7kW
(64)
C2 = 2200 pF selected.
Place a high-frequency pole at about five times the desired cross-over frequency and less than one-half the unity
gain bandwidth of the error amplifier:
C4 »
C4 >
1
1
=
= 56.74pF
10p ´ fL ´ R4 10p ´ 30kHz ´ 18.7kW
(65)
1
1
=
= 11.35pF
p ´ GBW ´ R4 p ´ 1.5MHz ´ 18.7kW
(66)
C4 = 47 pF selected.
8.2.2.11 R-C Oscillator
The R-C oscillator calculation as shown in Equation 31 substitutes 100 for CT and 600 for fSW. For a 600-kHz
switching frequency, a 100-pF capacitor is selected and a 262-kΩ resistor is calculated (261 kΩ selected).
8.2.2.12 Soft-Start Capacitor
Because VDD > 8 V, the soft-start capacitor is selected by using Equation 67 to calculate the value.
CSS = 20 ´ TSS ´ 10-6
(67)
For tSS = 12 ms, CSS = 240 nF, a 220-nF capacitor selected.
8.2.2.13 Regulator Bypass
A regulator bypass capacitor of 1.0-μF is selected per the recommendation.
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8.2.3 Application Curves
80
180
60
VIN = 8 V
VOUT = 24 V
IOUT = 2 A
135
90
20
45
Gain – dB
40
0
0
Gain
-20
-45
-40
-90
-60
-135
-80
100
GDRV
(5 V/ div)
Phase – °
Phase
FET Vds
(20 V/ div)
-180
1M
1000
10 k
100 k
fSW – Frequency – Hz
T – Time – 400 ns
Figure 32. FET Vds and Vgs Voltages vs Time
Figure 31. Gain and Phase vs Frequency
6
100
96
5
h – Efficiency – %
94
92
VIN = 12 V
90
88
VIN (V)
14
12
8
VIN = 14V
PLOSS – Power Loss – W
VIN (V)
14
12
8
98
VIN = 8 V
86
VIN = 8 V
4
VIN = 12 V
3
2
VIN = 14 V
84
1
82
80
0
0
0.5
1.0
1.5
2.0
ILOAD – Load Current – A
0
2.5
Figure 33. Efficiency vs Load Current
0.5
1.0
1.5
2.0
ILOAD – Load Current – A
2.5
Figure 34. Power Loss vs Load Current
24.820
VOUT – Output Voltage – V
24.772
24.724
VIN (V)
14
12
8
VIN = 8 V
24.676
24.628
24.580
VIN = 14 V
24.532
VIN = 12 V
24.484
24.436
24.388
24.340
0
0.5
1.0
1.5
2.0
ILOAD – Load Current – A
2.5
Figure 35. Output Voltage vs Load Current
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9 Power Supply Recommendations
All power (high-current) traces should be as thick and short as possible. The inductor and output capacitors
should be as close to each other as possible. This reduces EMI radiated by the power traces due to high
switching currents. In a two-sided PCB, TI recommends having ground planes on both sides of the PCB to help
reduce noise and ground loop errors. The ground connection for the input and output capacitors and IC ground
should connect to this ground plane. In a multi-layer PCB, the ground plane separates the power plane (where
high switching currents and components are) from the signal plane (where the feedback trace and components
are) for improved performance. Also, arrange the components such that the switching-current loops curl in the
same direction. Place the high-current components such that during conduction the current path is in the same
direction. This prevents magnetic field reversal caused by the traces between the two half-cycles, and helps
reduce radiated EMI. Route the feedback trace such that there is minimum interaction with any noise sources
associated with the switching components. The recommended practice is to ensure the inductor is placed away
from the feedback trace to prevent creating an EMI noise source. Do not locate the sensitive components and
their traces near any switching nodes or high-current traces.
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10 Layout
10.1 Layout Guidelines
•
•
•
•
•
•
•
34
The path formed from the input capacitor to the inductor and the switch node must have short trace length.
The same applies for the trace from the inductor to Schottky diode to the output capacitor.
Use a ceramic input capacitor located next to the VDD pin with a short return path to the "power" GND
copper. Locate input ceramic filter capacitors in close proximity to the VIN terminal. TI recommends surfacemount capacitors to minimize lead length and reduce noise coupling.
Use a low-EMI inductor with a ferrite-type shielded core. One can use other types of inductors; however, they
must have low-EMI characteristics and be located away from the low-power traces and components in the
circuit.
The VBP capacitor should be close to the BP pin with a short return path to the "power" GND copper.
All other analog components should be kept close to the IC such as those connected to RC, SS, COMP, FB,
and ISNS. It is recommend to isolate this ground return used for these components to create a "quiet" ground
minimizing any noise as shown in Figure 36.
Use foot print and vias pattern for the TPS40210 device as recommended towards the end of the data sheet.
The resistor divider for sensing the output voltage connects between the positive pin of the output capacitor
and the GND pin (IC signal ground).
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10.2 Layout Example
Switching Components (L1, D1, C12, C13, Q1, R11)
Minimize this loop area to reduce ringing
Feedback Components
Away from Power Path
(to avoid noise coupling)
VDD Decoupling Capacitor and
Current Sense Components
Placed Nearby
Figure 36. TPS40210 Top Layer
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Layout Example (continued)
Multiple vias connect the input and output to ground plane
Large ground plane to reduce noise
and ground-loop errors
Figure 37. TSP40210 Bottom Layer
36
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11 Device and Documentation Support
11.1 Device Support
11.1.1 Third-Party Products Disclaimer
TI'S PUBLICATION OF INFORMATION REGARDING THIRD-PARTY PRODUCTS OR SERVICES DOES NOT
CONSTITUTE AN ENDORSEMENT REGARDING THE SUITABILITY OF SUCH PRODUCTS OR SERVICES
OR A WARRANTY, REPRESENTATION OR ENDORSEMENT OF SUCH PRODUCTS OR SERVICES, EITHER
ALONE OR IN COMBINATION WITH ANY TI PRODUCT OR SERVICE.
11.2 Documentation Support
11.2.1 Related Documentation
These references may be found on the web at www.power.ti.com under Technical Documents. Many design
tools and links to additional references, may also be found at www.power.ti.com
1. Design and Application Guide for High Speed MOSFET Gate Drive Circuits, SEM 1400, 2001 Seminar
Series
2. Designing Stable Control Loops, SEM 1400, 2001 Seminar Series
3. PowerPAD™ Thermally Enhanced Package and PowerPAD™ Made Easy contain additional information on
PowerPAD packages.
4. QFN/SON PCB Attachment contains information on attaching these package types to a PCB.
11.3 Related Links
The following table lists quick access links. Categories include technical documents, support and community
resources, tools and software, and quick access to sample or buy.
Table 2. Related Links
PARTS
PRODUCT FOLDER
SAMPLE & BUY
TECHNICAL
DOCUMENTS
TOOLS &
SOFTWARE
SUPPORT &
COMMUNITY
TPS40210-Q1
Click here
Click here
Click here
Click here
Click here
TPS40211-Q1
Click here
Click here
Click here
Click here
Click here
11.4 Trademarks
PowerPAD is a trademark of Texas Instruments.
All other trademarks are the property of their respective owners.
11.5 Electrostatic Discharge Caution
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
11.6 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
Copyright © 2008–2020, Texas Instruments Incorporated
Product Folder Links: TPS40210-Q1 TPS40211-Q1
Submit Documentation Feedback
37
TPS40210-Q1, TPS40211-Q1
SLVS861F – AUGUST 2008 – REVISED JUNE 2020
www.ti.com
12 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the mostcurrent data available for the designated devices. This data is subject to change without notice and without
revision of this document. For browser-based versions of this data sheet, see the left-hand navigation pane.
38
Submit Documentation Feedback
Copyright © 2008–2020, Texas Instruments Incorporated
Product Folder Links: TPS40210-Q1 TPS40211-Q1
PACKAGE OPTION ADDENDUM
www.ti.com
23-Jul-2022
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
Samples
(4/5)
(6)
TPS40210QDGQRQ1
ACTIVE
HVSSOP
DGQ
10
2500
RoHS & Green NIPDAU | NIPDAUAG
Level-1-260C-UNLIM
-40 to 125
4210Q
Samples
TPS40211QDGQRQ1
ACTIVE
HVSSOP
DGQ
10
2500
RoHS & Green
Level-1-260C-UNLIM
-40 to 125
4211Q
Samples
NIPDAU
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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