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TPS43340-Q1
SLVSB16E – NOVEMBER 2011 – REVISED DECEMBER 2015
TPS43340-Q1 Low-IQ, 30-µA, High-VIN Quad-Output Power Supply
1 Features
2 Applications
•
•
•
1
•
•
•
•
•
•
•
•
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Qualified for Automotive Applications
AEC-Q100 Test Guidance With the Following
Results:
– Device Temperature Grade 1: –40°C to 125°C
Ambient Operating Temperature
– Device HBM ESD Classification Level H1C
– Device CDM ESD Classification Level C3B
Input Range up to 40 V, (Transients up to 60 V)
Input Voltage Range: 4 V to 40 V
– Transients up to 60 V
Dual-Output Synchronous Buck Controller
– Peak Gate Drive Current 0.6 A
– Automatic Low-Power Mode Operation
– Low-Power-Mode IQ : 30 µA (One Buck On),
35 µA (Two Bucks On)
Low Shutdown Current, Ish = 5 μA Typical
Single Synchronous Buck Regulator Converter
BUCK3
– Max Output Current 2 A
Linear Regulator LREG1
Separate Enable Inputs (EN1, EN2, EN3, EN4)
Internal Oscillator, Programmable via External
Resistor, 150 kHz to 600 kHz for Switching
Frequency fSW_BUCK1,2,3
Integrated PLL, External Synchronization
Frequency: 150 kHz to 600 kHz
Switch-Mode Regulators Operate With 180°
Phase-Shift
Reset Output for All Output Rails
Supply and Overvoltage Detection and Shutdown
Thermally Enhanced PowerPAD™ Package
– 48-Pin HTQFP (PHP)
•
•
Automotive Infotainment, Head Unit, Navigation,
Audio and Clusters
Advanced Driver Assistance System (ADAS)
Automotive and Industrial Multi-Rail DC Power
Distribution Systems
3 Description
The TPS43340-Q1 is a quad rail power supply
featuring two synchronous Buck Controller with 0.6-A
gate drive, one synchronous 2-A Buck Converter and
a 300-mA LDO with low quiescent current. The
device is designed to power the entire system
including MCU and DSP straight from the car battery
respectively input voltages up to 40 V. The device
features integrated short-circuit and overcurrent
protection on the gate-drive outputs for the buck
regulator controllers and independent currentfoldback control for each buck regulator supply during
regulator output short to ground. Each output supply
incorporates a soft start to ensure that on initial power
up these regulated outputs are not in current limit.
Implementation of reset delay on power up allows the
outputs of Buck1, Buck2, Buck3 and the linear
regulator to get to stable regulation. An external
capacitor sets the delay to a maximum range of 300
ms. Each power-supply output has adjustable output
voltage based on the external resistor-network
settings. The device has sequencing control during
power up and power down of the output rails, based
on the enable-and-disable control or soft start.
Device Information(1)
PART NUMBER
TPS43340-Q1
PACKAGE
HTQFP (48)
BODY SIZE (NOM)
7.00 mm × 7.00 mm
(1) For all available packages, see the orderable addendum at
the end of the data sheet.
Simplified Schematic
EN4
VLREG1
VOUT1
VOUT2
SYNC
EN3
EN2
EN1
TPS43340-Q1
VOUT3
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
TPS43340-Q1
SLVSB16E – NOVEMBER 2011 – REVISED DECEMBER 2015
www.ti.com
Table of Contents
1
2
3
4
5
6
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Pin Configuration and Functions .........................
Specifications.........................................................
6.1
6.2
6.3
6.4
6.5
6.6
7
1
1
1
2
4
7
Absolute Maximum Ratings ...................................... 7
ESD Ratings.............................................................. 8
Recommended Operating Conditions....................... 8
Thermal Information .................................................. 9
Electrical Characteristics........................................... 9
Typical Characteristics ............................................ 13
Detailed Description ............................................ 16
7.1
7.2
7.3
7.4
Overview .................................................................
Functional Block Diagram .......................................
Feature Description.................................................
Device Functional Modes........................................
16
16
16
20
8
Application and Implementation ........................ 26
8.1 Application Information............................................ 26
8.2 Typical Application .................................................. 26
9 Power Supply Recommendations...................... 31
10 Layout................................................................... 31
10.1
10.2
10.3
10.4
Layout Guidelines .................................................
Layout Example ....................................................
Power Dissipation .................................................
Thermal Considerations ........................................
31
32
32
34
11 Device and Documentation Support ................. 36
11.1
11.2
11.3
11.4
11.5
11.6
Device Support......................................................
Documentation Support ........................................
Community Resources..........................................
Trademarks ...........................................................
Electrostatic Discharge Caution ............................
Glossary ................................................................
36
36
36
36
36
36
12 Mechanical, Packaging, and Orderable
Information ........................................................... 36
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision D (July 2015) to Revision E
•
Update the HBM ESD ratings and added the MM ESD rating back in the ESD Ratings table ............................................. 8
Changes from Revision C (January 2013) to Revision D
•
Page
Page
Added ESD Ratings table, Feature Description section, Device Functional Modes, Application and Implementation
section, Power Supply Recommendations section, Layout section, Device and Documentation Support section, and
Mechanical, Packaging, and Orderable Information section ................................................................................................. 1
Changes from Revision B (April 2012) to Revision C
Page
•
Added bullets to top of Features list ....................................................................................................................................... 1
•
Appended missing "-Q1" to part number ................................................................................................................................ 1
•
Revised first-page schematic ................................................................................................................................................. 1
•
Added a sentence to the EXTSUP pin description................................................................................................................. 5
•
Changed "converter" to "controller" for pin SS2 ..................................................................................................................... 6
•
Deleted thermal characteristics from Recommended Operating Characteistics table ........................................................... 8
•
Added the Thermal Information table ..................................................................................................................................... 9
•
Multiple changes throughout Electrical Characteristics table ................................................................................................. 9
•
Appended missing "-Q1" to part number .............................................................................................................................. 16
•
Changed the recommended capacitor value........................................................................................................................ 17
•
Added a sentence to the second paragraph of the Gate-Driver Supply section .................................................................. 17
•
Added new sentence to Gate-Driver Supply section............................................................................................................ 17
•
Replaced the two paragraphs following Figure 15 with three new paragraphs ................................................................... 17
•
Modified Equation 2.............................................................................................................................................................. 20
•
Changed fSW-Trans-delay to tSW-Trans-delay ..................................................................................................................................... 20
•
Modified Equation 3.............................................................................................................................................................. 20
2
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Product Folder Links: TPS43340-Q1
TPS43340-Q1
www.ti.com
SLVSB16E – NOVEMBER 2011 – REVISED DECEMBER 2015
•
Renamed VBUCKx to VOUTx ................................................................................................................................................. 20
•
Added (Farads) to Equation 4 .............................................................................................................................................. 20
•
Changed "resistor" to "VOUT3" ............................................................................................................................................... 20
•
Revised Figure 18 ................................................................................................................................................................ 23
•
Changed "VBAT" to "VIN"...................................................................................................................................................... 24
•
Modified power-dissipation equations .................................................................................................................................. 27
•
Buck2 Component Selection, modified Equation 22, Equation 25, Equation 26, and Equation 29, Equation 31................ 28
•
Added Equation 27, Equation 28, Equation 30, and Equation 32 ........................................................................................ 28
•
Buck3 Component Selection, modified Equation 34, Equation 36, Equation 37, and Equation 38, .................................... 29
•
Added Equation 39, Equation 40, Equation 41, and Equation 43 ........................................................................................ 29
•
Modified several equations in Summary of Equations table................................................................................................. 33
•
Modified several equations in Summary of Equations table................................................................................................. 34
Changes from Revision A (January 2012) to Revision B
Page
•
Changed Feedback input to Supply sense input in Abs Max Ratings table........................................................................... 7
•
Inserted Input voltage for Buck 2 information in the Recommended Operating Conditions table.......................................... 8
•
Added VIN2SENSE = 4 V to 40 V in Electrical Characteristics table header. ....................................................................... 9
•
Changed Iq_LPM to Iq, changed LPM quiescent current to Quiescent current, and changed the conditions for EN in
the Electrical Characteristics table. ........................................................................................................................................ 9
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3
TPS43340-Q1
SLVSB16E – NOVEMBER 2011 – REVISED DECEMBER 2015
www.ti.com
5 Pin Configuration and Functions
BOOT1
EN4
LREG1
VSENSE4
RST4
VIN2SENSE
VLR1
VIN
EXTSUP
GPULL
VREG
BOOT2
48
47
46
45
44
43
42
41
40
39
38
37
PHP Package
48-Pin HTQFP With Thermal Pad
Top View
GU1
1
36
GU2
PH1
2
35
PH2
GL1
3
34
GL2
PGND1
4
33
PGND2
S2
5
32
S4
S1
6
31
S3
Thermal
Pad
24
RT
Rdelay
25
23
12
SYNC
PGND3
22
GND
EN1
26
21
11
EN2
VSUP
20
SS2
EN3
27
19
10
SLEW
SS1
18
RST2
COMP3
28
17
9
VSENSE3
RST1
16
COMP2
RST3
29
15
8
SS3
COMP1
14
VSENSE2
BOOT3
30
13
7
PH3
VSENSE1
Pin Functions
PIN
NAME
NO.
BOOT1
BOOT2
48
37
I/O
DESCRIPTION
I
A capacitor on this pin acts as the voltage supply for the high-side N-channel MOSFET gate-drive
circuitry in buck converter Buck1. When the buck is in a dropout condition, the device automatically
reduces the duty cycle of the high-side MOSFET to approximately 95% on every fourth cycle to
allow the capacitor to recharge.
I
A capacitor on this pin acts as the voltage supply for the high-side N-channel MOSFET gate-drive
circuitry in buck converter Buck2. When the buck is in a dropout condition, the device automatically
reduces the duty cycle of the high-side MOSFET to approximately 95% on every fourth cycle to
allow the capacitor to recharge.
BOOT3
14
I
A capacitor between BOOT3 and PH3 acts as the voltage supply for the high-side N-channel
MOSFET gate-drive circuitry in buck converter Buck3. When the buck is in a dropout condition, the
device automatically reduces the duty cycle of the high-side MOSFET to approximately 95% on
every fourth cycle to allow the capacitor to recharge.
COMP1
8
O
Error amplifier output of Buck1 and compensation node for voltage-loop stability. The voltage at this
node sets the target for the peak current through the respective inductor. Clamping this voltage on
the upper and lower ends provides current-limit protection for the external MOSFETs.
COMP2
29
O
Error amplifier output of Buck2 and compensation node for voltage-loop stability. The voltage at this
node sets the target for the peak current through the respective inductor. Clamping this voltage on
the upper and lower ends provides current-limit protection for the external MOSFETs.
4
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SLVSB16E – NOVEMBER 2011 – REVISED DECEMBER 2015
Pin Functions (continued)
PIN
NAME
NO.
I/O
DESCRIPTION
COMP3
18
O
Error amplifier output of Buck3 and compensation node for voltage loop stability. The voltage at this
node sets the target for the peak current through the respective inductor.
EN1
22
I
Enable input for Buck1. This input has an internal pullup with approximately 0.5 µA of current.
EN2
21
I
Enable input for Buck2. This input has an internal pullup with approximately 0.5 µA of current.
EN3
20
I
Enable input for Buck3. This input has an internal pullup with approximately 0.5 µA of current.
EN4
47
I
Enable input for LREG1 (active-high with an internal pullup current source). An input voltage higher
than VIH enables the regulator, whereas an input voltage lower than VIL disables the regulator. This
input has an internal pullup with approximately 0.5 µA of current.
EXTSUP
40
I
One can use EXTSUP to supply the VREG regulator from one of theTPS43340 buck regulator rails
to reduce power dissipation in cases where there is an expectation of high VIN. When EXTSUP is
open or lower than 4.6 V, VIN powers the regulator. If EXTSUP is unused, leave the pin open
without a capacitor installed.
GL1
3
O
External low-side N-channel MOSFET gate drive for buck regulator Buck1. The output provides high
peak currents to drive capacitive loads. VREG provides the voltage swing on this pin.
GL2
34
O
External low-side N-channel MOSFET for buck regulator This output can drive Buck2. The output
provides high peak currents to drive capacitive loads. VREG provides the voltage swing on this pin.
GND
26
O
Analog ground reference
GPULL
39
O
Gate-driver output to implement the reverse-battery protection by an external PMOS. See the
Application Information section for more details.
GU1
1
O
External high-side N-channel MOSFET gate drive for buck regulator Buck1. The output provides
high peak currents to drive capacitive loads. The gate-drive reference is a floating-ground reference
provided by PH1 and has a voltage swing provided by BOOT1.
GU2
36
O
This output can drive an external high-side N-channel MOSFET for buck regulator Buck2. The
output provides high peak currents to drive capacitive loads. The gate-drive reference is a floatingground reference provided by PH2 and has a voltage swing provided by BOOT2.
LREG1
46
O
Linear regulator output. Decouple with a low-ESR ceramic output capacitor in the range of 1 µF to
47 µF connected from this terminal to ground.
PGND1
4
O
Power ground connection for the GL1 driver. Connect to the source of the low-side N-channel
MOSFET of Buck1.
PGND2
33
O
Power ground connection to the source of the low-side N-channel MOSFETs of Buck2
PGND3
12
O
Buck3 power ground
PH1
2
O
Switching terminal of buck regulator Buck1, providing a floating ground reference for the high-side
MOSFET gate-driver circuitry and used to sense current reversal in the inductor when
discontinuous-mode operation is desirable.
PH2
35
O
Switching terminal of buck regulator Buck2, providing a floating ground reference for the high-side
MOSFET gate-driver circuitry and used to sense current reversal in the inductor when
discontinuous-mode operation is desirable.
PH3
13
O
Switching terminal of buck converter Buck3. Also provides a floating ground reference for the highside MOSFET gate-driver circuitry
Rdelay
24
O
The capacitor at the Rdelay pin sets the power-good delay interval used to de-glitch the outputs of
the power-good comparators. Leaving this pin open sets the power-good delay to an internal default
value of 20 μs, typical.
RST1
9
O
Open-drain power-good output for Buck1, with a 50-kΩ pullup resistor to S2. An internal power-good
comparator monitors the voltage at the feedback pin and pulls this output low when the output
voltage falls by RSTxth1 of the set value.
RST2
28
O
Open-drain power-good output for Buck2 with a 50 kΩ pullup resistor to S4. An internal power-good
comparator monitors the voltage at the feedback pin and pulls this output low when the output
voltage falls by RSTxth1 of the set value.
RST3
16
O
Open-drain power-good output for Buck3. An internal power-good comparator monitors the voltage
at the feedback pin and pulls this output low when the output voltage falls by RSTxth1 of the set
value.
RST4
44
O
Open-drain power-good indicator pin for LREG1, with a 50-kΩ pullup resistor to LREG1. An internal
power-good comparator monitors the voltage at the feedback pin and pulls this output low when the
output voltage falls by RSTxth1 of the set value.
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Pin Functions (continued)
PIN
NAME
NO.
I/O
DESCRIPTION
Connecting a resistor to analog ground on this pin sets the operating switching frequency of the
buck controllers and converter. Shorting this pin to ground or leaving it open defaults operation to
400 kHz for the buck controllers and the converter.
RT
25
O
S1
6
I
S2
5
I
S3
31
I
S4
32
I
SLEW
19
I
Slew rate (dV/dt) selector of the internal high-side switching MOSFET for Buck3. For details, see
the Application Information section.
O
Soft-start or tracking input for buck controller Buck1. The buck controller regulates the VSENSE1
voltage to the lower of 0.8 V or the SS1 pin voltage. An internal pullup current source of 1 μA is
present at the pin, and use of an appropriate capacitor connected here can set the soft-start ramp
duration. Alternatively, use of a resistor divider from another supply can provide a tracking input to
this pin.
SS1
10
High-impedance differential-voltage inputs from the current-sense element (sense resistor or
inductor DCR) for the buck controller. For details, see the Functional Description section.
SS2
27
O
Soft-start or tracking input for buck controller Buck2. The buck controller regulates the VSENSE2
voltage to the lower of 0.8 V or the SS2 pin voltage. An internal pullup current source of 1 μA is
present at the pin, and use of an appropriate capacitor connected here can set the soft-start ramp
interval. Alternatively, use of a resistor divider from another supply can provide a tracking input to
this pin.
SS3
15
O
Soft-start or tracking input for buck converter Buck3. The buck converter regulates the VSENSE3
voltage to the lower of 0.8 V or the SS3 pin voltage. An internal pullup current source of 1 μA is
present at the pin, and an appropriate capacitor connected here can set the soft-start ramp duration.
Alternatively, use of a resistor divider from another supply can provide a tracking input to this pin.
SYNC
23
I
PLL synchronization, low-power mode-control pin. If an external clock is present on this pin, the
device detects it and the internal PLL locks on to the external clock. This overrides the internal
oscillator frequency. The device can synchronize to frequencies from 150 kHz to 600 kHz. For
details, see the Application Information section.
VIN
41
I
Main Input pin. This is the buck controller and buck converter input pin. Additionally, it powers the
internal control circuits of the device. Connect a bypass capacitor to filter noise between this pin and
signal ground.
VIN2SENSE
43
I
Supply-voltage sense input for the current mode of Buck2. Connect to the drain of the high-sideFET of Buck2. Cascading Buck1 as the supply for the Buck2 configuration does not support LPM on
Buck2.
VLR1
42
I
The VLR1 terminal is the input voltage source for the linear regulator supply. This pin requires an
input capacitor to ground to filter any noise present on the line.
VREG
38
O
This pin requires an external capacitor to provide a regulated supply for the gate drivers of the buck
controllers and converter. The regulator can obtain power either from VIN or EXTSUP. This pin has
current limit-protection; do not use it to drive any other loads.
VSENSE1
7
I
Feedback voltage pin for Buck1. For details, see the Application Information section.
VSENSE2
30
I
Feedback voltage pin for Buck2. The buck controller regulates the feedback voltage to the internal
reference of 0.8 V. A suitable resistor divider network between the buck output and the feedback pin
sets the desired output voltage.
VSENSE3
17
I
Feedback voltage pin for Buck3. The buck controller regulates the feedback voltage to the internal
reference of 0.8 V. A suitable resistor divider network between the buck output and the feedback pin
sets the desired output voltage.
VSENSE4
45
I
Feedback voltage pin for linear regulator LREG1. LREG1 regulates the feedback voltage to the
internal reference. A suitable resistor divider network between the LDO output and the feedback pin
sets the desired output voltage. See the LREG1 parameters and the Application Information section.
VSUP
11
I
Power supply for the Buck3 regulator. Provide good decoupling to PGND3 with a ceramic capacitor
close to the pins.
6
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SLVSB16E – NOVEMBER 2011 – REVISED DECEMBER 2015
6 Specifications
6.1 Absolute Maximum Ratings
over operating free-air temperature range (unless otherwise noted) (1)
Supply inputs
MIN
MAX
UNIT
Input voltage
VIN
–0.3
60
V
Enable inputs
EN1, EN2
–0.3
60
V
Bootstrap supplies
BOOT1, BOOT2
–0.3
68
V
Bootstrap supplies
BOOT1-PH1, BOOT2-PH2,
BOOT3-PH3
–0.3
8.8
V
PH1, PH2
–1
60
V
PH1, PH2 (for 100 ns)
–2
Feedback inputs
VSENSE1, VSENSE2
–0.3
13
V
Error-amplifier outputs
COMP1, COMP2
–0.3
13
V
Peak output currents from external
MOSFET driver
GU1,GU2, GL1,GL2
1
A
Phase inputs
Buck controller
Buck1 and Buck2
GL1-PGND1,GL2-PGND2
–0.3
8.8
GU1-PH1,GU2-PH2
–0.3
8.8
Current-sense voltage
S1, S2, S3, S4
–0.3
13
V
Absolute differential voltage
|S1 – S2|, |S3 – S4|
2
V
Soft start
SS1, SS2
–0.3
13
V
Power-good outputs
RST1, RST2
–0.3
13
V
Switching-frequency oscillator
RT
–0.3
13
V
External input clock
SYNC
–0.3
13
V
External input supply for gate drive
EXTSUP
–0.3
13
V
Input supply
VSUP
–0.3
13
Slew-rate setting
SLEW
–0.3
13
Enable input
EN3
–0.3
13
Bootstrap supply
BOOT3
–1
20
PH3
–1
13
PH3 (for 100 ns)
–2
External MOSFET driver
Buck converter
Buck3
Linear regulator
LREG1
GPULL, Rdelay, VREG,
VIN2SENSE
Temperature
(1)
V
Phase inputs
Feedback input
VSENSE3
–0.3
13
Soft start
SS3
–0.3
13
Power-good output
RST3
–0.3
13
Error-amplifier output
COMP3
–0.3
13
Input voltage
VLR1
–0.3
60
Output voltage
LREG1
–0.3
7
Enable input
EN4
–0.3
60
Power-good output
RST4
–0.3
8.8
Feedback inputs
VSENSE4
–0.3
13
PMOS driver
GPULL
–0.3
Zener clamp current
GPULL
Internal regulator
VREG
Reset delay
Supply sense input
V
V
V
60
V
0.2
mA
–0.3
8.8
V
Rdelay
–0.3
8.8
V
VIN2SENSE
–0.3
60
V
Junction temperature: TJ
–40
150
Operating temperature: TA
–40
125
Storage temperature: TSTG
–55
165
°C
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended
Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
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6.2 ESD Ratings
VALUE
V(ESD)
Electrostatic
discharge
Human body model (HBM),
per AEC Q100-002 (1)
All pins except VLR1
±2000
VLR1
±1000
Charged device model (CDM),
per AEC Q100-011
Corner pins (1, 12, 13, 24, 25, 36, 37, and 48)
±750
Other pins
±500
All pins except RSTx
±200
RSTx
±100
Machine model (MM)
(1)
UNIT
V
AEC Q100-002 indicates HBM stressing is done in accordance with the ANSI/ESDA/JEDEC JS-001 specification.
6.3 Recommended Operating Conditions
over operating free-air temperature range (unless otherwise noted)
MIN
Supply inputs
Buck converter
Buck3
Linear regulator
LREG1
PMOS driver
Temperature ratings
8
MAX UNIT
VIN
4
40
V
Input voltage for Buck 2
VIN2SENSE
4
40
V
Enable inputs
EN1, EN2
0
40
V
Bootstrap inputs
BOOT1, BOOT2
4
48
V
–0.6
40
V
PH1, PH2
Phase inputs
Buck controller
Buck1 and Buck2
NOM
Input voltage
PH1, PH2 (for 50 ns)
–2
Feedback inputs
VSENSE1, VSENSE2
0
6
V
Error-amplifier outputs
COMP1, COMP2
0
6
V
Peak output currents from external
MOSFET driver
GU1,GU2, GL1,GL2
0.75
A
Current-sense voltage
S1, S2, S3, S4
0
11
V
Soft start
SS1, SS2
0
6
V
Power-good outputs
RST1, RST2
0
11
V
Switching-frequency setting
RT
0
1.2
V
External input clock
SYNC
0
9
V
External input supply for gate drive
EXTSUP
0
9
V
Input supply
VSUP
4
10
Slew-rate setting
SLEW
0
VREG
Enable input
EN3
0
6
Boot inputs
BOOT3
0
18
PH3
–1
11
PH3 (for 50 ns)
–2
Phase inputs
V
Feedback input
VSENSE3
0
6
Soft start
SS3
0
6
Power-good output
RST3
0
11
Error-amplifier output
COMP3
0
6
Input voltage
VLR1
4
40
Output voltage
LREG1
0.8
5.25
Enable input
EN4
0
40
Power-good output
RST4
0
5.25
Feedback inputs
VSENSE4
0
6
PMOS driver
GPULL
4
40
Internal regulator
VREG
0
6
–40
125
Operating temperature, TA
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V
V
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6.4 Thermal Information
TPS43340-Q1
THERMAL METRIC (1)
PHP (HTQFP)
UNIT
48 PINS
RθJA
Junction-to-ambient thermal resistance
26.3
°C/W
RθJC(top)
Junction-to-case (top) thermal resistance
12.2
°C/W
RθJB
Junction-to-board thermal resistance
7.2
°C/W
ψJT
Junction-to-top characterization parameter
0.2
°C/W
ψJB
Junction-to-board characterization parameter
7.1
°C/W
RθJC(bot)
Junction-to-case (bottom) thermal resistance
0.5
°C/W
(1)
For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report (SPRA953).
6.5 Electrical Characteristics
VIN = VLR1 = 8 V to 18 V, VSUP = 4 V to 10 V, VIN2SENSE = 4 V to 40 V, TJ = –40°C to 150°C (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
INPUT SUPPLY
Input voltage required for device on
initial start-up
VIN
6.5
Operating range after initial start-up
VIN
UV
Undervoltage lockout
4
VIN falling. After a reset, initial start-up conditions may
apply. (1)
3.5
VIN rising. After a reset, initial start-up conditions may
apply. (1)
Device operating range for linear
regulator
VLR1
Quiescent current
TA = 125°C
TA = 25°C
IVIN
3.6
3.8
V
3.8
4
V
40
V
EN1 = 1, LPM; EN2,3,4 = 0
30
40
EN2 = 1, LPM; EN1,3,4 = 0
30
40
EN4 = 1, LPM; EN1,2,3 = 0
48
60
EN1,2 = 1, LPM; EN3,4 = 0
35
45
4
4.5
EN1 = 1, LPM; EN2,3,4 = 0
40
50
EN2 = 1, LPM; EN1,3,4 = 0
40
50
EN4 = 1, LPM; EN1,2,3 = 0
52
60
EN1,2 = 1, LPM; EN3,4 = 0
40
45
EN3,4 = 1, EN1,2 = 0
Quiescent current
TA = 125°C
V
V
4
TA = 25°C
IQ
40
µA
mA
µA
EN3,4 = 1, EN1,2 = 0
5
mA
VIN = 13 V, Buck1: CCM, Buck2: off,
or
VIN = 13 V, Buck2: CCM, Buck1: off,
or
VIN = 13 V, Buck1 and Buck2: CCM
5
mA
Normal operation, SYNC = 5 V
5
VIN = 13 V, Buck1: CCM, Buck2: off
5
VIN = 13 V, Buck2: CCM, Buck1: off
5
VIN = 13 V, Buck1, 2: CCM
7
5
mA
IVIN-SD
Shutdown current at TA = 25°C
EN1,2,3,4 = 0: off, VIN = VLR1 = 13 V
10
µA
IVIN-SD
Shutdown current at TA = 125°C
EN1,2,3,4 = 0: off, VIN = VLR1 = 13 V
20
µA
IVLRI-SD
Shutdown current at TA = 125°C
EN1,2,3,4 = 0: off, VIN = VLR1 = 13 V
5
µA
V
INTERNAL SUPPLY VREG
VREG
VREG-EXTSUP
VEXTSUP-VREG
(1)
Internal regulated supply
VIN = 8 V to 18 V, EXTSUP = 0 V, SYNC = High
Load regulation
EXTSUP = 0 V, SYNC = High IVREG = 0 mA to 100 mA
5.5.
Internal regulated supply
EXTSUP = 8.5 V
Load regulation
EXTSUP = 8.5 V to 13 V, IVREG = 0 mA to 125 mA,
SYNC = High
EXTSUP switch-over voltage
IVREG = 0 mA to 100 mA, EXTSUP ramping positive
7.2.
4.4
5.8
6.1
0.2%
1%
7.5
7.8
0.2%
1%
4.6
4.8
V
V
If VBAT and VREG remain adequate, the buck can continue to operate if VIN is > 3.8 V
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Electrical Characteristics (continued)
VIN = VLR1 = 8 V to 18 V, VSUP = 4 V to 10 V, VIN2SENSE = 4 V to 40 V, TJ = –40°C to 150°C (unless otherwise noted)
PARAMETER
VEXTSUP-HYS
EXTSUP switch-over hysteresis
IREG-LIM
Current limit on VREG
IREG-EXTSUP-LIM
Current limit on VREG when using
EXTSUP
TEST CONDITIONS
MIN
MAX
UNIT
150
TYP
250
mV
EXTSUP = 0 V normal mode as well as LPM
100
400
mA
IVREG = 0 mA to 100 mA, EXTSUP = 8.5 V, SYNC =
High
125
400
mA
INPUT VOLTAGE VIN - OVERVOLTAGE LOCK OUT AND REVERSE POLARITY PROTECTION
VOVLO
Overvoltage shutdown
OVLOHys
Hysteresis
OVLOfilter
Filter time
VGD
Clamping voltage of ext. FET
RGPULL
Internal resistance to GND
VIN rising
45
46
47
V
VIN falling
43
44
45
V
1
2
3
5
VIN - GPULL
V
µs
17
V
500
kΩ
BUCK CONTROLLERS
VOUT1, VOUT2
Adjustable output voltage range
VREF
Internal reference voltage and
tolerance in normal mode
Measure VSENSEx pin
VREF, LPM
Internal reference voltage and
tolerance in low-power mode
Measure VSENSEx pin
VSENSE for forward-current limit in
CCM
VSENSEx = 0.75 V, duty cycles < 10%
VSENSE for reverse-current limit in
CCM
VSENSEx = 1 V
VI-Foldback
VSENSE for output short
VSENSEx = 0 V (foldback)
tdead
Shoot through delay, blanking time
VSENSE
0.9
0.792
0.8
–1%
0.784
11
V
0.808
V
1%
0.8
–2%
0.816
V
2%
60
75
90
mV
–65
–37.5
–23
mV
17
43.8
48
mV
High-side minimum on-time
20
ns
100
ns
DCNRM
Maximum duty cycle (digitally
controlled)
DCLPM
Duty cycle LPM
ILPM_Entry
LPM entry threshold load current as
fraction of maximum set load current
1%
VLPM_Exit
LPM exit threshold load current as
fraction of maximum set load current
10%
98.75%
80%
HIGH-SIDE EXTERNAL NMOS GATE DRIVERS FOR BUCK CONTROLLERS
IGUx_peak
Gate driver peak current
rDS(on)
Source and sink driver
VREG = 5.8 V, IGUx current = 200 mA
0.6
A
5
Ω
LOW-SIDE NMOS GATE DRIVERS FOR BUCK CONTROLLERS
IGLx_peak
Gate driver peak current
rDS(on)
Source and sink driver
VREG = 5.8V, IGLx current = 200 mA
0.6
A
5
Ω
INTERNAL OSCILLATOR (RT)
fSW
Buck switching frequency
RT pin: GND
360
400
440
kHz
fSW
Buck switching frequency
RT pin: 60 kΩ external resistor
360
400
440
kHz
fSW-adj
Buck adjustable range with external
resistor
RT pin: external resistor
150
600
kHz
fsync
Buck synch. range
External clock input on SYNC
150
600
kHz
VRT
Oscillator reference voltage
tSW-Prop dly
SYNC rising edge to PH rising edge
delay
tSW-Trans-delay
Last SYNC rising edge to return to
resistor mode if CLK is not present
on SYNC pin
10
1.2
0
20
20
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Electrical Characteristics (continued)
VIN = VLR1 = 8 V to 18 V, VSUP = 4 V to 10 V, VIN2SENSE = 4 V to 40 V, TJ = –40°C to 150°C (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
ERROR AMPLIFIER (OTA) FOR BUCK CONTROLLERS AND BUCK CONVERTER
IPULLUP_VSENSEx
gm
Pullup current at VSENSEx pins
VSENSEx = 0 V
50
100
200
nA
Forward transconductance
COMP1, COMP2 = 0.8 V; source/sink = 5 µA, Test in
feedback loop
0.7
0.9
1.35
mS
EXTERNAL CLOCK AND ENABLE INPUTS: SYNC. EN1, EN2, EN3, EN4
VIH
Higher threshold
VIN = 13 V
VIL
Lower threshold
VIN = 13 V
1.7
V
RIH
Pulldown resistance
VSYNC = 5 V
500
IIL_ENx
Pullup current
VENx = 0V
0.5
tdeglitch
Deglitch time, ENx
0.7
V
kΩ
2
µA
16
µs
0.8
5.25
V
–2.5%
2.5%
2
LINEAR REGULATOR LREG1
VLREG1
Regulated output range
IL = 10 µA to 300 mA
VREF
Internal reference voltage tolerance
Referred to 0.8-V VREF, measured at VSENSE4
Vline-reg
Line regulation
VIN = VLR1: 6 V
to 28 V, IOUT 4
= 10 mA,
Vload-reg
Load regulation
IOUT4 = 10 mA
to 300 mA, VIN
= 14 V
∆VOUT, VOUT = 5 V
15
∆VOUT, VOUT = 3.3 V
15
∆VOUT, VOUT = 1.5 V
15
∆VOUT, VOUT = 5 V
10
∆VOUT, VOUT = 3.3 V
10
∆VOUT, VOUT = 1.5 V
10
VIN = VLR1 = 4 V: IOUT = 250 mA
500
VIN = 9 V, VLR1 = 4 V: IOUT = 150 mA
300
VDropout
Drop out voltage
IOUT4
Output current
VOUT in regulation
0.01
300
ILREG1-CL
Output current limit
VOUT = 0 V
400
1000
dVLREG1 / dt
Output soft start slew rate
PSRR
Power supply ripple rejection
VTH-CP ONp
Charge-pump turnoff voltage, VIN
rising
ITH-CP-OFF
mV
mV
mA
mA
5
Vripple = 0.5
VPP, IOUT = 300
mA
Freq = 100 Hz
60
Freq = 150 kHz
25
V/ms
dB
9.4
Hysteresis
Low-load current-detection threshold
mV
V
0.18
IOUT4 falling
Low-load current-detection
hysteresis
V
2
mA
4
mA
SOFT START SSX
ISSx
Soft-start source current
SSx = 0 V
0.75
1
1.25
µA
RESET RSTx
RSTpullup
RST1 to S2, RST2 to S4, RST4 to
LREG1 internal pullups
RSTxth1
Reset threshold
RSTxhys
Hysteresis
50
VSENSEx falling
–5
–7
kΩ
–9.5
2
%VREF
%VREF
IRSTx = 5 mA
450
mV
IRSTx = 1 mA
100
mV
1
µA
RSTxdrop
Voltage drop
RSTxleak
Leakage
tdeglitch
Power-good deglitch time
tdelay
Reset release delay
External capacitor = 1 nF
tdelay_fix
Fixed reset delay
No external capacitor, Rdelay pin open
IOH
Activate current source (current to
charge external capacitor)
Current to charge external capacitor
IIL
Activate current sink (current to
discharge external capacitor)
Current to discharge external capacitor
VS2 = VS4 = VRSTx = 13 V, RST4 = 8 V
2
16
1
20
50
µs
30
40
50
µA
30
40
50
µA
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Electrical Characteristics (continued)
VIN = VLR1 = 8 V to 18 V, VSUP = 4 V to 10 V, VIN2SENSE = 4 V to 40 V, TJ = –40°C to 150°C (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
SYNCHRONOUS BUCK CONVERTER BUCK3
VSUP
Buck3 supply voltage
VSUP_UV
Buck3 undervoltage lockout
rDS(on)
10
V
VSUP falling
3.6
4
3.7
3.8
V
VSUP rising
3.7
3.8
3.9
V
0.14
0.28
Ω
0.15
0.28
Ω
High-side switch
VSUP = 9 V, VBoot3 –PH3 = 5.8 V
Low-side switch
VSUP = 9 V, VVREG-PGND3 = 5.8 V
IHS-Limit
High-side switch
ILS-Limit
Low-side switch, current into PH3
VSUPLkg
VSUP leakage current
VSUP = 10 V for high side, EN3 = Low. TJ = 100°C
IFB3
Current foldback
VSENSE3 = 0 V
fSW-adj
Buck3 switching frequency range
with external resistor
Using external resistor on RT/CLK
VSense
Feedback voltage
Internal ref = 0.8 V
2.5
A
2.38
A
DC3
Current loop transconductance
ΔIpeakPH3 / ΔVCOMP3
Minimum duty cycle
fSW = 400 kHz, SLEW = LOW or OPEN
Maximum duty cycle
In dropout operation
A
600
–1.5%
1.5%
2-times - frequency foldback entry
threshold, VSENSE3 falling
Gm3
µA
150
2-times - frequency foldback exit
threshold, VSENSE3 rising
fSW-f-back
1
1.9
kHz
0.65
V
0.6
V
5.4
S
10%
98.75%
TOT-BUCK3
Overtemperature sensor threshold,
leads to Buck3 FET deactivation
170
°C
TOT-BUCK3-HYS
Overtemperature sensor hysteresis
15
°C
170
°C
15
°C
THERMAL SHUTDOWN
Tshutdown
Junction temperature shutdown
threshold
Thys
Junction temperature hysteresis
12
150
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100
100
90
90
80
80
70
70
Efficiency (%)
Efficiency (%)
6.6 Typical Characteristics
60
50
40
30
60
50
40
30
20
20
10
10
0
0
1
10
100
1000
10000
High
Low
500
0
Output Current (mA)
VOUT = 3.3 V
VIN = 5 V
VIN = 14 V
Figure 1. Buck2 Efficiency versus Output Current
Continuous Mode
2000
VOUT = 3.3 V
2500
VSUP = 4 V
Figure 2. Buck3 Efficiency vs Output Current, 400 kHz 25°C
805
Regulated VSENSEx Voltage (mV)
60
50
VIN Shutdown Current
PA)
(
1000
1500
Current (mA)
40
30
20
10
804
803
802
801
800
799
798
797
796
795
0
5
-5
10
15
20
25
30
35
-40
-15
Input Voltage (V)
10
35
60
85
110
135
160
Temperature (°C)
Figure 3. VIN Shutdown Current vs VIN
Figure 4. Regulated VSENSEx Voltage vs Temperature
(Buck1 and Buck2)
100 mV/DIV
100 mV/DIV
VOUT AC-Coupled
VOUT AC-Coupled
2 A/DIV
IIND
2 A/DIV
IIND
VIN = 12 V
Inductor = 4.7 µH
Time = 50 µs/DIV
VOUTx = 5 V
RSENSE = 10 mΩ
Switching Frequency = 400 kHz
Figure 5. Buck1 and Buck2 Load Step: Low-Power-Mode
Entry (0.09 mA to 4 A at 2.5 A/µs)
VIN = 12 V
Inductor = 4.7 µH
Time = 50 µs/DIV
VOUTx = 5 V RSENSE = 10 mΩ
Switching Frequency = 400 kHz
Figure 6. Buck1 AND Buck2 Load Step: Low-Power-Mode
Exit (0.09 mA to 4 A AT 2.5 A/µs)
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Typical Characteristics (continued)
Peak Current Sense Voltage (mV)
80
Forced Continuous Mode (SYNC = 1), 200-mA Load
1 A/DIV
Discontinuous Mode (SYNC = 0), 200-mA Load
1 A/DIV
1 A/DIV
Low-Power Mode (SYNC = 0), 20-mA Load
70
60
50
40
30
20
10
0
0
Time = 2 µs/DIV
VOUTx = 5 V
RSENSE = 10 mΩ
Switching Frequency = 400 kHz
VIN = 12 V
Inductor = 4.7 µH
Sense Current (µA)
Buck3 Inductor Peak Current (A)
3
1.5
1
0.5
0
0
0.2
0.4
0.6
0.8
1
150°C
25°C
0
1
2
3
4
5
6
7
8
9
10 11 12
Output Voltage (V)
Figure 10. Current Sense Pins Input Current (Buck1 and
Buck2)
Figure 9. Foldback Current Limit (Buck3)
45
Peak Current Sense Voltage (mV)
80
40
GPULL Voltage (V)
0.8
0.9
0.8
0.7
0.6
0.5
0.4
0.3
0.2
0.1
0
-0.1
-0.2
-0.3
VSENSE Voltage (V)
35
30
25
20
15
10
5
70
60
VIN = 8 V
50
40
VIN = 12 V
30
20
10
0
0
0
10
20
30
40
50
60
70
0
10
20
30
40
50
60
70
80
90 100
Duty Cycle (%)
Input Voltage (V)
Figure 11. GPULL Voltage vs VIN
14
0.6
Figure 8. Foldback Current Limit (Buck1 and Buck2)
3.5
2
0.4
VSENSEx Voltage (V)
Figure 7. Inductor Currents (Buck1 and Buck2)
2.5
0.2
Figure 12. Current Limit vs Duty Cycle (Buck1 and Buck2)
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Typical Characteristics (continued)
Buck3 Peak Inductor Current (A)
3.5
3
2.5
2
1.5
1
4V
5V
6V
0.5
0
40%
60%
Duty Cycle
= 4, 5, OR 6 V
Inductor = 10 µH
Switching Frequency = 400 kHz
0
VSUP
20%
80%
100%
VSENSE3 = 0.75 V
Figure 13. Buck3 Maximum Peak Inductor Current vs Duty Cycle
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7 Detailed Description
7.1 Overview
The TPS43340-Q1 is a dual-buck regulator controller (Buck1, Buck2), single-buck regulator converter (Buck3)
and linear regulator (LREG1) designed for powering the Texas Instruments family of DSPs and microcontrollers
or general-market MCU products. The device features integrated short-circuit and overcurrent protection on the
gate-drive outputs for the buck regulator controllers and independent current-foldback control for each buck
regulator supply during regulator output short to ground. Each output supply incorporates a soft start to ensure
that on initial power up these regulated outputs are not in current limit. Implementation of reset delay on power
up allows the outputs of Buck1, Buck2, Buck3 and the linear regulator to get to stable regulation. An external
capacitor sets the delay to a maximum range of 300 ms. Each power-supply output has adjustable output voltage
based on the external resistor-network settings. The device has sequencing control during power up and power
down of the output rails, based on the enable-and-disable control or soft start.
42
41
40
39
BOOT2
43
VREG
44
GPULL
RST4
45
EXTSUP
VSENSE4
46
VIN
LREG1
47
VLR1
EN4
48
VSENSE
BOOT1
7.2 Functional Block Diagram
38
37
BUCK 1 - Channel
-
Thermal
sensor
+
V IN
3
-
4
VSENSE1
7
+
-
+
0.8V
PWM
comp
OTA
gm
+
+
PWM
comp
Slope
Comp
0.8V
VSUP
11
PGND3
12
-
+
-
Current
sense
Amp
+
-
gm
+
Buck3 Buck2
V REF
+
0.8 V
PH3
13
14
GU2
35
PH2
34
GL2
33
PGND2
32
S4
31
S3
30
VSENSE2
29
COMP2
28
RST2
27
SS2
26
GND
25
RT
Phasing
-
OTA
Buck1
Osc
Control Logic
15
16
17
18
COMP3
10
Filter timer
+
O -C
VSENSE3
V REF
36
-
RST3
9
-
SS3
RST1
PWM logic
+
Int Reg
8
BOOT3
COMP1
Filter timer
+
-
19
20
21
22
23
24
Rdelay
6
-
Control
SYNC/MPL
S1
+
SYNC
5
EN
Slope Comp
Current sense
Amp
S2
SS1
Int Reg supply
EN1
PGND1
Overcurrent
Int Reg
EN2
GL1
PWM logic
2
EN3
PH1
BUCK 2 –
Channel
(duplicate of
BUCK 1-Channel )
Internal
supply
SLEW
GU1
V IN
Filter
1
Figure 14. Internal Functional Blocks
7.3 Feature Description
7.3.1 Enable Inputs
The use of independent enable inputs at the EN1 through EN4 pins enables all the regulators. These pins have
internal pullup currents of 0.5 µA (typical). As a result, an open circuit on any of these pins enables its respective
regulator. EN1, EN2, and EN4 are high-voltage pins, which permits their connection directly to the battery for
self-bias. When all regulators are disabled, the device shuts down and consumes a current of 5 µA typical.
16
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Feature Description (continued)
7.3.2 Linear Regulator (LREG1)
The linear regulator is an NMOS output low-dropout regulator with output load current up to 300 mA. It can
operate directly from the battery. With EN4 tied high or open, LREG1 turns on its output following an internally
generated soft-start ramp. The regulation loop uses internal frequency compensation. If the output shorts to
ground, the device protects itself by limiting the current. For VIN lower than 9 V, LREG1 controls the internal
charge pump depending on VIN and the load current in accordance with Table 4. An internal voltage selector
selects the higher available supply, VIN or the charge pump voltage, for the error amplifier. The device monitors
the output voltage of the low-dropout regulator for undervoltage and signals its state on pin RST4.
7.3.3 Gate-Driver Supply (VREG, EXTSUP)
An internal linear regulator supplies the gate drivers of the buck controllers and the buck converter. The regulator
output (5.8 V typical) is available at the VREG pin and requires decoupling using a ceramic capacitor in the
range of 3.3 µF to 10 µF. This pin has an internal current-limit protection; do not use it to power any other
circuits.
Power for the VREG linear regulator comes from VIN by default when the EXTSUP voltage is lower than 4.6 V
(typical). Should there be an expectation of VIN going to high levels, there can be excessive power dissipation in
this regulator, especially at high switching frequencies and when using large external MOSFETs. In this case, it
is advantageous to power this regulator from the EXTSUP pin, connection to which can be to a supply lower than
VIN but high enough to provide the gate drive. The voltage on EXTSUP should not exceed 9 V. With EXTSUP
connected to a voltage greater than 4.6 V, the linear regulator automatically switches to EXTSUP. Efficiency
improvements are thus possible when using one of the switching regulator rails from the TPS43340-Q1 or any
other voltage available in the system to power the EXTSUP. If the EXTSUP supply is above 4.6 V but below 7.5
V, the EXTSUP-LDO acts as a pass element, providing EXTSUP voltage less a small dropout to VREG.
VIN
5.8 V
(typical)
LDO
VIN
EXTSUP
7.5 V
(typical)
LDO
EXTSUP
4.6 V (typical)
VREG
Figure 15. Internal Gate-Driver Supply
Using a voltage above 5.8 V (sourced by VIN) for EXTSUP is advantageous, as it provides a large gate drive
and hence better on-resistance of the external MOSFETs.
When using EXTSUP, always keep the buck rail supplying EXTSUP enabled. Alternatively, if it is necessary to
switch off the buck rail supplying EXTSUP, place a diode between the buck rail and EXTSUP.
During low-power mode, the EXTSUP functionality is not available. The internal regulator operates as a shunt
regulator powered from VIN and has a typical value of 7.5 V. Current-limit protection for VREG is available in
low-power mode as well. If EXTSUP is unused, leave the pin open without a capacitor installed.
7.3.4 External P-Channel Drive (GPULL) and Reverse Battery Protection
The TPS43340-Q1 includes a gate driver for an external P-channel MOSFET which can be used for reverse
battery protection. This is useful to reduce the voltage drop across the protection element compared to using a
series diode to VIN. The gate – source voltage of the external PMOS is clamped by an internal Zener diode to
17 V typical.
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Feature Description (continued)
VBAT ≤ VF → | VGS | = 0 V → FET and diode not conducting
VF ≤ VBAT ≤ VT (FET) → | VGS | = VBAT → FET NOT conducting and diode conducting
VT (FET) ≤ VBAT ≤ 17 V → | VGS | = VBAT → FET conducting
VBAT ≥ 17 V → | VGS | = 17 V → FET conducting
Spacer
VF
VBAT
VGS
GPULL
VIN
If the FET has significant leakage between
GPULL and VBAT, TI recommends an
external resistor in parallel to the internal
one (GPULL - GND).
17 V
Electrical Source of FET is at higher
potential (even though symbol indicates
Source on the right pin of the FET).
Voltages refer to VGS as shown.
500 k
TPS43340-Q1
GND
Figure 16. Internal Circuit of GPULL Output
NOTE
An implementation without the PMOS blocks the current coming from Buck-outputs
(improper OR-ing, and others), which may result in exceeding the absolute maximum
ratings.
7.3.5 Undervoltage Lockout and Overvoltage Protection
The TPS43340-Q1 starts up at a VIN voltage of 6.5 V (maximum). Once it has started up, the device operates
down to a VIN undervoltage lockout level of 3.6 V or until reaching a VREG undervoltage of 3.6 V. A voltage
above 46 V at VIN shuts down the device. In order to prevent transient spikes from shutting down the device, the
under- and overvoltage protection have filter times of 5 µs (typical). There is no support for overvoltage
protection in LPM.
When the voltages return to the normal operating region, the enabled regulators start up with a soft-start ramp.
7.3.6 Synchronous Buck Converter Buck3
This regulator operates with the switching frequency set on the RT terminal or an external clock input on the
SYNC terminal. The internal power FETs switch out of phase to regulate the output voltage, operating in a pulse
width modulation. The converter uses a peak-current mode-control loop with external frequency compensation.
The synchronous operation mode improves the overall efficiency.
18
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Feature Description (continued)
7.3.6.1 Soft Start and Foldback Functions
A capacitor on the SS3 terminal sets the converter soft start. Pulling the enable pin on EN3 high activates soft
start. During soft start or whenever the voltage on VSENSE3 falls below limits given by ƒSW-f-back, the converter
switches to a frequency foldback of ƒsw / 2 to help control the coil current. In addition to the frequency foldback,
implementation of current foldback reduces power dissipation to protect the converter against an output short to
ground. Like in the buck controllers, the current foldback reduces the maximum peak current limit depending on
the voltage on the VSENSE3 pin. Figure 9 shows the characteristic of current foldback.
7.3.6.2 Current-Mode Control and Current-Limit Protection
Measurement of the coil peak current is by use of the high-side integrated FET; peak-current regulation occurs in
each switching cycle in accordance with the voltage on the COMP3 pin. COMP3 is the output of a
transconductance error amplifier of the voltage feedback loop for Buck3, as COMP1 and COMP2 are for
controllers Buck1 and Buck2. COMP3 sets the target for the peak current comparator (inner current loop) and
serves as frequency compensation of the voltage loop using a type II compensation network.
Clamping the voltage on the COMP3 node realizes the positive current limit. The positive clamping level depends
on the voltage on the VSENSE3 pin, as described previously. The device also implements clamping for low
voltage on the COMP3 pin, thereby speeding up the transient response after output overshoot. For stability of the
current loop, during the switching cycle the internal slope compensation adjusts the current limit set by COMP3.
For correct operation of the slope compensation, the coil used for Buck3 must satisfy the following:
LBuck3 = 3.7 / ƒsw
where
•
•
LBuck3 is the inductance in henries
ƒsw is the switching frequency in hertz
(1)
Reaching the positive current limit during the high PWM phase resets the PWM. The high-side FET turns off and
the low part of the cycle is initiated. On detecting an overcurrent condition such as an output short to a supply
during the PWM low phase, the low-side FET turns off until the end of the given cycle, to allow the coil current to
flow through the body diode of the high-side FET.
7.3.6.3 Operation in Dropout and Undervoltage Protection
This converter is capable of operating with a low input-to-output voltage difference. In dropout operation, the
integrated high-side MOSFET stays on continuously. In every fourth clock cycle, the device limits the duty cycle
to 95% in order to charge the bootstrap capacitor at BOOT3. This allows a maximum duty cycle of 98.75% for
the buck converter. In this mode, the output tracks the input until initiation of the internal undervoltage lockout
due to low supply voltage on the VSUP pin.
Thermal shutdown monitors the virtual junction temperature of the integrated FETs. When TJ exceeds 170°C,
both the high- and low-side switches turn off. The converter returns to normal operation when the temperature
decreases to the acceptable level (typically TJ = 150°C)
7.3.6.4 Slew Rate Control (SLEW)
The setting on the SLEW terminal controlss the slew for Buck3. Setting the slew rate to logic high (slowest slew
rate) extends the minimum on-time of the buck converter by 5% of the clock period.
SLEW TERMINAL SETTING
tr (TYP) ns
tf (TYP) ns
SLEW > VREG – 0.2 V (low slew rate, logic
high)
24
7
SLEW pin open – medium slew rate
11
3
SLEW < 0.2 V (fast slew rate, logic low)
8
2
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7.4 Device Functional Modes
7.4.1 Buck Controllers: Normal Mode PWM Operation
7.4.1.1 Setting the Operating Frequency
The buck controllers operate using constant-frequency peak-current-mode control for optimal transient behavior
and ease of component choices. The switching frequency is programmable between 150 kHz and 600 kHz,
depending on the resistor value at the RT pin. Tying this pin to ground sets the default switching frequency to
400 kHz. A resistor connected to RT can also set the frequency according to the formula:
ƒsw = 24 x 109 / RT [Hz]
where
•
•
600 kHz requires 40 kΩ
150 kHz requires 160 kΩ
(2)
It is also possible to synchronize to an external clock at the SYNC pin in the same frequency range of 150 kHz to
600 kHz. The device detects clock pulses at this pin, and an internal PLL locks onto the external clock within the
specified range. The device can also detect a loss of clock at this pin, and detection of clock loss for tSW-Transdelaysets the switching frequency to the internal oscillator. The two buck controllers operate at the same switching
frequency, 180 degrees out of phase.
7.4.1.2 Feedback Inputs
Choose the resistor feedback divider networks connected to the VSENSEx (feedback) pins to set the output
voltages. Make the choice such that the regulated voltages at the VSENSEx pins equal 0.8 V. The VSENSEx
pins have 100-nA pullup current sources as a protection feature in case the pins open up as a result of physical
damage.
§
RTOP ·
VOUTx 0.8 ¨ 1
¸ V
¨ R BOTTOM ¸
©
¹
where
•
•
RTOP is the resistor from VOUTx to VSENSEx
RBOTTOM is the resistor from VSENSEx to ground.
(3)
7.4.1.3 Soft-Start Inputs
In order to avoid large inrush currents, both buck controllers have independent programmable soft-start timing.
The voltage at the SSx pins acts as the soft-start reference voltage. A 1-µA pullup current is available at the SSx
pins, and by choosing a suitable capacitor one can obtain a desired soft-start ramp speed. After start-up, the
pullup current ensures that pins SSx are higher than the internal reference of 0.8 V, which then becomes the
reference for the buck controllers. The required capacitor for ∆t, the desired soft-start time, is given by:
I u 't
CSS SS
(Farads)
'V
where
•
•
ISS = 1 µA (typical)
∆V = 0.8 V
(4)
Alternatively, one can use the soft-start pins as tracking inputs. In this case, connect the pins to the supply to be
tracked via a suitable divider network.
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Device Functional Modes (continued)
7.4.1.4 Current-Mode Operation
Peak current-mode control regulates the peak current through the inductor such that the output voltage maintains
its set value. The error between the feedback voltage at VSENSEx and the internal reference produces a signal
at the output of the error amplifier (COMPx) which serves as the target for the peak inductor current. This target
provides a comparison for the current through the inductor, sensed as a differential voltage at S1-S2 for Buck1
and S3-S4 for Buck2, and compared with this target during each cycle. A fall or rise in load current produces a
rise or fall in voltage at VSENSEx, causing COMPx to fall or rise, respectively, thus increasing or decreasing the
current through the inductor until the average current matches the load. In this way, the device maintains the
output voltage in regulation.
The high-side N-channel MOSFET turns on at the beginning of each clock cycle and remains on until the
inductor current reaches its peak value. Once this MOSFET turns off, and after a small delay (shoot-through
delay), the lower N-channel MOSFET turns on until the start of the next clock cycle. In dropout operation, the
high-side MOSFET stays on continuously. In every fourth clock cycle, the duty cycle is limited to 95% in order to
charge the bootstrap capacitor at BOOTx. This allows a maximum duty cycle of 98.75% for the buck regulators.
Thus, during dropout the buck regulators switch at one-fourth of the normal frequency.
7.4.1.5 Current Sensing and Current Limit With Foldback
Clamping the maximum value of COMPx is such as to limit the maximum current through the inductor to a
specified value. When the output of the buck regulator (and hence the feedback value at VSENSEx) falls to a low
value due to a short circuit or overcurrent condition, the clamping voltage at the COMPx successively decreases,
thus providing current foldback protection. This protects the high-side external MOSFET from excess current
(forward-direction current limit).
Similarly, if due to a fault condition the output shorts to a high voltage and turns the low-side MOSFET fully on,
the COMPx node drops low. The device holds COMPx at a low level as well in order to limit the maximum
current in the low-side MOSFET (reverse direction current limit).
An external resistor senses the current through the inductor. Choose the sense resistor such that the maximum
forward peak current in the inductor generates a voltage of 75 mV across the sense pins. This value specification
is at low duty cycles only. At typical duty cycle conditions around 40% (assuming 5-V output and 12-V input),
50 mV is a more reasonable value, considering the slope compensation and tolerances. The typical
characteristics in Figure 18 and Figure 12 provide a guide for using the correct current-limit sense voltage.
The current-sense pins Sx are high-impedance pins with low leakage across the entire output range. These pin
characteristics allow DCR current sensing using the dc resistance of the inductor for higher efficiency. Figure 17
shows DCR sensing. Here the series resistance (DCR) of the inductor serves as the sense element. Place the
filter components close to the device for noise immunity. Remember that while DCR sensing gives high
efficiency, it is less accurate due to the temperature sensitivity and a wide variation of the parasitic series
resistance of the inductor. Hence, it may often be advantageous to use the more-accurate sense resistor for
current sensing.
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Device Functional Modes (continued)
VIN
Current Loop (Inner Loop)
HS
DCR Sensing
+
L
PWM
RL
VPH
Logic
VOUT
Gate Drivers
±
COUT
LS
RDCR
CDCR
Summing
Comparator
S1
+
Current
Sensing
VSENSE
VSENSE,EXT
S2
±
VSLOPE
Slope
Compensation
±
R1
FB
+
gm
Error
Amplifier
Voltage Loop (Outer Loop)
R2
Figure 17. Overcurrent Sensing and Control
22
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Device Functional Modes (continued)
7.4.1.6 Slope Compensation
Optimal slope compensation which is adaptive to changes in input voltage and duty cycle allows stable currentmode operation in all conditions. For optimal performance of this circuit, satisfy the following condition in the
choice of inductor and sense resistor:
L=
200
¦ SW
u RS
where
•
•
•
L is the buck regulator inductor in henries
RS is the sense resistor in ohms
fsw is the buck regulator switching frequency in Hz
(5)
Peak Current-Sense Voltage (mV)
80
70
60
50
40
30
VIN = 12 V
Example With 40% DC, Max Sense Voltage |64 mV at VIN = 12 V, VOUT= 4.8 V
Average Current Ripple Current
Example With 70% DC, Max Sense Voltage |43 mV at VIN = 12 V, VOUT= 8.4 V
Average Current Ripple Current
20
10
0
0
10
20
30
40
50
60
70
80
90 100
Duty Cycle (%)
Figure 18. Peak Current-Sense Voltage versus Duty Cycle
7.4.1.7 Reset Outputs and Filter Delays
Each buck controller has an independent reset comparator monitoring the feedback voltage at the VSENSEx
pins and indicating whether the output voltage has fallen below the specified reset threshold. The reset indicator
is available as an open-drain output at the RSTx pins. An internal 50-kΩ pullup resistor to S2 or S4 is available,
or one can use an external resistor. When a buck controller shuts down, the device pulls down the power-good
outputs internally. Connecting the pullup resistor to a rail other than the output of that particular buck channel
causes a constant current flow through the resistor when the buck controller powers down.
In order to avoid triggering the power-good indicators due to noise or fast transients on the output voltage, the
device implements an internal delay of tdeglitch for de-glitching. The output voltage reaching its set value after a
start-up ramp or negative transient asserts the power-good indicator high (releases the open-drain pin) after a
delay of tdelay, at least tdelay_fix. A use of this is to delay the reset to the circuits being powered from the buck
regulator rail. Program the delay of this circuit by using a suitable capacitor at the Rdelay pin according to
Equation 6:
Power-Good Output Delay
tRdelay = 106 u CRdelay (seconds)
where
•
CRdelay is the capacitor value in farads on the Rdelay pin.
(6)
An open on the Rdelay pin sets the delay to a default value of 20 µs typical. The power-good delay timing is
common to all supply rails, but the power good comparators and outputs function independently.
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Device Functional Modes (continued)
7.4.1.8 Light-Load PFM Mode
An external clock or a high level on the SYNC pin or enabling Buck3 results in forced continuous-mode operation
of the bucks. Having the SYNC pin low or open allows the buck controllers to operate in discontinuous mode at
light loads by turning off the low-side MOSFET on detection of a zero-crossing in the inductor current.
In discontinuous mode, as the load decreases, the duration of the clock period when both the high-side and the
low-side MOSFETs are turned off increases (deep discontinuous mode). In case the duration exceeds 60% of
the clock period and VIN > 8 V, the buck controller switches to a low-power operation mode. The design ensures
that this typically occurs at 1% of the set full-load current if the choice of the inductor and the sense resistor is
appropriate as recommended in the slope compensation section.
Normal Mode
High Load
Time
0
Normal to DCM
Boundary Condition
0
tLS
Time
0
Time
Low-Power Mode
(LPM) Operation
Moves closer as
Load Increases
0
Time
tON
Entering Normal
Mode
0
Time
Figure 19. Modes of Operation
In low-power PFM mode, the buck controllers monitor the VSENSEx voltage and compare it with the 0.8 V internal
reference. Whenever the VSENSEx value falls below the reference, the high-side MOSFET turns on for a pulseduration inversely proportional to the difference across VIN-S2 for Buck 1 and VIN-S4 for Buck2. At the end of
this on-time, the high-side MOSFET turns off and the current in the inductor decays until it becomes zero. The
low-side MOSFET does not turn on. The next pulse occurs the next time VSENSEx falls below the reference value.
This results in a constant volt-second TON hysteretic operation with a total device quiescent current consumption
of 30 µA when a single buck channel is active and 35 µA when both channels are active.
As the load increases, the pulse become more and more frequent until the current in the inductor becomes
continuous. At this point, the buck controller returns to normal fixed-frequency current-mode control. Another
criterion for exit from the low-power mode is when VIN falls low enough to require a higher-than-80% duty cycle
of the high-side MOSFET.
24
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Device Functional Modes (continued)
The TPS43340-Q1 can support the full-current load during low-power mode until the transition to normal mode
takes place. The design ensures the low-power-mode exit occurs at 10% (typical) of full-load current if the
inductor and sense resistor choices are as recommended. Moreover, there is always a hysteresis between the
entry and exit thresholds to avoid oscillating between the two modes.
In the event that both buck controllers are active, low-power mode is only possible when both buck controllers
have light loads that are low enough for entry to low-power mode.
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8 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
8.1 Application Information
The TPS43340-Q1 multirail power supply operates with a supply voltage VIN of 4 V to 40 V for the Buck
controllers and the LDO. The TPS43340-Q1 Buck converter (Buck3) operates with a supply voltage VSUP of 4 V
to 10 V. For reducing power dissipation, TI strongly recommends using the output voltage of one of the buck
regulators as the input supply for the LDO regulator. To use the output voltage of the buck regulator this way, the
selected buck-regulator output voltage must be higher than the selected LDO-regulator output voltage. For
further efficiency improvements, the part offers a pin to control an external FET that can bypass the reversepolarity-protection diode (GPULL).
8.2 Typical Application
Power Switch
(PMOS)
6 V to 18 V
3.3 V, 1 W
Main Supply
VIN
VLREG1
TOP _SW 3
10 µF
VLREG1
Open or
Controller
10 kW
L1
5 V, 22.5 W
VOUT1
0.01 W
0.1 µF
49
38
37
VREG
40
BOOT2
41
GPULL
10 µF
VIN
42
VLR1
PH1
43
RST4
2
6W
44
VIN2SENSE
GU1
45
VSENSE4
1
46
EN4
BOOT1
Buck1TOPFET
47
LREG1
48
0.1 µF
VOUT1
EXTSUP
2 kW
255 kW
VIN
0.1 µF
10 kW
330 µF
80.6 kW
VIN or
other supply
GU2
36
PH2
35
Buck2TOPFET
L2
3.3 V, 15 W
VOUT2
0.01 W
8.2 µH
100 µF
CBuck1
8.2 µH
Buck1LOWFET
422 kW
3
GL1
4
PGND1
GL2
34
PGND2
33
5
6
S2
S4
32
S1
S3
7
VSENSE1
31
VSENSE2
30
8
COMP1
COMP2
29
100 µF
CBuck2
Buck2LOWFET
255 kW
TPS43340-Q1
47 pF
18 kW
SYNC
16
Rdelay
15
EN1
14
17
18
19
20
21
22
23
24
RST2
28
SS2
27
GND
26
RT
25
VOUT2
2.2 nF 12 kW
80.6 kW
2 kW
0.01 µF
100 kW
8.2 µH
0.01 µF
L3
0.1 µF
100 µF
CBuck3
4.7 nF
2 kW
180 pF
1.8 V, 4 W
VOUT3
13
EN2
PGND3
EN3
12
SLEW
VSUP
VSENSE3
11
COMP3
SS1
RST3
10 µF
RST1
BOOT3
0.01 µF
9
10
SS3
2 kW
VOUT3 input supply
68 pF
1.5 nF
VOUT1
PH3
80.5 kW
0.01 µF
12 kW
10 kW
10 kW
10 kW
102 kW
80.6 kW
Controller
Optional
Clock Input
L1, L2, L3: DR127-8R2-R (Coiltronics)
TOP_SW3: IRF7663TRPBF (International Rectifier)
TOP_SW1, BOT_SW2: Si4946BEY-T1-E3 (Vishay)
TOP_SW2, BOT_SW2: Si4946BEY-T1-E3 (Vishay)
CBUCK1, CBUCK2, CBUCK3:AVX- TPSD107K016R0060 (AVX)
Figure 20. Application Schematic
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Typical Application (continued)
8.2.1 Design Requirements
A few parameters must be known to begin the design process. Determination of these parameters is typically at
the system level.
The following example illustrates the design process and component selection for the TPS43340-Q1. Table 1
lists the design goal parameters.
Table 1. Application Example
PARAMETER
Input voltage, VIN
Buck1
Buck2
Buck3
6 V to 18 V
14 V, typical
6 V to 18 V
14 V, typical
4 V to 10 V
5 V, typical
Output ripple voltage
±0.2 V
±0.2 V
±0.1 V
Output voltage, VOUT
5 V ±2%
3.3 V ±2%
1.8 V ±2%
Maximum output current, IOUT
4.5 A
4.5 A
2.2 A
Minimum output current, IOUT
0.1 A
0.1 A
0.1 A
Load-step output tolerance,
∆VOUT + ∆VOUT(Ripple)
±0.3 V
±0.3 V
±0.15 V
Current-output load step, ∆IOUT
0.1 A to 4.5 A
0.1 A to 4.5 A
0.1 A to 2.2 A
Converter switching frequency,
fSW
400 kHz
400 kHz
400 kHz
125°C
125°C
125°C
Junction temperature, TJ
8.2.2 Detailed Design Procedure
8.2.2.1 High- and Low-Side Power NMOS Selection for the Buck Converters
An internal supply, which is 5.8 V typical under normal operating conditions, provides the gate-drive supply for
these MOSFETs. The output is a totem pole, allowing full voltage drive of VREG to the gate with a peak output
current of 0.6 A. The high-side MOSFET reference is the phase terminal (PHx), and the low-side MOSFET
referenced is the power ground (PGNDx) terminal. For a particular application, select these MOSFETs with
consideration for the following parameters rDS(on), gate charge Qg, drain-to-source breakdown voltage BVDSS,
maximum dc current IDC(maximum), and thermal resistance for the package.
Power dissipation on the high-side FET (PD_HS):
§ VIN u IOUT
3D _ HS
,OUT 2 u UDS(on)
7& u ' ¨
¨
2
©
·
¸ u Wr W f u ¦
SW
¸
¹
(7)
First term is conduction losses.
Second term is switching losses.
Power dissipation on the low-side FET (PD_LS):
3D _LS
,OUT
2
u UDS(on)
7& u
'
9f u ,OUT W dead u ¦ SW
(8)
The first term in the foregoing equation refers to conduction losses, and the second term covers the switching
losses in the FET body diode during the dead-time.
NOTE
rDS(on) has a positive temperature coefficient TC, which is typically 0.4%/°C.
Gate losses for high-side and low-side FETs:
PBuckX
GATE
= 2 × fsw × Qg × VREG
(9)
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8.2.2.2 Buck1 Component Selection
Duty Cycle
VOUT
D
VIN
5V
14 V
0.357
(10)
Selection of Current Sensing Resistor
0.075 V
RSENSE =
= 0.017
4.5 A
(11)
Use 10 mΩ to allow for ripple-current.
Inductor Selection L
0.01
L = 200 u
=5 +
400 kHz
(12)
Use 8.2 µH.
Inductor Ripple Current
5V
§
u 1
400 kHz u 8.2 + ¨©
'IL(RIPPLE)
Output Capacitor COUT
2 u 4.5 A
COUT
400 kHz u 0.2 V
5V ·
9 ¸¹
0.98 A
(13)
112 )
(14)
Use 100 µF.
'VOUT1
'IOUT2
'IOUT1 u ESR
u ¦C u &OUT1
IOUT1(Ripple)
VOUT1(Ripple)
u ¦SW u &OUT1
Input Capacitor CIN
0.25 u 4.5 A
CIN
400 kHz u 0.5 V
u
4.4 A
N+] u
IOUT1(Ripple) u ESR
)
u
4.4 A u 10 m:
264 mV
(15)
0.98 A
N+] u
)
0.98 A u 10 m: 12.8 mV
(16)
5.6 )
(17)
Use 10 µF, shared between Buck1 and Buck2.
High-Side MOSFET (Buck1TOPFET)
,OUT 2 u UDS(on)
3BuckTOPFET
CIN
0.25 u 4.5 A
400 kHz u 0.5 V
§ V uI ·
7& u ' ¨ IN OUT ¸ u Wr
2
©
¹
W f u ¦ SW
(18)
5.6 )
(19)
Low-Side MOSFET (Buck1LOWFET)
3BuckLOWERFET
,OUT
2
u UDS(on)
7& u
'
9F u ,OUT u
2
u W d u ¦SW
(4.5 A) u 0.009 u (1 0.4) u (1 0.357) 0.6 V u 4.5 A u 2 u 20 ns u 400 kHz
0.21 W
(20)
(21)
8.2.2.3 Buck2 Component Selection
Duty Cycle
VOUT
D
VIN
28
3.3 V
14 V
0.236
(22)
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Selection of Current-Sensing Resistor
0.075 V
RSENSE =
= 0.017
4.5 A
(23)
Use 10 mΩ to allow for ripple current.
Inductor Selection L
0.01
L = 200 u
=5 +
400 kHz
(24)
Use 8.2 uH.
Inductor Ripple Current
3.3 V
§ 3.3 V ·
'IL(RIPPLE)
u 1
400 kHz u 8.2 PH ¨©
14 V ¸¹
Output Capacitor COUT
2 u 4.5 A
COUT
400 kHz u 0.2 V
0.77 A
(25)
112 )
(26)
Use 100 µF.
'VOUT2
'I OUT2
'I OUT2 u ESR
u ¦C u &OUT2
IOUT2(Ripple)
VOUT2(Ripple)
u ¦SW u &OUT2
Input Capacitor CIN
0.25 u 4.5 A
CIN
400 kHz u 0.5 V
u
4.4 A
N+] u
IOUT2(Ripple) u ESR
)
u
4.4 A u 10 m:
264 mV
(27)
0.77 A
N+] u
0.77 A u 10 m: 10.1mV
)
(28)
5.6 )
(29)
Use 10 µF, shared between Buck1 and Buck2. For better line-transient immunity, use a larger value.
High-Side MOSFET (Buck2TOPFET)
,OUT 2 u UDS(on)
3BuckTOPFET
§ V uI ·
7& u ' ¨ IN OUT ¸ u Wr
2
©
¹
W f u ¦ SW
(30)
§ 14 V u 4.5 A ·
(4.5 A)2 u 0.009 : u (1 0.4) u 0.236 ¨
¸ u 20 ns 20 ns u 400 kHz
2
©
¹
0.56 W
(31)
Low-Side MOSFET (Buck2LOWFET)
3BuckLOWERFET
,OUT
2
u UDS(on)
7& u
'
9F u ,OUT u
u W d u ¦SW
(4.5 A)2 u 0.009 : u (1 0.4) u (1 0.236) 0.6 V u 4.5 A u (2 u 20 ns) u 400 kHz
(32)
0.24 W
(33)
8.2.2.4 Buck3 Component Selection
Duty Cycle
VOUT
D
VIN
1.8 V
5V
0.36
(34)
Inductor Selection LBuck3
3.7
LBUCK3 =
= 9.25 +
400 kHz
(35)
Use 8.2 µH.
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Inductor Ripple Current
1.8 V
§ 1.8 V ·
'IL(RIPPLE)
u 1
400 kHz u 8.2 PH ¨©
5 V ¸¹
Output Capacitor COUT
2 u 'IOUT3
COUT3 |
¦SW u '9OUT3
2 u 2.1 A
N+] u
9
0.46 A
(36)
70 )
(37)
Use 100 µF.
Input Capacitor CIN
0.25 u 2.2 A
C IN
400 kHz u 0.05 V
5.76 )
(38)
Use 10 µF.
'VOUT3
'IOUT3
u ¦C u &OUT3
'IOUT3 u ESR
I OUT3(Ripple)
VOUT3(Ripple)
u ¦SW u &OUT3
2.1 A
N+] u
u
I OUT3(Ripple) u ESR
)
u
2.1 A u 10 m: 126 mV
(39)
0.46 A
N+] u
0.46 A u 10 m:
)
6.03 mV
(40)
Internal High-Side MOSFET (Buck3TOPFET)
§ V uI ·
7& u ' ¨ IN OUT ¸ u Wr
2
©
¹
,OUT 2 u UDS(on)
3BuckTOPFET
(2.2 A)2 u 0.28
§ 5 V u 2.2 A ·
¨
¸ u
2
©
¹
u
QV
W f u ¦ SW
(41)
QV
u
00 kHz
0.58 W
(42)
Internal Low-Side MOSFET (Buck3LOWFET)
3BuckLOWERFET
2
(2.2 A) u 0.28
,OUT
2
u UDS(on)
u
7& u
u
$ u
'
9F u ,OUT u
u W d u ¦SW
[ 20 ns u 400 kHz
0.89 W
(43)
(44)
8.2.3 Application Curves
Figure 21. Load Step Buck 1
30
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Figure 22. Load Step Buck 2
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Figure 23. Load Step Buck 3
9 Power Supply Recommendations
The TPS43340-Q1 is designed to operate from an input voltage up to 40 V for the buck controllers (Buck1,
Buck2). The buck converter (Buck3) accepts input voltages up to 10 V; so in many applications, the output of
Buck1 or Buck2 is used to supply it. The linear regulator accepts up to 40 V; however, for power dissipation
reasons, TI advises using lower supply voltages. Ensure that the supply for all inputs is well regulated.
Furthermore, if the supply voltage in the application is likely to reach negative voltage (for example, reverse
battery), a forward diode must be placed at the input of the supply, where GPULL-pin can be used to bypass the
diode with an external FET to reduce the voltage drop and power dissipation. For the VIN pin, place a ceramic
capacitor or a set of ceramic capacitors close to the pin and add more capacitance as required. Consider
capacitance derating for aging, temperature, and DC bias. The PowerPAD package, which offers an exposed
thermal pad to enhance thermal performance, must be soldered to the copper landing on the PCB for optimal
performance.
• Connect a local decoupling capacitor close to the VSUP pin (supply for Buck3) for proper filtering.
• Connect a local decoupling capacitor close to the VLR1 pin (supply for LDO) for proper filtering.
• Connect a local decoupling capacitor close to the VREG-pin for proper filtering.
10 Layout
10.1 Layout Guidelines
Grounding and PCB Circuit Layout Considerations
1. Connect the drains of TOP_SW1 and TOP_SW2 together with the +ve terminal of the input capacitor COUT1.
The trace length between these terminals should be short.
2. The Kelvin-current sensing for the shunt resistor should have traces with minimum spacing, routed in parallel
with each other. Place any filtering capacitors for noise near the IC pins.
3. Connect the resistor divider for sensing output voltage between the +ve terminal of its respective output
capacitor CBuck1 or CBuck2 or CBuck3 and the IC signal ground. Do not route these components or their traces
near any switching nodes or high-current traces.
Other Considerations
1. Separate the IC signal ground and power ground terminals (GND and PGNDx) pins. Use a star-ground
configuration if connecting to a non-ground plane system. Use tie-ins for the EXTSUP capacitor,
compensation network ground, and voltage-sense feedback ground networks to this star ground.
2. Connect a compensation network between the compensation pins and IC signal ground. Connect the
oscillator resistor (frequency setting) between the RT pin and IC signal ground. Do not locate these sensitive
circuits near the dV/dt nodes; these include the gate drive outputs, phase pins, and boost circuits (bootstrap).
3. Reduce the surface area of the high-current-carrying loops to a minimum by ensuring optimal component
placement. Locate the bypass capacitors as close as possible to their respective power and ground pins.
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10.2 Layout Example
Power Switch
(PMOS)
Main Input Supply
TOP _ SW3
44
43
42
EN4
LREG1
VSENSE4
RST4
VIN2SENSE
VLR1
Buck1TOPFET
1
GU1
L1
2
PH1
Buck1LOWFET
3
GL1
4
PGND1
5
S2
40
39
38
37
GU2
36
PH2
35
GL2
34
PGND2
33
S4
32
TPS43340-Q1
Thermal Vias; use TI recommendation
for implementation
6
S1
7
VSENSE1
8
COMP1
9
RST1
10
SS1
11
VSUP
12
PGND3
S3
31
VSENSE2
30
COMP2
29
RST2
28
SS2
27
GND
26
RT
25
21
Rdelay
20
EN1
19
SYNC
18
EN3
17
EN2
16
SLEW
15
VSENSE3
14
COMP3
RST3
13
BOOT3
Exposed PAD Connected to Ground Plane for
electrical ground connection and thermal conduction
SS3
VReg 1
Input supply
41
VREG
45
BOOT2
46
GPULL
47
PH3
VOUT1
48
BOOT1
VLREG1
VIN
10 kW
VLREG1_Enable
EXTSUP
To Drain Node of
Top_SW2 MOSFET
1 kW
22
23
24
Buck2TOPFET
L2
VOUT2
Buck2LOWFET
VOUT3
10 kW
VOUT3_Enable
10 kW
VOUT2_Enable
Connection to backside of PCB through vias
10 kW
VOUT1_Enable
L3
Connection to topside of PCB through vias
Connection to ground plane of PCB through vias
Power bus
Voltage Output rails
Ground termination to ground plane or small
signal ground termination
10.3 Power Dissipation
The power dissipation depends on the MOSFET drive current and input voltage. The drive current is proportional
to the total gate charge of the external MOSFET.
10.3.1 Power Dissipation of Buck1 and Buck2 (VOUT1 and VOUT2)
PGate
drive
= Qg × VREG × fsw (Watts)
(45)
Assuming both high and low side MOSFETs are identical in a synchronous configuration, the total power
dissipation per buck is
PBuck1 = 2 × Qg × fsw × VREG (Watts)
(46)
10.3.2 Power Dissipation of Buck Converter Buck3 (VOUT3)
10.3.2.1 High-Side Switch
The power dissipation losses are applicable for positive output currents:
PHS-CON = IOUTr2 × rDS(on) × (VOUT / VIN) (Conduction losses)
PHS_SW = ½ × VSUP × IOUT × (tr + tf) × fSW (Switching losses)
PHS_Gate = 1 nC × fsw (Gate drive losses, valid at VREG = 5.8 V, VSUP = 4 V)
PHS_Total = PHS-CON + PH_SW + PHS_Gate
32
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(47)
(48)
(49)
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Power Dissipation (continued)
10.3.2.2 Low-Side Switch
The power dissipation losses are applicable for positive output currents.
PLS-CON = IOUTr2 × rDS(on) × ( 1 – VOUT / VIN) (Conduction losses)
PLS_SW = ½ × VSUP × IOUT × (tr + tf) × fSW (Switching losses)
PLS_Gate = 1 nC × fsw (Gate drive losses, valid at VVREG = 5.8 V, VSUP = 4 V)
PLS_DIODE = 2 × Vf × IOUT × fsw × tdead (Low-side body diode losses during dead time)
PLS_Total = PLS-CON + PL_SW + PLS_Gate + PLS_DIODE
(50)
(51)
(52)
(53)
(54)
10.3.2.3 Linear Regulator (LREG1)
PLREG1 = (VVLR1 – VLREG1) × IOUT
where
•
•
•
•
•
•
10.3.2.4
VOUT = Output voltage, VIN = Input voltage
IOUT = Output current, fSW = Switching frequency
tr = Rise time of switching node PH3
tf = Fall time of switching node PH3
VREG = FET gate drive voltage
Vf_diode = Low-side FET diode drop (conduction during dead time)
(55)
IC Power Consumption
PIC = Iq × VIN (Watts)
PTotal = PBuck1 and Buck2 + PHS_Total + PLS_Total + PLREG1+ PIC (Watts)
(56)
(57)
Table 2. Summary of Equations for Component Selection (1) (2)
PARAMETER OR
COMPONENT
Buck1 AND Buck2
Duty cycle D
D
Current-limit sense
resistor RS
Inductor selection L
Inductor ripple current
Output capacitor COUT
Input capacitor CIN
Soft-start CSS
Bootstrap capacitor CBoot
Compensation resistor for
GBW
(1)
(2)
RS
L=
VOUT
C IN
D
VIN
0.075
1.25 u IOUT max
200
u RS
¦SW
'IL(RIPPLE)
COUT
VOUT
Not Applicable
L=
'IOUT
4 u GBW u 'VOUT
0.25 u 'I OUT MAX
¦ SW u '9IN
1 $ u 'W
0.8
Qg
CBOOT =
'V
COUT
C IN
3.7
¦ SW
§
·
VOUT
V
u ¨ 1 OUT ¸
¦ SW u / ©
9IN ¹
GBW u 2S u COUT
R3
gm u K CFB u E
Buck3 is powered from Buck1 or Buck2.
Choose a current limit of 25% more than maximum
load.
Choose RS based on the current limit set for the
application.
Typically the ± inductor ripple current is 25% of
maximum load current.
'IOUT
Also consider that the ESR of the output capacitor
4 u GBW u 'VOUT influences the output-voltage ripple due to load steps.
0.25 u 'I OUT MAX Base the input-capacitor value on the input-voltage
¦ SW u '9IN
1 $ u 'W
0.8
Qg
CBOOT =
'V
CSS
COMMENTS
VIN
§
·
VOUT
V
u ¨ 1 OUT ¸ 'IL(RIPPLE)
¦ SW u / ©
9IN ¹
CSS
R3
Buck3
ripple desired.
Choose the soft-start time required, ∆t, and then
calculate CSS.
Choose based on the desired amount of ripple based
on FET gate charge and operating VIN.
GBW u 2S u COUT To determine resistor R3, assume GBW ≈ fsw / 5 to fsw
/ 20.
gm u Gm3 u E
KCFB = 0.125 / RSENSE
ß = VREF / VOUT
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Power Dissipation (continued)
Table 2. Summary of Equations for Component Selection()() (continued)
PARAMETER OR
COMPONENT
Buck1 AND Buck2
Compensation capacitor
for zero
C1 =
Buck3
COMMENTS
1
1
C1 can be also increased 2× for faster small-signal
C1 =
2 S u R3 u 0.1 u GBW
2 S u R3 u 0.1 u GBW settling at the expense of large step response (slew
rate on COMPx).
Compensation capacitor
for second pole
C2 =
Pole at low frequency with ¦
P1
high dc gain
Zero at control-loop pole
related to output filter LC
Second pole for type 2a
1
S u ¦SW u 5
1
2 S u C1 u ROUT _ OTA
C2 =
¦P1
1
S u ¦SW u 5
1
2 S u C1 u ROUT _ OTA
The value of C2 is also critical for buffering the noise
on the COMPx pin, and so the value of capacitance is
a trade-off between noise immunity and phase margin.
ROUT_OTA = 1 MΩ minimum
¦ Z1
1
¦ Z1
2 S u C1 u R3
Place zero at 0.05 to 0.1 × GBW (see comment on C1
1
above).
2 S u C1 u R3
¦PZ
1
¦PZ
2 S u C2 u R3
Place the second pole at or below half of the switching
1
frequency ƒsw, observing distance to GBW.
2 S u C2 u R3
Power Dissipation (W)
3
2
1
25
50
75
100
125
150
Ambient Temperature (qC)
Figure 24. Power Dissipation Derating Profile Based on High-K JEDEC PCB
10.4 Thermal Considerations
The TPS43340-Q1 is protected from overtemperature using an internal thermal shutdown circuit. If the die
temperature exceeds the thermal shutdown threshold (for example, due to fault conditions such as a short circuit
at the gate drivers or VREG), the device turns off, and restarts when the temperature has fallen by the
hysteresis.
Table 3. Low-Power-Mode Operation of the System
SETUP
Buck1 or Buck2 in LPM mode
Buck1 and Buck2 in LPM mode
Buck1 or Buck2 in PWM mode
Buck1 and Buck2 in PWM mode
LREG1
Low
High
N/A
LREG1 and Buck1 or Buck2 in LPM mode
LREG1 and Buck1 and Buck2 in LPM mode
34
SYNC
Low
QUIESCENT CURRENT
(TYP),
NO LOAD, 25°C
Approximately 30 µA
Approximately 35 µA
DESCRIPTION
Configuration for ignition-off applications with
standby functionality
Approximately 30-40 mA
Including switching currents
Approximately 30-40 mA
Including switching currents
Approximately 50 µA
Approximately 55 µA
Configuration for ignition-off applications with
standby functionality
Approximately 60 µA
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Thermal Considerations (continued)
Table 3. Low-Power-Mode Operation of the System (continued)
SETUP
QUIESCENT CURRENT
(TYP),
NO LOAD, 25°C
SYNC
LREG1 and Buck1 or Buck2 in PWM mode
LREG1 and Buck1 and Buck2 in PWM mode
High
DESCRIPTION
30-40 mA
Including switching currents
30-40 mA
Including switching currents
The synchronous buck converter Buck3 with the integrated FETs does not support LPM. Turning on Buck3
forces the system to operate in normal mode, and the quiescent current consumption increases.
Table 4. Input Voltage and Low-Power-Mode Operation
LOAD
BUCK
INPUT VOLTAGE
CHARGE PUMP
CURRENT OF
CONTROLLERS
AT VIN PIN
OF LREG1
LREG1
Buck1 AND Buck2
VIN > 9 V
7.5 V < VIN < 9 V
VIN < 7.5 V
VIN QUIESCENT
CURRENT (TYP),
NO LOAD, 25°C
N/A
OFF
LPM allowed
55 µA
< 2 mA
OFF
LPM allowed
55 µA
> 6 mA
ON
LPM allowed
N/A
ON
LPM not allowed
260 µA
2.6 mA
DESCRIPTION
Lowest current consumption of the
system at VIN (LREG1, Buck1 and
Buck2 enabled), typical ignition-off stayalive mode with up to three voltage rails
active
If VIN drops below 7.5 V, the buck
controllers Buck1 and Buck2 leave lowpower mode (LPM) and start PWM
operation, quiescent current of the
system increases. For applications that
use the LREG1 only as the standby
keep-alive supply, quiescent current is
still low.
Monitoring of the threshold for the charge pump of the low quiescent linear regulator LREG1 to be turned on
occurs at the VIN pin. If using LREG1 as post regulator with an input voltage VLR1 of less than 7.5 V, the charge
pump still stays off if operating within the required conditions for VIN and the load current. The sampling interval
for the foregoing voltage thresholds at the VIN pin is typically 60 µs.
10.4.1 Phase Configuration
The IC configuration has buck controller 1 and buck controller 2 switching 180 degrees out of phase. Buck
converter (Buck3) switches in phase with buck controller 1.
CONFIGURATION
Buck1
Buck2
Buck3
Phase
0º
180º
0º
DESCRIPTION
Buck1 and Buck2 out of phase, Buck1
and Buck3 in phase
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11 Device and Documentation Support
11.1 Device Support
11.1.1 Third-Party Products Disclaimer
TI'S PUBLICATION OF INFORMATION REGARDING THIRD-PARTY PRODUCTS OR SERVICES DOES NOT
CONSTITUTE AN ENDORSEMENT REGARDING THE SUITABILITY OF SUCH PRODUCTS OR SERVICES
OR A WARRANTY, REPRESENTATION OR ENDORSEMENT OF SUCH PRODUCTS OR SERVICES, EITHER
ALONE OR IN COMBINATION WITH ANY TI PRODUCT OR SERVICE.
11.2 Documentation Support
11.2.1 Related Documentation
For related documentation see the following:
TPS4334xEVM Evaluation Module, SLVU463
TPS43340-Q1 Family Design Checklist, SLVA614
11.3 Community Resources
The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective
contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of
Use.
TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration
among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help
solve problems with fellow engineers.
Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and
contact information for technical support.
11.4 Trademarks
PowerPAD, E2E are trademarks of Texas Instruments.
All other trademarks are the property of their respective owners.
11.5 Electrostatic Discharge Caution
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
11.6 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
12 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
36
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PACKAGE OPTION ADDENDUM
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10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
TPS43340QPHPRQ1
ACTIVE
HTQFP
PHP
48
1000
RoHS & Green
NIPDAU
Level-3-260C-168 HR
-40 to 125
43340Q1
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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