Typical Size
(6,3 mm x 6,4 mm)
TPS54110−Q1
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SLVS837 − JULY 2008
3-V TO 6-V INPUT, 1.5-A OUTPUT SYNCHRONOUS-BUCK PWM
SWITCHER WITH INTEGRATED FETs (SWIFT)
FEATURES
D Qualified for Automotive Applications
D Integrated MOSFET Switches for High
D
D
D
D
D
D
DESCRIPTION
As members of the SWIFT family of dc/dc regulators, the
TPS54110
low-input-voltage
high-output-current
synchronous-buck PWM converter integrates all required
active components. Included on the substrate with the
listed features are a true, high-performance, voltage error
amplifier that provides high performance under transient
conditions; an undervoltage-lockout circuit to prevent
start-up until the input voltage reaches 3 V; an internally
and externally set slow-start circuit to limit in-rush currents;
and a power-good output useful for processor/logic reset,
fault signaling, and supply sequencing.
Efficiency at 1.5-A Continuous Output Source
or Sink Current
0.9-V to 3.3-V Adjustable Output Voltage With
1% Accuracy
Externally Compensated for Design Flexibility
Fast Transient Response
Wide PWM Frequency: Fixed 350 kHz,
550 kHz, or Adjustable 280 kHz to 700 kHz
Load Protected by Peak Current Limit and
Thermal Shutdown
Integrated Solution Reduces Board Area and
Total Cost
The TPS54110 device is available in a thermally enhanced
20-pin TSSOP (PWP) PowerPAD package, which
eliminates bulky heatsinks. TI provides evaluation
modules and the SWIFT designer software tool to aid in
quickly achieving high-performance power supply designs
to meet aggressive equipment development cycles.
APPLICATIONS
D Low-Voltage, High-Density Systems With
D
D
D
Power Distributed at 5 V or 3.3 V
Point of Load Regulation for High
Performance DSPs, FPGAs, ASICs, and
Microprocessors
Broadband, Networking, and Optical
Communications Infrastructure
Portable Computing/Notebook PCs
Simplified Schematic
EFFICIENCY
vs
OUTPUT CURRENT
6.8 mH
Input
VIN
Output
PH
100
TPS54110
BOOT
10 mF
0.047 mF
95
100 mF
90
PGND
Efficiency − %
85
33 pF
COMP
19.1 kW
3.92 kW
VBIAS
VSENSE
0.1 mF
2700 pF
AGND
80
75
70
65
2.05 kW
2200 pF
60
55
3.92 kW
Compensation
Network
50
0
0.25
0.5
0.75
1
1.25
1.5
IO − Output Current − A
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments
semiconductor products and disclaimers thereto appears at the end of this data sheet.
PowerPAD and SWIFT are trademarks of Texas Instruments.
PRODUCTION DATA information is current as of publication date. Products
conform to specifications per the terms of Texas Instruments standard warranty.
Production processing does not necessarily include testing of all parameters.
Copyright 2008, Texas Instruments Incorporated
TPS54110−Q1
www.ti.com
SLVS837 − JULY 2008
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam during
storage or handling to prevent electrostatic damage to the MOS gates.
ORDERING INFORMATION(1)
PACKAGE(2)
TJ
−40°C to 125°C
HTSSOP − PW
ORDERABLE PART NUMBER
Reel of 2000
TPS54110QPWPRQ1
TOP-SIDE MARKING
54110Q
(1)
For the most current package and ordering information, see the Package Option Addendum at the end of this document, or see the TI web site
at http://www.ti.com.
(2)) Package drawings, thermal data, and symbolization are available at http://www.ti.com/packaging.
ABSOLUTE MAXIMUM RATINGS
over operating free-air temperature range unless otherwise noted(1)
I
Input
t voltage
lt
range, VI
O t t voltage
Output
lt
range, VO
S
Source
current,
t IO
Sink current
Voltage differential
VIN, SS/ENA, SYNC
−0.3 V to 7 V
RT
−0.3 V to 6 V
VSENSE
−0.3 V to 4 V
BOOT
−0.3 V to 17 V
VBIAS, PWRGD, COMP
−0.3 V to 7 V
PH
−0.6 V to 10 V
PH
Internally Limited
COMP, VBIAS
6 mA
PH
3.5 A
COMP
6 mA
SS/ENA,PWRGD
10 mA
AGND to PGND
±0.3 V
Continuous power dissipation
See Dissipation
Ratings Table
Operating virtual junction temperature range, TJ
−40°C to 150°C
Storage temperature, Tstg
−65°C to 150°C
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds
(1)
260°C
Stresses beyond those listed under “absolute maximum ratings” may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated under “recommended operating conditions” is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
RECOMMENDED OPERATING CONDITIONS
MIN
Input voltage range, VI
Operating junction temperature, TJ
DISSIPATION
(1)
2
MAX
UNIT
6
V
−40
125
°C
RATINGS(1) (2)
PACKAGE
THERMAL IMPEDANCE
JUNCTION-TO-AMBIENT
TA = 25°C
POWER RATING
TA = 70°C
POWER RATING
TA = 85°C
POWER RATING
20-pin PWP with solder
26.0°C/W
3.85 W(3)
2.12 W
1.54 W
20-pin PWP without solder
57.5°C/W
1.73 W
0.96 W
0.69 W
For more information on the PWP package, refer to TI technical brief, literature number SLMA002.
Test board conditions:
1. 3-in × 3-in, two layers, Thickness: 0.062 in
2. 1.5-oz copper traces located on the top of the PCB
3. 1.5-oz copper ground plane on the bottom of the PCB
4. Ten thermal vias (see recommended land pattern in application section of this data sheet)
(3) Maximum power dissipation may be limited by overcurrent protection.
(2)
NOM
3
TPS54110−Q1
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SLVS837 − JULY 2008
ELECTRICAL CHARACTERISTICS
TJ = −40°C to 125°C, VIN = 3 V to 6 V (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
fs = 350 kHz, SYNC ≤ 0.8 V, RT open
4.5
9.6
fs = 550 kHz, Phase pin open, SYNC 2.5 V,
RT open,
5.8
12.8
1
1.4
2.95
3
UNIT
SUPPLY VOLTAGE, VIN
VIN input voltage range
Quiescent current
3
Shutdown,
SS/ENA = 0 V
6
V
mA
UNDER VOLTAGE LOCK OUT
Start threshold voltage, UVLO
Stop threshold voltage, UVLO
2.70
Hysteresis voltage, UVLO
Rising and falling edge deglitch, UVLO(1)
2.80
V
0.12
V
2.5
µs
BIAS VOLTAGE
VO
Output voltage, VBIAS
I(VBIAS) = 0
2.70
2.80
Output current, VBIAS(2)
2.95
V
100
µA
CUMULATIVE REFERENCE
Vref
Accuracy
0.882
0.891
0.900
V
REGULATION
Line regulation(1) (3)
Load regulation (1) (3)
IL = 0.75 A,
fs = 350 kHz,
TJ = 85°C
0.05
IL = 0.75 A,
fs = 550 kHz,
TJ = 85°C
0.05
IL = 0 A to 1.5 A,
fs = 350 kHz,
TJ = 85°C
0.01
IL = 0 A to 1.5 A
fs = 550 kHz,
TJ = 85°C
0.01
SYNC ≤ 0.8 V,
RT open
265
350
440
SYNC ≥ 2.5 V,
RT open
415
550
680
RT = 180 kΩ (1% resistor to AGND)(1)
252
280
308
RT = 100 kΩ (1% resistor to AGND)
460
500
540
RT = 68 kΩ (1% resistor to AGND)(1)
663
700
762
%/V
%/A
OSCILLATOR
Internally set free-running
free running frequency range
Externally set free-running
free running frequency range
High-level threshold voltage, SYNC
2.5
Pulse duration,
Frequency range, SYNC(1)
V
700
kHz
ns
0.75
Ramp amplitude (peak-to-peak)(1)
V
1
Minimum controllable on time(1)
Maximum duty cycle
0.8
50
330
Ramp valley(1)
kHz
V
Low-level threshold voltage, SYNC
SYNC(1)
kHz
V
200
ns
90%
(1)
Specified by design
(2) Static resistive loads only
(3) Specified by the circuit used in Figure 9.
3
TPS54110−Q1
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SLVS837 − JULY 2008
ELECTRICAL CHARACTERISTICS (continued)
TJ = −40°C to 125°C, VIN = 3 V to 6 V (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
ERROR AMPLIFIER
Error-amplifier open loop voltage gain
1 kΩ COMP to AGND(1)
90
110
Error-amplifier unity gain bandwidth
Parallel 10 kΩ, 160 pF COMP to AGND (1)
3
5
Error-amplifier common-mode input voltage
range
Powered by internal LDO(1)
0
IIB
Input bias current, VSENSE
VSENSE = Vref
VO
Output voltage slew rate (symmetric),
COMP(1)
60
dB
MHz
VBIAS
V
250
nA
1.2
V/µs
PWM COMPARATOR
PWM comparator propagation delay time,
PWM comparator input to PH pin (excluding
dead time)
10 mV overdrive(1)
70
85
ns
1.20
1.40
V
SLOW-START/ENABLE
Enable threshold voltage, SS/ENA
0.82
Enable hysteresis voltage, SS/ENA(1)
Falling-edge deglitch, SS/ENA(1)
Internal slow-start time
Charge current, SS/ENA
SS/ENA = 0 V
Discharge current, SS/ENA
SS/ENA = 0.2 V,
VI = 2.7 V
0.03
V
2.5
µs
2.6
3.35
4.1
2.5
5
8
ms
µA
1.2
2.3
4
mA
POWER GOOD
Power-good threshold voltage
Power-good hysteresis
VSENSE falling
voltage(1)
Power-good falling-edge deglitch(1)
93
%Vref
3
%Vref
µs
35
Output saturation voltage, PWRGD
I(sink) = 2.5 mA
Leakage current, PWRGD
VI = 5.5 V
0.18
0.30
V
1
µA
CURRENT LIMIT
Current limit trip point
Current-limit
VI = 3 V, output shorted(1)
3.0
VI = 6 V, output shorted(1)
3.5
A
Current-limit leading edge blanking time
100
ns
Current-limit total response time
200
ns
THERMAL SHUTDOWN
Thermal-shutdown trip point(1)
Thermal-shutdown
135
hysteresis(1)
150
165
°C
°C
10
OUTPUT POWER MOSFETS
rDS(on)
(1)
Power MOSFET switches(3)
IO = 1.5 A,
VI = 6 V(2)
240
480
IO = 1.5 A,
VI = 3 V(2)
345
690
Specified by design
Matched MOSFETs, low side rDS(on) production tested, high side rDS(on) specified by design
(3) Includes package and bondwire resistance
(2)
4
mΩ
TPS54110−Q1
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SLVS837 − JULY 2008
PIN ASSIGNMENTS
PWP PACKAGE
(TOP VIEW)
AGND
VSENSE
COMP
PWRGD
BOOT
PH
PH
PH
PH
PH
1
2
3
4
5
6
7
8
9
10
20
19
18
17
16
15
14
13
12
11
RT
SYNC
SS/ENA
VBIAS
VIN
VIN
VIN
PGND
PGND
PGND
Terminal Functions
TERMINAL
NAME
DESCRIPTION
NO.
AGND
1
Analog ground—internally connected to the sensitive analog-ground circuitry. Connect to PGND and PowerPAD.
BOOT
5
Bootstrap input. 0.022-µF to 0.1-µF low-ESR capacitor connected from BOOT to PH generates floating drive for the
high-side FET driver.
COMP
3
Error amplifier output. Connect compensation network from COMP to VSENSE.
PGND
11−13
Power ground. High current return for the low-side driver and power MOSFET. Connect PGND with large copper areas to the
input and output supply returns, and negative terminals of the input and output capacitors. Connect to AGND and
PowerPAD.
PH
6−10
Phase input/output. Junction of the internal high and low-side power MOSFETs, and output inductor.
PWRGD
4
Power-good open drain output. High when VSENSE ≥ 93% Vref, otherwise PWRGD is low. Note that output is low when
SS/ENA is low or internal shutdown signal active.
RT
20
Frequency setting resistor input. Connect a resistor from RT to AGND to set the switching frequency, fs.
SS/ENA
18
Slow-start/enable input/output. Dual-function pin that provides logic input to enable/disable device operation and capacitor
input to externally set the start-up time.
SYNC
19
Synchronization input. Dual-function pin that provides logic input to synchronize to an external oscillator or pin select
between two internally set switching frequencies. When used to synchronize to an external signal, a resistor must be
connected to the RT pin.
VBIAS
17
Internal bias regulator output. Supplies regulated voltage to internal circuitry. Bypass VBIAS pin to AGND pin with a high
quality, low ESR 0.1-µF to 1.0-µF ceramic capacitor.
14−16
Input supply for the power MOSFET switches and internal bias regulator. Bypass VIN pins to PGND pins close to device
package with a high quality, low ESR 1-µF to 10-µF ceramic capacitor.
VIN
VSENSE
2
Error amplifier inverting input.
5
TPS54110−Q1
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SLVS837 − JULY 2008
FUNCTIONAL BLOCK DIAGRAM
VBIAS
AGND
VIN
Enable
Comparator
SS/ENA
Falling
Edge
Deglitch
1.2 V
Hysteresis: 0.03 V
2.5 µs
VIN UVLO
Comparator
VIN
2.95 V
Hysteresis: 0.16 V
REG
VBIAS
SHUTDOWN
VIN
ILIM
Comparator
Thermal
Shutdown
150°C
3−6V
Leading
Edge
Blanking
Falling
and
Rising
Edge
Deglitch
100 ns
BOOT
2.5 µs
SS_DIS
SHUTDOWN
Internal/External
Slow-start
(Internal Slow-start Time = 3.35 ms
PH
+
−
R Q
Error
Amplifier
Reference
VREF = 0.891 V
S
PWM
Comparator
CO
Adaptive Dead-Time
and
Control Logic
VIN
OSC
PGND
Powergood
Comparator
PWRGD
VSENSE
Falling
Edge
Deglitch
0.93 Vref
TPS54110
Hysteresis: 0.03 Vref
VSENSE
6
COMP
RT
SYNC
SHUTDOWN
LOUT
35 µs
VO
TPS54110−Q1
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SLVS837 − JULY 2008
TYPICAL CHARACTERISTICS
IO = 1.5 A
0.5
VI = 5 V
0.4
0.3
0.2
0.1
IO = 1.5 A
0.3
0.2
0.1
0
−40
0
−40
0
25
85
125
0
125
85
TJ − Junction Temperature − °C
TJ − Junction Temperature − °C
Figure 1
600
RT = 100 k
500
400
RT = 180 k
300
25
85
125
0.893
0.891
0.889
0.887
SYNC ≤ 0.8 V
350
250
−40
0
0
Figure 4
25
85
125
0.8910
0.8890
fS = 350 kHz
0.8870
3
3.80
Phase
−80
−100
60
−120
40
Gain
20
−140
−160
0
−180
−20
−200
10 k 100 k 1 M 10 M
100
1k
f − Frequency − Hz
Figure 7
Phase − Degrees
−40
Internal Slow-Start Time − ms
−20
−60
80
6
INTERNAL SLOW-START TIME
vs
JUNCTION TEMPERATURE
0
100
4
5
VI − Input Voltage − V
Figure 6
ERROR AMPLIFIER
OPEN LOOP RESPONSE
120
125
0.8930
Figure 5
RL= 10 kΩ,
CL = 160 pF,
TA = 25°C
85
TA = 85°C
TJ − Junction Temperature − °C
140
25
0.8850
0.885
−40
TJ − Junction Temperature − °C
Gain − dB
450
0.8950
VO − Output Voltage Regulation − V
Vref − Voltage Reference − V
700
10
550
OUTPUT VOLTAGE REGULATION
vs
INPUT VOLTAGE
0.895
0
SYNC ≥ 2.5 V
Figure 3
RT = 68 k
0
650
TJ − Junction Temperature − °C
VOLTAGE REFERENCE
vs
JUNCTION TEMPERATURE
800
200
−40
750
Figure 2
EXTERNALLY SET OSCILLATOR
FREQUENCY
vs
JUNCTION TEMPERATURE
f − Externally Set Oscillator Frequency − kHz
25
f − Internally Set Oscillator Frequency −kHz
0.4
Drain-Source On-State Resistance − Ω
Drain-Source On-State Resistance − Ω
0.6
VI = 3.3 V
INTERNALLY SET OSCILLATOR
FREQUENCY
vs
JUNCTION TEMPERATURE
DRAIN-SOURCE ON-STATE RESISTANCE
vs
JUNCTION TEMPERATURE
DRAIN-SOURCE ON-STATE RESISTANCE
vs
JUNCTION TEMPERATURE
3.65
3.50
3.35
3.20
3.05
2.90
2.75
−40
0
25
85
TJ − Junction Temperature − °C
125
Figure 8
7
TPS54110−Q1
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SLVS837 − JULY 2008
APPLICATION INFORMATION
Figure 9 shows the schematic diagram for a typical TPS54110 application. The TPS54110 can provide up to 1.5 A of output
current at a nominal output voltage of 3.3 V. For proper thermal performance, the exposed PowerPAD under the device
must be soldered down to the printed-circuit board.
R7
10 kW
VIN (4.5 − 5.5 V)
C8
2200 pF
PWRGD
+
R4
71.5 kW
C1
10 mF
U2
TPS54110PWP
20
19
18
17
16
15
14
C5
.047 mF
C4
0.1 mF
C9
10 mF
13
12
11
RT
AGND
SYNC
VSENSE
SS/ENA
COMP
VBIAS PWRGD
VIN
PH
VIN
PH
PGND
PH
PGND
PH
C6
2700 pF
2
3
4
5
BOOT
VIN
R3
19.1 kW
1
R5
2.05 kW
R1
10.7 kW
C7
33 pF
R2
3.92 kW
C3 0.047 mF
6
7
8
L1
6.8 mH
9
10
PGND
PH
PwrPd
1
2
3.3 V at 1.5 A
C2
100 mF
21
Figure 9. Application Schematic
DESIGN PROCEDURE
The following design procedure can be used to select
component values for the TPS54110. Alternately, the
SWIFT Designer Software can be used to generate a
complete design. The SWIFT Designer Software uses an
iterative design procedure to access a comprehensive
database of components when generating a design. This
section presents a simplified discussion of the design
process.
SWITCHING FREQUENCY
The switching frequency is set within the range of 280 kHz
to 700 kHz by connecting a resistor from the RT pin to
AGND. Equation (1) is used to determine the proper RT
value.
RT(kW) + 100 500 kHz
ƒ
s(kHz)
DESIGN PARAMETERS
The required parameters to begin the design process and
values for this design example are listed in Table 1.
Table 1. Design Parameters
8
As an additional constraint, the design is set up to be small
size and low component height.
DESIGN PARAMETER
EXAMPLE VALUE
Input voltage range
4.5 to 5.5 V
(1)
In this example, the timing-resistor value chosen for R4 is
71.5 kΩ, setting the switching frequency to 700 kHz.
Alternately, the TPS54110 can be set to preprogrammed
switching frequencies of 350 kHz or 550 kHz by
connecting pins RT and SYNC as shown in Table 2.
Output voltage
3.3 V
Input ripple voltage
100 mV
Output ripple voltage
30 mV
FREQUENCY
RT
SYNC
Output current rating
1.5 A
350 kHz
Float
Float or AGND
Operating frequency
700 kHz
550 kHz
Float
2.5 V
Table 2. Selecting the Switching Frequency
TPS54110−Q1
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SLVS837 − JULY 2008
INPUT CAPACITORS
The TPS54110 requires an input decoupling capacitor
and, depending on the application, a bulk input capacitor.
The minimum value for the decoupling capacitor, C9, is
10 µF. A high-quality ceramic type X5R or X7R with a
voltage rating greater than the maximum input voltage is
recommended. A bulk input capacitor may be needed,
especially if the TPS54110 circuit is not located within
approximately 2 inches from the input voltage source. The
capacitance value is not critical, but the voltage rating must
be greater than the maximum input voltage including ripple
voltage. The capacitor must filter the input ripple voltage to
acceptable levels.
Input ripple voltage can be approximated by equation 2:
DVIN +
I OUT(MAX)
C BULK
ǒ
0.25
) I OUT(MAX)
ƒsw
Ǔ
ESR MAX
KIND is a coefficient that represents the amount of inductor
ripple current relative to the maximum output current. For
designs using low-ESR capacitors such as ceramics, use
KIND = 0.2. When using higher ESR output capacitors,
KIND = 0.1 yields better results. If higher ripple currents can
be tolerated, KIND can be increased allowing for a smaller
output-inductor value.
This example design uses KIND = 0.2, yielding a minimum
inductor value of 6.29 uH. The next-higher standard value
of 6.8 uH is chosen for this design. If a lower inductor value
is desired, a larger amount of ripple current must be
tolerated.
The RMS-current and saturation-current ratings of the
output filter inductor must not be exceeded. The RMS
inductor current can be found from equation 5:
(2)
I
Where
IOUT(MAX) is the maximum load current,
ƒSW is the switching frequency,
CBULK is the bulk capacitor value and
ESR MAX is the maximum series resistance of the
bulk capacitor.
Worst-case RMS ripple current is approximated by
equation 3:
I
I
CIN
+
OUT(MAX)
2
(3)
In this case the input ripple voltage is 66 mV with a 10-uF
bulk capacitor. Figure 15 shows the measured ripple
waveform. The RMS ripple current is 0.75 A. The
maximum voltage across the input capacitors is VINMAX +
∆VIN/2. The bypass capacitor and input bulk capacitor are
each rated for 6.3 V and a ripple-current capacity of 1.5 A,
providing some margin. It is very important that the
maximum ratings for voltage and current are not exceeded
under any circumstance.
OUTPUT FILTER COMPONENTS
Two components, L1 and C2, are selected for the output
filter. Since the TPS54110 is an externally-compensated
device, a wide range of filter-component types and values
are supported.
Inductor Selection
Use equation 4 to calculate the minimum value of the
output inductor:
V
L
MIN
+
V
OUT
IN(MAX)
ǒVIN(MAX) * VOUTǓ
K
IND
I
OUT
F
L(RMS)
(4)
Ǹ
1
)
OUT(MAX) 12
ǒ
V
V
ǒVIN(MAX)–VOUTǓ
OUT
L
IN(MAX)
F
OUT
SW
0.8
Ǔ
2
(5)
The peak inductor current is determined from equation 6:
I L(PK) + I OUT(MAX) )
VOUT
1.6
V
ǒVIN(MAX) * VOUTǓ
IN(MAX)
L
OUT
F
SW
(6)
For this design, the RMS inductor current is 1.503 A and
the peak inductor current is 1.673 A. The inductor chosen
is a Coilcraft DS3316P-682 6.8 µH. It has a saturationcurrent rating of 2.8 A and an RMS current rating of 2.2 A,
easily meeting these requirements.
Capacitor Selection
The important design parameters for the output capacitor
are dc voltage, ripple current, and equivalent series
resistance (ESR). The dc-voltage and ripple-current
ratings must not be exceeded. The ESR rating is important
because along with the inductor current it determines the
output ripple voltage level. The actual value of the output
capacitor is not critical, but some practical limits do exist.
Consider the relationship between the desired closed-loop
crossover frequency of the design and LC corner
frequency of the output filter. In general, it is desirable to
keep the closed-loop crossover frequency at less than
one-fifth of the switching frequency. With high switching
frequencies such as the 700 kHz frequency of this design,
internal circuit limitations of the TPS54110 limit the
practical maximum crossover frequency to about 100 kHz.
To allow adequate phase gain in the compensation
network, set the LC corner frequency to approximately one
decade below the closed-loop crossover frequency. This
limits the minimum capacitor value for the output filter to:
C
SW
+
I2
OUT(MIN)
+
1
LOUT
ǒ
K
2pƒCO
Ǔ
2
(7)
9
TPS54110−Q1
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SLVS837 − JULY 2008
where K is the frequency multiplier for the spread between
fLC and fCO. K should be between 5 and 15, typically 10 for
one decade of difference.
For a desired crossover of 60 kHz, K=10 and a 6.8 µH
inductor, the minimum value for the output capacitor is
100 µF. The selected output capacitor must be rated for a
voltage greater than the desired output voltage plus one
half the ripple voltage. Any derating factors must also be
included. The maximum RMS ripple current in the output
capacitor is given by equation 8:
ICOUT(RMS) + 1
Ǹ12
ȡ VOUT ǒVIN(MAX)–VOUTǓ ȣ
ȧVIN(MAX) LOUT FSW NCȧ
Ȣ
Ȥ(8)
where NC is the number of output capacitors in parallel.
The maximum ESR of the output capacitor is determined
by the allowable output ripple specified in the initial design
parameters. The output ripple voltage is the inductor ripple
current times the ESR of the output filter so the maximum
specified ESR as listed in the capacitor data sheet is given
by equation 9:
ESR MAX + N
ǒ
V IN(MAX)
C
V
OUT
L
OUT
F
SW
0.8
ǒVIN(MAX)–VOUTǓ
Ǔ
DV p–p(MAX)
(9)
For this design example, a single 100 µF output capacitor
is chosen for C2. The calculated RMS ripple current is
80 mA and the maximum ESR required is 87 mΩ. An
example of a suitable capacitor is the Sanyo Poscap
6TPC100M, rated at 6.3 V with a maximum ESR of 45
milliohms and a ripple-current rating of 1.7 A.
Other capacitor types work well with the TPS54110,
depending on the requirements of the application.
A number of considerations apply when designing
compensation networks for the TPS54110. The
compensated error-amplifier gain must not be limited by
the open-loop amplifier gain characteristics and must not
produce excessive gain at the switching frequency. Also,
the closed-loop crossover frequency must be set less than
one-fifth of the switching frequency, and the phase margin
at crossover must be greater than 45 degrees. The general
procedure outlined here meets these requirements
without going into great detail about the theory of loop
compensation.
First, calculate the output filter LC corner frequency using
equation 10:
ƒ
There are a number of different ways to design a
compensation network. This procedure outlines a
relatively simple procedure that produces good results
with most output filter combinations. Use the SWIFT
Designer Software for designs with unusually high
closed-loop crossover frequencies; with low-value,
low-ESR output capacitors such as ceramics; or if you are
unsure about the design procedure.
10
+
Ǹ
2p L
1
C
OUT OUT
(10)
For the design example, fLC = 6103 Hz.
Choose a closed-loop crossover frequency greater than
fLC and less than one-fifth of the switching frequency. Also,
keep the crossover frequency below 100 kHz, as the error
amplifier may not provide the desired gain at higher
frequencies. The 60-kHz crossover frequency chosen for
this design provides comparatively wide loop bandwidth
while still allowing adequate phase boost to ensure
stability.
Next, the values for the compensation components that
set the poles and zeros of the compensation network are
calculated. Assuming an R1 value > than R5 and a C6
value > C7, the pole and zero locations are given by
equations 11 through 14:
ƒ
Z1
+
1
2pR3C6
(11)
ƒ
Z2
+
1
2pR1C8
(12)
ƒ
P1
+
1
2pR5C8
(13)
ƒ
P2
+
1
2pR3C7
(14)
COMPENSATION COMPONENTS
The external compensation used with the TPS54110
allows for a wide range of output-filter configurations. A
large range of capacitor values and dielectric types are
supported. The design example uses type 3 compensation
consisting of R1, R3, R5, C6, C7 and C8. Additionally, R2
and R1 form a voltage-divider network that sets the output
voltage. These component reference designators are the
same as those used in the SWIFT Designer Software.
LC
Additionally there is a pole at the origin, which has unity
gain at a frequency:
ƒ
INT
+
1
2pR1C6
(15)
This pole is used to set the overall gain of the compensated
error amplifier and determines the closed loop crossover
frequency. Since R1 is given as 10 kΩ and the crossover
frequency is selected as 60 kHz, the desired fINT is
calculated from equation 16:
ƒ
INT
+
10*0.74
ƒ
CO
2
And the value for C6 is given by equation 17:
(16)
TPS54110−Q1
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SLVS837 − JULY 2008
C6 +
1
2pR1ƒ
INT
(17)
Since C6 is calculated to be 2900 pF, and the location of
the integrator crossover frequency is important in setting
the overall loop crossover, adjust the value of R1 so that
C6 is a standard value of 2700 pF, using equation 18:
R1 +
1
2pC6ƒ
LC
(18)
The value for R1 is 10.7 KΩ
The first zero, fZ1 is located at one half the output filter LC
corner frequency, so R3 is calculated from:
1
R3 +
pC6ƒ
LC
(19)
1
2pR1ƒ
LC
(20)
The first pole, fP1 is located to coincide with output filter
ESR zero frequency. This frequency is given by:
ƒ
ESR0
+
1
2pR
C
ESR OUT
(21)
where RESR is the equivalent series resistance of the
output capacitor.
In this case, the ESR zero frequency is 35.4 kHz, and R5
is calculated from:
R5 +
C7 +
1
2pC8 ƒ
1
8pR3ƒ
CO
(23)
Finally, calculate the R2 resistor value for the output
voltage of 3.3 V using equation 24:
R2 +
The second zero, fZ2 is located at the output filter LC corner
frequency, so C8 is calculated from:
C8 +
The final pole is placed at a frequency high enough above
the closed-loop crossover frequency to avoid causing an
excessive phase decrease at the crossover frequency
while still providing enough attenuation so that there is little
or no gain at the switching frequency. The fP2 pole location
for this circuit is set to 4 times the closed-loop crossover
frequency and the last compensation component value C7
is derived:
R1 0.891
V
–0.891
OUT
(24)
For this TPS54110 design, use R1 = 10.7 kΩ instead of
10.0 kΩ. R2 is then 3.92 kΩ.
Since capacitors are only available in a limited range of
standard values, the nearest standard value was chosen
for each capacitor. The measured closed-loop response
for this design is shown in Figure 19.
BIAS AND BOOTSTRAP CAPACITORS
Every TPS54110 design requires a bootstrap capacitor
(C3), and a bias capacitor (C4). The bootstrap capacitor
must be between 0.022 µF and 0.1 µF. This design uses
0.047 µF. The bootstrap capacitor is located between the
PH pins and BOOT. The bias capacitor is connected
between the VBIAS pin and AGND. Recommended
values are 0.1 µF to 1.0 µF. This design uses 0.1 µF. Use
high-quality ceramic capacitors with X7R or X5R grade
dielectric for temperature stability. Place them as close to
the device pins as possible.
ESR
(22)
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SLVS837 − JULY 2008
GROUNDING AND PowerPAD LAYOUT
The TPS54110 has two internal grounds (analog and
power). Inside the TPS54110, the analog ground connects
all noise-sensitive signals, while the power ground
connects the noisier power signals. The PowerPAD must
be tied directly to AGND. Noise injected between the two
grounds can degrade the performance of the TPS54110,
particularly at higher output currents. However, ground
noise on an analog ground plane can also cause problems
with some of the control and bias signals. For these
reasons, separate analog and power ground planes are
recommended. Tie these two planes together directly at
the IC to reduce noise between the two grounds. The only
components that tie directly to the power-ground plane are
the input capacitor, the output capacitor, the input voltage
decoupling capacitor, and the PGND pins of the
TPS54110. The layout of the TPS54110 evaluation
module represents recommended layout for a 2-layer
board. Documentation for the TPS54110 evaluation
module is obtained from the Texas Instruments web site
under the TPS54110 product folder and in the application
note, TI literature number SLVA109.
6 PL ∅ 0.0130
4 PL ∅ 0.0180
Connect Pin 1 to Analog Ground Plane
in This Area for Optimum Performance
0.0227
0.0600
0.0400
0.2560
0.2454
0.0400
0.0600
Minimum Recommended Top
Side Analog Ground Area
LAYOUT CONSIDERATIONS FOR THERMAL
PERFORMANCE
For operation at full rated load current, the analog ground
plane must provide adequate heat dissipation area. A
3-inch-by-3-inch plane of 1-ounce copper is
recommended, though not mandatory, depending on
ambient temperature and airflow. Most applications have
larger areas of internal ground plane available. Connect
the PowerPAD to the largest area available. Additional
areas on the top or bottom layers also help dissipate heat.
Use any area available when 1.5-A or greater operation is
desired. Connect the exposed area of the PowerPAD to
the analog ground-plane layer with 0.013-inch-diameter
vias to avoid solder wicking through the vias. An adequate
design includes six vias in the PowerPAD area with four
additional vias located under the device package. The size
of the vias under the package, but not in the exposed
thermal pad area, can be increased to 0.018. Additional
vias in areas not under the device package enhance
thermal performance.
Minimum Recommended Thermal Vias: 6 × .013 dia.
Inside Powerpad Area 4 × .018 dia. Under Device as Shown.
Additional .018 dia. Vias May be Used if Top Side Analog
Ground Area is Extended.
ÓÓÓ
ÓÓÓ
ÓÓÓ
ÓÓÓ
ÓÓÓ
ÓÓÓ
0.0150
0.06
0.1010
0.0256
0.1700
0.1340
0.0620
0.0400
Minimum Recommended Exposed
Copper Area For Powerpad. 5mm
Stencils may Require 10 Percent
Larger Area
Figure 10. Recommended Land Pattern for 20-Pin PWP PowerPAD
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PERFORMANCE GRAPHS
All performance data shown for VI = 5 V, VO = 3.3 V, fs = 700 kHz, TA = 25°C, Figure 9
EFFICIENCY
vs
OUTPUT CURRENT
POWER DISSIPATION
vs
OUTPUT CURRENT
1.2
100
LOAD REGULATION
vs
OUTPUT CURRENT
0.05
95
Efficiency − %
85
80
75
70
65
60
1
Output Voltage Varistion − %
P D − Power Dissipation − W
0.04
90
0.8
0.6
0.4
0.2
0.03
0.02
0.01
0
−0.01
−0.02
−0.03
−0.04
55
50
0
0.25
0.5
0.75
1
1.25
1.5
−0.05
0
0
0.25
0.5
0.75
1
1.25
0
1.5
0.25
IO − Output Current − A
IO − Output Current − A
Figure 11
Figure 12
LINE REGULATION
vs
INPUT VOLTAGE
0.5
0.75
1
1.25
1.5
IO − Output Current − A
Figure 13
OUTPUT VOLTAGE RIPPLE
INPUT VOLTAGE RIPPLE
VI = 50 mV/div (AC)
0.015
VO = 10 mV/div (AC)
0.01
IO = 0.75 A
0.005
0
V(phase)= 2 V/div
V(phase)= 2 V/div
IO = 0 A
−0.005
IO = 1.5 A
−0.01
−0.015
−0.02
4.5
4.75
5
5.25
5.5
Time = 500 ns/div
Time = 500 ns/div
VI − Input Voltage − V
OUTPUT VOLTAGE TRANSIENT
RESPONSE
Figure 15
Figure 16
MEASURED LOOP RESPONSE
START UP WAVEFORM
60
VO= 10 mV/div (AC)
VI = 2 V/div
180
150
50
Phase
40
60
20
IO= 1 V/div
VO = 1 V/div
10
Time = 5 ms/div
30
Gain
0
0
−10
−30
−20
−60
−30
−90
−40
−120
−50
−150
−180
1M
−60
Time = 200 ms/div
120
90
30
Phase − Degrees
Figure 14
Gain − dB
Output Voltage Varistion − %
0.02
100
1k
10 k
100 k
f − Frequency − Hz
Figure 17
Figure 18
Figure 19
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SLVS837 − JULY 2008
Very−small form-factor application
R7
10 kW
VIN
C8
560 pF
PWRGD
+
R4
71.5 kW
C1
OPEN
U2
TPS54110PWP
20
19
18
17
16
15
14
C5
OPEN
C4
0.1 mF
C9
10 mF
13
12
11
RT
AGND
SYNC
SS/ENA
VSENSE
COMP
VBIAS PWRGD
VIN
BOOT
VIN
PH
VIN
PH
PGND
PH
PGND
PH
R3
1.74 kW
1
C6
1000 pF
2
3
4
5
R5
432 W
R1
10.0 kW
R2
14.7 kW
C7
47 pF
C3 0.047 mF
6
7
8
10
PGND
PH
PwrPd
21
L1
1 mH
9
1
2
1.5 V at 1.5 A
C2
10 mF
Figure 20. Small Form-Factor Reference Design
Figure 20 shows an application schematic for a TPS54110 application designed for extremely small size. To achieve this
goal, the design procedure given in the previous application circuit is modified. For example, to use a small-footprint
Coilcraft DO3314-103MX inductor, the maximum-allowable inductor ripple current was increased above that normally
specified. A small 0805 10-µF ceramic capacitor is used in the output filter. All the additional components are 0402 case
size.
14
TPS54110−Q1
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SLVS837 − JULY 2008
All performance data shown for VI = 5 V, VO = 1.5 V, FS = 700 kHz, TA = 25_C, Figure 20
POWER DISSIPATION
vs
OUTPUT CURRENT
EFFICIENCY
vs
OUTPUT CURRENT
0.1
95
0.08
85
VI = 5 V
80
75
70
65
60
1.2
Output Voltage Varistion − %
P D − Power Dissipation − W
100
90
Efficiency − %
LOAD REGULATION
vs
OUTPUT CURRENT
1
0.8
0.6
VI = 5 V
0.4
VI = 3.3 V
0.2
55
0
0
0.25
0.5
0.75
1
1.25
0
1.5
0.04
0.02
VI = 3.3 V
0
−0.02
−0.04
VI = 5 V
−0.06
−0.08
VI = 3.3 V
50
0.06
0.25
0.5
0.75
1
1.25
1.5
−0.1
0
0.25
IO − Output Current − A
IO − Output Current − A
Figure 21
0.5
0.75
1
1.25
IO − Output Current − A
Figure 22
LINE REGULATION
vs
INPUT VOLTAGE
1.5
Figure 23
OUTPUT VOLTAGE RIPPLE
INPUT VOLTAGE RIPPLE
0.02
Output Voltage Varistion − %
VI = 50 mV/div (AC)
VO = 20 mV/div (AC)
0.015
0.01
IO = 0.75 A
0.005
0
IO = 0 A
V(phase)= 2 V/div
V(phase)= 2 V/div
−0.005
−0.01
IO = 1.5 A
−0.015
−0.02
3
3.5
4
4.5
5
5.5
6
Time = 500 ns/div
Time = 500 ns/div
VI − Input Voltage − V
Figure 26
Figure 25
Figure 24
OUTPUT VOLTAGE TRANSIENT
RESPONSE
VO= 20 mV/div (AC)
START UP WAVEFORM
VI = 1 V/div
VO = 500 mV/div
IO= 1 V/div
Time = 200 ms/div
Figure 27
Time = 5 ms/div
Figure 28
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SLVS837 − JULY 2008
TWO-OUTPUT SEQUENCED-STARTUP APPLICATION
PWRGD_3P3
VI 5 V
R7
C8
560 pF
10 kΩ
+
C1
470 µF
R4
71.5 kΩ
U1
TPS54110PWP
20
19
18
17
16
C4
0.1 µF
C9
10 µF
15
14
13
12
11
RT
AGND
SYNC VSENSE
SS/ENA COMP
VBIAS PWRGD
VIN
BOOT
VIN
PH
VIN
PH
PGND
PH
PH
PGND
PGND
PH
PWPD
21
1
2
3
4
5
6
C6
R3
1.74 kΩ 1000 pF
R1
10 kΩ
R2
3.74 kΩ
C7
47 pF
C3 0.047 µF
7
8
9
L1
1 µH
10
1
C2
10 µF
PWRGD_1P5
C13
560 pF
10 kΩ
U2
TPS54110PWP
20
19
18
17
16
C10
0.1 µF
C15
10 µF
15
14
13
12
11
RT
AGND
SYNC VSENSE
SS/ENA COMP
VBIAS PWRGD
VIN
BOOT
VIN
PH
VIN
PH
PGND
PH
PH
PGND
PGND
PH
PWPD
21
VOUT1
2
3.3 V at 1.5 A
R8
R9
71.5 kΩ
R5
432 Ω
1
C5
R6
1.74 kΩ 1000 pF
2
C11
3
47 pF
4
5 C14 0.047 µF
6
7
8
9
10
R12
432 Ω
R11
10 kΩ
R10
14.7 kΩ
L2
1 µH
1
VOUT2
2
1.5 V at 1.5 A
C12
10 µF
Figure 29. TPS54110 Sequencing Application Circuit
In Figure 29, the power-good output of U1 is used as a
sequencing signal in a two-output design. Connecting the
PWRGD pin of U1 to the SS/ENA pin of U2 causes the
1.5-V output to ramp up after the 3.3-V output is within
regulation. Figure 30 shows the startup waveforms
associated with this circuit.
When VIN reaches the UVLO-start threshold, the U1
output ramps up towards the 3.3-V set point. After the
output reaches 90 percent of 3.3 V, the U1 asserts the
power-good signal driving the U2 SS/ENA input high. The
output of U2 then ramps up towards the final output set
point of 1.5 V.
VIN − 5 V/div
U1 − VOUT1
3.3 − 2 V/div
U1 PWRGD
− 5 V/div
U2 − VOUT2 1.5 − 2 V/div
Figure 30. Sequencing Start Up Waveforms
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TPS54110−Q1
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SLVS837 − JULY 2008
DETAILED DESCRIPTION
Under Voltage Lock Out (UVLO)
The TPS54110 incorporates an under voltage lockout
circuit to keep the device disabled when the input voltage
(VIN) is insufficient. During power up, internal circuits are
held inactive until VIN exceeds the nominal UVLO
threshold voltage of 2.95 V. Once the UVLO start threshold
is reached, device start-up begins. The device operates
until VIN falls below the nominal UVLO stop threshold of
2.8 V. Hysteresis in the UVLO comparator, and a 2.5-µs
rising and falling edge deglitch circuit reduce the likelihood
of shutting the device down due to noise on VIN.
Slow-Start/Enable (SS/ENA)
The slow-start/enable pin provides two functions; first, the
pin acts as an enable (shutdown) control by keeping the
device turned off until the voltage exceeds the start
threshold voltage of approximately 1.2 V. When SS/ENA
exceeds the enable threshold, device start up begins. The
reference voltage fed to the error amplifier is linearly
ramped up from 0 V to 0.891 V in 3.35 ms. Similarly, the
converter output voltage reaches regulation in
approximately 3.35 ms. Voltage hysteresis and a 2.5-µs
falling edge deglitch circuit reduce the likelihood of
triggering the enable due to noise.
The second function of the SS/ENA pin provides an
external means of extending the slow-start time with a
low-value capacitor connected between SS/ENA and
AGND. Adding a capacitor to the SS/ENA pin has two
effects on start-up. First, a delay occurs between release
of the SS/ENA pin and start up of the output. The delay is
proportional to the slow-start capacitor value and lasts until
the SS/ENA pin reaches the enable threshold. The
start-up delay is approximately:
td + C
(SS)
1.2 V
5 mA
(25)
Second, as the output becomes active, a brief ramp-up at
the internal slow-start rate may be observed before the
externally set slow-start rate takes control and the output
rises at a rate proportional to the slow-start capacitor. The
slow-start time set by the capacitor is approximately:
t
(SS)
+C
(SS)
0.7 V
5 mA
(26)
The actual slow-start is likely to be less than the above
approximation due to the brief ramp-up at the internal rate.
VBIAS Regulator (VBIAS)
The VBIAS regulator provides internal analog and digital
blocks with a stable supply voltage over variations in
junction temperature and input voltage. A high-quality,
low-ESR, ceramic bypass capacitor is required on the
VBIAS pin. X7R or X5R grade dielectrics are
recommended because their values are more stable over
temperature. Place the bypass capacitor close to the
VBIAS pin and returned to AGND. External loading on
VBIAS is allowed, with the caution that internal circuits
require a minimum VBIAS of 2.70 V, and external loads on
VBIAS with ac or digital switching noise may degrade
performance. The VBIAS pin may be useful as a reference
voltage for external circuits.
Voltage Reference
The voltage reference system produces a precise Vref
signal by scaling the output of a temperature stable
bandgap circuit. During manufacture, the bandgap and
scaling circuits are trimmed to produce 0.891 V at the
output of the error amplifier, with the amplifier connected
as a voltage follower. The trim procedure adds to the high
precision regulation of the TPS54110, since it cancels
offset errors in the scale and error amplifier circuits.
Oscillator and PWM Ramp
The oscillator frequency can be set to internally fixed
values of 350 kHz or 550 kHz using the SYNC pin as a
static digital input. If a different frequency of operation is
required for the application, the oscillator frequency can be
externally adjusted from 280 kHz to 700 kHz by connecting
a resistor from the RT pin to ground and floating the SYNC
pin. The switching frequency is approximated by the
following equation, where R is the resistance from RT to
AGND:
SWITCHING FREQUENCY + 100 kW
R
500 kHz
(27)
External synchronization of the PWM ramp is possible
over the frequency range of 330 kHz to 700 kHz by driving
a synchronization signal into SYNC and connecting a
resistor from RT to AGND. Choose an RT resistor that sets
the free-running frequency to 80% of the synchronization
signal. Table 3 summarizes the frequency selection
configurations.
Table 3. Summary of the Frequency Selection
Configurations
SWITCHING
FREQUENCY
SYNC PIN
RT PIN
350 kHz, internally
set
Float or AGND
Float
550 kHz, internally
set
≥ 2.5 V
Float
Externally set 280
kHz to 700 kHz
Float
R = 68 k to 180 k
Externally
synchronized
frequency
Synchronization
signal
R = RT value for 80% of
external synchronization
frequency
Error Amplifier
The high-performance, wide-bandwidth, voltage error
amplifier sets the TPS54110 apart from most dc/dc
converters. The user is given the flexibility to use a wide
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TPS54110−Q1
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SLVS837 − JULY 2008
range of output L- and C-filter components to suit the
particular application needs. Type-2 or type-3
compensation can be employed using external
compensation components.
PWM Control
Signals from the error-amplifier output, oscillator, and
current-limit circuit are processed by the PWM control
logic. Referring to the internal block diagram, the control
logic includes the PWM comparator, OR gate, PWM latch,
and portions of the adaptive dead-time and control-logic
block. During steady-state operation below the
current-limit threshold, the PWM-comparator output and
oscillator pulse train alternately reset and set the PWM
latch. Once the PWM latch is set, the low-side FET
remains on for a minimum duration set by the oscillator
pulse duration. During this period, the PWM ramp
discharges rapidly to its valley voltage. When the ramp
begins to charge back up, the low-side FET turns off and
high-side FET turns on. As the PWM ramp voltage
exceeds the error-amplifier output voltage, the PWM
comparator resets the latch, thus turning off the high-side
FET and turning on the low-side FET. The low-side FET
remains on until the next oscillator pulse discharges the
PWM ramp.
During transient conditions, the error amplifier output
could be below the PWM ramp valley voltage or above the
PWM peak voltage. If the error-amplifier output is high, the
PWM latch is never reset and the high-side FET remains
on until the oscillator pulse signals the control logic to turn
the high-side FET off and the low-side FET on. The device
operates at its maximum duty cycle until the output voltage
rises to the regulation set-point, setting VSENSE to
approximately the same voltage as Vref. If the
error-amplifier output is low, the PWM latch is continually
reset and the high-side FET does not turn on. The low-side
FET remains on until the VSENSE voltage decreases to
a range that allows the PWM comparator to change states.
The TPS54110 is capable of sinking current continuously
until the output reaches the regulation set-point.
If the current-limit comparator remains tripped longer than
100 ns, the PWM latch resets before the PWM ramp
exceeds the error-amplifier output. The high-side FET
turns off and low-side FET turns on to decrease the energy
in the output inductor, and consequently the output
current. This process is repeated each cycle that the
current-limit comparator is tripped.
Dead-Time Control and MOSFET Drivers
Adaptive dead-time control prevents shoot-through
current from flowing in both N-channel power MOSFETs
during the switching transitions by actively controlling the
18
turn-on times of the MOSFET drivers. The high-side driver
does not turn on until the gate-drive voltage to the low-side
FET is below 2 V. The low-side driver does not turn on until
the voltage at the gate of the high-side MOSFETs is below
2 V. The high-side and low-side drivers are designed with
300-mA source and sink capability to quickly drive the
power MOSFETs gates. The low-side driver is supplied
from VIN, while the high-side driver is supplied from the
BOOT pin. A bootstrap circuit uses an external BOOT
capacitor and an internal 2.5-Ω bootstrap switch
connected between the VIN and BOOT pins. The
integrated bootstrap switch improves drive efficiency and
reduces external-component count.
Overcurrent Protection
Cycle-by-cycle current limiting is achieved by sensing the
current flowing through the high-side MOSFET and
differential amplifier and comparing it to the preset
overcurrent threshold. The high-side MOSFET is turned
off within 200 ns of reaching the current-limit threshold. A
100-ns leading-edge blanking circuit prevents false
tripping of the current limit. Current-limit detection occurs
only when current flows from VIN to PH when sourcing
current to the output filter. Load protection during
current-sink operation is provided by thermal shutdown.
Thermal Shutdown
The device uses the thermal shutdown to turn off the power
MOSFETs and disable the controller if the junction
temperature exceeds 150°C. The device is released from
shutdown when the junction temperature decreases to
10°C below the thermal-shutdown trip point, and starts up
under control of the slow-start circuit. Thermal shutdown
provides protection when an overload condition is
sustained for several milliseconds. In a persistent-fault
condition, the device cycles continuously; starting up
under control of the soft-start circuit, heating up due to the
fault, and then shutting down upon reaching the
thermal-shutdown point.
Power Good (PWRGD)
The power-good circuit monitors for undervoltage
conditions on VSENSE. If the voltage on VSENSE is 7%
below the reference voltage, the open-drain PWRGD
output is pulled low. PWRGD is also pulled low if VIN is
less than the UVLO threshold, or SS/ENA is low, or if
thermal shutdown asserts. When VIN = UVLO threshold,
SS/ENA = enable threshold, and VSENSE > 93% of Vref,
the open-drain output of the PWRGD pin is high. A
hysteresis voltage equal to 3% of Vref and a 35-µs
falling-edge deglitch circuit prevent tripping of the
power-good comparator due to high-frequency noise.
TPS54110−Q1
www.ti.com
SLVS837 − JULY 2008
PCB LAYOUT CONSIDERATIONS
pins, and minimize the area of the conductor to prevent
excessive capacitive coupling. Connect the boot capacitor
between the phase node and the BOOT pin as shown.
Keep the boot capacitor close to the IC and minimize the
conductor trace lengths. Connect the output-filter
capacitor(s) as shown between the VOUT trace and
PGND. It is important to keep the loop formed by the PH
pins, Lout, Cout, and PGND as small as is practical. Place
the compensation components from the VOUT trace to the
VSENSE and COMP pins. Do not place these
components too close to the PH trace. Due to the size of
the IC package and the device pin-out, they must be
somewhat closely routed while maintaining as much
separation as possible, yet keeping the layout compact.
Connect the bias capacitor from the VBIAS pin to analog
ground using the isolated analog ground trace. If a
slow-start capacitor or RT resistor is used, or if the SYNC
pin is used to select 350-kHz operating frequency, connect
them to this trace as well.
The VIN pins are connected together on the printed board
(PCB) and bypassed with a low-ESR ceramic bypass
capacitor. Minimize the loop area formed by the bypass
capacitor connections, the VIN pins, and the TPS54110
ground pins. The recommended bypass capacitor is 10-µF
(minimum) ceramic with X5R or X7R dielectric. The
optimum placement is closest to the VIN pins and the
AGND and PGND pins. See Figure 31 for a layout
example. It has an area of ground on the top layer directly
under the IC, with an exposed area for connection to the
PowerPAD. Use vias to connect this ground area to any
internal ground planes. Use additional vias at the ground
side of the input and output filter capacitors as well. Tie the
AGND and PGND pins to the PCB ground area under the
device as shown. Use a separate wide trace for the
analog-ground path, connecting the voltage set-point
divider, timing resistor RT, slow-start capacitor, and
bias-capacitor grounds. Tie the PH pins together and route
to the output inductor. Since the PH connection is the
switching node, locate the inductor very close to the PH
ANALOG GROUND TRACE
FREQUENCY SET RESISTOR
AGND
RT
SYNC
VSENSE
COMPENSATION
NETWORK
COMP
SS/ENA
PWRGD
BOOT
CAPACITOR
BOOT
SLOW START
CAPACITOR
VBIAS
Exposed
Powerpad
Area
BIAS CAPACITOR
VIN
PH
VIN
PH
VIN
PH
PGND
PH
PGND
PH
PGND
VIN
VOUT
LOUT
OUTPUT INDUCTOR
PH
COUT
OUTPUT
FILTER
CAPACITOR
INPUT
BYPASS
CAPACITOR
INPUT
BULK
FILTER
PGND
TOPSIDE GROUND AREA
VIA to Ground Plane
Figure 31. PC Board Layout Example
19
PACKAGE OPTION ADDENDUM
www.ti.com
10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
TPS54110QPWPRQ1
ACTIVE
HTSSOP
PWP
20
2000
RoHS & Green
NIPDAU
Level-3-260C-168 HR
-40 to 125
54110Q
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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